JP4162010B2 - System for coherent demodulation of binary phase shift keying (BPSK) signals - Google Patents

System for coherent demodulation of binary phase shift keying (BPSK) signals Download PDF

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JP4162010B2
JP4162010B2 JP2006053187A JP2006053187A JP4162010B2 JP 4162010 B2 JP4162010 B2 JP 4162010B2 JP 2006053187 A JP2006053187 A JP 2006053187A JP 2006053187 A JP2006053187 A JP 2006053187A JP 4162010 B2 JP4162010 B2 JP 4162010B2
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bpsk
ilo
phase
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JP2006246476A (en
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マルティ ホアン アイトール オソリオ
モリンズ ジョセップ イグナシ カイロ
コルドバ ホセ ハビエル シエイロ
ビリェガス ホセ マリア ロペス
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セイコーエプソン株式会社
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    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/18Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/22Demodulator circuits; Receiver circuits
    • H04L27/227Demodulator circuits; Receiver circuits using coherent demodulation
    • H04L27/2275Demodulator circuits; Receiver circuits using coherent demodulation wherein the carrier recovery circuit uses the received modulated signals
    • H04L27/2276Demodulator circuits; Receiver circuits using coherent demodulation wherein the carrier recovery circuit uses the received modulated signals using frequency multiplication or harmonic tracking

Description

  The present invention relates to a system for demodulation of binary phase shift keying (BPSK) signals.
  A general field of application of the invention is digital communication, in particular wireless digital communication.
  Digital phase shift keying (PSK) of sinusoidal signals is one of the most effective modulation techniques in terms of both noise immunity and required bandwidth. Nevertheless, demodulation of the PSK signal requires a complex demodulator system. Therefore, other digital modulation schemes that are inefficient, such as frequency shift keying (FSK) or amplitude shift keying (ASK), are usually preferred for simple demodulation.
  The simplest PSK signal is a binary PSK (BPSK) signal. In this case, the carrier phase is shifted according to the bitstream between two possible states: 0 ° and 180 °. A BPSK signal can be easily obtained by multiplying the carrier by +1 (0 ° phase state) or -1 (180 ° phase state). From the receiver's point of view, it is impossible to know if the phase of the incoming BPSK signal corresponds to a 0 ° state or a 180 ° state. The reason is due to the fact that the actual propagation path from the emitter to the receiver is usually unknown. To avoid this uncertainty, the information to be transmitted is encoded as a transition between phase states instead of being encoded as a fixed phase value. Thus, if a logic “1” is to be transmitted, the phase of the carrier signal is shifted while the phase is not changed for logic “0” and vice versa. The signal encoded in this way is known as differential BPSK (DBPSK). Note that there is no difference between BPSK and DBPSK from a signal perspective. The only difference between them is pre-processing (on the transmitter side) or post-processing (on the receiver side) of the baseband signal. FIG. 1 shows a BPSK or DBPSK signal generated as the product of a baseband signal (derived from a bitstream or derived from a processed bitstream) and a sine wave carrier of the desired frequency.
  The normal procedure for demodulating a BPSK signal is a coherent demodulation procedure. Basically, the demodulation process consists of multiplying the received signal by a reference signal having the same frequency as the original carrier wave.
The BPSK signal can be expressed mathematically as follows:
BPSK = ± Acos (ωt + ψ) (1)
  Here, the sign of + corresponds to the 0 ° phase state, and the sign of − corresponds to the 180 ° phase state. A is the amplitude of the received signal, and ψ is an arbitrary phase due to signal propagation.
The reference signal, S, is obtained as follows (the amplitude is set to 1 for simplicity):
S = cos (ωt) (2)
The product, or P, can be expressed as:
P = ± Acos (ωt + ψ) · cos (ωt) = ± A / 2cos (ψ) ± A / 2cos (2ωt + ψ) (3)
Ultimately, the following baseband representation is obtained by low pass filtering P.
P LPF = ± A / 2 cos (ψ) (4)
The result is a signal that reproduces the original modulation (±), ie, P LPF . From (4), if the propagation phase ψ is 0 ° or 180 °, the efficiency of the demodulation process reaches a maximum (regardless of phase uncertainty). On the other hand, if ψ = ± 90 °, the efficiency of the demodulation process is zero. This fact points to the first drawback of coherent demodulation of PSK signals, which is a propagation phase uncertainty. The second and most important drawback is the availability of a reference signal with exactly the same frequency as the original carrier.
  The usual way to overcome both problems is by using a carrier recovery circuit. Carrier recovery is obtained by using a synchronization loop. The square loop and the Costas loop are the most widely used. The characteristics and operation of the square loop and the Costas loop are shown in FIGS. 2 and 3, respectively.
  As shown in FIG. 2, the square loop is composed of a square block and a band pass filter (BPF), which makes it possible to generate a reference signal having a frequency twice the original carrier frequency from the BPSK input signal. Occurs without phase modulation. A phase locked loop (PLL) consisting of a phase / frequency detector, a loop filter and a voltage controlled oscillator (VCO) is used to recover a double frequency carrier. The original carrier is finally recovered using a divider that divides the frequency by two. Demodulation is accomplished by multiplying the recovered carrier by the incoming BPSK signal.
The Costas loop circuit consists of two mixers that generate the product of the incoming signal and two reference quadrature signals (0 ° / 90 °). A third mixer acting as a phase detector generates an error signal as the product of the low pass filtered outputs of both previous mixers. The error signal is finally passed through a loop filter (ie, an integrator) to generate a voltage controlled oscillator (VCO) control signal. When combined with a 90 ° phase shifter, the VCO generates a reference quadrature signal and closes the loop. If the frequency of the reference orthogonal signal is equal to the original carrier frequency, the error signal is zero. Moreover, the VCO output reference signal (in-phase signal) has the same propagation phase ψ of the carrier wave or is 180 ° different from it. That is, if the error function is zero in the synchronized state, the Costas loop acts as a demodulation of the BPSK signal. In practice, the baseband modulator signal is detected at the output of the first low-pass filter (LPF 1 in FIG. 3) (regardless of the code uncertainty).
  The main advantage of coherent demodulation performed by both of the previous schemes is input signal tracking. This allows correction of frequency deviations, for example due to relative movement between the emitter and receiver in a mobile system. Moreover, no previous information (ie bit period) about the modulated signal is required. However, the synchronization time is generally large and results in data loss at the beginning of communication or does not function properly in burst mode transmission. Another important drawback of the synchronization loop is the need for a loop filter that is difficult to implement in a monolithic form.
  As an example, a coherent demodulation procedure based on a Costas loop (as shown in FIG. 3) that detects the demodulator tuning state (phase tuning and accurate demodulation of the input signal) or pseudo-tuning state (incorrect modulation) Some use a coherent demodulation procedure supplemented by a series of additional components (see Patent Document 1).
  Some have proposed other procedures to obtain a phase reference that allows demodulation (see Patent Document 2). In this case, an average of the received phasors is performed, thereby obtaining a phase reference estimate. In order to demodulate the signal, each received phasor is compared to a reference and then used to narrow down the phase reference estimate. This procedure has the advantage that a discontinuously received signal can be accurately demodulated without loss of information associated with tuning time. The disadvantage is the high complexity of the demodulator system, a potential requirement to know the modulation signal bit period in order to perform phasor averaging.
  Another feasible demodulation procedure has also been proposed for signals using digital phase modulation (see Patent Document 3). This method can be applied to digital phase modulated signals that contain changes only between adjacent phase states. Basically, the operating principle consists of multiplying a signal received in one period by the signal received in the previous period. The time difference is obtained by using delay components and is adjusted to be equal to the bit time. In order to generate a DC component of the obtained signal, the result of this multiplication is filtered by a low-pass filter. There is a change in the value of the DC component only when there is a phase change during the bit period. In this case, demodulation is directly performed without requiring synchronization. The basic disadvantage is that the modulation signal bit period needs to be known in advance.
US Pat. No. 5,347,228 U.S. Pat. No. 4,613,486 U.S. Pat.
  With respect to the background described, the present invention has the advantage of coherent demodulation (input signal tracking and modulation signal bit period) without requiring explicit use of frequency and phase locked loops (PLL or Costas loop). Independent demodulation processing).
  The basic operating principle of the present invention is to synchronize the resonant circuit by injecting superharmonics in order to recover the carrier wave of the BPSK signal. In this way, carrier recovery is achieved by injection of superharmonics that synchronize the oscillator without the need for an external feedback path. As a result, no loop filter is required and the resulting structure is suitable for monolithic integration.
  The invention refers to a system for demodulation of a binary phase shift keying (BPSK) signal according to claim 1 and a method according to claim 6. Preferred embodiments of the system and method are defined in the dependent claims.
A first aspect of the invention relates to a system for coherent demodulation of a binary phase shift keying (BPSK) signal having a frequency f. This system for demodulation is
Means for recovering a carrier signal (C) of frequency 2f from the BPSK signal;
A injection locking oscillator ILO, which have approximately equal natural resonant frequency f r to the frequency f, (θ e -k) / 2 phase shift, wherein, θ e = arcsin ((f r -f) / (αA i f)), α and k are parameters that depend on the type of dominant nonlinearity in the injection locked oscillator ILO, and A i is the amplitude of the recovered carrier signal at a frequency of 2f. the injection locking oscillator ILO for generating a differential output O p and O n signal to recover the original carrier with a means for injecting said signal having a frequency 2f,
In order to generate a demodulated signal (DEMOD), it has a copy of the incoming BPSK signal, and means for combining the differential output O p and O n signals.
If f r is not approximately equal to f, the yield of the coherent demodulator is lower than if f r ≈f, but this demodulator is also useful.
The operating principle of the present invention, both the injection locking oscillator ILO frequency and phase of when the signal having a frequency close to the second harmonic of the natural resonant frequency f r is injected, or the division circuit according to the second argument It is a synchronization phenomenon. According to what has been established and verified by the inventor, this argument (frequency and phase) synchronization phenomenon is due to a nonlinear response that is more or less present in the components used in the ILO circuit.
The following can be pointed out as general sources of nonlinearity.
a) When a varactor diode is used, the capacitance of the varactor diode to which a bias voltage is applied varies.
b) If a bipolar transistor is used, the capacitance variation of the base-emitter and base-collector coupling of the bipolar transistor.
c) If MOSFET transistors are used, the gate transistor's gate-source, gate-drain and gate-substrate capacitance variations.
d) The drain current for MOSFET transistors and the base-collector current for bipolar transistors, depending on the polarization voltage according to the law of squares or higher.
Non-linearity is responsible for mixing harmonics and then generating new spectral components. When a signal having a frequency 2f close to 2f r (where f r is the ILO natural resonance frequency) is injected into the ILO, the non-linearity (especially the second-order non-linearity) becomes the frequency 2f−f r. ≒ to f r (voltage and / or current) leads to additional components. Since this component is added to components already present at the same frequency, the ILO resonance characteristics are changed. In analytical it is that it can represent the change in ILO operating conditions as variation Delta] f r of the resonance frequency is also revealed experimentally. This amount of change Δf r is,
Δf r = αA i fSin (θ) (5)
Given by. Where α is a parameter that depends on the type of dominant nonlinearity, A i is the amplitude of the input signal at a frequency of 2f, and the angle θ is
θ = 2φ (t) −Φ + k (6)
Can be expressed as Here, Φ and φ (t) are the phases of the input and output signals, respectively, and t is time. The value of k also depends on the dominant nonlinearity in the circuit. For example, if the nonlinearity is due to a current that varies with the bias voltage, k = 0, and if the nonlinearity is due to a variable capacitor, k = π / 2.
Further, the O p and O n output from the ILO,
O p = Bcos (2πft + φ (t)), O n = O p + π (7)
Can be expressed as Where B is the amplitude of the output signal and φ (t) is
[Equation 1]
Prove that.
Combining (5) and (6) with (8) provides a differential equation that determines the dynamic response of the ILO to the injected input signal. If dφ / dt = 0, a stable state (lock-in state) is obtained. Or, if the frequency of the output signal is exactly half the frequency of the input signal, and therefore Δf r = f−f r , the same thing is said in another way.
By substituting this condition in (5), two possible values of equilibrium are obtained for the angle θ. These values are expressed as θ e = arcsin ((f r −f) / (αA i f)) and θ m = π−θ e (9)
Can be expressed as
This indicates that the first possibility, ie, θ e corresponds to a stable equilibrium state, and the second possibility, ie, θ m, is a metastable equilibrium state. If the input signal has a frequency close to twice the natural resonance frequency of the ILO, the stable equilibrium angle θ e will be shorter.
  From (6) it can be inferred that there is an uncertainty of π radians that the synchronization condition is not unique to the output phase φ and is only a mathematical result of argument division by 2 performed by the ILO circuit. .
And a copy of the incoming BPSK signal, means for combining the differential output O p and O n signals,
A copy of the incoming BPSK signal, and the same frequency, multiplied very similar signals i 1, i 3 the injection locking oscillator ILO differential output O p and O n signal having an amplitude and phase, Means Mix 1 , Mix 2 for generating output IF 1 and IF 2 signals, respectively;
Means LPF 1 , LPF 2 for low-pass filtering the output IF 1 and IF 2 signals to generate baseband signals BB p , BB n , respectively;
Means for subtracting the baseband signal to generate the demodulated signal DEMOD.
  Preferably, the means for recovering the carrier signal C having the frequency 2f includes a squaring circuit.
  Preferably, the system for demodulation has a band pass filter block connected between the squaring circuit block and the injection locked oscillator (ILO).
BPSK = ± Acos (2πft + ψ) A general BPSK signal of frequency f that can be expressed as (10) is squared and bandpass filtered to obtain carrier C of frequency 2f. Carrier C is
C = (A 2/2) cos (4πft + 2ψ) (11)
Given by.
Considering equation (6), if replaced by 2ψ the [Phi, it is possible to obtain the relationship between the lock-in condition below between the phase ψ of ILO phase phi e and the input BPSK signal output O p.
φ e = ψ + (θ e −k) / 2 + nπ; n = 0, 1, 2,... (12)
That, ILO output O p (Similarly, O n) recovers the original carrier with the uncertainty (θ e -k) / 2 phase shift and π phase.
According to the phase relationship of (12), the outputs IF 1 and IF 2 of Mix 1 and Mix 2 are
IF 1 = ± ABcos (2πft + ψ) · cos (2πft + φ e) (13)
IF 2 = ± ABcos (2πft + ψ) · cos (2πft + φ e + π) (14)
Can be obtained after low pass filter processing,
BB p = ± AB / 2 cos [(θ e −k) / 2 + nπ] (15)
BB n = ± AB / 2 cos [(θ e −k) / 2 + (n + 1) π] (16)
Can be obtained.
It should be noted that either BB p or BB n is a binary signal (complementary to each other) and that the sign change has already reproduced the phase change of the input BPSK signal. However, due to mismatch or asymmetry, these signals can be affected by common mode offsets that can affect the normal operation of the next stage (ie saturate the baseband amplifier or signal regenerator). . To avoid this problem, both signals are subtracted to generate the final demodulated output DEMOD. This output DEMOD is
DEMOD = ± ABcos [(θ e −k) / 2 + nπ] (17)
Can be expressed as
The maximum efficiency of the demodulation process corresponds to the case of θ e = k. Under these conditions, DEMOD = ± AB · (± 1).
Depending on the dominant non-linearity, two distinct cases can be distinguished: a) and b).
a) Non-linear current (k = 0).
In this case, the maximum efficiency of the demodulation process can be obtained when θ e = 0. From (9), this corresponds to f = f r. This is also a requirement for the maximum sensitivity of the synchronization process (ie, minimum injection power is required to phase lock the ILO).
b) Non-linear capacitance (k = π / 2).
Here, the maximum efficiency is obtained when θ e = π / 2. However, according to (9), this is, the frequency f, corresponding to f r in the synchronization limits. That is, (f r −f) / (αA i f) = 1. For example, the deviation from the original value of the natural resonant frequency f r due to noise or drift in the component characteristics, the synchronization to zero. If you expect maximum synchronization sensitivity (ie, θ e = 0) instead of maximum demodulation efficiency,
[Equation 2] DEMOD = ± AB · (√2 / 2),
That is, 70% of the maximum efficiency. Therefore, a tradeoff should be established between maximum demodulation efficiency and maximum synchronization sensitivity, or a delay path is included to compensate for phase deviation for optimal synchronization and optimal demodulation. The delay block can be located anywhere in the chain from i 2 to C in FIG. 4 or simultaneously in the paths i 1 and i 3 . In the first case, the delay path generates a phase shift of π / 2 (half of this value when connected before the square stage), and in the next case a phase shift of −π / 2. There is a need.
  A second aspect of the invention relates to a method for coherent demodulation of a BPSK signal of frequency f, based on synchronization of an oscillator by injection of a signal having a frequency of 2f.
When injecting a signal having a frequency of 2f, and synchronizes the oscillator when a natural resonant frequency f r substantially equal oscillator f.
The method for coherent demodulation of a BPSK signal of frequency f is
Recovering a carrier signal (C) of frequency 2f from the BPSK signal;
θ e = arcsin ((f r −f) / (αA i f)), α and k are parameters depending on the type of dominant nonlinearity in the injection locked oscillator (ILO), and A i is In order to recover the original carrier having a phase shift of (θ e −k) / 2, which is the amplitude of the recovered carrier signal having a frequency of 2f, the injection-locked oscillator (ILO) is supplied with the signal having the frequency 2f. Injecting, and
In order to generate a demodulated signal (DEMOD), it has a BPSK signals that have entered, and a step of coupling the differential output (O p, O n) and a signal.
  The basic operating principle of the present invention is to recover the carrier wave of the BPSK signal by synchronizing the resonance circuit by superharmonic injection, thus synchronizing the oscillator without requiring an external feedback path. Since carrier recovery is achieved by superharmonic injection, the result is that no loop filter is required and the resulting structure is suitable for monolithic integration.
The present invention refers to a system for demodulation of binary digital phase shift keying (BPSK) signals. FIG. 4 shows one possible implementation of the demodulator system. This demodulator system can be divided into the following sections:
(A) Output divider PDIV. The input of the output divider PDIV is a BPSK phase modulation signal of frequency f, where f is the frequency of the carrier signal. This output divider generates output signals i 1 , i 2 , i 3 having the same frequency f as the input signal. Moreover, i 1 and i 3 have equal amplitude and are in the same phase state. This phase state can be the same as the input signal, or the phase state can have a particular phase imbalance or delay that is both the same. The amplitude and phase state of signal i 2 can be the same as in signals i 1 and i 3 , or the amplitude and phase state can have a specific amplitude and / or phase imbalance. This output divider can be either passive or active.
(B) A square circuit block. This squaring circuit block can be implemented using an active or passive circuit having a quadratic term in the transfer function from input to output. Specific examples of these circuits are mixers that operate as full-wave diode rectifiers or analog multipliers.
(C) Band pass filter (BPF). The band pass filter selects an appropriate component of the frequency 2f from the output of the square circuit block as necessary.
(D) Injection locked oscillator (ILO). The ILO acts as a division analog argument divider by 2, (when there is no injection signal) natural resonant frequency is f r. The ILO is the differential output signal O p of the frequency f, to generate O n. The differential output phase is fixed by the signal C having the frequency 2f according to the equation (6).
(E) Two mixers Mix 1 and Mix 2 . These mixers Mix 1 and Mix 2 are passive or active and are coupled with low-pass filters LPF 1 and LPF 2 to downconvert the BPSK input signal to baseband signals BB p and BB n .
(F) Subtractor. This subtractor is either passive or active and generates a DEMOD output from the baseband signals BB p , BB n .
FIG. 5 shows the BPSK input signal measured, a temporal relationship between the output signal S 2 of the squaring circuit block. In this case, a commercial frequency doubler circuit has been used to generate the S 2 signal.
The injection locked oscillator (ILO) of FIG. 4 can be implemented in several ways. FIG. 6 includes a preferred but not limited embodiment of an ILO circuit. As already described, the principle based on the frequency division processing is the frequency and phase synchronization phenomenon of the resonance circuit when a signal having a frequency close to the second harmonic of the fundamental frequency is injected. This circuit consists of the following sections:
(A) Bias T circuit BT. The purpose of this bias T circuit BT is to couple the injected signal i of frequency 2f with the continuous DC bias required for resonant circuit operation.
(B) Inverter transformer T1. This inverter transformer T1 has primary and secondary windings connected at one end to the bias network output and connected at the other end to varactor diodes V1, V2.
(C) The described varactor diodes V1, V2. The anodes of these varactor diodes V1, V2 are connected to the control voltage Vc.
(D) Two cross-coupled transistors Q1, Q2.
(E) Differential output Op, On.
(F) Current source S1. This current source S1 ensures accurate transistor polarization.
  The frequency / phase synchronization process that is characteristic of this type of divider circuit is faster than the process associated with a square or Costas loop because the frequency / phase synchronization process is inherent in the component, It must be noted that this is because it is not inherent in the synchronous circuit as a whole.
The transformer and the two varactor diodes form a resonant tank circuit, and the resonant frequency of the resonant tank circuit is fixed by the value of the control voltage Vc. These varactor diodes can be replaced by fixed value capacitors, in which case the possibility of controlling the resonant frequency is lost. The purpose of the cross-coupled transistor pairs (these are MOSFETs in FIG. 6, but they can be bipolar) is to obtain sufficient gain to compensate for resonant tank circuit losses, the vibration of constant amplitude in order to generate at the resonant frequency f r. If the injection signal has sufficient power, the tank's resonance characteristics will change. This is due to the non-linear behavior of the varactor diode response and / or the amplifier stage transistors. The new resonant frequency is tuned to half the frequency of the injected signal and the phase is adjusted to one of two possible values with a 180 ° difference.
  FIG. 7 shows a measured spectrum of one of the ILO outputs (Op or On) before the injection (free-running) and in the locked-in state (Locked) after injecting an input signal having a frequency of 506 MHz. Note that the natural frequency of 255.5 MHz is shifted by -2.5 MHz due to synchronization.
  FIG. 8 shows a measured time-domain waveform of an input waveform C of 2f ILO and one of f ILO outputs (Op or On). Note the phase relationship of the lock-in state between the fundamental vibration of f and the second harmonic of 2f.
  FIG. 9 shows measured time-domain waveforms of the BPSK signal having the frequency f and the differential outputs Op (thick lines) and On (dashed lines) of the ILO having the frequency f. It should be noted that the BPSK signal is in phase with the Op output before the 180 ° phase change and then in phase with the On output.
  FIG. 10 shows the BPSK input signal together with the demodulated output DEMOD. In the example shown, the fall time of the DEMOD signal is about 15-20 ns, which means a maximum demodulation rate of about 50-60 Mbits / s.
  FIG. 11 shows a DEMOD output corresponding to a BPSK input signal that changes the phase by 180 ° every 500 ns.
FIG. 3 is a diagram schematically showing generation of a BPSK signal. It is a diagram which shows a square loop. It is a diagram which shows a Costas loop. FIG. 2 shows a preferred BPSK demodulator according to the present invention. And BPSK signal of the frequency f, a measured time domain waveforms of the S 2 signal frequency 2f. FIG. 2 is a diagram illustrating a preferred embodiment of an injection locked oscillator (ILO) using a non-linear varactor diode. It is a measured spectrum of one (Op or On) of the output of the ILO before injection (free-running) of the 2f input signal C and the locked-in state (Locked). It is a time-domain waveform measured with an input waveform C of 2f ILO and one of the ILO outputs of f (Op or On). It is the time-domain waveform which measured the BPSK signal of frequency f, and the differential outputs Op (thick line) and On (dashed line) of ILO of frequency f. It is a graph which shows a BPSK input signal with demodulated output DEMOD. It is a graph which shows the DEMOD output corresponding to the BPSK input signal which changes a 180 degree phase every 500 ns.
Explanation of symbols
PDIV power dividers i 1, i 2, i 3 the output signal S 2 output signals of the squaring circuit block C carrier signal O p, O n differential output signal BPF bandpass filter ILO injection-locked oscillator Mix 1, Mix 2 mixer IF 1 , IF 2 mixer output LPF 1 , LPF 2 low-pass filter BB p , BB n baseband signal DEMOD demodulated signal

Claims (5)

  1. A system for coherent demodulation of a binary phase shift keying (BPSK) signal of frequency f, comprising:
    Means for recovering a carrier signal (C) of frequency 2f from the BPSK signal;
    A injection locking oscillator having a substantially equal natural resonant frequency f r to the frequency f (ILO), (θ e -k) / 2 phase shift, wherein, θ e = arcsin ((f r -f) / ( αA i f)), α and k are parameters that depend on the type of dominant nonlinearity in the injection locked oscillator (ILO), and A i is the amplitude of the recovered carrier signal at a frequency of 2f. said differential output to recover the original carrier with a phase shift (O p, O n) for injection locking oscillator for generating a signal (ILO), and means for injecting said signal having a frequency 2f is,
    In order to generate a demodulated signal (DEMOD), a copy of the incoming BPSK signal and the differential output (O p, O n) means for combining the signals,
    Having a system.
  2. A system for demodulation according to claim 1, and a copy of the BPSK signals that have the entered, the differential output (O p, O n) wherein the means for coupling the signal,
    A copy of the incoming BPSK signal having the same frequency and very similar amplitude and phase (i 1 , i 3 ) is converted to a differential output signal (O p ) of the injection locked oscillator (ILO). and means for a means for multiplying the O n) (Mix 1, Mix 2), and generates output (IF 1, IF 2) signals, respectively,
    Means (LPF 1 , LPF 2 ) for low-pass filtering the output (IF 1 , IF 2 ) signal to generate baseband signals (BB p , BB n ), respectively;
    Means for subtracting the baseband signal to generate a demodulated signal (DEMOD);
    Having a system.
  3. A system for demodulation according to claim 1 or 2, wherein the means for recovering the carrier signal (C) of frequency 2f comprises a squaring circuit.
  4. 4. The system for demodulation according to claim 3, further comprising a band-pass filter block connected between the square circuit block and the injection locked oscillator (ILO).
  5. A system for demodulation according to any one of claims 2 to 4, a system means (Mix 1, Mix 2) is equal to the multiplying.
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