JP4155152B2 - AC rotating electrical equipment - Google Patents

Description

  The present invention relates to an AC rotating electrical machine device, and more particularly to a quiet AC rotating electrical machine device with low magnetic noise.

  In recent years, electric vehicles, hybrid vehicles, fuel cell vehicles, and the like have become practical or development levels. In these automobiles, a high-output AC rotating electric machine is a main element for generating driving power, but such a high-output AC rotating electric machine has a problem that its magnetic noise is large. In addition, silent AC motors are desired in various applications, and various techniques for reducing magnetic noise in AC rotating electrical machines have been proposed. For example, Patent Document 1 proposes to reduce torque pulsation and noise caused by superimposing a voltage component that cancels a harmonic component included in an induced voltage of an electric motor on an output voltage of an inverter.

In addition, for example, Patent Document 2 proposes to reduce the torque pulsation by actively superimposing the harmonic component on the fundamental frequency component of the energization current.
Japanese Patent No. 2928594 Japanese Patent Laid-Open No. 11-55986

  However, in spite of the conventional harmonic superposition type magnetic noise reduction technology described above, it has not been said that the reduction in noise of the AC rotating electric machine has reached a satisfactory level. The reason for this will be described below.

  Since noise is generated by mechanical vibration of the vibrating body (here, the core of an AC rotating electrical machine), if vibration energy having the same frequency, opposite phase, and equal amplitude as the vibration energy of the frequency to be damped is added, It is clear that vibration at frequency can be canceled. In addition, as described above, various knowledge has been asserted for a long time as the magnetic noise can be reduced by adding some harmonic component to the fundamental frequency component of the armature current.

  However, in these prior art documents, in order to reduce magnetic noise (acoustic energy) of a predetermined frequency of an AC rotating electric machine, what frequency, phase and amplitude should be superimposed on the armature coil, that is, reduce it. The quantitative relationship between power sound (vibration energy) and energization current is not specifically described, and it is unclear what current should be passed to reduce magnetic noise at a specific frequency. Furthermore, the relationship between this magnetic noise and the current for suppressing it should vary depending on the structure of the AC rotating electrical machine and its usage. As a result, the above harmonic superposition type magnetic noise reduction technology is the philosophy. Although it was known as, nothing was put into practical use yet.

  In view of this point, the present applicant has succeeded in clarifying the relationship between magnetic noise and its suppressed harmonics according to Japanese Patent Application No. 2002-303650, and thereby AC that can significantly suppress magnetic noise than before. Succeeded in realizing a rotating electrical machine.

  However, in this magnetic noise reduction technology, since it is necessary to superimpose a harmonic component of a predetermined order, a predetermined phase, and a predetermined amplitude on the fundamental frequency component of the phase current for generating the necessary torque, a complicated calculation function Therefore, it is necessary to provide a feedback circuit system having a complicated structure of the control circuit. In addition, the complexity of the control circuit configuration leads to calculation errors and calculation failures due to a decrease in reliability due to approach noise, etc., and the fundamental frequency component of the phase current itself deviates from the original waveform, so that the required torque cannot be obtained. The possibility that.

  The present invention has been made in view of the above problems, and an object of the present invention is to provide a quiet AC rotating electric machine while suppressing the complexity of the control system and the decrease in reliability.

The AC rotating electrical machine apparatus according to claim 1 is an AC rotating electrical machine having an armature winding having a phase winding of m (m is a positive integer of 3 or more) phase, and a predetermined basic frequency for each phase winding. And a motor current control means for individually energizing each phase current having an amplitude,
Each of the phase windings is divided into a first phase winding group and a second phase winding group each composed of a plurality of the phase windings connected so as to be capable of passing a phase current, and the motor current control means includes: A first inverter that forms a basic phase current of each phase mainly including a fundamental frequency component of the phase current and energizes the first phase winding group for each phase; and for reducing magnetic noise caused by the phase current And a second inverter that forms a harmonic current and energizes the second phase winding group for each phase.

  That is, in the present invention, since the basic frequency component of the phase current for torque generation and the harmonic current for magnetic noise reduction are employed in an armature coil configuration that supplies current to different phase winding groups, The fundamental frequency component and the harmonic component can be controlled by separate inverters. Therefore, each inverter only needs to control a current of a single frequency or a small frequency, and the control system of each inverter can be simplified and the operation reliability can be improved.

  According to a second aspect of the present invention, in the AC rotating electrical machine apparatus according to the first aspect, the second inverter is a radial vibration that is a vibration generated radially about the axis of the rotating shaft of the AC rotating electric machine. The radial direction of each phase mainly composed of the n + 1st order (times) harmonic component of the fundamental frequency component of the phase current at the phase and amplitude for attenuating the nth order (times) harmonic component of the fundamental frequency component A vibration-reducing harmonic current is supplied to the second-phase winding group for each phase. This has made it possible for the first time to satisfactorily reduce the magnetic noise of AC rotating electrical machines of various sizes and arbitrary output states.

  This will be described in more detail below.

  The magnetic noise is caused by vibration (also referred to as magnetic vibration) formed by the magnetic force (magnetic excitation force) of the iron core of the AC rotating electric machine, and this magnetic vibration is a combined vibration of the circumferential vibration and the radial vibration.

  The circumferential vibration of the iron core causes torque ripple, but the stator core or rotor core has a substantially cylindrical or columnar shape. Therefore, even if these iron cores vibrate periodically in the circumferential direction, The vibration of the air in contact with the magnet, that is, the magnetic noise, is small. On the other hand, the radial vibration of the iron core causes the radial vibration of the outer peripheral surface or inner peripheral surface of the stator core or rotor core, but the outer peripheral surface or inner peripheral surface is in contact with air, so the stator core Alternatively, the outer peripheral surface or the inner peripheral surface vibrates in the radial direction due to the radial vibration of the rotor iron core, generating a large magnetic noise. That is, torque pulsation is mainly reduced by reducing the circumferential component of the magnetic excitation force, and magnetic noise is mainly reduced by reducing the radial component of the magnetic excitation force.

  In view of this knowledge, in the present invention, a harmonic of a predetermined order of a radial component (also referred to as a radial magnetic excitation force) of a magnetic excitation force that is normally formed by a rotor magnetomotive force and a stator current (fundamental frequency component). In order to reduce the wave component, the amplitude of the vector sum with the harmonic component of this magnetic excitation force decreases (becomes smaller than the original state), and in order to add a magnetic excitation force having a phase and amplitude, A harmonic vibration reducing harmonic current having an order larger by 1 than the order of the magnetic excitation force (radial magnetic vibration component) is superimposed on the stator current (multiphase AC current). In this way, it was found that magnetic noise can be reduced satisfactorily.

  In addition, by superimposing this radial vibration reducing harmonic current, in addition to the radial component of the magnetic excitation force formed by the rotor magnetomotive force and the stator current (fundamental frequency component), other radial directions of the AC rotating electrical machine Vibrations such as radial vibrations input from the outside can also be reduced.

  That is, according to the present invention, the radial vibration reducing harmonic current having a frequency of n + 1 times the fundamental frequency is energized separately from the energizing current for torque formation, so that the frequency is n times the fundamental frequency of the energizing current. It has been found that the harmonic component of radial vibration with can be reduced. The detailed reason will be described later.

  According to a third aspect of the present invention, in the AC rotating electrical machine apparatus according to the second aspect of the invention, the second inverter has the diameter mainly including an odd-order harmonic component having a seventh or higher order with respect to a fundamental frequency component of a three-phase current. A harmonic current for reducing directional vibration is formed.

  As a result, even-order radial vibration (magnetic noise) of 6 or more can be satisfactorily reduced with respect to the fundamental frequency component of the stator current of the three-phase AC rotating electric machine that makes the main magnetic noise.

  According to a fourth aspect of the present invention, in the AC rotating electrical machine apparatus according to the third aspect of the present invention, the second inverter forms the radial vibration reducing harmonic current mainly including a seventh harmonic component. Yes. As a result, it has been found that the sixth-order radial vibration (magnetic noise), which is the largest magnetic noise component in hearing in a three-phase AC rotating electric machine, can be satisfactorily reduced.

  According to a fifth aspect of the present invention, in the AC rotating electrical machine apparatus according to the third aspect, the second inverter further forms the radial vibration reducing harmonic current mainly including a 13th harmonic component. Yes. Thus, it has been found that the 12th order radial vibration (magnetic noise), which is an auditory main magnetic noise component in the three-phase AC rotating electric machine, can be satisfactorily reduced.

  According to a sixth aspect of the present invention, in the AC rotating electrical machine apparatus according to the third aspect of the present invention, the second inverter forms the radial vibration reducing harmonic current mainly including seventh and thirteenth harmonic components. It is characterized by that. As a result, it was found that the 6th and 12th order radial vibrations (magnetic noise), which are the major magnetic noise components in hearing in the three-phase AC rotating electric machine, can be satisfactorily reduced.

Structure of claim 7 further in AC rotary electric machine according to claim 2, wherein said second inverter, a vibration reduction operation mode for forming said radial vibration reducing harmonic current at the time of small output less than the predetermined value The second phase winding is formed by forming a basic phase current of each phase mainly including a fundamental frequency component of the phase current in place of the radial vibration reducing harmonic current when operated and having a large output of a predetermined value or more. It is characterized by being operated in an output priority mode in which a line group is energized for each phase. In this way, for example, in an AC rotating electrical machine for a vehicle, it is possible to perform a silent operation at the time of partial load operation of 50% or less that occupies most of the operation time, and to cope with a large output operation. The increase in size of the rotating electrical machine and the motor current control means can be suppressed.

  In a preferred aspect (Claim 8), the second inverter superimposes the 19th-order radial vibration reducing harmonic current to thereby convert the harmonic component of the radial vibration having an 18th-order frequency into the harmonic component. Attenuate more than when no superposition is performed. Thereby, a quiet operation of the rotating electrical machine is possible.

  In a preferred aspect (Claim 9), the second inverter superimposes the 25th-order harmonic vibration-reducing harmonic current, thereby generating a harmonic component of the radial vibration having a 24th-order frequency. Attenuate more than when no superposition is performed. Thereby, a quiet operation of the rotating electrical machine is possible.

  In a preferred embodiment, when I1 is the amplitude of the fundamental frequency, Im is the amplitude of the harmonic current, and t, x, and y are predetermined phase angles, the fundamental frequency component of the polyphase alternating current is: The first phase fundamental frequency component Iu1 (= I1sin (θ)), the second phase fundamental frequency component Iv1 (= I1sin (θ−x)), and the third phase fundamental frequency component Iw1 (= I1sin (θ−y)) are at least In addition, the harmonic vibration-reducing harmonic current having a frequency of m (= n + 1) times is a first-phase harmonic component Ium (= I1sinm (θ + t)) and a second-phase fundamental frequency component Ivm (= I1sinm (θ + t)) -X) and at least a third phase fundamental frequency component Iwm (= I1sinm (θ + t) -y), and the first phase harmonic component Ium is added to the first fundamental frequency component Iu1 and the second phase harmonic component Ivm. Is the second fundamental frequency component Iv1 and the third phase harmonic component Iwm is It is superimposed on the serial third fundamental frequency component Iw1. In this way, the rotation order of the fundamental frequency component of each phase, that is, the phase order matches the rotation order of the harmonic component for reducing the (n + 1) th order vibration of each phase. Can be reduced.

In the above description, in order to reduce radial vibration of m-1 times the frequency of the fundamental frequency component of the stator current, the harmonic phase current for the frequency of m times the phase current of the fundamental frequency component. It was explained that is superimposed. Of course, a harmonic current having a lower order than that may be superimposed in order to further reduce the radial vibration of a different order.
(Modification)
Hereinafter, various aspects of the present invention will be described.

  In a preferred embodiment, harmonic components of sixth and twelfth radial vibrations and harmonics of sixth and twelfth radial vibrations by a fundamental wave and harmonic currents for reducing seventh and thirteenth radial vibrations. The phase and amplitude of the seventh-order and thirteenth-order radial vibration reducing harmonic currents are set so that the amplitude of the vector sum with the component is not more than a predetermined value. The phase and amplitude of the seventh-order and thirteenth-order radial vibration reducing harmonic currents may be obtained by mathematical calculation, may be calculated by a finite element method, or may be experimentally determined. In each of the above-described configurations, the reduction in radial vibration has been described. However, a harmonic current for reducing the circumferential magnetic excitation force (torque ripple) may be supplied to the second inverter. As an AC rotating electric machine, an induction machine can be adopted in addition to a synchronous machine having various configurations, and either an electric mode or a power generation mode may be used. The harmonic current for reducing radial vibration may be applied to the second inverter in all rotation ranges, and in particular, the harmonic current for reducing radial vibration is supplied to the second inverter only in the rotation range where magnetic noise is a problem. May be. A predetermined one-order radial vibration reducing harmonic current may be reduced by applying a predetermined one-order radial vibration reducing harmonic current to the second inverter, and a plurality of orders of radial vibration reducing harmonics may be reduced. A plurality of orders of radial vibration may be reduced by passing a current through the second inverter. Depending on the rotation range, the energization of the radial vibration reducing harmonic current to the second inverter and the energization of the torque ripple reducing harmonic current to the second inverter may be switched.

  N + 1, which is the order of the harmonic current for reducing radial vibration described above (that is, the magnification of the frequency of the harmonic current for reducing radial vibration with respect to the frequency of the fundamental frequency component) is the manufacturing tolerance of the harmonic current generating circuit. Of course it can be included. For example, n + 1 may be in the range of (n + 1) −0.1 to (n + 1) +0.1.

  In each of the above-described configurations, reduction of radial vibration has been described, but at the same time, a harmonic current for reducing circumferential magnetic excitation force (torque ripple) can be further superimposed.

Hereinafter, preferred embodiments of the present invention will be described with reference to the drawings.
(Principle explanation)
Hereinafter, the principle when the present invention is applied to a multiphase AC rotating electrical machine will be described.

  FIG. 1 is a diagram schematically showing a magnetic circuit for one phase of an N-phase AC rotating electric machine, and FIG. 2 is an equivalent magnetic circuit diagram of FIG. In the synchronous machine, the magnetic flux φ is formed by the magnetic pole of the rotor (formed by a coil or a permanent magnet), the rotor magnetomotive force Fmag is the magnetomotive force of the magnetic pole of the rotor in the magnetic circuit, ie, the magnetic field strength, and the stator magnetomotive force Fcoil is This is the magnetomotive force, that is, the magnetic field strength formed in the magnetic circuit by the current. Rg is the magnetoresistance of the gap between the stator and the rotor.

  From FIG. 1 and FIG. 2, the magnetic flux φ, magnetic energy W, and radial magnetic excitation force f per phase are

Defined by That is, the magnetic excitation force f is defined as the sum of the square of the rotor magnetomotive force, the square of the stator magnetomotive force, and the product of the rotor magnetomotive force and the stator magnetomotive force.

  In the above figures and equations, Icoil is the stator current (phase current of the armature), x is the gap width, S is the gap facing area, μ0 is the air permeability, N is the number of turns of each phase coil of the armature, N Is the number of phases.

  The rotor magnetomotive force Fmag and the stator current (phase current) Icoil of the first (U in three phases), second (V in three phases), and N (W in three phases) phases of this multiphase AC rotating electric machine The

Shown in The rotor magnetomotive force is described by a trigonometric function to represent the rotor rotation. Since those skilled in the art can easily understand, the description of the rotor magnetomotive force Fmag and the stator current Icoil of the remaining phases in the case of more than three phases is omitted. Naturally, in the three phases, x is 360 / N = 120 degrees.

  Note that Fi (i is a subscript) is the amplitude of the i-order component of the rotor magnetomotive force, Ii is the amplitude of the i-order component of the stator current, θ is the rotor rotation angle, α, β, γ, δ, s, t, and u are phase angles, and j, k, L, m, and n are integer values.

  In these equations, the rotor magnetomotive force Fmag has only harmonic components of j, k and L as harmonics, and the stator current Icoil has only harmonic components of m and n as harmonics. Obviously, other harmonics may be present. For simplification, when limiting to a three-phase AC rotating electrical machine, Equations 4, 5, and 6 are as follows.

  The U-phase excitation force fu is obtained from Equation 10 obtained by substituting Equation 7 into Equation 3.

  The V-phase excitation force fv is obtained from Equation 11 obtained by substituting Equation 8 into Equation 3.

  The W-phase excitation force fw is obtained from Equation 12 obtained by substituting Equation 9 into Equation 3.

  In Equations 4 to 6, θ is the angle of the fundamental wave. Of course, θ is equal to ωt when the angular velocity of the fundamental wave is ω, and is equal to 2πft when the frequency (fundamental frequency) of the fundamental wave is f. Further, in the equations 4 to 6, the numerical values 360, 120, and 240 can be easily read as 2π, 2π / 3, and 4π / 3, respectively, in actual calculation.

  The magnetic sound has a positive correlation with the vector sum of each phase excitation force, but each phase excitation force is a linear function (sum) of a number of terms as shown in Equations 10, 11, and 12. Also, the difference formula). The sum of each phase excitation force is a linear function (sum or difference) of a term (hereinafter also referred to as a vector addition term) obtained by vector addition of each term of each phase separately for each same order (for each frequency). Formula). Magnetic sound is a problem when the terms constituting the vector addition term are strengthened in the same phase (similar phase) in each vector addition term, and when the phase of each term constituting each vector addition term is greatly different Since the amplitude of the vector addition term is small, there is little problem.

  That is, when Equation 10, Equation 11, and Equation 12 are added, the terms constituting the vector addition term are in phase, and the amplitude of the vector addition term is significantly larger than the amplitude of each term. (M-1) order, (n-1) order, and (nm) order.

  From Equations 10, 11, and 12, when the m-order harmonic current component and the n-order harmonic current component are superimposed on the stator current Icoil, the (m−1) th order, the (n−1) th order, (n− m) It can be seen that the following excitation force component is remarkably generated.

  This means that an (x-1) th order excitation force can be generated by superimposing an xth order harmonic current component having a certain phase.

  Therefore, in order to cancel the (x-1) th order component of the magnetic sound that causes the currently generated magnetic sound, an excitation having an opposite phase and equal amplitude to the (x-1) th order component of this magnetic sound. The magnetic sound can be canceled by superimposing the x-order harmonic current that generates a force. Further, even if the phases are not completely opposite to each other or the amplitudes are not completely equal, it is possible to significantly reduce the amplitude by summing up the vector-added sums.

  Further, in order to change (increase or decrease) the (x-1) order component of the magnetic sound that causes the currently generated magnetic sound, the phase opposite to the (x-1) order component of this magnetic sound, etc. The magnetic sound can be changed (increased or decreased) by superimposing an x-order harmonic current that generates an excitation force having an amplitude on the fundamental frequency component of the stator current. In other words, the amplitude of the sum of the vectors can be increased or decreased.

  In order to increase or decrease the (x-1) th order magnetic sound of an AC rotating electrical machine, the knowledge that the xth order harmonic current should be superimposed with a suitable phase and a suitable amplitude has not been known in the past. It is very important in the development of low noise motors in the future.

  Similarly, from (10), (11), and (12), the (m−1) th order and the (n−1) th order are obtained by superimposing the mth order harmonic current component and the nth order harmonic current component on the stator current Icoil. , (Nm) next magnetic excitation force components can be simultaneously changed (increased or decreased). However, in this case, the (m−1) th order, the (n−1) th order, the (n−m) order, and the phase and amplitude of the mth order harmonic current component to be superimposed and the phase and amplitude of the nth order harmonic current component. ) Since the next excitation force is generated, the order of these excitation forces and the (m−1) th order, (n−1) th order, and (nm) th order magnetic sound that originally exist are It is not easy for all vector sums to be zero. However, the phase and amplitude of the superimposed current can be adjusted so that the amplitude of the vector sum of the respective excitation forces becomes as small as possible or a desired magnitude.

  In order to simultaneously change (reduce or increase) the (m-1) th order, (n-1) th order, and (nm) th order magnetic sound of AC rotating electricity, the mth order and nth order harmonic currents are suitable. The knowledge that the phase and the preferred amplitude should be added has not been known so far, and is very important in the future development of a low noise motor.

  Of the terms represented by Equation 10, Equation 11, and Equation 12, the following simplified equations are obtained by eliminating the terms for which the vector sum is zero.

  As for the harmonics of the rotor magnetomotive force Fmag of a three-phase AC rotating electric machine, although it depends on the number of poles and the number of status lots, generally the third harmonic component, the fifth harmonic component and the seventh harmonic component are Since it is much more dominant than the harmonic component of the order, it has a fundamental component, a third harmonic component, a fifth harmonic component, and a seventh harmonic component of the rotor magnetomotive force Fmag, for changing the magnetic sound. The magnetic sound (magnetic sound) that occurs when the harmonic current is not superimposed will be described below.

  Since the integer values (order values) are j = 3, k = 5, and L = 7, when substituting these into Equations 7 to 9, the following equation is obtained.

  Substituting these Equations 16, 17, and 18 into Equation 3 and not superimposing harmonic currents, if m = 0 (Im = 0) and n = 0 (In = 0), the following equation is obtained. .

  In the equations (19), (20), and (21), the terms for which the vector sum is 0 are deleted, and the terms that strengthen each other in the same phase and the DC component terms are extracted to obtain the following equation.

  Therefore, the sum of each phase excitation force obtained by adding Expression 22, Expression 23, and Expression 24 is expressed by Expression 25 below.

  In Equation 25, if the number surrounded by ○ is a round numeral, the term indicated by the round numeral 1 is the direct current component term of the total excitation force, and the term indicated by the round numeral 2 is the third order of the rotor magnetomotive force Fmag. The sixth harmonic component generated by the harmonics, the term indicated by the round numeral 3 is the sixth harmonic component generated by the first and fifth harmonics of the rotor magnetomotive force Fmag, and the term indicated by the round numeral 4 is the rotor The sixth harmonic component generated by the first and seventh harmonics of the magnetomotive force Fmag, and the term indicated by the circled number 5 is the twelfth harmonic component generated by the fifth and seventh harmonics of the rotor magnetomotive force Fmag. The term indicated by the round numeral 6 is the sixth harmonic component generated by the fifth harmonic of the rotor magnetomotive force Fmag and the primary current component (fundamental wave) of the stator current, and the term indicated by the round numeral 7 is the rotor origin. 7th harmonic of the magnetic force Fmag and 1 of the stator current The sixth-order harmonic component generated by the current component (the fundamental wave).

  That is, the magnetic sound of the three-phase AC rotating electric machine is converted into sixth and twelfth magnetic sound components by the fundamental wave component, the third harmonic component, the fifth harmonic component, and the seventh harmonic component of the rotor magnetomotive force Fmag. It can be seen from Equation 25 that this is caused.

  Accordingly, a case where the seventh harmonic component and the thirteenth harmonic component are superimposed on the stator current Icoil for changing (reducing or increasing) the sixth and twelfth magnetic sound components will be described below.

  In this case, in Equations 7, 8, and 9, if j = 3, k = 5, L = 7, m = 7, and n = 13, the following equation is obtained.

  When Equations 10, 11, and 12 are calculated using Equations 26, 27, and 28, the following equations are obtained.

  In Equations 29, 30, and 31, the terms for which the vector sum is 0 are eliminated, and the terms that strengthen each other in the same phase and the DC component terms are extracted to obtain the following equation.

  If these formulas are arranged in the same manner as Equation 25, the sum of the respective phase excitation forces is as follows.

In this number 35,
The term indicated by the circled number 1 is a DC component term,
The term indicated by the round numeral 2 is a sixth harmonic component generated by the third harmonic of the rotor magnetomotive force Fmag,
The term indicated by the round numeral 3 is a sixth-order harmonic component generated by the first-order and fifth-order harmonics of the rotor magnetomotive force Fmag,
The term indicated by the circled number 4 is a sixth harmonic component generated by the first and seventh harmonics of the rotor magnetomotive force Fmag,
The term indicated by the circled number 5 is the 12th harmonic component generated by the 5th and 7th harmonics of the rotor magnetomotive force Fmag,
The term indicated by the round numeral 6 is the sixth harmonic component generated by the fifth harmonic of the rotor magnetomotive force Fmag and the primary current component (fundamental wave) of the stator current,
The term indicated by the round numeral 7 is a sixth harmonic component generated by the seventh harmonic of the rotor magnetomotive force Fmag and the primary current component (fundamental wave) of the stator current.

The term indicated by the circled number 8 is the first-order component (fundamental wave) of the rotor magnetomotive force Fmag and the sixth-order harmonic component generated by the seventh-order harmonic of the stator current,
The term indicated by the round numeral 9 is the 12th-order harmonic component generated by the first-order component (fundamental wave) of the rotor magnetomotive force Fmag and the 13th-order harmonic of the stator current,
The term indicated by the round numeral 10 is the 12th harmonic component generated by the 5th harmonic of the rotor magnetomotive force Fmag and the 7th harmonic of the stator current,
The term indicated by the round numeral 11 is the 18th harmonic component generated by the 5th harmonic of the rotor magnetomotive force Fmag and the 13th harmonic of the stator current,
The term indicated by the round numeral 12 is the sixth harmonic component generated by the seventh harmonic of the rotor magnetomotive force Fmag and the 13th harmonic of the stator current,
The term indicated by the circled number 13 is a sixth-order harmonic component generated by the first-order component (fundamental wave) and the seventh-order component of the stator current,
The term indicated by the round numeral 14 is a 12th-order harmonic component generated by the first-order component (fundamental wave) and the 13th-order component of the stator current,
The term indicated by the circled numeral 15 is a sixth-order harmonic component generated by the seventh-order component and the thirteenth-order component of the stator current.

  Eventually, the sum of the respective phase excitation forces shown in Equation 35 becomes the sixth and twelfth harmonics, so the terms of the circled numbers 2, 3, 4, 6, 7, 8, 12, 13, 15 above. If the phase angle and each amplitude are set, the vector sum of the above-mentioned circle numbers 2, 3, 4, 6, 7, 8, 12, 13, and 15 is reduced to 0, increased, or increased. It is possible to cancel or reduce (or increase) the sixth-order magnetic noise.

  Similarly, if the phase angle and each amplitude of the term of the round numbers 5, 9, 10, 14 are set, the vector sum of the terms of the round numbers 5, 9, 10, 14 is reduced to 0 or smaller, Alternatively, it can be increased, and the 12th order noise of the magnetic sound can be canceled or reduced (or increased).

  In other words, by adjusting the phase and amplitude of the seventh-order harmonic current component and the thirteenth-order harmonic current component in Equation 35, the sum of terms other than the DC term shown in Equation 35 is set to 0 to achieve three-phase AC rotation. It is possible to cancel or reduce (or increase) the sixth-order harmonic component and the twelfth-order harmonic component of the magnetic sound that are most important in an electric machine.

  The cancellation condition for the sixth harmonic of the magnetic sound is shown in the following equation.

  In Equation 36, the sum or difference of each term indicated by a broken line represents the vector sum of the sixth harmonics of the radial magnetic excitation force serving as the magnetic sound component, and the sum or difference of each term indicated by the solid line is for cancellation. The vector sum of the 6th harmonic of the exciting force by the harmonic current of is shown. Therefore, in Equation 36, the phase angle and the amplitude may be determined so that the vector sum of these two becomes zero. The cancellation condition for the 12th harmonic of the magnetic sound is shown in the following equation.

  In Expression 37, the sum or difference of each term indicated by a broken line represents the vector sum of the 12th harmonics serving as a magnetic sound component, and the sum or difference of each term indicated by a solid line represents excitation by a harmonic current for cancellation. The vector sum of the 12th harmonic of the force is shown. Therefore, in Equation 37, the phase angle and the amplitude may be determined so that the vector sum of these two becomes zero.

  Since θ as a function of the number of rotations changes every moment in the above Equations 35, 36, and 37, the total sum of the magnetic sounds shown in Equation 35 as the rotational speed, phase, and amplitude of the fundamental wave change with time. The amplitude and phase of the harmonic current for setting the current to a predetermined level changes every moment. Similarly, the amplitude and phase of the harmonic current that satisfies Equations 36 and 37 also change every moment. For this reason, the amplitude and phase of the superimposed harmonic current are calculated every predetermined time according to the frequency, phase and amplitude of the fundamental current.

  Next, an example of a motor control circuit that realizes the above-described magnetic sound reduction (or increase) will be described below.

(Circuit configuration example 1)
FIG. 3 shows an example of a circuit that generates the above harmonic current. This motor control circuit is an embodiment that performs feedback control of the motor current.

  1 is a battery, 2 is a smoothing capacitor, 4 is a first inverter that is a three-phase inverter, 5 is a second inverter that is a three-phase inverter, 6 is a first motor controller, 7 is a second motor controller, and 8 is permanent This is a magnet rotor type three-phase synchronous motor. The motor 8 may be a generator or a generator motor. The motor 8 has two three-phase star-connected armature coils 81 and 82 that are independent of each other. The coil 81 is a first phase winding group referred to in the present invention, and the coil 82 is a second phase winding referred to in the present invention. It constitutes a group.

  DC power is supplied from the battery 1, the first inverter 4 controlled by the motor controller 6 applies the first three-phase armature voltage to the coil 81, and the second inverter 5 controlled by the second motor controller 7 is A second three-phase armature voltage is applied to the coil 82. The motor controller 6 receives a current signal from a current detection sensor (not shown) that detects a three-phase alternating current that is passed through the coil 81, and receives a rotation angle signal from a rotation angle detector (not shown). Based on the above, the first inverter 4 is controlled. The motor controller 7 receives a current signal from a current detection sensor (not shown) that detects a three-phase alternating current that is passed through the coil 82 and a rotation angle signal from the rotation angle detector. The second inverter 5 is controlled based on the above.

  An example of the motor controllers 6 and 7 will be described with reference to FIG. In FIG. 4, reference numeral 9 denotes first motor current control means including the first inverter 4 and the motor controller 6, and reference numeral 10 denotes motor current control means including the second inverter 5 and the motor controller 7.

  Reference numeral 100 denotes an amplitude / phase command circuit block that indicates the amplitude and phase of a current command value (three-phase AC coordinate system) corresponding to the fundamental wave. Reference numeral 101 denotes an amplitude / phase command circuit block for instructing the amplitude and phase of a harmonic current (three-phase AC coordinate system) of a predetermined order (seventh order here).

  The amplitude / phase command circuit block 100 determines the amplitude and phase based on a current command (fundamental wave) received from an external control device such as a vehicle control ECU. Moreover, the circuit block 100 may be comprised by this vehicle control ECU. This external control device calculates the current command value as the fundamental wave based on the rotation angle signal (rotation position signal) of the three-phase synchronous machine 8 and the torque command.

  The circuit block 101 inputs in advance the frequency, amplitude, and phase of the current command (fundamental wave) current into the above-described Expression 35, Expression 36, Expression 37, Expression 13, or Expression 14 or Expression 15 to calculate in advance. The frequency, amplitude, and phase of a predetermined predetermined order harmonic current are determined, and an amplitude / phase command that indicates them is output. Other constants of these mathematical expressions are set in advance according to the purpose. For example, when reducing or canceling the 6th and 12th magnetic sounds, the 7th or 13th harmonic current is reduced so that the calculated value of the formulas 36 and 37 is equal to or less than a predetermined value or 0. Determine amplitude and phase. The other constants are preset as numerical values specific to the AC rotating electric machine. In the case of canceling only the sixth-order magnetic sound, the amplitude and phase of the harmonic current are determined so that the calculated value of Expression 36 is equal to or less than a predetermined value or zero. In the case of canceling only the 12th-order magnetic sound, the amplitude and phase of the harmonic current are determined so that the calculated value of Expression 37 is equal to or less than a predetermined value or 0. In any case, in these equations, by adjusting the phase and / or amplitude of the superimposed 7th and / or 13th harmonic currents, the 6th / 12th magnetic sound, that is, the majority of the magnetic sound is amplified. Can be reduced, reduced, or canceled. Instead of calculating the above formulas, the frequency, phase, and amplitude of the fundamental frequency component are substituted into a map or table corresponding to these formulas in advance, and the phase and amplitude of the seventh and / or thirteenth harmonic currents are calculated. You may search for values. Further, the circuit block 100 calculates and outputs the current value based on the calculated amplitude and phase of the fundamental frequency component of the stator current, and the circuit block 101 based on the calculated amplitude and phase of the harmonic current. Thus, the current value of the harmonic current can be calculated and output.

  The commands related to the fundamental wave current and the harmonic current are coordinate-converted into the dq axis system by the coordinate axis conversion circuit blocks 103 and 103 ′, and the detected values (dq axis) are subtracted by the subtracters 104 and 104 ′. ), And the difference between them is adjusted by the current amplifiers 400 and 400 ′, and then output to the three-phase AC current value by the coordinate axis conversion circuit blocks 105 and 105 ′.

  The three-phase alternating current values output from the circuit blocks 105 and 105 ′ are converted into three-phase PWM control voltages by the circuit blocks 106 and 106 ′, and each of the three-phase inverters 4 and 5 is converted by these three-phase PWM control voltages. The switching element is intermittently controlled, and the output voltage of the inverter 4 is applied to the first phase winding group 81 and the output voltage of the inverter 5 is applied to the second phase winding group 82.

  Each conductor of the first phase winding group 81 and each conductor of the second phase winding group 82 are accommodated in the same slot, and the first phase winding group 81 and the second phase winding group 82 Of course, the phase difference may be zero, or the two conductors may be shifted by a predetermined slot pitch in the circumferential direction to give the phase difference. As a result, the fundamental wave current flows through the first phase winding group 81 and the seventh harmonic current flows through the second phase winding group, for example.

  The three-phase synchronous machine 8 has a built-in rotation angle sensor, and the speed / position signal processing circuit block 107 extracts a speed signal and a position signal from a rotation position signal of a rotor output from a rotation angle sensor (not shown). These are output to the circuit blocks 103, 103 ′, 105, 105 ′. The phase currents of the first phase winding group 81 and the second phase winding group 82 are detected by the current sensors 109 and 109 ′, and the d axis current detection values are detected by the coordinate axis conversion circuit blocks 108 and 108 ′. And the q-axis current detection value are input to the subtracters 104 and 104 ′ to perform feedback control for converging the current value to the command value. Reference numeral 202 denotes an order generation circuit block having a function of instructing the order of harmonic components to be generated in the coordinate conversion blocks 103 'and 105'.

  By doing so, the fundamental frequency component and the harmonic component of the motor current applied to the three-phase synchronous machine 8 can be controlled separately, so that the circuit reliability is improved and, in particular, the fundamental frequency component for realizing the torque. While improving the control response and control reliability, it is possible to realize a very quiet motor by reducing magnetic noise. In addition, since the basic phase current control can be performed at a low frequency, the circuit load can be reduced.

(Circuit configuration example 2)
Another circuit example for generating the above harmonic current is shown in FIG. This circuit is obtained by changing the circuit shown in FIG. 4 to open control.

  The commands related to the fundamental wave current and the harmonic current output from the fundamental wave circuit block 100 and the harmonic wave circuit block 101 are coordinate-converted into the dq axis system by the coordinate axis conversion circuit blocks 103 and 103 ′, and the current amplifier 400. , 400 ′, the gain is adjusted, and then output to the three-phase alternating current value by the coordinate axis conversion circuit blocks 105 and 105 ′.

  The circuit blocks 106 and 106 'generate a three-phase PWM control voltage corresponding to the input three-phase AC current value, and the switching elements of the three-phase inverters 4 and 5 are intermittently controlled by these three-phase PWM control voltages. The three-phase synchronous machine 8 has a built-in rotation angle sensor, and the speed / position signal processing circuit block 107 extracts a speed signal and a position signal from the rotation position signal output from the rotation angle sensor, and performs coordinate conversion. Are output to the circuit blocks 103, 103 ′, 105, 105 ′. Thereby, the same effects as those of the circuit configuration example 1 can be obtained.

(Circuit configuration example 3)
Another circuit example for generating the above harmonic current is shown in FIG. This circuit example is an embodiment in which the motor current is feedback controlled in a three-phase AC coordinate system.

  Reference numeral 100 denotes an amplitude / phase command circuit block for instructing the amplitude and phase as a current command value (three-phase AC coordinate system) corresponding to the fundamental wave. Reference numeral 101 denotes an amplitude / phase command circuit block for instructing the amplitude and phase as a harmonic current (three-phase AC coordinate system) of a predetermined order. The function of these circuit blocks is the same as in the case of FIG. 3, and the harmonic block 101 outputs the frequency, phase, and amplitude output from the circuit block as in Equation 35, Equation 36, Equation 37, Equation 13 or Equation 14, or Substituting into Equation 15, the amplitude and phase of the harmonic are determined, or substantially the same calculation processing is performed using a map or table.

  The amplitude / phase commands output from the circuit blocks 100 and 101 are input to the circuit blocks 102 and 102 '. The circuit block 102 forms a fundamental wave current command value (three-phase AC coordinate system) based on the input amplitude / phase command of the fundamental wave and the detected rotational position signal, and the circuit block 102 ′ A harmonic current command value (three-phase AC coordinate system) is formed on the basis of the amplitude / phase command of the harmonic of the predetermined order and the detected rotational position signal.

  The subtractor 300 obtains a difference between the detected fundamental wave U-phase current detection value iu ′ and the fundamental wave current command value iu, and outputs this difference to the circuit block 302 constituting the current controller. The subtractor 301 obtains a difference between the detected fundamental wave V-phase current detection value iv ′ and the fundamental wave current command value iv, and outputs this difference to the circuit block 302 constituting the current controller. The circuit block 302 forms a U-phase voltage and a V-phase voltage that eliminate the difference, and the circuit block 105 calculates and outputs the U-phase and V-phase PWM voltages corresponding to the U-phase voltage and the V-phase voltage. Further, the subtracting and inverting circuit 303 calculates the analog inverted signal of the difference between the U phase voltage and the V phase voltage as the W phase voltage, and the circuit block 105 calculates and outputs the PWM voltage of the W phase voltage. The first inverter 4 is intermittently controlled according to the duty corresponding to these three-phase PWM voltages.

The subtractor 300 ′ obtains a difference between the detected harmonic U-phase current detection value iu ′ and the harmonic current command value iu, and outputs this difference to the circuit block 302 ′ constituting the current controller. The subtractor 301 ′ obtains the difference between the detected harmonic V-phase current detection value iv ′ and the harmonic current command value iv, and outputs this difference to the circuit block 302 ′ constituting the current controller. The circuit block 302 ′ forms a U-phase voltage and a V-phase voltage that eliminate the difference, and the circuit block 105 ′ calculates and outputs the U-phase and V-phase PWM voltages corresponding to the U-phase voltage and the V-phase voltage. Further, the subtraction / inversion circuit 303 ′ calculates an analog inversion signal of the difference between the U-phase voltage and the V-phase voltage as the W-phase voltage, and the circuit block 105 ′ calculates and outputs the PWM voltage of the W-phase voltage. The second inverter 5 is intermittently controlled according to the duty corresponding to these three-phase PWM voltages.
(Modification)
In addition to the circuit examples described above, various known motor control circuits can naturally be employed. Further, the fundamental wave forming motor controller 6 and the harmonic forming controller 7 may be completely different circuits, or may have different clock frequencies. For example, the motor controller 6 may be a high-precision feedback circuit, and the motor controller 7 may be an open control circuit.
(Modification)
A modification of each circuit example will be described below with reference to a flowchart shown in FIG. In this modification, the operation mode of the second inverter is changed based on the required current (required torque) and the rotational speed.

First, a torque command value (or current command value) and a rotation speed detection value input from the outside are read (steps S100 and S102). When these command values are less than a predetermined level and the rotation speed is less than a predetermined value, for example, by changing the output order of the order generation circuit block 202 shown in FIG. Drive to generate. Of course, at this time, the amplitude and phase of the harmonic current are determined so that the n-th radial vibration is reduced. When these command values are equal to or higher than a predetermined level or the rotation speed is equal to or higher than a predetermined value, for example, by changing the output order of the order generation circuit block 202 shown in FIG. (Step S104). Since magnetic noise becomes a problem in the low rotation region, the effects of loss reduction, large torque output, and magnetic noise reduction can be realized in a balanced manner.
(Modification)
Note that the equations of Equations 35, 36, and 37 show examples in which the harmonic component of the rotor magnetomotive force Fmag is canceled or changed by superimposing the harmonic current on the fundamental current of the stator current. In PWM control of an AC rotating electrical machine, harmonics are inevitably superimposed on the fundamental current due to switching of the inverter, resulting in magnetic noise. In order to further reduce this magnetic noise, the harmonic current obtained by subtracting the harmonic current inevitably superimposed on the stator current by switching of this inverter from the harmonic current for changing the magnetic sound obtained by the above formula is fundamental. A calculation to be superimposed on the current may be performed.

(Experimental example)
The experiment for reducing the magnetic noise was performed using a three-phase synchronous machine (8 poles, 24 slots, IPM) 14 shown in FIG. The fundamental current was 43 A, and the rotor phase angle was controlled to a value that maximized the torque.

  FIG. 9 shows a three-phase current waveform when the motor is driven without superimposing a harmonic current for reducing magnetic noise on the stator current of the synchronous machine. Each phase current includes a relatively small harmonic component in addition to its fundamental frequency component. FIG. 10 shows a three-phase current waveform when the motor is driven by superimposing a harmonic current for reducing magnetic noise on the stator current shown in FIG. 9 and 10, the rotation speed was 17000 rpm. The harmonic current for magnetic noise reduction was superimposed by a feedback method using the circuit of FIG.

  FIG. 11 shows the frequency spectrum A of the U-phase current shown in FIG. 9 and the frequency spectrum B of the harmonic component superimposed U-phase current for reducing magnetic noise shown in FIG. In FIG. 11, the left side (displayed in light gray) of the pair of bars shown for each order shows the spectrum A of the U-phase current shown in FIG. 9, and the pair of bars shown for each order The right side (displayed in dark gray) shows the spectrum B of the U-phase current shown in FIG. In FIG. 9, the seventh current (seventh harmonic component) has an amplitude that is 3% of the amplitude of the primary current (fundamental frequency component). In FIG. 10, the seventh current (seventh harmonic component) has an amplitude of 12% of the amplitude of the primary current (fundamental frequency component).

  The three adjacent teeth of the stator when the current of FIG. 9 is applied (hereinafter referred to as a magnetic noise non-reduction mode) and when the current of FIG. 10 is applied (hereinafter referred to as a magnetic noise reduction mode). FIG. 12 shows changes in the applied radial direction (radial direction) excitation force depending on the rotor rotation angle. The larger period change waveform C of the two period change waveforms shown in FIG. 12 shows the change in the radial excitation force in the magnetic noise non-reduction mode shown in FIG. 9, and the smaller period change waveform D shows the change in the period change waveform D in FIG. Changes in the radial excitation force in the magnetic noise non-reduction mode shown in FIG. E represents the average value of the radial excitation force in the magnetic noise non-reduction mode. The radial excitation force has a direct current component due to the attraction of the permanent magnet of the rotor.

  In FIG. 13, F indicates the frequency spectrum of the radial excitation force (large periodic variation waveform) C when magnetic noise is not reduced, and G indicates the radial excitation force (small periodic variation waveform) D when magnetic noise is reduced. The frequency spectrum of is shown. The frequency spectrum F is indicated by a bar on the left side (displayed in light gray) among a pair of bars indicated for each order. The frequency spectrum G is indicated by a bar on the right side (displayed in dark gray) of a pair of bars indicated for each order.

  FIG. 14 shows the measurement result (during electric operation) of the sixth-order magnetic sound (frequency 6θ) when the rotation speed in the experimental machine is changed. From FIG. 14, it can be seen that the 6th-order component of magnetic noise can be satisfactorily reduced in a high rotation range (about 1500 rpm or more) in which magnetic noise is annoying. FIG. 15 shows the measurement results (during power generation) of the sixth-order magnetic sound (frequency 6θ) when the rotation speed in the experimental machine is changed. FIG. 15 shows that the sixth-order component of magnetic noise can be reduced over almost the entire rotation range. It can also be seen that the sixth-order magnetic noise can be reduced by 20 db by superimposing the harmonic current for reducing the magnetic noise in the rotation region where the magnetic noise is maximum. 14 and 15, the sixth-order magnetic noise increases in some rotation regions due to the superposition of the seventh-order component of the harmonic current for magnetic noise reduction. This is because it is not realized, and is not essential. Further, the reversal of the magnetic noise occurs at a level where the sixth-order magnetic noise level is small, and the absolute value thereof is small, so that there is no problem. Of course, in this synchronous machine, the superimposition of the seventh harmonic current for magnetic noise reduction can be stopped in the rotation range where the reduction in magnetic noise cannot be expected so much.

(Modification)
In addition, by using the harmonic superposition technique by the circuit of the above embodiment, that is, by superimposing a harmonic current having a higher order than the circumferential vibration on the fundamental frequency component, the magnetic noise reduction is stopped and the torque is adjusted as necessary. It can also be switched to reduce ripple. Specifically, the amplitude and phase angle of the harmonic current may be changed.

(Modification)
Equations 35, 36, and 37 show examples in which the harmonic component of the rotor magnetomotive force Fmag is canceled or changed by superimposing the harmonic current on the fundamental current of the stator current. In the PWM control of an electric machine, harmonics are inevitably superimposed on the fundamental current due to switching of the inverter, resulting in magnetic noise. In order to further reduce this magnetic noise, the harmonic current obtained by subtracting the harmonic current inevitably superimposed on the stator current by switching of this inverter from the harmonic current for changing the magnetic sound obtained by the above formula is fundamental. A calculation to be superimposed on the current may be performed.

(Modification)
A control operation when the magnetic noise reduction type rotating electrical machine is mounted on an engine traveling vehicle will be described with reference to a flowchart shown in FIG. In addition, this kind of rotary electric machine is widely used as, for example, a generator motor or an electric air conditioner for a hybrid vehicle or a torque assist vehicle, and is well known.

  First, in step S400, it is checked whether the engine is stopped and the electric operation is being performed, or whether the power is being generated by regenerative braking. If so, the second inverter for reducing radial vibration is operated (S402). Otherwise, the second inverter is not operated (S404). The second inverter superimposes the 7th and 13th harmonic currents with a predetermined phase and a predetermined amplitude, respectively. The phase and amplitude of the seventh-order harmonic current cancels the sixth-order harmonic component of the original radial magnetic excitation force (magnetic sound) of the three-phase AC rotating electric machine based on the above formula. Set to size. This reduces the magnetic noise during regenerative braking (power generation) when the engine is stopped, where there is no engine noise and the magnetic noise of the generator motor tends to be harsh, or when the engine speed is low and the engine noise tends to be harsh. Silence can be improved. Further, during other operations, the harmonic current superimposing control circuit can be suspended.

  In step S404, the second inverter may be instructed to generate a fundamental wave in cooperation with the first inverter. In this way, since the current amount of one inverter is halved, it is possible to reduce the inverter loss and improve the efficiency.

  Although preferred embodiments of the present invention have been described in detail above, various modifications and changes can be made by those skilled in the art, and the scope of the claims falls within the true spirit and scope of the present invention. It is to be understood by those skilled in the art that all such modifications and changes are included in the scope of the present invention.

(Reference example)
A control reference example of the superimposed harmonic current will be described with reference to FIGS. FIG. 17 is a block circuit diagram showing the motor control device of this embodiment, FIG. 18 is a block circuit diagram showing an example of the coordinate conversion circuit 2, and FIG. 19 is a waveform diagram showing signal waveforms (rotation coordinate system display) of each part of the circuit. FIG. 20 is a waveform diagram showing signal waveforms (stationary coordinate system display) of each part of the circuit.

  This motor control device is an embodiment for performing feedback control of motor current, wherein 1 is a fundamental wave command value generation circuit, 2 is a harmonic command value generation circuit, 3 and 4 are adders, 5 and 6 are subtractors, 7 and 8 are PI amplifiers (proportional-integral circuits), 9 is a coordinate conversion circuit, 10 is a PWM voltage generation circuit, 11 is a three-phase inverter, 12 is two current sensors (phase current detection elements), and 13 is three-phase synchronous. A motor generator (synchronous AC rotating machine for vehicles), 14 is a resolver (rotation angle detecting element), 15 is a position signal processing circuit, 16 is a delay compensation circuit, and 17 is a coordinate conversion circuit.

  Of the above components 1 to 17, the components other than the three-phase synchronous motor generator 13 constitute a motor control device according to the present invention, and among the components 1 to 17, a current sensor (phase current detection). The component (circuit) except for the element 12, the three-phase synchronous motor generator (synchronous AC rotating electric machine for vehicle) 13 and the resolver (rotation angle detecting element) constitutes a motor current control element or motor control means in the present invention. is doing.

  The component (circuit) 17 constitutes a phase current detection value coordinate system conversion element referred to in the present invention, and the component (circuit) 1 constitutes a fundamental wave command value output element referred to in the present invention. ) 2 constitutes a harmonic command value output element referred to in the present invention, components (circuits) 3 to 6 constitute current deviation calculation elements referred to in the present invention, and components (circuits) 7 to 11 represent the present invention. This constitutes the phase voltage control element. Needless to say, the three-phase inverter 11 is fed from a DC power source to generate a three-phase AC voltage.

  A fundamental wave command value generation circuit (fundamental wave command value output element) 1 sets a target value of a fundamental wave current corresponding to an input torque command value and a rotational speed command value, and a d-axis fundamental wave that is a d-axis current component. This is a known circuit that converts the command value Id1 * and the q-axis fundamental wave command value Iq1 * that is the q-axis current component thereof. The torque command value is input from, for example, an external control device such as a vehicle control ECU, and the fundamental wave command value generation circuit 1 generates a d-axis fundamental wave command value Id1 * and a q-axis fundamental wave command value Iq1 * based on the torque command value. decide. If necessary in determining the d-axis fundamental wave command value Id1 * and the q-axis fundamental wave command value Iq1 *, in addition to the torque command value, the voltage of the three-phase inverter 11, the output signal of the resolver 14 and the like are further used as the fundamental wave command value. Input to the generation circuit 1.

  The harmonic command value generation circuit 2 (harmonic command value output element) is set to a preset target value of 6k + 1 (k is an integer, k of the fundamental frequency component is 0) and the d-axis current component. Is a d-axis harmonic command value Id6k + 1 * and a q-axis harmonic command value Iq6k + 1 * which is the q-axis current component. Furthermore, the harmonic command value generation circuit 2 is a circuit for generating a harmonic current command value that reduces the radial vibration of the three-phase synchronous motor generator 13.

  A specific example of the harmonic command value generation circuit 2 will be described with reference to a block diagram shown in FIG. In FIG. 21, 21 is a seventh current command value generation circuit, 22 is a 13th current command value generation circuit, 24 and 25 are coordinate conversion circuits, and 27 and 28 are adders. However, in this embodiment, the harmonic command value generation circuit 2 generates only the 7th and 13th harmonic command values to reduce the 6th and 12th radial vibrations. A value may be generated and superimposed in the adders 27 and 28 in the same manner.

  The seventh-order current command value generation circuit 21 receives the d-axis command value Id * and q-axis command value Iq * input from the fundamental wave command value generation circuit 1, and the seventh-order harmonic command value for canceling the sixth-order radial vibration. Is a table describing the relationship between the amplitude I7 * and the phase angle β7 *. That is, the amplitude I7 * and the phase angle β7 * of the seventh harmonic command value are function values having the d-axis command value Id * and the q-axis command value Iq * on the fundamental wave rotation coordinate system as variables. Here, the amplitude I7 * and the phase angle β7 * of the seventh harmonic command value are values on the fundamental wave rotation coordinate system, but are also the same values on the stationary coordinate system.

  Similarly, the 13th-order current command value generation circuit 22 includes the d-axis command value Id * and the q-axis command value Iq * input from the fundamental wave command value generation circuit 1 and the 13th-order harmonic for canceling the 12th-order radial vibration. It is a table which describes the relationship between the amplitude I13 * of the command value and the phase angle β13 *. That is, the amplitude I13 * and the phase angle β13 * of the 13th harmonic command value are function values having the d-axis command value Id * and the q-axis command value Iq * on the fundamental wave rotation coordinate system as variables. Here, the amplitude I13 * and the phase angle β13 * of the 13th harmonic command value are values on the fundamental wave rotation coordinate system, but are the same values on the stationary coordinate system. These data I7 *, β7 *, I13 *, β13 * are stored in ROM (not shown) of the seventh and thirteenth current command value generation circuits 21 and 22. These data I7 * and β7 * obtained by substituting the d-axis command value Id * and the q-axis command value Iq * into the circuits 21 and 22 are sent to the coordinate conversion circuit 24, and these data I13 * and β13 * are the coordinates. It is output to the conversion circuit 25.

  The coordinate conversion circuit 24 rotates the fundamental wave based on the amplitude I7 * of the seventh harmonic current input from the seventh current command value generation circuit 21 and the phase angle (determined based on the phase angle θ of the fundamental wave) β7 *. The d-axis harmonic command value Id7 *, which is the d-axis component of the seventh-order harmonic current command value displayed in the coordinate system (also referred to as the dq-axis coordinate system or the fundamental wave dq coordinate system), and its q-axis component The q-axis harmonic command value Iq7 * is calculated.

  The coordinate conversion circuit 25 rotates the fundamental wave based on the amplitude I13 * of the 13th harmonic current input from the 13th current command value generation circuit 22 and the phase angle (determined based on the phase angle θ of the fundamental wave) β13 *. The d-axis harmonic command value Id13 *, which is the d-axis component of the 13th harmonic current command value displayed in the coordinate system (also referred to as the dq-axis coordinate system or the fundamental wave dq coordinate system), and the q-axis component thereof The q-axis harmonic command value Iq13 * is calculated. This calculation is performed by the following calculation 38.

  In Equation 38, θv is a phase compensation rotation angle signal obtained by phase compensation of the motor rotation angle θ output from a delay compensation circuit (phase compensation circuit) 16 described later.

  However, in the above description, the circuits 21 and 22 store a table of the d-axis command value Id *, the q-axis command value Iq *, and the amplitude and phase angle of the harmonic command value to be output. Store a table of angle, voltage, and current, and the amplitude and phase angle of the harmonic command value to be output. Substitute the rotation angle, voltage, and current detection values into this table and output the harmonic command. The amplitude and phase angle of the value may be calculated.

  Next, the seventh-order d-axis harmonic command value Id7 * and the thirteenth-order d-axis harmonic command value Id13 * are added by the adder 27 to obtain the d-axis harmonic command value Id6n + 1 *, and the seventh-order q-axis harmonic is obtained. Command value Iq7 * and 13th-order q-axis harmonic command value Iq13 * are added by adder 28 to obtain q-axis harmonic command value Iq6n + 1 *. Of course, higher order harmonic command values such as the 19th order harmonic command value may be formed and added in the same way by the adders 27 and 28.

  The d-axis harmonic command value Id6n + 1 * thus obtained is added to the d-axis fundamental wave command value Id1 * by the adder 3 to obtain the d-axis command value Id *. Similarly, the q-axis harmonic command value Iq6n + 1 * Is added to the q-axis fundamental wave command value Iq1 * by the adder 4 to obtain the q-axis command value Iq *. Thereby, the noise cancellation current command value can be determined by a simple calculation. That is, each order harmonic command value is simplified because only a single frequency component needs to be calculated as shown in Equation 38.

The position signal processing circuit 15 calculates the rotation angle θ of the stationary coordinate system based on the rotation angle signal from the resolver 14 and outputs it to the delay compensation circuit 16 and the coordinate conversion circuit 17. The delay compensation circuit 16 is a phase compensation circuit, and outputs the phase-compensated rotation angle θv to the coordinate conversion circuits 24 and 25 and a coordinate conversion circuit 9 to be described later to compensate for calculation delays of these circuits. The coordinate conversion circuit 17 performs a coordinate conversion process on the U-phase current Iu and the V-phase current Iv detected by the current sensor 12, thereby detecting the d-axis detection value Id and the q-axis detection as current detection values in the rotating coordinate system display. The value Iq is output.

  The subtractor 5 subtracts the d-axis detected value Id from the d-axis command value Id * obtained by the above calculation to obtain a deviation ΔId, and the subtractor 6 calculates the q-axis detected value Iq from the q-axis command value Iq *. To obtain a deviation ΔIq. The PI amplifier 7 amplifies the deviation ΔId by PI (proportional-integral) to converge the deviation ΔId to 0 and outputs a corresponding d-axis voltage Vd, and the PI amplifier 8 performs the deviation ΔIq to converge the deviation ΔIq to 0. Is amplified by PI (proportional-integral) and the corresponding q-axis voltage Vq is output.

The coordinate conversion circuit 9 converts these voltages Vd and Vq into the three-phase AC voltages Vu, Vv and Vw of the rotating coordinate system using the input phase compensation rotation angle signal θv, and the PWM voltage generation circuit 10 The AC voltages Vu, Vv, Vw are converted into PWM signal voltages Uu, Uv, Uw, and the three-phase inverter 11 intermittently controls the six built-in switching elements based on the input PWM signal voltages Uu, Uv, Uw. A three-phase AC voltage is created and applied to each phase terminal of the three-phase synchronous motor generator 13. The motor control circuit described above is the same as the normal motor control system except for the harmonic command value generation circuit 2, and this type of PWM feedback control itself is already well known, and thus detailed description thereof is omitted. Thereby, the level of the 6th and 12th harmonic noise, which is the main cause of noise in the motor, can be greatly reduced.
(Modification)
In the explanation using the above formula, the 6th and 12th magnetic sounds generated by the rotor magnetomotive force Fmag including the 3rd, 5th and 7th harmonics are canceled or reduced by superimposing the 7th and 13th harmonic current components. Shown when to do. As a result of performing the same calculation as described above, by superimposing a harmonic current of a certain order, magnetic noise having an order lower than this order by at least 1 or magnetic noise having an order lower by 1 than this order and other orders It was found that the magnetic noise can be reduced or canceled.

  For example, the 18th-order magnetic sound can be canceled or reduced by superimposing the 19th-order harmonic current component and the 24th-order magnetic sound can be canceled or reduced by superimposing the 25th-order harmonic current component by the same calculation as described above. In addition, in the above-described formula, only the 6th-order magnetic sound (magnetic sound) can be canceled or reduced, and only the 12th-order magnetic sound can be canceled or reduced. It can be easily implemented by setting 0 to 0. As a result, it is possible to satisfactorily reduce the 6th, 12th, 18th, and 24th order magnetic noise, which is the dominant magnetic noise in the three-phase rotating electric machine.

  Further, in the explanation using the above mathematical formulas, only the fundamental frequency component is assumed as the original stator (armature current) for easy understanding of the theoretical explanation, and m-order and n-order magnetic noise reduction is applied to this fundamental frequency component. It was thought that the harmonic current for use would be superimposed. However, in general (when the control of this embodiment is not performed), since the actual stator current includes a harmonic component as well as the rotor magnetomotive force, the stator currents of Equations 16, 17, and 18 are examples. When the j-th order, the k-th order, and the L-th order are included, they are expressed as Expression 39, Expression 40, Expression 41. Therefore, if this is expanded by the above-described mathematical formula, it is clear that the excitation force due to the harmonic current originally contained is generated in the same manner as the excitation force generated by the harmonic component of the rotor magnetomotive force. Therefore, it is natural that the harmonic current superimposed in the present invention can be reduced or canceled by determining the order, amplitude, and phase based on this. (These mathematical expressions are omitted because they can be applied to the above-mentioned expressions.)

  When the order of the harmonic current included in the original armature current is one order higher than the order of the harmonic component of the radial vibration to be reduced, the harmonic for magnetic noise reduction calculated by the above formula is used. It is preferable to subtract the harmonic current contained in the original armature current from the wave current and superimpose it on the fundamental frequency component of the original armature current.

  Further, even in a multi-phase AC rotating electric machine different from a three-phase AC rotating electric machine, it is possible to reduce a magnetic sound (magnetic sound) of an order lower than this order by superimposing a harmonic current of a certain order. It will be understood that the above knowledge is established by changing each of the above equations in accordance with the phase change.

  Further, it is natural that this embodiment can be applied to a multi-slot configuration called distributed winding (for example, a configuration having two or more slots per pole per phase).

  In the above description, the stator current Icoil is described with reference to the stationary coordinate axis (angle θ). However, it is naturally possible to display the stator current Icoil with reference to the rotating coordinate system (d, q axes).

  In addition, instead of calculating the harmonic current for magnetic noise reduction to be superimposed on the armature current by the above formula, the set of the variable parameter of the above formula and the amplitude and phase of the harmonic current of the predetermined order for magnetic noise reduction Can be stored in a table, and the amplitude and phase of the harmonic current for magnetic noise reduction can be determined by substituting variable parameters into this table. Of course, the above-described processing for reducing magnetic noise can be obtained using software as well as dedicated hardware.

It is the figure which illustrated typically the magnetic circuit for one phase of a polyphase alternating current rotating electrical machine. FIG. 2 is an equivalent magnetic circuit diagram of FIG. 1. It is a block circuit diagram showing an example of a magnetic noise reduction type motor control circuit of the present invention. It is a circuit diagram which shows the specific example of the motor controller of FIG. It is a circuit diagram which shows the specific example of the motor controller of FIG. It is a circuit diagram which shows the specific example of the motor controller of FIG. It is a flowchart which shows a modification. It is a model radial direction cross section of the three-phase synchronous machine used for experiment. It is a wave form diagram (at the time of magnetic noise non-reduction) of each phase current of the three-phase synchronous machine of FIG. It is a wave form diagram (at the time of magnetic noise reduction) of each phase current of the three-phase synchronous machine of FIG. FIG. 9 is a frequency spectrum diagram of phase currents of the three-phase synchronous machine of FIG. 8 (when magnetic noise is not reduced and when magnetic noise is reduced). It is a wave form diagram (at the time of magnetic noise non-reduction and magnetic noise reduction) of the radial direction excitation force of the three-phase synchronous machine of FIG. FIG. 9 is a frequency spectrum diagram of radial excitation force of the three-phase synchronous machine of FIG. 8 (when magnetic noise is not reduced and when magnetic noise is reduced). It is a figure which shows the measurement result (at the time of electric drive) of the magnetic noise (magnetic sound) of the three-phase synchronous machine of FIG. It is a figure which shows the measurement result (at the time of electric power generation) of the magnetic noise (magnetic sound) of the three-phase synchronous machine of FIG. It is a flowchart which shows the control operation | movement which reduces the magnetic noise intrinsic | native in the three-phase alternating current rotating electrical machine mounted in the engine traveling vehicle on suitable conditions. It is a block circuit diagram which shows a reference example. FIG. 18 is a block circuit diagram showing an example of the circuit shown in FIG. 17. It is a wave form diagram which shows each part signal waveform (fundamental wave rotation coordinate system) in FIG. It is a wave form diagram which shows each part signal waveform (stationary coordinate system) in FIG.

Explanation of symbols

4 1st inverter 5 2nd inverter 6 1st motor controller 7 2nd motor controller 8 AC rotating electrical machine 81 1st phase winding group 82 2nd phase winding group

Claims (9)

AC rotating electric machine having an armature winding having a phase winding of m (m is a positive integer of 3 or more) phase;
Motor current control means for individually supplying each phase current having a predetermined fundamental frequency and amplitude to each phase winding;
In a motor control device comprising:
Each phase winding is
Divided into a first phase winding group and a second phase winding group each composed of a plurality of phase windings connected so as to be capable of energizing a phase current;
The motor current control means includes
A first inverter that forms a basic phase current of each phase mainly including a fundamental frequency component of the phase current and energizes the first phase winding group for each phase;
A second inverter that forms a harmonic current for reducing magnetic noise caused by the phase current and energizes the second phase winding group for each phase;
An AC rotating electrical machine device comprising: In the AC rotating electrical machine apparatus according to claim 1,
The second inverter is
In the phase and amplitude for attenuating the nth-order (multiple) harmonic component of the fundamental frequency component of radial vibration that is radially generated around the axis of the rotating shaft of the AC rotating electric machine, A harmonic current for reducing radial vibration of each phase mainly including an n + 1-order (times) harmonic component of a fundamental frequency component of a phase current is supplied to the second phase winding group for each phase. Magnetic noise reduction method for AC rotating electrical machines. In the AC rotating electrical machine apparatus according to claim 2,
The second inverter is
An AC rotating electrical machine apparatus characterized by forming the radial vibration reducing harmonic current mainly including an odd-numbered or higher harmonic component of the seventh or higher order with respect to a fundamental frequency component of a three-phase phase current. In the AC rotating electrical machine apparatus according to claim 3,
The second inverter is
An AC rotating electrical machine apparatus characterized by forming the harmonic current for radial vibration reduction mainly including a seventh harmonic component. In the AC rotating electrical machine apparatus according to claim 3,
The second inverter is
An AC rotating electrical machine apparatus that forms the radial vibration reducing harmonic current mainly including a 13th harmonic component. In the AC rotating electrical machine apparatus according to claim 3,
The second inverter is
An AC rotating electrical machine apparatus characterized by forming the harmonic current for radial vibration reduction mainly including seventh and thirteenth harmonic components. In the AC rotating electrical machine apparatus according to claim 2 ,
The second inverter is
It is operated in a vibration reduction operation mode that forms the radial vibration reduction harmonic current at a small output less than a predetermined value, and instead of the radial vibration reduction harmonic current at a large output of a predetermined value or more, An AC rotating electrical machine apparatus that is operated in an output priority mode in which a basic phase current of each phase mainly including a fundamental frequency component of a phase current is formed and the second phase winding group is energized for each phase. In the AC rotating electrical machine apparatus according to claim 3,
The second inverter is
An AC rotation characterized in that a harmonic component of the radial vibration having an 18th-order frequency is attenuated by superimposing the 19th-order radial vibration reducing harmonic current than when the superposition is not performed. Electric equipment. In the AC rotating electrical machine apparatus according to claim 3,
The second inverter is
An AC rotation characterized in that by superimposing the 25th-order radial vibration reducing harmonic current, the harmonic component of the radial vibration having a 24th-order frequency is attenuated as compared with the case where the superposition is not performed. Electric equipment.

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