JP4134730B2 - Diversity receiver - Google Patents

Description

[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a diversity receiver, and more particularly to a diversity receiver for reducing the influence of frequency selective fading due to multipath.
[0002]
[Prior art]
Conventionally, in a diversity receiver, a weighting factor is determined based on a correlation coefficient between signals received by each antenna. In the diversity receiver of Patent Document 1, the weighting coefficient is determined based on the correlation coefficient between the combined signal and the signal received by each antenna. As a result, even when the reception level of a specific antenna is extremely lowered, it is possible to reliably synthesize a desired wave and obtain a stable reception signal.
[0003]
[Patent Document 1]
Japanese Patent Laid-Open No. 2001-156689
[Problems to be solved by the invention]
Generally, when the main wave and the delayed wave are received simultaneously, the frequency characteristics of the received signal are distorted. The period of distortion depends on the delay time of the delayed wave, and the longer the delay time, the shorter the period of distortion on the frequency axis. In the diversity receiver described in Patent Document 1, all the bands of received signals are weighted and combined in a lump. Therefore, if the delay time of the delay wave is short and the distortion of the frequency characteristics is slight, it is possible to synthesize the signals in the entire band in the same phase by synthesizing based on the maximum ratio synthesis. The quality of can be greatly improved. However, if the delay time of the delayed wave becomes long, it becomes difficult to perform weighting so that signals in all the bands have the same phase even if weighting is performed based on the maximum ratio combining. That is, in the diversity receiving device described in Patent Document 1, when the delay time becomes long, the reception quality improvement effect may be reduced.
[0005]
The present invention solves the above-described problem, and even when the delay time of a delayed wave is long, diversity reception that can reduce the influence of frequency selective fading and improve the reception quality improvement effect. An object is to provide an apparatus.
[0006]
[Means for Solving the Problems]
A diversity receiving apparatus according to claim 1 of the present application is a diversity receiving apparatus in communication using multicarrier modulation, wherein a plurality of antennas and a division received by each of the plurality of antennas are divided into a plurality of frequency bands. And a combining means for weighting each divided signal and outputting a combined signal for each frequency band, and a combining means for aligning and combining the phases of the output combined signals for each frequency band. The combining means includes correlation coefficient calculating means for obtaining a correlation coefficient between the combined signals of adjacent frequency bands, and combining the combined signals by aligning phases based on the calculated correlation coefficients. And
[0010]
[Action and effect of the invention]
The diversity receiver of the present invention divides a received signal into a plurality of frequency bands and performs weighting for each frequency band to synthesize the signals. The received signal is stochastically degraded at a certain frequency component due to frequency selective fading due to multipath. This phenomenon occurs independently for each antenna. The synthesizing means for each frequency band can calculate a more accurate weighting coefficient in which the influence of the deteriorated signal component is reduced, and the signal level for each frequency band is leveled. As described above, the diversity receiver of the present invention can reduce the influence of frequency selective fading due to multipath.
[0011]
DETAILED DESCRIPTION OF THE INVENTION
Embodiments of the present invention will be described below with reference to the drawings.
(First embodiment)
FIG. 1 is a block diagram showing the diversity receiving apparatus according to the first embodiment of the present invention. The diversity receiver of this embodiment includes antennas 11, 12, 13, and 14, receivers 21, 22, 23, and 24, filters 31 and 32, combiners 41 and 42, a combiner 50, and a demodulator 60. . Each operation will be described below.
[0012]
The signals received by the antennas 11 to 14 are converted into frequencies suitable for signal processing by the receivers 21 to 24, respectively, A / D converted and orthogonally demodulated, and then into two types of digital filters (filters 31 and 32). Entered. The filters 31 and 32 are composed of a low-pass filter and a high-pass filter having a characteristic of dropping by 6 dB at the exact center of the signal frequency band. This is because when the combined signals of the respective bands are combined, the signal level of the overlapping portion on the frequency axis is restored.
[0013]
Thereafter, the combiners 41 and 42 combine the signals of the respective antennas in the respective frequency bands with maximum ratio combining or equal gain combining. In the present embodiment, maximum ratio synthesis is performed. This synthesizer can use the means described in Patent Document 1. As a result, a weighting factor that improves the S / N ratio for each frequency band is multiplied. That is, even if a certain frequency component of a certain antenna element is deteriorated due to frequency selective fading by multipath, the influence can be reduced.
[0014]
The synthesized signals synthesized by the synthesizers 41 and 42 are input to the combiner 50. The combiner 50 obtains a signal correlation between two adjacent synthesized signals. In the present embodiment, since there are originally two combined signals, the signal correlation between the two combined signals is obtained. The phases are aligned based on the obtained signal correlation. The phase in this case is the phase of the signal where the two signals overlap on the frequency axis. If this phase is not aligned, in the worst case (that is, the opposite phase), the signal component of the overlapping frequency due to coupling Are offset, so the phases are aligned.
[0015]
Strictly speaking, since the purpose of this coupling is to align the phases of the central portions, the overlapping portion on the frequency axis should be narrower, that is, it is more preferable for the filter to have as steep characteristics as possible.
[0016]
The combined signal is input to the demodulator 60. Increasing the number of divisions of the frequency band only increases the number of filters and synthesis units, and the operation principle is the same as described above.
[0017]
Here, it will be described that the circuit scale of the filter can be greatly reduced when the frequency band is divided into two as in the present embodiment. The filter is realized by a digital filter.
[0018]
In the case of two divisions, one kind of low-pass filter coefficient may be prepared. Generally speaking, the filter is considered in the real signal region, that is, in the positive frequency region, but here it is extended to the negative frequency region. That is, in this case, the low-pass filter with the cutoff F (Hz) is a band-pass filter with −F to F (Hz) (FIG. 2). By changing the filter coefficient of this filter, if it is moved + F (Hz) on the frequency axis, it becomes a band pass filter of 0-2F (Hz), and conversely, if it is moved -F (Hz), -2F-0 ( Hz) band-pass filter.
[0019]
When a digital filter is configured with a circuit having an operating frequency Fd (Hz), a low-pass filter of F = Fd / 4 (Hz) is prepared. The filter coefficients are real values {KR1, KR2,..., KRn}, which can be easily extended to the complex domain, and the (complex) filter coefficients in this case are {KR1 + 0 * j = K1, KR2 = 0 * j = K2,..., KRn + 0 * j = Kn} (where j * j = −1). When the (complex) filter coefficient is multiplied by the i-th power of the rotation coefficient exp (2πjφ / 360) by Ki, the frequency characteristic of the filter is −Fd / 4 + Fd * φ / 360 to Fd / 4 + Fd * φ / 360 (Hz ) Band pass characteristics. The filter characteristics when φ = 90 and −90 are a bandpass filter (upper filter) of 0 to 2 Fd (Hz) and a bandpass filter (lower filter) of −2 Fd to 0 (Hz), respectively.
[0020]
Here, φ = ± 90 is very important in considering the circuit scale. Since the filter is a process of accumulating the product of the signal {SRi + j * SIi} and the filter coefficient {KRi + j * KIi} with respect to i, originally four multiplications (SRi * KRi, SRi * Kii, SIi * KRI, SIi * KIi) is required. However, if φ = ± 90, either KRi or KIi becomes zero. In other words, only two multipliers per coefficient are required. Furthermore, since the sign of KIi is only partially inverted between the coefficient of the upper filter and the coefficient of the lower filter, the value after multiplication can be used for both the upper filter and the lower filter. Therefore, the filter can be configured with a multiplication amount that is 1/4 of the originally required multiplication amount.
[0021]
On the other hand, if the frequency band of the demodulated signal is set to −Fs to Fs (Fs <Fd / 2), for example, in the case of terrestrial digital broadcasting, Fs is about 3 MHz. In this system, when Fd is about 32 MHz, if the filter is designed as described above, the terrestrial digital broadcast signal output by the upper filter is a signal of 0 to 3 MHz, and the lower output is a signal of -3 to 0 MHz. Then, the signals of the respective bands are synthesized.
[0022]
(Second Embodiment)
FIG. 3 shows a block diagram of the diversity receiver according to the second embodiment. In the present embodiment, each of the filter and the combiner includes three (filters 31 to 33 and combiners 41 to 43). That is, the frequency band is divided into three. The operation of each component of the present embodiment is the same as that of the first embodiment. Here, the configuration of each filter will be described.
[0023]
In the case of the present embodiment as well, one type of low-pass filter coefficient may be prepared as in the first embodiment. A low-pass filter of F = Fd / 8 is prepared (FIG. 4). The (complex) filters obtained by multiplying the filter coefficients by three types of rotation coefficients φ = −90, 0, 90 are −3 Fd / 8 to −Fd / 8, −Fd / 8 to Fd / 8, and Fd, respectively. It becomes a bandpass filter of / 8 to 3Fd / 8. If this is the case, -3 to 3 MHz cannot be divided into three. Therefore, 0 is inserted between these filter coefficients. If 0 is inserted m by m, the frequency characteristic of the filter becomes 1 / (m + 1). Therefore, a new filter coefficient obtained by inserting m = 3 pieces of 0s between the band pass filter coefficients divides -3 to 3 MHz exactly into three.
[0024]
(Third embodiment)
Next, FIG. 5 shows a block diagram of the diversity receiver according to the third embodiment. In the present embodiment, each of the filter and the combiner includes five (filters 31 to 35 and combiners 41 to 45). That is, the frequency band is divided into five. The operation of each component is the same as that in the first embodiment. Here, the configuration of each filter will be described.
[0025]
In the case of the present embodiment as well, one type of low-pass filter coefficient may be prepared as in the first embodiment. A low-pass filter of F = Fd / 16 is prepared. The (complex) filters obtained by multiplying this filter coefficient by three types of rotation coefficients φ = −45, 0, and 45 are −3 Fd / 16 to −Fd / 16, −Fd / 16 to Fd / 16, and Fd, respectively. / 16 to 3Fd / 16 band-pass filter. As in the second embodiment, 0 is inserted between these filter coefficients. In the case of the present embodiment, 0 is inserted two by two, and the new filter coefficient that can be obtained is divided into −2 to 2 MHz by three. For the remaining −3 to −2 and 2 to 3 MHz, a filter multiplied by a rotation coefficient of φ = −45 and 45 becomes a bandpass filter of −6 to −2 and 2 to 6 MHz (Fd = 32 MHz). This may be applied as it is (FIG. 6).
[0026]
Here, φ = ± 45 is also very important in considering the circuit scale. Since the filter is a process of accumulating the product of the signal {SRi + j * SIi} and the filter coefficient {KRi + j * KIi} with respect to i, when i is an even number, it is the same as when φ = ± 90. When i is an odd number, the multiplication result for each coefficient is {√2 / 2 * KRi (± SRi ± SIi)}. Therefore, the coefficient {√2 / 2 * KRi} is obtained after addition / subtraction of the input signal. Multiply the number of multipliers by two. Further, since the signs of the upper filter coefficient and the lower filter coefficient are only partially inverted, the value after multiplication can be used for both the upper filter and the lower filter. Therefore, even when φ = ± 45, the filter can be configured with a multiplication amount that is a quarter of the originally required multiplication amount.
[0027]
FIG. 7 shows an example of the effect of the present invention. FIG. 7 shows the reception rate when mobile reception of terrestrial digital broadcasting is performed. The horizontal axis in FIG. 7 represents the delay time of the main wave and the delayed wave, and the vertical axis represents the proportion of time that has been successfully received. From FIG. 7, it can be seen that the degradation of the reception characteristics with respect to the delay time of the delayed wave can be reduced by increasing the number of divisions of the frequency band.
[0028]
It should be noted that the diversity receiving apparatus of the present invention is not limited to the above-described illustrated examples, and it is needless to say that various modifications can be made without departing from the gist of the present invention.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a configuration of a first exemplary embodiment of the present invention.
FIG. 2 is a diagram showing an operation of a filter according to the first embodiment of the present invention.
FIG. 3 is a block diagram showing a configuration of a second exemplary embodiment of the present invention.
FIG. 4 is a diagram showing an operation of a filter according to a second embodiment of the present invention.
FIG. 5 is a block diagram showing a configuration of a third exemplary embodiment of the present invention.
FIG. 6 is a diagram showing an operation of a filter according to a third embodiment of the present invention.
FIG. 7 is a graph showing the effect of the present invention.

Claims (1)

In a diversity receiver in communication using multicarrier modulation,
Multiple antennas,
Dividing means for dividing the signals received by the plurality of antennas into a plurality of frequency bands, respectively;
Combining means for weighting each of the divided signals and outputting a combined signal for each frequency band;
Have a coupling means for coupling aligns the phase of the composite signal for each of the output frequency band,
The combining means includes a correlation coefficient calculating means for obtaining a correlation coefficient between synthesized signals in adjacent frequency bands,
A diversity receiving apparatus characterized in that the combined signals are combined with the same phase based on the calculated correlation coefficient .

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