JP2004221808A - Diversity receiver - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a diversity receiver capable of enhancing a reception quality improvement effect by reducing the effect of frequency selective fading even when the delay time of a delayed wave gets longer. <P>SOLUTION: Signals received by antennas 11 to 14 are converted into signals with frequencies proper to signal processing respectively by receivers 21 to 24. The resulting signals are subjected to analog / digital conversion and orthogonal demodulation and given to two kinds of digital filters (31, 32). Synthesizers 41, 42 thereafter apply maximum ratio synthesis or equi-gain synthesis to the signals of each antenna for the respective frequency bands. A coupler 50 receives a synthesis signal synthesized by the synthesizers 41, 42 and the coupler 50 obtains signal correlation between two adjacent synthesis signals. The signals are synthesized with matched phases on the basis of the obtained signal correlation. <P>COPYRIGHT: (C)2004,JPO&NCIPI

Description

[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a diversity receiver, and more particularly, to a diversity receiver for reducing the influence of frequency selective fading due to multipath.
[0002]
[Prior art]
Conventionally, a diversity receiver determines a weighting factor based on a correlation coefficient between signals received by each antenna. In the diversity receiver of Patent Literature 1, a weight coefficient is determined based on a correlation coefficient between a combined signal and a signal received by each antenna. As a result, even when the reception level of a specific antenna is extremely lowered, it is possible to reliably synthesize a desired wave, and a stable reception signal can be obtained.
[0003]
[Patent Document 1]
JP 2001-156689 A
[Problems to be solved by the invention]
In general, when the main wave and the delayed wave are received at the same time, the frequency characteristics of the received signal are distorted. The period of the distortion depends on the delay time of the delay wave, and the longer the delay time, the shorter the period of the distortion on the frequency axis. In the diversity receiver described in Patent Literature 1, the entire band of the received signal is collectively weighted and combined. Therefore, if the delay time of the delay wave is short and the distortion of the frequency characteristic is slight, it is possible to combine the signals in all the bands in the same phase by combining based on the maximum ratio combining, and to combine the received signals. Quality can be greatly improved. However, when the delay time of the delayed wave becomes longer, even if weighting is performed based on the maximum ratio combining and combining, it is difficult to perform weighting so that signals in all bands have the same phase. That is, in the diversity receiving apparatus described in Patent Literature 1, when the delay time increases, the reception quality improvement effect may decrease.
[0005]
SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problem, and provides a diversity reception that can reduce the influence of frequency selective fading even when the delay time of a delayed wave is long, and improve the reception quality improvement effect. It is intended to provide a device.
[0006]
[Means for Solving the Problems]
A diversity receiving apparatus according to claim 1 of the present application is a diversity receiving apparatus in communication using multicarrier modulation, in which a plurality of antennas and a signal received by the plurality of antennas are divided into a plurality of frequency bands. Means, weighting the divided signals to output a synthesized signal for each frequency band, and coupling means for aligning and combining the phases of the output synthesized signals for each frequency band. It is characterized by the following.
[0007]
Further, the diversity receiving apparatus according to claim 2 of the present application is characterized in that the weighting factor used for weighting the combining means is determined based on maximum ratio combining or equal gain combining.
[0008]
Also, in the diversity receiving apparatus according to claim 3 of the present application, the coupling means has a correlation coefficient calculating means for calculating a correlation coefficient between synthesized signals of adjacent frequency bands, and based on the calculated correlation coefficient. It is characterized in that the combined signals are combined with the same phase.
[0009]
Further, the diversity receiving apparatus according to claim 4 of the present application is characterized in that the dividing means divides the frequency band into two, three, or five.
[0010]
Function and effect of the present invention
The diversity receiving apparatus of the present invention divides a received signal into a plurality of frequency bands, and weights and synthesizes each frequency band. The received signal is stochastically degraded in a certain frequency component due to frequency selective fading due to multipath. This phenomenon occurs independently for each antenna. In each frequency band synthesizing means, it is possible to calculate a more accurate weight coefficient in which the influence of the degraded signal component is reduced, and the signal level for each frequency band is leveled. As described above, the diversity receiving apparatus of the present invention can reduce the influence of frequency selective fading due to multipath.
[0011]
BEST MODE FOR CARRYING OUT THE INVENTION
Embodiments of the present invention will be described below with reference to the drawings.
(First Embodiment)
FIG. 1 is a block diagram showing a configuration of the diversity receiver according to the first embodiment of the present invention. The diversity receiver according to the present embodiment includes antennas 11, 12, 13, 14, receivers 21, 22, 23, 24, filters 31, 32, combiners 41, 42, combiner 50, and demodulator 60. . Hereinafter, each operation will be described.
[0012]
The signals received by the antennas 11 to 14 are converted into frequencies suitable for signal processing by the receivers 21 to 24, A / D converted and quadrature demodulated, and are converted into two types of digital filters (filters 31 and 32). Is entered. The filters 31 and 32 are composed of a low-pass filter and a high-pass filter having a characteristic of dropping 6 dB at the center of the frequency band of the signal. This is because when combining the synthesized signals of the respective bands, the signal level of the overlapping portion on the frequency axis is restored.
[0013]
Thereafter, the combiners 41 and 42 combine the signals of the respective antennas in the respective frequency bands with the maximum ratio combination or equal gain combination. In the present embodiment, maximum ratio combining is performed. For this synthesizer, the means described in Patent Document 1 can be used. As a result, a weighting factor that maximizes the S / N ratio is applied for each frequency band. That is, even if a certain frequency component of a certain antenna element is deteriorated due to frequency selective fading due to multipath, the effect can be reduced.
[0014]
Then, the combined signal combined by the combiners 41 and 42 is input to the combiner 50. The combiner 50 obtains a signal correlation between two adjacent synthesized signals. In the present embodiment, since there are two synthesized signals in the first place, the signal correlation between the two synthesized signals is obtained. The phases are aligned based on the obtained signal correlation. The phase in this case is the phase of the signal in the portion where the two signals overlap on the frequency axis. If the phases are not aligned, in the worst case (ie, the opposite phase), the signal components of the overlapping frequency are combined. Are canceled out, so the phases are aligned.
[0015]
Strictly speaking, since the purpose of this coupling is to make the phases of the central portions uniform, it is better that the overlapping portion on the frequency axis is narrower, that is, it is more preferable that the filter has as steep a characteristic as possible.
[0016]
The combined signal is input to demodulator 60. When the number of divisions of the frequency band is increased, only the number of filters and the number of synthesis units are increased, and the operation principle is the same as described above.
[0017]
Here, it will be described that the circuit size of the filter can be significantly reduced when the frequency band is divided into two as in the present embodiment. The filter is realized by a digital filter.
[0018]
In the case of division into two, one kind of low-pass filter coefficient may be prepared. Generally speaking, the filter is considered in the real signal domain, that is, in the positive frequency domain. Here, the filter is extended to the negative frequency domain. That is, in this case, the low-pass filter having the cutoff F (Hz) is a band-pass filter having a frequency of -F to F (Hz) (FIG. 2). By transforming the filter coefficient of this filter, a band pass filter of 0 to 2 F (Hz) is obtained by moving the filter coefficient on the frequency axis by + F (Hz), and conversely, by moving it by -F (Hz), -2 F to 0 ( Hz).
[0019]
When a digital filter is configured by a circuit having an operating frequency of Fd (Hz), a low-pass filter of F = Fd / 4 (Hz) is prepared. The filter coefficient is a real value {KR1, KR2,..., KRn}, which can be easily extended to the complex domain. In this case, the (complex) filter coefficient is {KR1 + 0 * j = K1, KR2 = 0 * j = K2,..., KRn + 0 * j = Kn｝ (where j * j = −1). When this (complex) filter coefficient is multiplied by Ki to the i-th power of a rotation coefficient exp (2πjφ / 360), the frequency characteristic of the filter is −Fd / 4 + Fd * φ / 360 to Fd / 4 + Fd * φ / 360 (Hz ). The filter characteristics when φ = 90 and −90 are a bandpass filter (upper filter) of 0 to 2 Fd (Hz) and a bandpass filter (lower filter) of −2 Fd to 0 (Hz), respectively.
[0020]
Here, it is very important that φ = ± 90 when considering the circuit scale. Since the filter is a process of accumulating the product of the signal {SRi + j * SIi} and the filter coefficient {KRi + j * KIi} with respect to i, four multiplications (SRi * KRi, SRi * Kii, SIi * KRI, SIi * KIi) is required. However, if φ = ± 90, either KRi or KIi becomes 0. That is, the number of multipliers per coefficient is two. Further, since the sign of KIi is only partially inverted between the coefficient of the upper filter and the coefficient of the lower filter, the value after the multiplication can be used for both the upper filter and the lower filter. Therefore, the filter can be configured with a multiplication amount that is one fourth of the originally required multiplication amount.
[0021]
On the other hand, if the frequency band of the demodulated signal is -Fs to Fs (Fs <Fd / 2), for example, in the case of terrestrial digital broadcasting, Fs is about 3 MHz. In this system, when Fd is set to about 32 MHz, if the filter is designed as described above, the terrestrial digital broadcast signal output by the upper filter becomes a signal of 0 to 3 MHz, and the lower output becomes a signal of -3 to 0 MHz. Then, the signals of the respective bands are combined.
[0022]
(Second embodiment)
FIG. 3 shows a block diagram of the diversity receiver according to the second embodiment. In the present embodiment, three filters and three combiners (filters 31 to 33 and combiners 41 to 43) are provided. That is, the frequency band is divided into three. The operation of each component of this embodiment is the same as that of the first embodiment. Here, the configuration of each filter will be described.
[0023]
Also in the case of the present embodiment, one type of low-pass filter coefficient may be prepared as in the case of the first embodiment. What is prepared is a low-pass filter of F = Fd / 8 (FIG. 4). The (complex) filters obtained by multiplying the filter coefficients by three kinds of rotation coefficients of φ = −90, 0, 90 are −3Fd / 8 to −Fd / 8, −Fd / 8 to Fd / 8, Fd, respectively. / 8 to 3Fd / 8. In this state, -3 to 3 MHz cannot be divided into three. Therefore, 0 is inserted between these filter coefficients. If 0s are inserted every m, the frequency characteristic of the filter becomes 1 / (m + 1). Therefore, a new filter coefficient obtained by inserting m = 3 0s between the coefficients of the band-pass filter is obtained by exactly dividing -3 to 3 MHz into three.
[0024]
(Third embodiment)
Next, a block diagram of a diversity receiver according to the third embodiment is shown in FIG. In the present embodiment, each of the filter and the combiner is composed of five filters (filters 31 to 35 and combiners 41 to 45). That is, the frequency band is divided into five. The operation of each component is the same as in the first embodiment. Here, the configuration of each filter will be described.
[0025]
Also in the case of the present embodiment, one type of low-pass filter coefficient may be prepared as in the case of the first embodiment. What is prepared is a low-pass filter of F = Fd / 16. (Complex) filters obtained by multiplying these filter coefficients by three kinds of rotation coefficients of φ = −45, 0, and 45 are −3Fd / 16 to −Fd / 16, −Fd / 16 to Fd / 16, and Fd, respectively. / 16 to 3Fd / 16. As in the second embodiment, 0 is inserted between these filter coefficients. In the case of the present embodiment, 0s are inserted two by two, and a new filter coefficient created by this is to divide -2 to 2 MHz into three. For the remaining -3 to -2 and 2 to 3 MHz, the filters multiplied by the rotation coefficients of φ = −45 and 45 become band-pass filters of −6 to −2 and 2 to 6 MHz (Fd = 32 MHz). This can be applied as it is (FIG. 6).
[0026]
Here, φ = ± 45 is also very important in considering the circuit scale. Since the filter is a process for accumulating the product of the signal {SRi + j * SIi} and the filter coefficient {KRi + j * KIi} with respect to i, when i is an even number, the same as when φ = ± 90. When i is an odd number, since the multiplication result for each coefficient is {2/2 * KRi (± SRi ± SIi)}, the coefficient {2/2 * KRi} after adding / subtracting the input signal. , The number of multipliers can be reduced to two. Further, since the sign of the coefficient of the upper filter and the coefficient of the lower filter are only partially inverted, the value after the multiplication can be used for both the upper filter and the lower filter. Therefore, even when φ = ± 45, the filter can be configured with a multiplication amount that is の of the originally required multiplication amount.
[0027]
FIG. 7 shows an example of the effect of the present invention. FIG. 7 shows the reception rate when mobile reception of digital terrestrial broadcasting is performed. The horizontal axis of FIG. 7 represents the delay time of the main wave and the delayed wave, and the vertical axis represents the ratio of the time during which the signal can be received normally. From FIG. 7, it can be seen that by increasing the number of divisions of the frequency band, the deterioration of the reception characteristics with respect to the delay time of the delayed wave can be reduced.
[0028]
It should be noted that the diversity receiving apparatus of the present invention is not limited to the illustrated example described above, and it is needless to say that various changes can be made without departing from the gist of the present invention.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a configuration of a first embodiment of the present invention.
FIG. 2 is a diagram showing an operation of a filter according to the first embodiment of the present invention.
FIG. 3 is a block diagram showing a configuration according to a second embodiment of the present invention.
FIG. 4 is a diagram illustrating an operation of a filter according to a second embodiment of the present invention.
FIG. 5 is a block diagram showing a configuration according to a third embodiment of the present invention.
FIG. 6 is a diagram illustrating an operation of a filter according to a third embodiment of the present invention.
FIG. 7 is a graph showing the effect of the present invention.

Claims (4)

In a diversity receiver in communication using multi-carrier modulation,
Multiple antennas,
Dividing means for dividing the signals received by the plurality of antennas for each of a plurality of frequency bands,
Combining means for weighting each of the divided signals to output a combined signal for each frequency band,
A diversity receiving device comprising: coupling means for aligning and combining the phases of the output synthesized signals for each frequency band. The diversity receiving apparatus according to claim 1, wherein a weighting coefficient used for weighting of the combining unit is determined based on maximum ratio combining or equal gain combining. The coupling means has a correlation coefficient calculation means for obtaining a correlation coefficient between the synthesized signals of adjacent frequency bands,
3. The diversity receiver according to claim 1, wherein the combined signals are combined with the phases thereof being adjusted based on the calculated correlation coefficient. The diversity receiving apparatus according to any one of claims 1 to 3, wherein the dividing unit divides the frequency band into two, three, or five.

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