JP4095632B2 - Interrogator - Google Patents

Description

  The present invention relates to an interrogator that captures response information from a responder using wireless communication, and in particular, transmits only a carrier following transmission of a signal modulated with a carrier, and backscatter modulates the carrier to obtain information. The present invention relates to an interrogator that receives a modulated signal from a responder that sends back.

  In recent years, an RFID (Radio Frequency Identification) system has been known as an automatic recognition technology using radio. The RFID system performs wireless communication between an interrogator called a reader / writer or the like and a responder called an RFID tag or the like. The interrogator transmits information to the responder using the modulated radio signal, and continues to transmit the unmodulated signal after the transmission of the information. The transponder transmits information from the transponder to the interrogator by performing backscatter modulation by changing the reflection amount of the unmodulated signal from the interrogator. The interrogator receives and demodulates the backscatter modulated wave, and reads the information of the responder.

As a configuration of such an interrogator, conventionally, on the transmission side, information is modulated by a modulator, amplified by an amplifier and transmitted from an antenna, and on the reception side, input from an antenna to a reception mixer, and from a high-frequency signal by a reception mixer It has been known that a baseband signal is extracted, demodulated by a demodulator, and information is extracted. Furthermore, a configuration using an orthogonal demodulator in which a receiving mixer and a demodulator are integrated has been known (see, for example, Patent Document 1).
JP 2004-021776 A

  By the way, in the RFID system, in order to increase the communication distance between the interrogator and the responder, it is necessary to increase the transmission power from the interrogator. For this purpose, it is necessary to increase the output from the amplifier on the transmission side. However, as the output from the transmission side increases, the power of the signal entering the reception side from the transmission side within the interrogator also increases. When the power of the signal entering from the transmission side to the reception side increases, the low noise amplifier and the quadrature demodulator on the reception side are saturated with the incoming signal. If the low-noise amplifier or the like is saturated, the minute signal that is the backscatter modulated signal from the responder cannot be reproduced, and the interrogator cannot receive information from the responder.

  The present invention has been made based on such circumstances, and the object of the present invention is to provide an interrogator that can accurately receive information from the responder by reducing the unmodulated signal that enters the receiving side from the transmitting side. Is to provide.

The present invention provides a transmission processing unit that outputs a high-frequency signal modulated by transmission information and an unmodulated high-frequency signal, and radiates the high-frequency signal output from the transmission processing unit as a radio wave and converts the received radio wave into a high-frequency signal. And an antenna unit for outputting and a reception processing unit for inputting and processing the high-frequency signal output from the antenna unit, and responding to information by reflecting the radio wave received based on the signal processed by the transmission processing unit A phase shifter that divides the input high-frequency signal into two signals having a phase difference of 90 degrees, adjusts the amplitude of each of the branched signals, and combines them . a branching device for branching a high-frequency signal output from the transmission processing unit in the direction of the direction and the phase shifter of the antenna unit, the signal which is input from the transmission processing unit via the antenna unit A combiner which inputs the output signal from the phase vessel synthesis and outputs a combined signal to the reception processing unit, a first power detector for detecting power of a signal input to the combiner via the antenna unit A second power detector for detecting the power of the combined signal output from the combiner, and the first power detector when an unmodulated high-frequency signal is transmitted from the transmission processing unit. The amplitude adjustment amount of the signal of the phase shifter so that the phase is inverted with the same amplitude as the signal input to the combiner via the antenna unit from the power and the power detected by the second power detector And a control means for controlling the phase shifter according to the calculation result .

  According to the present invention in which such measures are taken, it is possible to provide an interrogator that can reduce the unmodulated signal that wraps around from the transmission side to the reception side, and that can improve the accuracy of receiving wireless information from the responder.

The best mode for carrying out the present invention will be described below with reference to the drawings.
[First Embodiment]
First, an RFID system according to the present invention will be described.

  FIG. 2 is a schematic configuration diagram of the RFID system. The RFID system includes one interrogator 1 and a plurality of RFID tags 2a, 2b, and 2c. The interrogator 1 includes an antenna 3. The antenna 3 radiates a high-frequency signal as a radio wave during transmission and functions to convert the radio wave into a high-frequency signal during reception. The radio waves radiated from the antenna 3 reach the RFID tags 2a, 2b, and 2c, and are received by the RFID tags 2a, 2b, and 2c. Each RFID tag 2a, 2b, 2c stores a unique identification number.

  When the interrogator 1 wirelessly transmits an inquiry signal, each RFID tag 2a, 2b, 2c operates corresponding to the inquiry signal. For example, when the inquiry signal includes only data addressed to the identification number corresponding to the RFID tag 2a, only the RFID tag 2a returns a response, and the RFID tags 2b and 2c do not respond.

  For example, when the interrogator 1 transmits an inquiry signal to the RFID tag 2a, the interrogator 1 radiates a high-frequency signal modulated by the signal to be transmitted from the antenna 3. When the signal to be transmitted is completed, it continues to transmit an unmodulated high-frequency signal using only a carrier wave. After receiving the high-frequency signal modulated by the interrogator 1, the RFID tag 2 a performs backscatter modulation that changes the amount of reflection of the carrier according to the response information, and sends the response information signal back to the interrogator 1. The interrogator 1 receives a backscatter-modulated signal while transmitting a carrier wave, and receives information from the RFID tag 2a. When the inquiry signal is transmitted to the other RFID tags 2b and 2c, the interrogator 1 and the RFID tags 2b and 2c perform the same operation. Here, the RFID tags 2a, 2b, and 2c constitute a responder that responds to information by reflecting received radio waves.

  The transmission / reception time between the interrogator 1 and the RFID tags 2a, 2b, 2c will be described with reference to FIG. The horizontal axis is a time axis, and advances to the right side of the axis as time passes. When transmission of a radio wave modulated by command data from the interrogator 1 is started, the RFID tags 2a, 2b, 2c generate a voltage from the received radio wave and activate it to receive a command. When the interrogator 1 finishes transmitting the command, it transmits only the unmodulated carrier wave. The RFID tags 2a, 2b, and 2c operate in accordance with the received command, and return a response by performing backscatter modulation on the unmodulated carrier wave. Interrogator 1 receives the backscatter modulated response and stops transmitting the unmodulated carrier wave. The RFID tags 2a, 2b, and 2c stop operating when the radio wave that generates the starting voltage is stopped.

Next, a first embodiment of the present invention will be described.
FIG. 1 is a block diagram showing a main configuration of an interrogator 1 according to the first embodiment. The interrogator 1 includes a control unit 11, a PLL (Phase Locked Loop) circuit 12, a modulator 13, a transmission side mixer 14, a transmission side amplifier 15, a branching unit 16, a power amplifier 17, a circulator 18, a phase shifter 19, and a synthesis. 20, a low noise amplifier 21, a receiving side mixer 22, an LPF (Low Pass Filter) 23, a demodulator 24, and the like.

  The control unit 11 performs an operation of sending out digital transmission data and receiving digital reception data. Also, the PLL circuit 12 is set. Although not shown, the control unit 11 has a function of performing wired communication with a host device such as a personal computer. The PLL circuit 12 generates a high-frequency sine wave signal. In this embodiment, the frequency of the high frequency signal generated by the PLL circuit 12 is the same frequency as the unmodulated carrier signal.

  The modulator 13 modulates the digital data sent from the control unit 11 and outputs it to the transmission side mixer 14. When the interrogator 1 transmits an unmodulated high frequency signal, the modulator 13 does not operate. The transmission side mixer 14 outputs a signal obtained by adding the modulation signal generated by the modulator 13 and the high frequency signal generated by the PLL circuit 12 to the transmission side amplifier 15. Therefore, when information is transmitted from the interrogator 1, a signal obtained by adding the modulation signal and the high-frequency signal is output to the transmission-side amplifier 15. On the other hand, when the interrogator 1 transmits an unmodulated high-frequency signal, a signal having the same frequency as the high-frequency signal generated by the PLL circuit 12 is output to the transmission-side amplifier 15. The transmission-side amplifier 15 increases the amplitude of the high-frequency signal output from the transmission-side mixer 14 and outputs it to the branching device 16. Here, the modulator 13, the transmission-side mixer 14, and the transmission-side amplifier 15 constitute a transmission processing unit.

The branching device 16 branches the high-frequency signal output from the transmission-side amplifier 15 to the power amplifier 17 and the phase shifter 19 and outputs it. The power amplifier 17 is called PA (Power Amplifier), and further increases the amplitude of the high-frequency signal input via the branching device 16 and outputs it to the circulator 18. The circulator 18 has a characteristic that a signal input from the power amplifier 17 side is output to the antenna 3 side, and a signal input from the antenna 3 side is output to the combiner 20 side. If the characteristics of the circulator 18 are ideal, the high-frequency signal input from the power amplifier 17 side is not transmitted to the synthesizer 20 side. However, actually, the high frequency signal input from the power amplifier 17 side is attenuated by about 10 to 30 dB and enters the synthesizer 20 side. In addition, there is a signal transmitted from the antenna 3 side to the combiner 20 side due to signal reflection due to impedance mismatch between the circulator 18 and the antenna 3. Here, the circulator 18 constitutes an antenna unit together with the antenna 3.

The phase shifter 19 shifts the phase of the high-frequency signal input via the branching device 16 and outputs it to the combiner 20. The synthesizer 20 outputs a signal obtained by adding the high frequency signal output from the circulator 18 and the high frequency signal output from the phase shifter 19 to the low noise amplifier 21. Here, the phase of the high frequency signal at the input terminal on the phase shifter 19 side of the synthesizer 20 is determined by the length of the wiring from the branching device 16. Therefore, the phase shifter 19 can be created only by shifting the phase of the input high-frequency signal according to the length of the wiring, that is, the line length.

  The low noise amplifier 21 is a low noise amplifier called LNA (Low Noise Amplifier), amplifies the high frequency signal output from the synthesizer 20, and outputs the amplified signal to the reception-side mixer 22. The reception-side mixer 22 combines the high-frequency signal output from the low-noise amplifier 21 with the high-frequency signal from the PLL circuit 12 to convert it into a baseband signal, and outputs it to the LPF 23. The LPF 23 is a filter that does not pass high frequencies but passes low frequencies, and allows only the baseband signal output from the reception-side mixer 22 to pass. The demodulator 24 demodulates the baseband signal that has passed through the LPF 23, converts it to digital data, and outputs the digital data to the control unit 11. Here, the low-noise amplifier 21, the reception-side mixer 22, the LPF 23, and the demodulator 24 constitute a reception processing unit.

As described above, the interrogator 1 according to the present embodiment radiates, as radio waves, a transmission processing unit that outputs a high-frequency signal modulated by transmission information and an unmodulated high-frequency signal, and a high-frequency signal output from the transmission processing unit. In addition to the antenna unit that converts the received radio wave into a high-frequency signal and outputs it, and the reception processing unit that inputs and processes the high-frequency signal output from this antenna unit, the phase of the input high-frequency signal is shifted and output A phase shifter 19, a branching device 16 for branching a high-frequency signal output from the transmission processing unit in the direction of the antenna unit and the phase shifter 19, and a signal output from the antenna unit to the reception processing unit side and the shift. An output signal from the phase shifter 19 is input and combined, and a combiner 20 that outputs the combined signal to the reception processing unit is provided.

The signal synthesis performed by the synthesizer 20 will be described with reference to the waveform diagram of FIG. The horizontal axis of the waveform diagram is the time axis and shows one cycle of the carrier wave. The vertical axis indicates the amplitude of the signal. The signal S 1 is a signal input from the circulator 18 to the combiner 20, and the signal S 2 is a signal input from the phase shifter 19 to the combiner 20. In the synthesizer 20, the signal S3 is generated and output by combining the signal S1 and the signal S2. The phase shifter 19 outputs an unmodulated high frequency signal output from the combiner 20 when the signal S2 and the signal S1 of the unmodulated high frequency signal component input from the antenna unit are combined in the combiner 20. A signal S2 set so that the amplitude of the signal S3 of the signal component is smaller than the amplitude of the signal S1 of the unmodulated high-frequency signal component input from the antenna unit is input to the combiner 20.

In this manner, the signal S1 input from the circulator 18 to the combiner 20 and the signal S2 input from the phase shifter 19 to the combiner 20 are combined by the combiner 20 and output from the combiner 20. The amplitude of the carrier component of the high-frequency signal can be reduced. When the amplitude of the carrier component of the high-frequency signal output from the synthesizer 20 is reduced, the output signal from the synthesizer 20 is not saturated even when amplified by the low noise amplifier 21.

Normally, the signal that is input from the power amplifier 17 through the circulator 18 to the combiner 20 has the same frequency as the signal that is input from the branching unit 16 through the phase shifter 19 and input to the combiner 20. The only difference is the amplitude component. On the other hand, since the RFID tags 2a, 2b, and 2c perform backscatter modulation on the radio waves radiated from the antenna 3, the modulation signal received by the antenna 3 includes a frequency other than the carrier wave. However, this modulated signal is at a very low level compared to the signal that enters the combiner 20 from the power amplifier 17. However, in the synthesizer 20, only the frequency component of the high frequency signal entering from the power amplifier 17 is removed by the signal input from the phase shifter 19, so that the modulation signal from the RFID tags 2 a, 2 b, 2 c is not attenuated.

  Therefore, according to the present embodiment, the amplification factor for the modulation signal can be increased in the circuit of the reception processing unit such as the low noise amplifier 21, so that the distance between the interrogator 1 and the RFID tags 2a, 2b, and 2c is increased. Even when the power of the response signal from the RFID tags 2a, 2b, 2c received by the interrogator 1 is reduced, the carrier signal component entering from the transmission side to the reception side can be reduced, so the interrogator 1 correctly demodulates the response signal. be able to. As a result, information from the RFID tags 2a, 2b, and 2c can be received with high accuracy.

When the amplitude of the signal input from the power amplifier 17 through the circulator 18 to the synthesizer 20 and the signal input from the phase shifter 19 to the synthesizer 20 are the same, and the phases are inverted and synthesized, the synthesizer 20 Output becomes “0”. However, in practice, it is difficult to set the output of the synthesizer 20 to “0”, and it should be used as close to “0” as possible.

By the way, as described above, the phase of the high frequency signal at the input terminal on the phase shifter 19 side of the synthesizer 20 is determined by the length of the wiring from the branching device 16, so in this embodiment, the phase shifter. 19 is created only by the length of the wiring. Therefore, an effect that the configuration is simple can be achieved.

In the present embodiment, the phase shifter 19 is not limited to this. Another configuration example of the phase shifter 19 is shown in the block diagram of FIG. The phase shifter 19 includes a 3 dB hybrid coupler 31, a pair of fixed attenuators 32 and 33, and a combiner 34.

  The hybrid coupler 31 has four input / output terminals. When a high frequency signal is input to one input / output terminal, a signal attenuated by 3 dB from the input power is output to the two input / output terminals. In this case, there is a characteristic that the phases of the signals output from both input / output terminals are shifted by 90 degrees. The remaining one input / output terminal ideally has no output power. An output signal from the branching device 16 is input to a terminal described as IN. The terminal connected to the terminal is a terminal with no output power, and is generally terminated with a terminal resistance of 50 ohms.

  The input / output terminal described as 0 ° of the hybrid coupler 31 is connected to one fixed attenuator 32, and is 90 degrees out of phase with the signal output from the input / output terminal described as 0 °. The input / output terminal described as 90 ° from which the signal is output is connected to the other fixed attenuator 33. The pair of fixed attenuators 32 and 33 has a function of attenuating each input high frequency signal by a fixed attenuation amount and outputting the attenuated signal to the synthesizer 34. The synthesizer 34 adds the high-frequency signals output from the pair of fixed attenuators 32 and 33 and outputs the sum to the synthesizer 20.

If the input signal of one fixed attenuator 32 is sin θ, the input signal of the other fixed attenuator 33 can be expressed as cos θ because the phase is shifted by 90 degrees. In this case, the output signal of one fixed attenuator 32 is A × sin θ, and the amplitude A is determined by the set attenuation amount. Similarly, the output signal of the other fixed attenuator 33 is B × cos θ, and the amplitude B is determined by the set attenuation amount. When the output signal Asinθ of one fixed attenuator 32 and the output signal Bcosθ of the other fixed attenuator 33 are added together by the synthesizer 34, the output signal is expressed by the following [Equation 3 ].

  Therefore, each fixed attenuator 32, so that a signal whose amplitude of the carrier wave component is the same as that of the signal input to the combiner 20 through the circulator 18 from the power amplifier 17 and whose phase is inverted is output from the combiner 34. By setting the amount of attenuation 33, the same effects as in the present embodiment can be obtained.

[Second Embodiment]
Since the phase shift amount in the phase shifter 19 is fixed, the first embodiment is effective when a signal entering the circuit of the reception processing unit from the power amplifier 17 is determined. However, the signal that enters the circuit of the reception processing unit from the power amplifier 17 may differ depending on the antenna to be connected. Therefore, a second embodiment that is effective in such a case will be described next.

FIG. 6 is a block diagram showing the configuration of the main part of the interrogator 1 in the second embodiment, and the same reference numerals are given to the parts common to FIG. 1 used in the first embodiment. That is, the second embodiment is different from the first embodiment in that the power detector 41 is interposed between the combiner 20 and the low noise amplifier 21 and the phase shifter is branched. The phase shifter 42 that changes the phase and amplitude of the high-frequency signal input through the device 16 and outputs the same to the combiner 20 and the control unit 43 shifts the phase according to the detection output of the power detector 41. It has a function as a control means for controlling the device 42.

The power detector 41 outputs a voltage corresponding to the power of the high frequency signal input from the combiner 20 to the control unit 43 and transmits the input high frequency signal to the low noise amplifier 21. The control unit 43 controls the phase shifter 42 so as to reduce the power detected by the power detector 41 when only the carrier wave is output from the power amplifier 17.

One structural example of the phase shifter 42 is shown in the block diagram of FIG. The phase shifter 42 includes a 3 dB hybrid coupler 51, a pair of variable attenuators 52 and 53, and a combiner 54. The hybrid coupler 51 has the same function as the hybrid coupler 31 shown in FIG. 5, and an output signal from the branching device 16 is input to an input / output terminal indicated as IN.

  Each of the pair of variable attenuators 52 and 53 has a function of attenuating the input high-frequency signal and outputting the attenuated high-frequency signal to the synthesizer 54, and the amount of attenuation can be changed according to the change of the control signals CNT1 and CNT2. It is. The control unit 43 controls the attenuation amount of the variable attenuator 52 by outputting the control signal CNT1 to the variable attenuator 52. Further, by outputting the control signal CNT2 to the variable attenuator 53, the attenuation amount of the variable attenuator 53 is controlled.

  The synthesizer 54 adds the high-frequency signals output from the pair of variable attenuators 52 and 53 and outputs the sum to the synthesizer 20.

If the input signal of one variable attenuator 52 is sin θ, the input signal of the other variable attenuator 53 is 90 degrees out of phase and can be expressed as cos θ. In this case, the output signal of one variable attenuator 52 is A × sin θ, and the amplitude A is determined by the amount of attenuation controlled by the control unit 43. Similarly, the output signal of the other variable attenuator 53 is B × cos θ, and the amplitude B is determined by the amount of attenuation controlled by the control unit 43. When the output signal Asinθ of one variable attenuator 52 and the output signal Bcosθ of the other variable attenuator 53 are added together by the synthesizer 54, the output signal is expressed by the above [Equation 3 ].

Since the amplitude A is changed by changing the attenuation amount of one of the variable attenuator 52, the amplitude B varied by varying the attenuation of the other of the variable attenuator 53, a phase shifter 42 The amplitude and phase of the input high frequency signal can be controlled and output.

In the second embodiment including the power detector 41, the phase shifter 42, and the control unit 43 having such a configuration, when only the carrier wave is output from the power amplifier 17, the control unit 43 The phase shifter 42 is controlled so as to reduce the power detected by the detector 41. By doing so, the high frequency signal that enters from the power amplifier 17 through the circulator 18 and the high frequency signal output from the phase shifter 42 are added together by the synthesizer 20, and the high frequency signal input to the low noise amplifier 21 is added. Of these, the carrier wave, which is the component of the high-frequency signal that does not contain information, is reduced. Further, not only the carrier wave, but also signal distortion generated by the transmission side mixer 14 and the transmission side amplifier 15 is reduced by the synthesizer 20. Therefore, even when the signal entering the circuit of the reception processing unit from the power amplifier 17 differs depending on the connected antenna or the like, unnecessary signal components input to the low noise amplifier 21 can be reduced, and the reception sensitivity can be improved. Play.

In the second embodiment, the phase shifter 42 is not limited to that shown in FIG. 7. Even if the variable attenuators 52 and 53 are replaced with variable amplifiers, the amplitude and phase of the high-frequency signal are changed. Can be controlled. Incidentally, when the amplification factor of the power amplifier 17 is low, a variable attenuator may be used for the phase shifter 42. When the amplification factor of the power amplifier 17 is high, a variable amplifier is used for the phase shifter 42. That's fine.

  In addition, the variable phase amount can be changed by using a circuit that outputs signals having a phase difference of 180 degrees instead of the 3 dB hybrid coupler 51 that outputs signals having a phase difference of 90 degrees.

  Further, the branching device 16 is disposed between the transmission side amplifier 15 and the power amplifier 17, but the same effect can be obtained when the branching device 16 is disposed between the power amplifier 17 and the circulator 18.

[Third Embodiment]
Next, a third embodiment that is effective even in an environment where an object that affects antenna characteristics passes in the vicinity of the antenna 3 of the interrogator 1 will be described.

FIG. 9 is a block diagram showing the configuration of the main part of the interrogator 1 in the third embodiment, and the same reference numerals are given to the parts common to FIG. 6 used in the second embodiment. That is, the third embodiment is different from the second embodiment in that the first power detector 61 is interposed between the circulator 18 and the synthesizer 20, The control unit 63 controls the phase shifter 42 in accordance with the point where the second power detector 62 is interposed between the noise amplifier 21 and the detection outputs of the first and second power detectors 61 and 62. It is the point which had the function to do. Further, the control unit 63 stores the set value and output value of the phase shifter 42.

  The high-frequency signal output from the power amplifier 17 is output from the circulator 18 to the antenna 3 side, but a part of the first power detection is performed from the antenna 3 side due to signal reflection due to impedance mismatch between the circulator 18 and the antenna 3. Is transmitted to the device 61 side. In this case, since the antenna 3 is affected by objects in the vicinity, the amount of reflection of the high-frequency signal supplied from the circulator 18 to the antenna 3 varies depending on the presence or absence of an object that affects the vicinity.

Therefore, the first power detector 61 outputs a voltage corresponding to the power of the high-frequency signal input from the circulator 18 to the control unit 63 and transmits the input high-frequency signal to the synthesizer 20. The synthesizer 20 adds and outputs the high frequency signal output from the first power detector 61 and the high frequency signal output from the phase shifter 42. The second power detector 62 outputs a voltage corresponding to the power of the high frequency signal input from the combiner 20 to the control unit 63 and transmits the input high frequency signal to the low noise amplifier 21.

The schematic control operation of the control unit 63 in the third embodiment will be described with reference to the flowchart of FIG. The control unit 63 transmits a command to the responder at P1. When the command transmission is completed, the transmission of only the carrier wave, that is, the unmodulated high frequency signal is controlled at P2. In P3, the phase and amplitude of the high-frequency signal output from the phase shifter 42 are controlled. Thereafter, at P4, a response is received from the responder. When the response reception from the responder is completed, the transmission of the carrier wave is stopped at P5.

As described above, the setting change of the phase shifter 42 is performed as follows in the period from the start of transmission of only the P3 carrier to the start of response reception from the P4 responder.

First, since the power of the carrier wave input from the circulator 18 is detected by the first power detector 61 and a signal corresponding to the detected power is input to the control unit 63, the control unit 63 transfers the transfer stored in advance. from the set value of the phase vessel 42 and the output value, and sets the phase shifter 42 so that the output power of the power and the phase shifter 42 of the first power detector 61 are the same.

Here, when the phase shifter 42 shown in FIG. 7 is used, the first variable attenuator 52 and the second variable attenuator 53 have the same attenuation, and the output power of the combiner 54 is The first and second variable attenuators 52 and 53 are set so as to be the same as the power detected by the first power detector 61.

The amplitude of the signal input from the first power detector 61 to the combiner 20 is C, the phase is φ, and the signal input from the first power detector 61 to the combiner 20 is C × sin (ωt + φ ). The signal input from the phase shifter 42 to the synthesizer 20 is set so that the amplitude is C, and the frequency is the same as the signal input from the first power detector 61 to the synthesizer 20. Since only the phases are different, C × sin [ωt + tan ⁻¹ (B / A)], and the signal input to the second power detector 62 is expressed by the following equation ( 4 ). Become.

Here, when the amplitude of the signal input to the second power detector 62 is D, the amplitude of the equation represented by [Equation 4 ] can be represented by the following [Equation 5 ].

振幅が最小となるＤ＝０にするためには、φ−tan^-1(B/A)=πとすればよい。そこで制御部６３は、第１の電力検出器６１の出力と第２の電力検出器６２の出力から、第１の可変減衰器５２と第２の可変減衰器５３の出力が、それぞれ次の［数７］，［数８］で示されるように、可変減衰器５２，５３を制御すればよい。

Here, since A = B is set, when the equation shown in [Equation 5 ] is modified, the following equation shown in [Equation 6 ] is obtained.

In order to set D = 0 at which the amplitude is minimized, φ−tan ⁻¹ (B / A) = π may be set. Therefore, the control unit 63 determines that the outputs of the first variable attenuator 52 and the second variable attenuator 53 from the outputs of the first power detector 61 and the second power detector 62 are the following [ The variable attenuators 52 and 53 may be controlled as shown in [Equation 7 ] and [Equation 8 ].

  Actually, the resolution at which the attenuation amount can be set is finite or there is a variation in each part. Therefore, after performing the above-described control, as a fine adjustment, the settings of the amplitude A and the amplitude B are changed little by little. The power detection value of the second power detector 62 is minimized.

By doing so, the setting of the phase shifter 42 can be completed in a short time from the transmission of only the carrier wave to the start of reception of the response signal.

  Thus, also in the third embodiment, the power of the carrier wave output from the synthesizer 20 can be reduced, and the low noise amplifier 21 is not attenuated without attenuating the modulated signals received from the radio tags 2a, 2b, 2c. Can be entered. A good reception sensitivity can be obtained.

Further, the phase shifter 42 is controlled between the command transmission P1 and the response reception P4. However, only the carrier wave for starting the RFID tag is transmitted before the command transmission P1 is transmitted. Even if the phase shifter 42 is controlled during transmission of only the carrier wave, the same effect can be obtained.

Note that the present invention is not limited to the above-described embodiments as they are, and can be embodied by modifying the constituent elements without departing from the spirit of the invention in the implementation stage.
For example, the configurations of the transmission processing unit and the reception processing unit are not limited to those of the present embodiment, and it is needless to say that the transmission processing unit and the reception processing unit can be modified as appropriate according to the application.

  In addition, various inventions can be formed by appropriately combining a plurality of constituent elements disclosed in the respective embodiments. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, the constituent elements over different embodiments may be combined.

The block diagram which shows the principal part structure of the interrogator which is the 1st Embodiment of this invention. 1 is a schematic diagram showing a schematic configuration of an RFID system according to the present invention. The schematic diagram which shows the relationship of the transmission / reception time of the interrogator and RFID tag in the RFID system. The waveform diagram explaining the synthesis | combination of the signal performed with a combiner | synthesizer. The block diagram which shows the other structural example of the phase shifter in 1st Embodiment. The block diagram which shows the principal part structure of the interrogator which is the 2nd Embodiment of this invention. The block diagram which shows the example of 1 structure of the phase shifter in the said 2nd Embodiment. The flowchart which shows the outline of the control procedure which the control part of the interrogator which is the 3rd Embodiment of this invention performs. The block diagram which shows the principal part structure of the interrogator which is the said 3rd Embodiment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Interrogator, 2a, 2b, 2c ... RFID tag (responder), 3 ... Antenna, 11, 43, 63 ... Control part, 13 ... Modulator, 16 ... Branch device, 18 ... Circulator, 19, 42 ... Transfer phase vessel, 20 ... synthesizer, 24 ... demodulator, 41,61,62 ... power detector.

Claims (5)

A transmission processing unit that outputs a high-frequency signal modulated by transmission information and an unmodulated high-frequency signal, radiates the high-frequency signal output from the transmission processing unit as a radio wave, converts the received radio wave into a high-frequency signal, and outputs the signal A responder comprising an antenna unit and a reception processing unit that inputs and processes a high-frequency signal output from the antenna unit, and responds to information by reflecting a radio wave received based on the signal processed by the transmission processing unit In the interrogator that captures response information from
A phase shifter that divides the input high-frequency signal into two signals having a phase difference of 90 degrees, adjusts the amplitude of each of the branched signals, and then synthesizes ;
A branching device that branches the high-frequency signal output from the transmission processing unit in the direction of the antenna unit and the direction of the phase shifter;
A combiner that inputs and combines a signal input from the transmission processing unit via the antenna unit and an output signal from the phase shifter and outputs a combined signal to the reception processing unit ;
A first power detector that detects power of a signal input to the combiner via the antenna unit;
A second power detector for detecting the power of the combined signal output from the combiner;
From the power detected by the first power detector and the power detected by the second power detector when the unmodulated high frequency signal is transmitted from the transmission processing unit, the combiner Control means for calculating an amplitude adjustment amount of the signal of the phase shifter so that the phase is inverted with the same amplitude as that of the signal input through the antenna unit, and controlling the phase shifter according to the calculation result ; An interrogator characterized by comprising. 2. The interrogator according to claim 1 , wherein the control means controls the phase shifter so that the power detected by the second power detector becomes smaller . The control means is configured to receive the radio wave reflected from the responder after the unmodulated high-frequency signal is output following the high-frequency signal modulated by the transmission information. 3. The interrogator according to claim 2 , wherein the phase shifter is controlled so that the power detected by the second power detector is small . The antenna unit radiates a high-frequency signal as a radio wave, converts the received radio wave into a high-frequency signal, and outputs the high-frequency signal output from the transmission processing unit to the antenna, and is input from the antenna interrogator according to any one of claims 1 to 3, characterized in that it consists of a circulator for outputting a signal to a receiver processor. When the amplitude corresponding to the power detected by the first power detector is C and the amplitude corresponding to the power detected by the second power detector is D, the control means The amplitude B of the other signal with a phase difference of 90 degrees in the phase shifter is calculated by the equation [Equation 2] so that the amplitude A of the branched signal becomes the value calculated by the equation [Equation 1]. The interrogator according to claim 1 , wherein the phase shifter is controlled so as to be a value calculated by:

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