JP4031215B2 - Sensor signal processing circuit - Google Patents

Sensor signal processing circuit Download PDF

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JP4031215B2
JP4031215B2 JP2001184428A JP2001184428A JP4031215B2 JP 4031215 B2 JP4031215 B2 JP 4031215B2 JP 2001184428 A JP2001184428 A JP 2001184428A JP 2001184428 A JP2001184428 A JP 2001184428A JP 4031215 B2 JP4031215 B2 JP 4031215B2
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voltage
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JP2003004482A (en
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誉 増田
康秀 吉川
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Azbil Corp
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Azbil Corp
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Description

【０００１】
【発明の属する技術分野】
本発明は、センサ部からの信号を被測定物理量の関数として取り出すセンサ信号処理回路に関し、特に、温度変化等に基づく素子誤差およびオペアンプのオフセット等の回路誤差を信号処理によって除去する手段を備えたセンサ信号処理回路に関する。
【０００２】
【従来の技術】
圧力計測の分野では、機械式から電子式への置き換えが急速に進んでいる。電子式圧力計を大別すると、感圧ダイアフラムの応力変化を電気抵抗変化に変換する抵抗式と、感圧ダイアフラムの変位を静電容量変化に変換する静電容量式とに分類できる。この中で、静電容量式圧力センサには微圧計測に優れているという特徴がある。
【０００３】
静電容量式圧力センサは台座基板およびダイアフラムなどによって囲まれた空間が容量性リアクタンスを構成することを利用して、加圧する圧力を計測する原理を有するため、微小圧力の場合にはリアクタンスの変化も微小なものになる。
【０００４】
そのため、リアクタンスの変化を電気信号に変えるには、電気信号もいきおい微小なものになり、種々のノイズが重畳するため、信号対雑音比を高め計測精度を上げることが難しかった。そのため、従来は容量式リアクタンスを用いた積分回路を設けて、そのリアクタンスを計測する方法がとられてきた（特許 第1500174号 容量式変位変換方式）。しかし、積分動作による方法では、電荷蓄積のためにある程度の時間が必要なため、応答性能に問題があった。
【０００５】
近年、電子部品集積化技術の進歩により、交流的信号を容量性リアクタンス素子に印加して小信号演算技術を用いて、直流的な熱的ドリフトなどのノイズを除去する方法が開発されてきている。そこでは、最終的に被測定対象の物理量を直流的信号成分に復調しなければならないために、交流的信号から信号分を検波し復調する方法が取られる必要があり、ダイオードによる検波方式が一般的であった（USPatentNo.5942692 CapacitivePressureSensingMethod）。しかし、ダイオードによる半波または全波整流の検波方式では、ダイオード特性の熱的変化などが影響して精度に難がある、また信号対雑音比を高めるために誘導性インダクタンス素子を加えることなどが必要となり、回路が複雑となっていた。
【０００６】
本出願人による特許出願でも、交流的小信号演算を施して、センサ容量に並列に抵抗分が加わったような場合にもそのノイズ成分を除去するために、交流発振信号に同期して信号を検波する基本的な方法が提案されている（特願平11-210471）。しかし、この方式でも、検波回路内にあるスイッチのOFF期間の漏れ電流などによるノイズ電圧の影響を受ける場合があり、それを解決することが必要である。
【０００７】
【発明が解決しようとする課題】
本発明は前記のような応答性能の問題、回路構成素子の熱などの要素による計測雑音の課題を解決するためになされたものであり、センサ部、演算部および検波部の回路素子変動による雑音成分をスイッチング構成により相殺的に消去し、当該センサで得られた被測定物理量を得るセンサの測定精度向上を目的とする。
【０００８】
【課題を解決するための手段】
このような目的を達成するために、本発明は、センサを駆動する交流電源を適宜な期間で切り換えて差動演算部で増幅された演算出力信号に対して、前記演算出力信号の交流波を前記交流電源周波数に同期して１/２周期ごとに分割し、分割信号に対して積分機能により直流成分を抽出するために平均値を取得し、さらに平均後の二信号の差を求める同期検波回路を構成する。
【０００９】
すなわち同期検波回路は、演算出力の信号を、当該発振波形の一周期の間に交流信号から直流信号に変換する方法を提供するものである。これは、交流信号の振幅ピーク値、又は平均値を求めれば交流波形に変調された直流的信号値が得られる原理に基づくものである。
【００１０】
本発明では、同期検波回路は始めの１/２周期にて、片側の積分器またはローパスフィルタ（以下「LPF」と記す）に接続して、交流に変調された演算出力信号の半周期を平均化することで、信号の直流分であるところの平均値を得る。後の１/２周期にては、もう一方のLPFに演算出力信号を接続することで負極性の平均値を得る。一方、それぞれのLPFに、演算出力信号を直結しないところの残りの１ / ２周期には、演算出力信号を正負極性反転させたものを入力するようにして、 1 周期の直流分の平均値を求める。
【００１１】
一周期経過の時点で、それぞれのLPFの出力である演算出力信号の全波平均値の差を差動増幅器で求める。LPFの出力は、理想的には極性が反転した同一絶対値の演算出力であるため、差分により２倍の信号レベルが得られる。この結果、演算出力にバイアスされた直流的ノイズ電圧のみならず、それぞれのLPF自体に重畳したオフセット電圧までもが差分により相殺されることになる。このように、先に述べた温度変化等に基づく素子誤差およびオペアンプのオフセット等の回路誤差を信号処理によって除去することができ信号対雑音比の向上を図る。
【００１２】
【発明の実施の形態】
以下、図面を参照して、本発明の実施の形態について詳細に説明する。共通事項として、本発明が適用されるセンサは、例えば圧力を測定するものが考えられ、容量を検出する手段は台座基板とダイアフラムで囲まれた空間である容量室の容量キャパシタンス（Ｃ）を測定するものである。キャパシタンスのパラメータとして、容量室の空気誘電率ε、容量室に設けられた電極間ギャップｄ、感圧ダイアフラムの圧力感度変位△ｄ、および対向面積Ｓがある。
【００１３】
（１）第 1 の発明の実施例
回路の例を図1に示し、各構成部分を破線で示してある。そこでは、例えば圧力により電極間が変化する変化容量CXとそれとは対称に変化する変化容量CYの差を計測する場合である。

Figure 0004031215
Figure 0004031215
通常は単体容量を計測するだけだと、誘電率εの影響を受けてしまうので、下記式（３）のように差容量と和容量の比を計測することで、圧力感度変位△ｄを求める。
【００１４】
Figure 0004031215
以下に概略動作をしめす。そこでは、計測する期間を切り替え（それら期間を「象限」とも云う）、4つの期間で同期検波動作を行い、そこで求まった検波後の値として電位ＶＤを利用して式（３）に相当する圧力変位を求めるものである。
【００１５】
第1の計測として、SW1をa側、SW2もa側に接続し、CX、CYともに駆動電源として振幅E、周波数ｆの正弦波を入力する。
そのときのA点の電圧は下式（４）であらわせる。
Figure 0004031215
SW3は駆動電源の正弦波に同期させて、１/２周期ごとにON,OFFさせる。このときの動作状況の例として、図７に同期検波回路内のＬＰＦ手前の波形を示す。このようにB点、C点の電圧は其々ONしているときはVA1の値を計測し、OFFしているときはその値を保持するので、下式となる。
Figure 0004031215
Figure 0004031215
式（５）および（６）におけるVQbおよびVQcはスイッチのOFFにより発生するチャージインジェクション、クロックフィードスルーおよび漏れ電流により発生する誤差電圧である。
D点の電圧はB点の電圧とC点の電圧の差を増幅するので、下式となる。
Figure 0004031215
ここでNは差動増幅器の増幅率、Voffは差動増幅器のオフセット電圧である。
【００１６】
次に第2の計測として、SW1をｂ側、SW2もｂ側に接続し、CX、CYともに駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。Lは1であることが望ましいが、完全な1をつくることは難しいので、ここではあえてL倍とする。
そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００１７】
次に第3の計測として、SW1をa側、SW2をｂ側に接続し、CXには駆動電源として振幅E、周波数ｆの正弦波を入力する。CYには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式（１３）となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式（１５）となる。
Figure 0004031215
【００１８】
次に第4の計測として、SW1をb側、SW2をa側に接続し、CXには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力するCYには駆動電源として振幅E、周波数ｆの正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式（１９）となる。
Figure 0004031215
【００１９】
４つの計測の結果として最後に、式（７）、（１１）、（１５）および（１９）から計測した4つの信号に下記演算を施すと下記となる。
Figure 0004031215
結果として正確な計測ができることとなる。
【００２０】
図1に示した回路にて正確な計測ができるが、その前提条件として、各計測におけるVQbおよびVQcが共通であるとしている。共通でない場合は正確な測定ができなくなる。
その第 1 の発明にある問題を解決する発明を次に示す。
【００２１】
（２）第２の発明の実施例
回路構成は図2に示す。
第1の計測として、SW1をa側、SW2もa側に接続し、CX、CYともに駆動電源として振幅E、周波数ｆの正弦波を入力する。
そのときのA点の電圧は下式であらわせる。
Figure 0004031215
SW3は駆動電源の正弦波に同期させてON,OFFさせる。このときの動作状況の例として、図８に同期検波回路内のＬＰＦ手前の波形を示す。B点、C点の電圧は半周期は信号に接続され、もう半周期はGNDに接続されることとなるので、下式となる。ここで第 1 の発明例（図1）と異なり、ＬＰＦの入力がOFFしている状態がないので、VQb、VQcは発生しない。
Figure 0004031215
Figure 0004031215
D点の電圧はB点の電圧とC点の電圧の差を増幅するので、下式となる。
Figure 0004031215
ここでNは差動増幅器の増幅率、Voffは差動増幅器のオフセット電圧である。
【００２２】
次に第2の計測として、SW1をｂ側、SW2もｂ側に接続し、CX、CYともに駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。
そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００２３】
次に第3の計測として、SW1をa側、SW2をｂ側に接続し、CXには駆動電源として振幅E、周波数ｆの正弦波を入力する。CYには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００２４】
次に第4の計測として、SW1をb側、SW2をa側に接続し、CXには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力するCYには駆動電源として振幅E、周波数ｆの正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００２５】
最後に、式（２４）、（２８）、（３２）および（３６）から計測した4つの信号に下記演算を施すと下記となる。
Figure 0004031215
結果としてVQb、VQcにかかわらず正確な計測ができることとなる。
【００２６】
（３）第３の発明の実施例（請求項 1 に係る発明）
第２の発明の例（図2）においては各計測におけるD点の電圧（ＶＤ）が図1の場合に対して半分に減少している。これを解決する発明を図3に示し、以下に概略動作を示す。
【００２７】
第1の計測として、SW1をa側、SW2もa側に接続し、CX、CYともに駆動電源として振幅E、周波数ｆの正弦波を入力する。
そのときのA点の電圧は下式であらわせる。
Figure 0004031215
SW3は駆動電源の正弦波に同期させてON,OFFさせる。このときの動作状況の例として、図９に同期検波回路内のＬＰＦ手前の波形を示す。B点、C点の電圧は半周期は信号に接続され、もう半周期はＭ倍で反転した信号に接続されることとなるので、下式となる。ここで図1と異なりOFFしている状態がないので、VQb、VQcは発生しない。またM＝1であることが望ましいが、完全な1をつくることは困難であるので、あえてMとする。
Figure 0004031215
Figure 0004031215
D点の電圧はB点の電圧とC点の電圧の差を増幅するので、下式となる。
Figure 0004031215
【００２８】
次に第2の計測として、SW1をｂ側、SW2もｂ側に接続し、CX、CYともに駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。
そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００２９】
次に第3の計測として、SW1をa側、SW2をｂ側に接続し、CXには駆動電源として振幅E、周波数ｆの正弦波を入力する。CYには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００３０】
次に第4の計測として、SW1をb側、SW2をa側に接続し、CXには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力するCYには駆動電源として振幅E、周波数ｆの正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００３１】
最後に、式（４１）、（４５）、（４９）および（５３）から計測した4つの信号に下記演算を施すと下記となる。
Figure 0004031215
結果としてVQb、VQcにかかわらず、またD点の電圧は第２の発明の例（図２）の場合のＶＤのゲインの約2倍となり、信号対雑音比率が2倍なるより精度の良い計測ができることとなる。
【００３２】
（４）センサ構成が異なる場合の実施例
次に圧力により電極間が変化する変化容量CXと、圧力により変化しない基準容量CYを持つ構造のセンサにより物理量を計測する場合である。
Figure 0004031215
Figure 0004031215
通常は単体容量を計測するだけだと、誘電率εの影響を受けてしまうので、下記式（５７）のような差容量と単体容量の比を計測することで、圧力変位を求める。
Figure 0004031215
このような計測をする、回路例を図４に示すが、これは第 1 の発明が適用される。
前述のセンサ構成の場合と同様に４つの期間で切り換えて測定する方法を用いる。
【００３３】
第1の計測として、SW1をa側としCXの駆動電源として振幅E、周波数ｆの正弦波を入力する。SW2はc側に接続し、CYはGNDを入力する。
そのときのA点の電圧は下式であらわせる。
Figure 0004031215
SW3は駆動電源の正弦波に同期させてON,OFFさせる。電位の形状は図７と同様である。
B点、C点の電圧はONしているときはVA1の値を計測し、OFFしているときはその値を保持するので、下式となる。
Figure 0004031215
Figure 0004031215
ここでVQbおよびVQcはスイッチのOFFにより発生するチャージインジェクション、クロックフィードスルーおよび漏れ電流により発生する誤差電圧である。
D点の電圧はB点の電圧とC点の電圧の差を増幅するので、下式となる。
Figure 0004031215
【００３４】
第2の計測として、SW1をｂ側としCXの駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。SW2はc側に接続し、CYはGNDを入力する。
そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００３５】
次に第3の計測として、SW1をa側、SW2をｂ側に接続し、CXには駆動電源として振幅E、周波数ｆの正弦波を入力する。CYには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００３６】
次に第4の計測として、SW1をb側、SW2をa側に接続し、CXには駆動電源として振幅E、周波数ｆをL倍で反転させた正弦波を入力するCYには駆動電源として振幅E、周波数ｆの正弦波を入力する。そのときのA点の電圧は下式であらわせる。
Figure 0004031215
B点、C点の電圧は、下式となる。
Figure 0004031215
Figure 0004031215
D点の電圧は、下式となる。
Figure 0004031215
【００３７】
最後に、式（６１）、（６５）、（６９）および（７３）から計測した4つの信号に下記演算を施すと下記となる。
Figure 0004031215
【００３８】
図4に示すようにセンサ構成が異なる場合にも、図1のセンサ構成の場合と同様にVQb、VQcの問題点を解決するための第２の発明（図２）および第３の発明（図３）に示す同期検波回路を構成することで、図5および6に示す構成にて解決できる。それぞれ請求項２および３に対応するものである。計算式による説明は省略する。
【００３９】
以上では説明の簡素化のために、積分器相当のローパスフィルタ（ＬＰＦ）で一周期の間に信号を検波することを基本に述べてきた。
【００４０】
しかし、一次フィルタによる簡易な回路では三角関数波を整流しリップル分を除去ために、ある程度大きな時定数を持つフィルタを構成する必要があるが、これでは逆に信号の変化に対する応答性が悪くなることがある。図１０のａとｂに図３の場合のＢ点のＬＰＦの前後の電位の例を示す。そこでは象限、すなわち第１から第４までの計測期間の切り替え時に、信号が安定するまで信号周波数の数周期分の時間がかかる様子を示している。
【００４１】
（５）第４の発明の実施例（請求項 2 に係る発明）
第４の発明として、純粋なる積分器をもって構成することが望ましい場合にとる回路構成例を、図１１に示す。そこでは、まず初期値として積分器をゼロ値リセットしておき、ＳＷ５とＳＷ６は其々信号側に接続する。その状態で信号の三角関数波に同期させたタイミングで積分リセット信号を解除し、積分を開始する。図１２のａ、ｂ、およびｃにあるように、積分が開始され停止するまでの間に図１１のＥ点の電位（図12a）について、図１２のｄに示すように、全波整流された信号を積分することで、B点とC点の電位を求め、検波できる。
【００４２】
( 削除 )
【００４３】
【発明の効果】
本発明では、演算部における適当なる切り換えスイッチによる信号演算を繰り返す処理（複数期間（象限）レシオメトリック演算）の過程で発生する回路誤差および素子誤差の両方が同期検波回路によって除去され、その結果として信号対雑音比の高い測定結果が、簡潔な回路構成によって得られる。したがって、各種の物理量測定に利用できる。
【図面の簡単な説明】
【図１】 第 1 の発明の実施例およびセンサ信号処理回路における同期検波回路の位置付け（点線ブロック図（以降共通のため省略する））を表すものである。
【図２】 第２の発明の、１/２周期ごとに演算信号およびグランド（接地）を切り換える実施例を示す。
【図３】 第３の発明の、１/２周期ごとに演算信号および反対極性信号を切り換える実施例を示す。
【図４】 異なるセンサ構成の場合（変化容量と固定容量による場合）の第 1 の発明の実施例を示す。
【図５】 異なるセンサ構成の場合（変化容量と固定容量による場合）の第２の発明の実施例を示す。
【図６】 異なるセンサ構成の場合（変化容量と固定容量による場合）の第３の発明の実施例を示す。
【図７】 第 1 の発明における同期検波回路内のＢ点、Ｃ点のＬＰＦ手前電位を示す。
【図８】 第２の発明における同期検波回路内のＢ点、Ｃ点のＬＰＦ手前電位を示す。
【図９】 第３の発明における同期検波回路内のＢ点、Ｃ点のＬＰＦ手前電位を示す。
【図10】 ローパスフィルタ（ＬＰＦ）で実現した同期検波回路の信号波形を示す。
10a 第 3 の発明に係る（図3）回路におけるＢ点のＬＰＦ手前の電位
10b 第 3 の発明に係る（図3）回路におけるＢ点（ＬＰＦ後）の電位
【図11】 第４の発明の積分器にて構成した実施例を示す。
【図12】 図11の回路における検波動作を示す各点の電位波形を示す。
12a Ｅ点における電位
12b 積分リセット信号
12c 積分開始停止信号
12d Ｂ点における電位
【符号の説明】
１ センサ信号処理回路におけるセンサ部
２ センサ信号処理回路における発振部
３ センサ信号処理回路における演算部
４ センサ信号処理回路における同期検波回路
１０ 三角関数波を励起する交流電源部
１１ ゲインL倍の反転増幅器
１２ 差動増幅器（OPアンプ）
１３ ゲインＮ倍の差動増幅器
１４ ゲインM倍の反転増幅器
２１，２２ ローパスフィルタ（ＬＰＦ）
２３，２４ 積分器
Cx,Cy, センサ・キャパシタンス
Cf フィードバック演算用キャパシタンス
SW1,SW2 センサ電源印加・象限切り換え用スイッチ
SW3,SW4 同期検波用信号切り換えスイッチ
SW5,SW6 積分開始停止切り替えスイッチ[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a sensor signal processing circuit that extracts a signal from a sensor unit as a function of a physical quantity to be measured, and particularly includes means for removing a circuit error such as an element error and an operational amplifier offset based on a temperature change or the like by signal processing. The present invention relates to a sensor signal processing circuit.
[0002]
[Prior art]
In the field of pressure measurement, the replacement from mechanical to electronic is rapidly progressing. Electronic pressure gauges can be roughly classified into a resistance type that converts a stress change of a pressure-sensitive diaphragm into an electric resistance change, and a capacitance type that converts a displacement of the pressure-sensitive diaphragm into a capacitance change. Among these, the capacitance type pressure sensor has a feature that it is excellent in fine pressure measurement.
[0003]
Capacitive pressure sensors have the principle of measuring the pressure to be applied by utilizing the fact that the space surrounded by the base substrate and diaphragm constitutes capacitive reactance. Will be very small.
[0004]
For this reason, in order to change the change in reactance into an electric signal, the electric signal also becomes very small and various noises are superimposed, so it is difficult to increase the signal-to-noise ratio and increase the measurement accuracy. For this reason, conventionally, an integration circuit using capacitive reactance has been provided and the reactance is measured (Patent No. 1500174, capacitive displacement conversion method). However, the method based on the integration operation has a problem in response performance because a certain amount of time is required for charge accumulation.
[0005]
In recent years, due to advances in electronic component integration technology, a method for removing noise such as DC thermal drift has been developed by applying an AC signal to a capacitive reactance element and using a small signal arithmetic technique. . In this case, since the physical quantity to be measured must be finally demodulated into a DC signal component, it is necessary to take a method of detecting and demodulating the signal component from the AC signal, and a diode detection method is generally used. (US Patent No. 5942692 CapacitivePressureSensingMethod). However, half-wave or full-wave rectification detection methods using diodes have difficulty in accuracy due to the influence of thermal changes in diode characteristics, etc., and inductive inductance elements may be added to increase the signal-to-noise ratio. It was necessary and the circuit was complicated.
[0006]
Even in a patent application by the present applicant, in order to remove the noise component even when a resistance component is added in parallel to the sensor capacitance by performing an alternating small signal calculation, the signal is synchronized with the alternating oscillation signal. A basic method for detection has been proposed (Japanese Patent Application No. 11-210471). However, even this method may be affected by noise voltage due to leakage current during the OFF period of the switch in the detection circuit, and it is necessary to solve it.
[0007]
[Problems to be solved by the invention]
The present invention has been made to solve the above-described problem of response performance and the problem of measurement noise due to factors such as the heat of circuit components, and the noise caused by circuit element fluctuations in the sensor unit, arithmetic unit and detector unit. The purpose is to improve the measurement accuracy of a sensor that cancels out components by a switching configuration and obtains a measured physical quantity obtained by the sensor.
[0008]
[Means for Solving the Problems]
In order to achieve such an object, the present invention switches an AC power source for driving a sensor in an appropriate period and applies an AC wave of the operation output signal to an operation output signal amplified by a differential operation unit. Synchronous detection that divides every 1/2 period in synchronization with the AC power supply frequency, obtains an average value to extract a DC component from the divided signal by an integration function, and further calculates the difference between the two signals after averaging Configure the circuit.
[0009]
In other words, the synchronous detection circuit provides a method for converting a calculation output signal from an AC signal to a DC signal during one cycle of the oscillation waveform. This is based on the principle that a DC signal value modulated into an AC waveform can be obtained by obtaining an amplitude peak value or an average value of the AC signal.
[0010]
In the present invention, the synchronous detection circuit is connected to an integrator or a low-pass filter (hereinafter referred to as “LPF”) on one side in the first half cycle, and the half cycle of the arithmetic output signal modulated in alternating current is averaged. To obtain an average value which is the direct current component of the signal. In the subsequent 1/2 cycle, the negative output average value is obtained by connecting the calculation output signal to the other LPF. On the other hand, each of the LPF, the remaining 1/2 period where no directly connected operation output signal, so as to enter the ones of the operation output signal obtained by a polarity inversion, the average value of the DC component of one period Ask.
[0011]
At the time of one period elapses, determining the difference between the total wave average value of each of which is an output operation output signal of the LPF in the differential amplifier. Since the output of the LPF is ideally a calculation output of the same absolute value with the polarity reversed, a double signal level can be obtained by the difference. As a result, not only the DC noise voltage biased to the calculation output but also the offset voltage superimposed on each LPF itself is canceled by the difference. As described above, the circuit error such as the element error and the operational amplifier offset based on the temperature change described above can be removed by the signal processing, and the signal-to-noise ratio is improved .
[0012]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. As a common matter, the sensor to which the present invention is applied may be, for example, a sensor that measures pressure, and the means for detecting the capacitance measures the capacitance capacitance (C) of the capacitance chamber, which is a space surrounded by the base substrate and the diaphragm. To do. Capacitance parameters include air permittivity ε of the capacity chamber, gap d between the electrodes provided in the capacity chamber, pressure sensitivity displacement Δd of the pressure sensitive diaphragm, and facing area S.
[0013]
(1) Examples of the embodiment circuit of the first invention shown in FIG. 1, there is shown a respective component by a broken line. In this case, for example, the difference between the change capacity CX in which the distance between the electrodes changes due to pressure and the change capacity CY in which it changes symmetrically is measured.
Figure 0004031215
Figure 0004031215
Normally, if only a single capacitance is measured, it will be affected by the dielectric constant ε, so the pressure sensitivity displacement Δd is obtained by measuring the ratio between the differential capacitance and the total capacitance as shown in the following equation (3). .
[0014]
Figure 0004031215
The general operation is as follows. In this case, the measurement periods are switched (the periods are also referred to as “quadrants”), the synchronous detection operation is performed in four periods, and the potential VD is used as the post-detection value found there, which corresponds to Equation (3). The pressure displacement is obtained.
[0015]
As a first measurement, SW1 is connected to the a side and SW2 is also connected to the a side, and a sine wave having an amplitude E and a frequency f is input as a driving power source for both CX and CY.
The voltage at point A at that time is expressed by the following equation (4).
Figure 0004031215
SW3 is synchronized with the sine wave of the drive power supply and turned on and off every 1/2 cycle. As an example of the operation situation at this time, FIG. 7 shows a waveform before the LPF in the synchronous detection circuit. Thus, when the voltages at points B and C are ON, the value of VA1 is measured, and when the voltage is OFF, the value is held.
Figure 0004031215
Figure 0004031215
In equations (5) and (6), VQb and VQc are error voltages generated by charge injection, clock feedthrough and leakage current generated by turning off the switch.
Since the voltage at point D amplifies the difference between the voltage at point B and the voltage at point C, the following equation is obtained.
Figure 0004031215
Here, N is the amplification factor of the differential amplifier, and Voff is the offset voltage of the differential amplifier.
[0016]
Next, as a second measurement, SW1 is connected to the b side and SW2 is also connected to the b side, and a sine wave obtained by inverting the amplitude E and the frequency f by L times is input to both CX and CY as drive power. L is preferably 1, but it is difficult to make a perfect one.
The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0017]
Next, as a third measurement, SW1 is connected to the a side and SW2 is connected to the b side, and a sine wave having an amplitude E and a frequency f is input to CX as a drive power source. A sine wave obtained by inverting the amplitude E and the frequency f by L times is input to CY as a drive power source. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltages at points B and C are expressed by the following equation (13).
Figure 0004031215
Figure 0004031215
The voltage at point D is expressed by the following equation (15).
Figure 0004031215
[0018]
Next, as the fourth measurement, SW1 is connected to the b side and SW2 is connected to the a side, and the drive power supply is input to CX as a drive power supply to CY. A sine wave of amplitude E and frequency f is input. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is given by the following equation (19).
Figure 0004031215
[0019]
As a result of the four measurements, the following calculation is performed on the four signals measured from the equations (7), (11), (15), and (19).
Figure 0004031215
As a result, accurate measurement can be performed.
[0020]
Although accurate measurement can be performed with the circuit shown in FIG. 1, it is assumed that VQb and VQc in each measurement are common as a prerequisite. If it is not common, accurate measurement will not be possible.
Following the invention to solve the problems in the first invention.
[0021]
(2) The circuit configuration of the second embodiment is shown in FIG.
As a first measurement, SW1 is connected to the a side and SW2 is also connected to the a side, and a sine wave having an amplitude E and a frequency f is input as a driving power source for both CX and CY.
The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
SW3 is turned on and off in synchronization with the sine wave of the drive power supply. As an example of the operation situation at this time, FIG. 8 shows a waveform before the LPF in the synchronous detection circuit. The voltage at point B and point C is connected to the signal for half the period and is connected to GND for the other half period. Unlike wherein the first invention example (FIG. 1), the input of the LPF is no state in which the OFF, VQB, Vqc does not occur.
Figure 0004031215
Figure 0004031215
Since the voltage at point D amplifies the difference between the voltage at point B and the voltage at point C, the following equation is obtained.
Figure 0004031215
Here, N is the amplification factor of the differential amplifier, and Voff is the offset voltage of the differential amplifier.
[0022]
Next, as a second measurement, SW1 is connected to the b side and SW2 is also connected to the b side, and a sine wave obtained by inverting the amplitude E and the frequency f by L times is input to both CX and CY as drive power.
The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0023]
Next, as a third measurement, SW1 is connected to the a side and SW2 is connected to the b side, and a sine wave having an amplitude E and a frequency f is input to CX as a drive power source. A sine wave obtained by inverting the amplitude E and the frequency f by L times is input to CY as a drive power source. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0024]
Next, as the fourth measurement, SW1 is connected to the b side and SW2 is connected to the a side, and the drive power supply is input to CX as a drive power supply to CY. A sine wave of amplitude E and frequency f is input. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0025]
Finally, the following calculation is performed on the four signals measured from the equations (24), (28), (32), and (36).
Figure 0004031215
As a result, accurate measurement is possible regardless of VQb and VQc.
[0026]
(3) Embodiment of the third invention ( Invention according to claim 1 )
In the example of the second invention (FIG. 2), the voltage (VD) at point D in each measurement is reduced by half compared to the case of FIG. The invention for solving this is shown in FIG.
[0027]
As a first measurement, SW1 is connected to the a side and SW2 is also connected to the a side, and a sine wave having an amplitude E and a frequency f is input as a driving power source for both CX and CY.
The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
SW3 is turned on and off in synchronization with the sine wave of the drive power supply. As an example of the operation situation at this time, FIG. 9 shows a waveform before the LPF in the synchronous detection circuit. The voltage at point B and point C is connected to the signal for half a period, and the other half period is connected to a signal inverted by a factor of M, so the following equation is obtained. Here, unlike FIG. 1, since there is no OFF state, VQb and VQc are not generated. Although it is desirable that M = 1, it is difficult to make a perfect 1, so M is dared.
Figure 0004031215
Figure 0004031215
Since the voltage at point D amplifies the difference between the voltage at point B and the voltage at point C, the following equation is obtained.
Figure 0004031215
[0028]
Next, as a second measurement, SW1 is connected to the b side and SW2 is also connected to the b side, and a sine wave obtained by inverting the amplitude E and the frequency f by L times is input to both CX and CY as drive power.
The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0029]
Next, as a third measurement, SW1 is connected to the a side and SW2 is connected to the b side, and a sine wave having an amplitude E and a frequency f is input to CX as a drive power source. A sine wave obtained by inverting the amplitude E and the frequency f by L times is input to CY as a drive power source. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0030]
Next, as the fourth measurement, SW1 is connected to the b side and SW2 is connected to the a side, and the drive power supply is input to CX as a drive power supply to CY. A sine wave of amplitude E and frequency f is input. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0031]
Finally, the following calculation is performed on the four signals measured from the equations (41), (45), (49), and (53).
Figure 0004031215
As a result VQB, regardless Vqc, and the voltage of the point D becomes about twice the gain of VD in the case of the example of the second invention (Fig. 2), good measurement accuracy than the signal-to-noise ratio is 2 times Will be able to.
[0032]
(4) Example in which sensor configuration is different Next, a physical quantity is measured by a sensor having a structure having a change capacity CX in which the distance between electrodes changes due to pressure and a reference capacity CY that does not change in accordance with pressure.
Figure 0004031215
Figure 0004031215
Normally, if only the single capacity is measured, it is affected by the dielectric constant ε. Therefore, the pressure displacement is obtained by measuring the ratio between the differential capacity and the single capacity as shown in the following equation (57).
Figure 0004031215
To such measurement, it shows a circuit example in FIG. 4, which is applied first aspect of the present invention.
As in the case of the sensor configuration described above, a method of switching and measuring in four periods is used.
[0033]
As a first measurement, SW1 is set to the a side, and a sine wave having an amplitude E and a frequency f is input as a driving power source for CX. Connect SW2 to the c side and input GND to CY.
The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
SW3 is turned on and off in synchronization with the sine wave of the drive power supply. The shape of the potential is the same as in FIG.
When the voltage at point B and point C is ON, the value of VA1 is measured, and when it is OFF, the value is held.
Figure 0004031215
Figure 0004031215
Here, VQb and VQc are error voltages generated by charge injection, clock feedthrough and leakage current generated when the switch is turned OFF.
Since the voltage at point D amplifies the difference between the voltage at point B and the voltage at point C, the following equation is obtained.
Figure 0004031215
[0034]
As a second measurement, a sine wave having SW1 as the b side and inverting the amplitude E and the frequency f by L times is input as a driving power source for CX. Connect SW2 to the c side and input GND to CY.
The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0035]
Next, as a third measurement, SW1 is connected to the a side and SW2 is connected to the b side, and a sine wave having an amplitude E and a frequency f is input to CX as a drive power source. A sine wave obtained by inverting the amplitude E and the frequency f by L times is input to CY as a drive power source. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0036]
Next, as the fourth measurement, SW1 is connected to the b side and SW2 is connected to the a side, and the drive power supply is input to CX as a drive power supply to CY. A sine wave of amplitude E and frequency f is input. The voltage at point A at that time is expressed by the following equation.
Figure 0004031215
The voltage at point B and point C is as follows.
Figure 0004031215
Figure 0004031215
The voltage at point D is as follows.
Figure 0004031215
[0037]
Finally, the following calculation is performed on the four signals measured from the equations (61), (65), (69), and (73).
Figure 0004031215
[0038]
As shown in FIG. 4, when the sensor configuration is different, the second invention (FIG. 2) and the third invention (FIG. 2) for solving the problems of VQb and VQc as in the case of the sensor configuration of FIG. The configuration shown in FIGS. 5 and 6 can be solved by configuring the synchronous detection circuit shown in 3). These correspond to claims 2 and 3, respectively. The explanation by the calculation formula is omitted.
[0039]
For the sake of simplicity, the above description has basically been based on detecting a signal during one cycle with a low pass filter (LPF) corresponding to an integrator.
[0040]
However, in a simple circuit using a first-order filter, it is necessary to construct a filter having a somewhat large time constant in order to rectify the trigonometric function wave and remove the ripple component. However, this makes the response to changes in the signal worse. Sometimes. FIGS. 10a and 10b show examples of potentials before and after the LPF at point B in the case of FIG. There, it is shown that it takes time corresponding to several cycles of the signal frequency until the signal becomes stable when switching between quadrants, that is, the first to fourth measurement periods.
[0041]
(5) Embodiment of the fourth invention ( invention according to claim 2 )
As a fourth invention, FIG. 11 shows a circuit configuration example taken when it is desirable to configure with a pure integrator. First, the integrator is reset to zero as an initial value, and SW5 and SW6 are respectively connected to the signal side. In this state, the integration reset signal is canceled at the timing synchronized with the trigonometric wave of the signal, and integration is started. As shown in FIGS. 12a, 12b, and 12c, the potential at point E in FIG. 11 (FIG. 12a) is full-wave rectified as shown in FIG. By integrating the obtained signals, the potentials at points B and C can be obtained and detected.
[0042]
( Delete )
[0043]
【The invention's effect】
In the present invention, both the circuit error and the element error generated in the process of repeating the signal calculation by the appropriate changeover switch in the calculation unit (multiple period (quadrant) ratiometric calculation) are removed by the synchronous detection circuit, and as a result A measurement result with a high signal-to-noise ratio can be obtained with a simple circuit configuration. Therefore, it can be used for various physical quantity measurements.
[Brief description of the drawings]
1 is intended to represent the positioning (dotted line block diagram (omitted common for later)) of the synchronous detection circuit in the examples and the sensor signal processing circuit of the first invention.
FIG. 2 shows an embodiment in which the calculation signal and the ground (ground) are switched every 1/2 cycle of the second invention.
FIG. 3 shows an embodiment in which a calculation signal and an opposite polarity signal are switched every 1/2 cycle according to the third invention.
Figure 4 shows an embodiment of the first invention in the case of different sensor configurations (if due to changes capacitance and a fixed capacitance).
FIG. 5 shows an embodiment of the second invention in the case of different sensor configurations (in the case of changing capacitance and fixed capacitance).
FIG. 6 shows an embodiment of the third invention in the case of different sensor configurations (in the case of changing capacitance and fixed capacitance).
[7] B points in the synchronous detection circuit in the first aspect of the present invention, showing the LPF before the potential at point C.
FIG. 8 shows potentials before point LPF at points B and C in the synchronous detection circuit according to the second invention.
FIG. 9 shows potentials before the LPF at points B and C in the synchronous detection circuit according to the third invention.
FIG. 10 shows a signal waveform of a synchronous detection circuit realized by a low-pass filter (LPF).
10a according to the third aspect of the invention (FIG. 3) LPF before the potential at point B in the circuit
10b according to the third aspect of the invention (FIG. 3) shows an embodiment constructed by the integrator potential [11] A fourth aspect of the point B in the circuit (after LPF).
12 shows a potential waveform at each point showing a detection operation in the circuit of FIG.
12a Potential at point E
12b Integration reset signal
12c Integration start / stop signal
12d Potential at point B [Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 Sensor part in sensor signal processing circuit 2 Oscillation part in sensor signal processing circuit 3 Arithmetic part in sensor signal processing circuit 4 Synchronous detection circuit 10 in sensor signal processing circuit AC power supply part 11 for exciting trigonometric function wave Inverting amplifier of gain L times 12 Differential amplifier (OP amplifier)
13 Differential amplifier with N times gain 14 Inverting amplifiers 21 and 22 with M times gain Low pass filter (LPF)
23, 24 integrator
Cx, Cy, sensor capacitance
Cf Capacitance for feedback calculation
SW1, SW2 Sensor power application / quadrant switch
SW3, SW4 Signal selector switch for synchronous detection
SW5, SW6 Integration start / stop switch

Claims (2)

第１および第２の静電容量型センサ素子を有するセンサ部と、A sensor unit having first and second capacitive sensor elements;
極性の異なる第１および第２の電源を有する電源部と、A power supply unit having first and second power supplies having different polarities;
前記電源部の出力側と前記センサ部の入力側との間に接続されかつ前記第１および第２の電源と前記第１および第２のセンサ素子との接続を４通りの組合せで切り換えるスイッチング部と、A switching unit that is connected between the output side of the power supply unit and the input side of the sensor unit and switches the connection between the first and second power sources and the first and second sensor elements in four combinations. When,
前記センサ部の出力側に接続されかつ前記第１の電源が前記第１および第２の両センサ素子に接続されたときに得られる第１の信号、前記第２の電源がそれぞれ前記第１および第２の両センサ素子に接続されたときに得られる第２の信号、前記第１および第２の電源がそれぞれ前記第１および第２のセンサ素子に接続されたときに得られる第３の信号、および前記第１および第２の電源がそれぞれ前記第２および第１のセンサ素子に接続されたときに得られる第４の信号が入力される演算部とを備え、A first signal obtained when the first power source is connected to the output side of the sensor unit and the first power source is connected to the first and second sensor elements, and the second power source is the first and second power sources, respectively. A second signal obtained when connected to both second sensor elements, and a third signal obtained when the first and second power sources are connected to the first and second sensor elements, respectively. And a calculation unit to which a fourth signal obtained when the first and second power sources are connected to the second and first sensor elements, respectively,
前記演算部は、前記スイッチング部により指定された所定期間について、前記第１乃至第４の信号のうち一つの信号を取り込んで、当該信号を前記電源の発振信号の周期と同期してその１／２周期ごとに二つの積分器のうちの一つに接続し、当該信号の正負極性を反転させた信号を他方の積分器に接続して、少なくとも１周期の間は極性反転した信号をそれぞれ二つの積分器に接続する同期検波部と、The arithmetic unit captures one of the first to fourth signals for a predetermined period specified by the switching unit, and synchronizes the signal with the period of the oscillation signal of the power source. Connected to one of the two integrators every two cycles, and connected to the other integrator a signal obtained by inverting the positive / negative polarity of the signal. A synchronous detector connected to two integrators;
前記二つの積分器出力間の差分を求める差動増幅器とを有することと、Having a differential amplifier for determining a difference between the two integrator outputs;
および前記第１乃至前記第４の信号に基づいて求めた前記差動増幅器の出力の四値を用いて所定の数式により前記センサ素子の容量変化を求める手段を備えることとを特徴とするセンサ信号処理回路。And a sensor signal comprising means for obtaining a change in capacitance of the sensor element by a predetermined mathematical expression using four values of the output of the differential amplifier obtained based on the first to fourth signals. Processing circuit. 前記演算部が前記二つの積分器に対して、積分リセット、積分開始および積分停止を行う回路を備えることを特徴とする請求項１に記載のセンサ信号処理回路。The sensor signal processing circuit according to claim 1, wherein the arithmetic unit includes a circuit that performs integration reset, integration start, and integration stop with respect to the two integrators.
JP2001184428A 2001-06-19 2001-06-19 Sensor signal processing circuit Expired - Fee Related JP4031215B2 (en)

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