JP3858754B2 - Motor drive device for laundry equipment - Google Patents

Description

【００３１】
記憶手段６２には、ｓｉｎθとｃｏｓθのデータを記憶しているので、電気角データに対応したデータを呼び出して積和演算を行うことにより、ｄ軸電流Ｉｄとｑ軸電流Ｉｑに分解できる。電気角θの検知とモータ電流瞬時値の検出はキャリヤ信号に同期して行うもので、後述するフローチャートに従い、詳細な説明を行う。
【００３２】
回転数検知手段６６は、ロータ位置検出手段４ａの出力基準信号Ｈ１よりモータ回転数を検知し、回転数信号を設定変更手段６５、回転数制御手段６７に加える。設定変更手段６５は、モータ４の起動制御と回転数の設定、回転数優先制御あるいは電流優先制御の切り換え、および回転数とｑ軸電流設定値Ｉｑｓに応じたｄ軸電流Ｉｄｓの演算を行い、回転数制御手段６７に回転数設定信号Ｎｓ、あるいは、減速制動時には電流優先制御信号Ｉｑｏを加え、モータ電流制御手段６８にｄ軸設定信号Ｉｄｓを加える。
【００３３】
回転数制御手段６７は、検知回転数ｎと設定回転数Ｎｓを比較する回転数比較手段６７ａと、検知回転数ｎと設定回転数Ｎｓとの誤差信号Δｎと、回転数の変化率（加速度）に応じてｑ軸電流設定値Ｉｑｓを制御するトルク電流設定手段６７ｂと、ｑ軸電流設定値Ｉｑｓをトルク電流設定手段６７ｂからそのまま出力するか、あるいは、設定変更手段６５からの指令により電流優先制御にするかを切り換えるｑ軸電流切り換え手段６７ｃより構成される。
【００３４】
回転数制御の場合には、誤差信号Δｎに応じてｑ軸電流設定値Ｉｑｓを制御する、いわゆる、回転数制御電流マイナーループ制御を行い、減速制動時には電流制御とし、設定変更手段６５からｑ軸電流設定値Ｉｑｏが設定され、Ｉｑｓ＝Ｉｑｏとなる。
【００３５】
モータ電流制御手段６８は、３相／２相ｄｑ変換手段６１の出力信号Ｉｑ、Ｉｄと設定信号Ｉｑｓ、Ｉｄｓをそれぞれ比較して制御電圧信号Ｖｑ、Ｖｄを出力するもので、ｑ軸電流比較手段６８ａ、ｑ軸電圧設定手段６８ｂ、ｄ軸電流比較手段６８ｃ、ｄ軸電圧設定手段６８ｄより構成し、ｑ軸電流とｄ軸電流をそれぞれ制御する電圧信号Ｖｑ、Ｖｄを生成する。
【００３６】
ｄ軸電流設定値Ｉｄｓは、設定変更手段６５からモータ電流制御手段６８に信号が加えられるもので、埋め込み磁石モータの場合には回転数に応じてｄ軸電流設定値Ｉｄｓを増加させて弱め界磁制御を行う。表面磁石モータの場合には、通常、Ｉｄｓは零に設定し、高回転数駆動の場合にＩｄｓを増加させる。
【００３７】
２相／３相ｄｑ逆変換手段６３は、電圧信号Ｖｑ、Ｖｄより３相モータ駆動制御電圧ｖｕ、ｖｖ、ｖｗを（数２）より演算するもので、キャリヤ信号に同期して、電気角検知手段６０により検知した電気角θに対応した正弦波状の信号をＰＷＭ制御手段に加える。記憶手段６２に記憶したｓｉｎθ、ｃｏｓθの積和演算の方法は、３相／２相ｄｑ変換手段６１の演算とほぼ同じである。
【００３８】
【数２】

【００３９】
図７は、回転駆動時と制動時のモータ電流をｑ軸電流Ｉｑとｄ軸電流に分解したベクトル図で、Ｉ１は正転、あるいは反転駆動時の電流ベクトルで、Ｉｑ１、−Ｉｄ１はそのときのｑ軸電流Ｉｑとｄ軸電流であり、Ｉ２は減速制動時の電流ベクトルで、ｑ軸電流、ｄ軸電流をそれぞれ負に設定し（−Ｉｑ２、−Ｉｄ２）、Ｉ３は制動停止直前の電流ベクトルで、ｑ軸電流を負、ｄ軸電流を正に設定（−Ｉｑ２、Ｉｄ３）している。
【００４０】
モータ回転数が高い場合、減速制動時のｄ軸電流をそれぞれ負に設定して弱め界磁制御することにより、電力回生を抑えることができ、モータ逆起電力はモータ抵抗で消費される。モータ回転数が低下するとモータ逆起電力は減少してモータ巻線抵抗でエネルギーが消費されて回生しなくなり、ｄ軸電流を正にする強め界磁制御を行うことによりモータ制動トルクが増加して停止し易くなる。
【００４１】
ｑ軸電流Ｉｑとｄ軸電流に分解するベクトル制御の場合には、ｑ軸電流Ｉｑとｄ軸電流をそれぞれ設定する必要があるが、減速制動時のみモータ印加電圧ベクトルを制御して電流制御しても構わない。また、洗濯兼脱水槽内に水がない状態で撹拌する場合、例えば、洗濯乾燥機の乾燥行程における乾燥撹拌行程においては、撹拌翼が急速に停止して、回生電力が急速低下するので、モータ印加電圧ベクトルを２４０度進角させるだけでよい。
【００４２】
上記構成において図８、図９、図１０、図１１、図１２、図１３および図１４を参照しながら動作を説明する。図８は撹拌行程のフローチャートで、洗濯機の洗い行程、あるいは、縦型洗濯乾燥機の洗い行程と、乾燥行程における撹拌翼による衣類の撹拌に使用される。
【００４３】
図９は撹拌行程のタイムチャートで、時間経過により回転数がどのように変化するかを示し、正転、反転のモータ駆動時間をｔｏｎ、モータ駆動後の減速制動期間をｔｂ、モータ駆動停止期間をｔｏｆｆとしている。
【００４４】
図８において、ステップ１００より撹拌行程を開始し、ステップ１０１にて洗い撹拌行程の各種初期設定を行い、つぎにステップ１０２に進んで正反転制御設定を行い、前回のモータ駆動が正転ならば次回は反転とし、前回のモータ駆動が反転ならば次回は正反転と設定する。つぎに、ステップ１０３に進んで、図１１に示すモータ駆動サブルーチンを実行し、所定回転数となるようにモータ電流をベクトル制御する。つぎに、ステップ１０４に進んでモータ駆動時間が設定値ｔｏｎ以上かどうか判定し、設定値ｔｏｎ以上ならばステップ１０５に進み、ｔｏｎに達しなければステップ１０３に戻る。
【００４５】
ステップ１０５にて減速制動設定を行い、ステップ１０６に進んでｑ軸電流設定値、ｄ軸電流設定値をそれぞれ負に設定する。つぎに、ステップ１０７のモータ駆動サブルーチンを実行する。このとき、モータ駆動サブルーチンはステップ１０５により減速制動に設定されているので、負のトルク電流制御となる。
【００４６】
減速制御するために、電流制御ではなく、回転数制御にして設定回転数を零に設定しても減速制動は可能であるが、そのときには、ｑ軸電流、ｄ軸電流設定値に上下限値を設けて制御する必要がある。すなわち、減速制動時には、ｑ軸電流を零から負の最大値−Ｉｑｍａｘに設定し、ｄ軸電流は、負の最大値−Ｉｄｍａｘから正の最大値＋Ｉｄｍａｘに設定し、正の最大値＋Ｉｄｍａｘを小さくしないと電力回生作用によりインバータ直流電圧が異常上昇する。また、制動トルクが大きくなって、負荷が軽い場合には減速機構と撹拌翼の遊びにより、カタカタ音が発生する。よって、負荷量に応じて制動トルク最大値を制御する必要があり、実質電流制御と同じとなる。
【００４７】
つぎに、ステップ１０８に進んでモータ回転数ｎがほぼ零に達したかどうか判定し、ほぼ零ならばステップ１０９に進んで所定時間モータを停止させる。つぎに、ステップ１１０に進んで撹拌行程が終了したかどうか判定し、終了ならばステップ１１１に進んでサブルーチンをリターンし、終了でなければステップ１０２に進んで撹拌行程をつづける。
【００４８】
図１０は、減速制動時のｑ軸電流設定値（−Ｉｑｓ）と正反転駆動時のｑ軸電流の制御特性を示すもので、撹拌翼を設定回転数で回転数制御すると、負荷量が多いほどモータ駆動時のｑ軸電流が増加するので、ｑ軸電流により負荷量が判定できるので、負荷量に応じて減速制動時の負のトルク電流設定値（−Ｉｑｓ）を制御するものである。負荷量が大きい程ブレーキ電流を増加させて制動時間を短縮でき、負荷量が少なければ、ブレーキ電流を減らして撹拌翼の急速停止により発生する減速機構と撹拌翼の遊びによるカタカタ音を減らすことができる。
【００４９】
負荷量判定の他の方法として、撹拌駆動停止後の惰性回転数、あるいは、ロータ位置検出手段の位置検出パルス数により判別する方法が考えられる。また、撹拌駆動時のトルク電流を一定にして回転数上昇からも判別できる。また、撹拌駆動時のｑ軸電流と、撹拌駆動停止後の惰性回転数の２つの信号により負荷量検知すると検知精度がさらに向上する。惰性回転数検知時、減速制動させると負荷量検知精度が悪くなるので、負荷量検知においては減速制動を設けないようにする。
【００５０】
つぎに、図１１に示すモータ駆動サブルーチンのフローチャートについて説明する。ステップ２００よりモータ駆動サブルーチンが開始する。ステップ２０１はサブルーチン実行の最初に判断する初期判定で、起動あるいは制動初期を判定し、起動あるいは制動初期であればステップ２０２に進み、各種初期設定を行い、メインルーチンからのパラメータの受け渡しと各種設定を実行する。
【００５１】
つぎに、ステップ２０３に進んで回転起動制御あるいは制動初期制御を行う。ステップ２０２、２０３は最初に一回だけ実行する。起動制御は、回転数フィードバック制御ができない起動時に、所定のモータ印加電圧に設定して１２０度通電するものであり、低いモータ印加電圧から高い電圧まで時間経過とともに電圧を上昇させるソフトスタートを行う。制動運転の場合には負のｄ軸電流を増やして、負のｑ軸電流を減らし、高回転数での電力回生を防止し、急激なブレーキトルクが加わらないようなソフトブレーキを行う。
【００５２】
つぎに、ステップ２０４に進んでキャリヤ信号割込の有無を判定する。キャリヤ信号割込とは、ＰＷＭ制御手段６５のキャリヤカウンタがオーバーフローすると発生する割込信号ｃｋにより実行するもので、ステップ２０５に進んでキャリヤ信号割込サブルーチンを実行する。
【００５３】
図１２は、キャリヤ信号割込サブルーチンの詳細フローチャートを示し、ステップ３００よりキャリヤ信号割込サブルーチンを開始し、ステップ３０１にて割込信号ｃｋをカウントする。つぎに、ステップ３０２に進んでロータ位置電気角θを演算する。ロータ位置信号θは、別途求めたキャリヤ信号１周期当たりの電気角Δθとキャリヤカウンタのカウント値ｋを掛けた値ｋ・Δθを、ロータ位置検出手段４ａより検知できる６０度ごとの電気角φを加えることで推定演算する。
【００５４】
モータ４を８極、キャリヤ周波数を１５．６ｋＨｚ、回転数を９００ｒ／ｍｉｎとするとモータ駆動周波数は６０Ｈｚとなり、電気角６０度内のキャリヤカウンタカウント値ｋは約４３となる。よって、Δθは約１．４度となる。モータ回転数が低い程、電気角６０度内のカウント値ｋは高くなり、演算上の電気角検知分解能は向上するので、回転数が低く精度が要求される場合でも問題はないことがわかる。
【００５５】
つぎに、ステップ３０３に進んでモータ電流Ｉｕ、Ｉｖを検出する。電流検出１回ではノイズが含まれる可能性があるので、ステップ３０４に進んで再度検出し、ステップ３０５にて平均値を求めてノイズを除去し、Ｉｗ＝−（Ｉｕ＋Ｉｖ）よりモータ電流Ｉｗを演算する。
【００５６】
つぎに、ステップ３０６に進んで電気角θとモータ電流より、（数１）に示した演算を行い、３相／２相ｄｑ変換を行い、ｄ軸電流Ｉｄ、ｑ軸電流Ｉｑを求める。つぎにステップ３０７に進んでＩｄ、Ｉｑをメモリし、別途回転数制御データとして用いる。
【００５７】
つぎに、ステップ３０８に進んでｄ軸制御電圧Ｖｄ、ｑ軸制御電圧Ｖｑを呼び出し、ステップ３０９に進んで（数２）に従い２相／３相ｄｑ逆変換を行い、３相制御電圧ｖｕ、ｖｖ、ｖｗを求める。この逆変換は、ステップ３０６と同じように記憶手段６２の電気角に対応したｓｉｎθ、ｃｏｓθデータを用い、積和演算を高速で行う。つぎに、ステップ３１０に進んで３相制御電圧ｖｕ、ｖｖ、ｖｗに対応したＰＷＭ制御を行い、ステップ３１１に進んでサブルーチンをリターンする。
【００５８】
ＰＷＭ制御は、図２でも説明したように、Ｕ相、Ｖ相、Ｗ相各相に対応して、鋸歯状波（または三角波）のキャリヤ信号と制御電圧ｖｕ、ｖｖ、ｖｗを比較してインバータ回路３のＩＧＢＴオンオフ制御信号を発生させ、モータ４を正弦波駆動するもので、上アームトランジスタと下アームトランジスタの信号は逆転された波形で、上アームトランジスタの導通比を増加すると出力電圧は正電圧が増加し、下アームトランジスタの導通比を増加させると出力電圧は負電圧が増加する。導通比を５０％にすると出力電圧は零となる。
【００５９】
電気角θに対応して制御電圧を正弦波状に変化させると正弦波状の電流が流れる。正弦波駆動の場合、トランジスタの導通比を最大値１００％にしたとき、出力電圧は最大となり変調度Ａｍは１００％で、導通比の最大値を５０％にしたとき、出力電圧は最低となり変調度Ａｍは０％と呼ぶ。
【００６０】
モータ電流をベクトル制御するための、３相／２相ｄｑ変換と２相／３相ｄｑ逆変換をキャリヤ毎に高速で実行するので、高速の電流制御が可能となり、さらに、キャリヤ毎にトルク電流Ｉｑを検出するので負荷量が瞬時に判定できる特徴がある。また、キャリヤ毎にベクトル制御することにより常に一定トルクで駆動でき、減速制動時に負の定トルク制御をすれば、ブレーキトルクを一定にして過電流を防止できる。
【００６１】
図１１に戻って、キャリヤ信号割込サブルーチンを実行した後、ステップ２０６に進み、位置信号割込の有無を判定する。位置信号Ｈ１、Ｈ２、Ｈ３のいずれかの信号が変化すると割込信号が発生し、ステップ２０７に進んで、図１３に示す位置信号割込サブルーチンを実行する。図２に示すように、電気角６０度ごとに割込信号が発生する。
【００６２】
ここで、位置信号割込サブルーチンについて説明する。ステップ４００より位置信号割込サブルーチンを開始し、ステップ４０１に進んで位置信号Ｈ１、Ｈ２、Ｈ３を入力し位置検出を行い、つぎに、ステップ４０２に進んで位置信号よりロータ電気角θｃを検出する。つぎに、ステップ４０３に進んでキャリヤ信号割込サブルーチンでカウントしているカウント値ｋをｋｃにメモリし、ステップ４０４に進んでカウント値ｋをクリヤし、ステップ４０５に進み、電気角６０度間のキャリヤカウンタカウント値ｋｃより１キャリヤの電気角Δθを演算する。
【００６３】
つぎに、ステップ４０６に進んで基準位置信号Ｈ１による割込信号かどうかを判定し、基準位置信号割込ならばステップ４０７に進んで回転周期測定タイマーＴのカウント値Ｔを周期Ｔｏとしてメモリーし、ステップ４０８に進んでタイマーＴをクリヤし、ステップ４０９に進んでモータ回転数ｎを演算する。つぎに、ステップ４１０に進んで回転周期測定タイマーのカウントを開始させ、ステップ４１１に進んでサブルーチンをリターンする。
【００６４】
回転周期測定タイマーの検知分解能を８ｂｉｔ精度にすると、クロックは６４μｓとなり、キャリヤ信号をクロックに使用できるが、回転制御性能を向上するためには回転周期検知分解能を向上させる必要があり、クロックの周期は１〜１０μｓに設定する必要がある。この場合には、マイクロコンピュータのシステムクロックを分周してクロックに使用する。
【００６５】
以上に説明した回転数検知方法は、位置信号Ｈ１の周期から求める方法を示したが、位置信号Ｈ１、Ｈ２、Ｈ３をすべて使用してもよい。また、キャリヤ信号を三角波にすると、キャリヤカウンタタイマーの周期は鋸歯状波の２倍となるので、三角波のオーバーフロー信号をクロックにすると分解能が向上するので三角波タイマーのオーバーフロー信号をクロックにしてもよい。
【００６６】
つぎに、図１１において、位置信号割込サブルーチン２０７を実行した後、ステップ２０８に進み回転数制御フラグの有無を判定し、回転数制御設定ならば、ステップ２０９に進んで回転数制御サブルーチンを実行し、減速制動設定ならば、ステップ２１０に進んで、回転数制御サブルーチンの中に含まれる減速制動サブルーチンを実行する。回転数制御サブルーチンの詳細は図１４に示す。
【００６７】
図１４において、ステップ５００より回転数制御サブルーチンを開始し、ステップ５０１にてモータ回転数ｎを呼び出し、ステップ５０２に進んで通常駆動か減速制動かのフラグ判定する。
【００６８】
通常駆動ならばステップ５０３に進み設定回転数と検知回転数の誤差によりｑ軸電流設定値Ｉｑｓを制御してトルク制御を行い、つぎに、ステップ５０４に進んでｑ軸電流設定値Ｉｑｓを上限値Ｉｑｍａｘと比較し、大ならばステップ５０５に進んでＩｑｓをＩｑｍａｘとして、ｑ軸電流Ｉｑが上限値Ｉｑｍａｘ以上とならないようにする。
【００６９】
減速制動ならばステップ５０６に進んで負のトルク制御、すなわち、ブレーキトルク制御のためにｑ軸電流設定値を−Ｉｑｓに設定する。つぎに、ステップ５０７に進んで回転数に応じてｄ軸電流設定値Ｉｄｓを制御する。回転数が高い場合には、電力回生しないようにｄ軸電流設定値Ｉｄｓを負にして弱め界磁とし、回転数が低下するとＩｄｓを正にして強め界磁にする。
【００７０】
つぎに、ステップ５０８にてｄ軸電流Ｉｄを呼び出し、ステップ５０９にてＩｄとＩｄｓの大小比較判定を行い、ｄ軸電流Ｉｄが設定値Ｉｄｓよりも大きければステップ５１０に進んでｄ軸制御電圧Ｖｄを減らし、ｄ軸電流Ｉｄが設定値Ｉｄｓよりも小さければステップ５１１に進んでｄ軸制御電圧Ｖｄを増やす。
【００７１】
つぎに、ステップ５１２に進んで３相／２相ｄｑ変換手段６１より求めたｑ軸電流Ｉｑを呼び出し、ステップ５１３にてＩｑとＩｑｓの大小比較判定を行い、ｑ軸電流Ｉｑが設定値Ｉｑｓよりも大きければステップ５１４に進んでｑ軸制御圧Ｖｑを減らし、ｑ軸電流Ｉｑが設定値Ｉｑｓよりも小さければステップ５１５に進んでｑ軸制御電圧Ｖｑを増やす。
【００７２】
つぎに、ステップ５１６に進んで演算されたｄ軸制御電圧Ｖｄ、ｑ軸制御電圧Ｖｑをそれぞれメモリし、ステップ５１７に進んでサブルーチンをリターンする。
【００７３】
ｄ軸電流Ｉｄ、ｑ軸電流Ｉｑは、基本的にはキャリヤ信号ごとに変換するので、トルクリップルも含めて変動が大きい。変換したｄ軸電流Ｉｄ、ｑ軸電流Ｉｑと設定値Ｉｄｓ、Ｉｑｓをキャリヤごとに比較判断制御すると変動要素が大きく制御が安定しないので、平均化するなどの積分要素を加える必要がある。よって、回転数制御サブルーチンは、図１１に示すように、キャリヤ信号割込サブルーチン、あるいは、位置信号割込サブルーチンの中で実行せず、モータ駆動制御の中で独立に実行させる。ただし、回転制御の応答速度を速めるために、位置信号割込サブルーチンの中で行う方法も考えられるが、回転数が低い場合には逆に応答が遅くなる欠点がある。
【００７４】
（実施例２）
上記実施例１は、減速制動時にモータ電流をベクトル制御するものであるが、撹拌行程において、減速制動トルクの細かな制御は不必要なので、モータ印加電圧ベクトルを制御するだけで十分な場合があり、本実施例は、モータ印加電圧ベクトルを制御するものであり、図１５および図１６を参照しながら説明する。
【００７５】
図１５に示すように、制御手段６’は、上記実施例１の制御手段６を構成する２相／３相ｄｑ逆変換手段６３と設定変更手段６５に若干の変更を加えたものである。２相／３相ｄｑ逆変換手段６３’は、モータ電流制御手段６８の出力信号Ｖｑ、Ｖｄを（数２）に従い３相モータ電圧に変換するだけではなく、設定変更手段６５’からのｑ軸電圧設定信号Ｖｑｓ、ｄ軸電圧設定信号Ｖｄｓを直接モータ印加電圧に変換する。すなわち、通常モータ駆動時には、モータ電流をベクトル制御するが、減速時には、設定変更手段６５’からの設定信号によりｑ軸電圧制御信号Ｖｑ、ｄ軸電圧制御信号Ｖｄを制御してモータ印加電圧を制御するものである。他の構成は上記実施例１と同じである。
【００７６】
制御フローチャートは、上記実施例１とほとんど同じであり、図１６に示すように、回転数制御サブルーチンのフローチャートを一部変更することにより実現できる。
【００７７】
図１６において、ステップ５００’より回転数サブルーチンが開始し、ステップ５０１にてモータ４の検知回転数ｎを呼び出し、つぎにステップ５０２に進んで通常撹拌駆動か、減速制御かを判定する。通常駆動の場合のフローチャートは図１４と同じなので説明を省略する。減速駆動の場合には、ステップ５１８に進んでｑ軸電圧設定信号Ｖｑｓを負に設定し、つぎに、ステップ５１９に進み、モータ回転数ｎに応じてｄ軸電圧設定信号Ｖｄｓを制御する。
【００７８】
電力回生させないようにするには、ｄ軸電圧設定信号Ｖｄｓを負に設定して弱め界磁とし、回転数が低下してからＶｄｓを正にして強め界磁にすることが望ましい。しかし、撹拌時の慣性モーメントが少ない場合には、制動開始時点でＶｄｓを正にし、回転数に応じてＶｄｓを変化させなくても構わない。
【００７９】
特に、縦型乾燥洗濯機の乾燥撹拌行程においては、水なしで衣類を撹拌させて乾燥させるので慣性モーメントは少なく、電圧ベクトルを一定にしても電力回生は起こらない。ただし、電圧ベクトルを大きくすると制動トルクが大きくなるので、電圧ベクトルを減らして予め制動トルクを小さく設定する必要がある。
【００８０】
上記実施例１に示したように、負荷量に応じて電圧ベクトルを制御すると、より最適な減速制動制御が可能となることは明らかである。
【００８１】
ステップ５１９を実行した後、ステップ５１６に進んでｑ軸電圧信号Ｖｑ、ｄ軸電圧信号Ｖｑをメモリし、サブルーチンをリターンする。回転数制御サブルーチンは、図１１に示すモータ駆動サブルーチンに戻るので、モータ駆動サブルーチン内の図１２に示すキャリヤ信号割り込みサブルーチンにて、２相／３相ｄｑ逆変換が実行されてモータ印加電圧が制御され、モータ電流が制御される。
【００８２】
以上述べた如く本発明の特徴は、撹拌翼の正反転制御のモータ停止前に、モータ誘起電圧と逆位相のモータ電流を制御して減速制動させるもので、撹拌翼の正反転制御の減速制動トルクの制御を容易にして最適な制動トルクを得るとともに、減速機構の遊びにより発生するカタカタ音を減らすことができ、モータ４が停止してから反転させる休止時間を短縮できるので、正反転時間を短縮させることができ、洗い撹拌時の洗浄性能を向上することができる。
【００８３】
また、縦型洗濯乾燥機の乾燥工程においては、衣類に温風が均一に吹き付けられるように水なし撹拌を行う必要があり、衣類が乾燥するに従い最適なトルク制御が必要となるが、本発明によれば、撹拌駆動時と減速制動トルクをそれぞれ最適に制御できるので、布痛みの少ない水なし撹拌が可能となる。
【００８４】
さらに、制動時に発生する回生エネルギーの制御を容易にして発電エネルギーをモータ４の内部抵抗に消費させ、インバータ回路３の直流電圧の異常上昇を防止することができ、モータ４の減速制動時の停止を確実にすることができる。
【００８５】
【発明の効果】
以上のように本発明の請求項１に記載の発明によれば、交流電源と、前記交流電源に接続された整流回路と、前記整流回路の直流電力を交流電力に変換するインバータ回路と、前記インバータ回路により駆動され撹拌翼を駆動するモータと、前記モータのロータ位置を検出するロータ位置検出手段と、前記インバータ回路を制御する制御手段とを備え、前記制御手段は、前記モータを正反転制御して撹拌行程を実行するとともに、前記撹拌行程における前記撹拌翼の正反転制御の減速制動時に、モータ電圧位相をモータ誘起電圧から１８０度以下に進角させ電力回生しないようにしたから、モータ電流、あるいはモータ電流位相を制御することにより制動トルクを容易に制御できるので、減速制動時の最適制動トルク制御が可能となり、撹拌駆動時の駆動トルクを大きくして、正反転時間を短縮することができる。
【００８６】
また、請求項２に記載の発明によれば、制御手段は、モータの減速制動時に、前記モータの回転数が低下するとモータ電圧位相をモータ誘起電圧に対して、２４０度近傍まで進角させるようにしたから、モータ電流位相を１８０度未満進角させることにより発電エネルギーをモータの内部抵抗に消費させてインバータ回路の直流電圧の異常上昇を防止することができ、低速回転時に１８０度以上進角させることにより強め界磁となり回転停止時間を短縮できる。
【００８７】
また、請求項３に記載の発明によれば、交流電源と、前記交流電源に接続された整流回路と、前記整流回路の直流電力を交流電力に変換するインバータ回路と、前記インバータ回路により駆動され撹拌翼を駆動するモータと、前記モータのロータ位置を検出するロータ位置検出手段と、前記モータ電流を検出する電流検出手段と、前記インバータ回路を制御する制御手段とを備え、前記制御手段は、前記モータを正反転制御して撹拌行程を実行するとともに、前記モータ電流を磁束に対応した電流成分とトルクに対応した電流成分に分解し、前記撹拌行程における前記撹拌翼の正反転制御の減速制動時に、前記トルクに対応した電流成分と磁束に対応した電流成分をそれぞれ負に設定し、前記トルクに対応した電流成分を負荷量に応じて設定制御するようにしたから、モータ電流を磁束に対応した電流成分とトルクに対応した電流成分に分解して減速制動することで、ベクトル制御により制動トルクに対応した電流成分を制御することにより、最適制動トルクを得ることができ、撹拌駆動時の駆動トルクを大きくして、正反転時間を短縮することができる。
【００８８】
また、請求項４に記載の発明によれば、制御手段は、モータの減速制動時に、制動トルクに対応した電流成分を撹拌駆動時のトルクに対応した電流に応じて設定制御するようにしたから、撹拌駆動時のトルク電流により負荷量が判定できるので、負荷量に応じて減速制動時のトルク電流成分を制御することによりブレーキ時間を短縮して撹拌反転時間を短縮することができ、負荷量が軽い場合には制動トルク電流を減らすことができるので、撹拌翼と駆動軸の遊びによるカタカタ音を減らすことができる。
【図面の簡単な説明】
【図１】 本発明の第１の実施例のランドリー機器のモータ駆動装置の一部ブロック化した回路図
【図２】 同ランドリー機器のモータ駆動装置の撹拌駆動時のタイムチャート
【図３】 同ランドリー機器のモータ駆動装置の減速制動時のタイムチャート
【図４】 同ランドリー機器のモータ駆動装置の撹拌駆動時のモータ電圧電流ベクトル図
【図５】 同ランドリー機器のモータ駆動装置の減速制動時のモータ電圧電流ベクトル図
【図６】 同ランドリー機器のモータ駆動装置の低回転数における減速制動時のモータ電圧電流ベクトル図
【図７】 同ランドリー機器のモータ駆動装置のモータ電流をｄ軸電流とｑ軸電流に分解した電流ベクトル図
【図８】 同ランドリー機器のモータ駆動装置の撹拌行程のフローチャート
【図９】 同ランドリー機器のモータ駆動装置の撹拌行程の制御タイムチャート
【図１０】 同ランドリー機器のモータ駆動装置の負荷量と負のトルク電流設定値の制御特性図
【図１１】 同ランドリー機器のモータ駆動装置のモータ駆動サブルーチンのフローチャート
【図１２】 同ランドリー機器のモータ駆動装置のキャリヤ信号割込サブルーチンのフローチャート
【図１３】 同ランドリー機器のモータ駆動装置の位置信号割込サブルーチンのフローチャート
【図１４】 同ランドリー機器のモータ駆動装置の回転数制御サブルーチンのフローチャート
【図１５】 本発明の第２の実施例のランドリー機器のモータ駆動装置の一部ブロック化した回路図
【図１６】 同ランドリー機器のモータ駆動装置の回転数制御サブルーチンのフローチャート
【符号の説明】
１ 交流電源
２ 整流回路
３ インバータ回路
４ モータ
４ａ ロータ位置検出手段
６ 制御手段[0001]
BACKGROUND OF THE INVENTION
  The present invention relates to a motor drive device of a laundry machine that drives a motor by an inverter circuit.
[0002]
[Prior art]
  In recent years, it has been proposed to reduce noise and improve cleaning power by driving a motor of a washing machine with an inverter device.
[0003]
  Conventionally, this type of washing machine has been configured as shown in JP-A-4-129594. That is, at the time of forward reversal control of the stirring blade, the stirring blade is decelerated and braked by a short-circuit brake to rapidly stop the stirring blade to shorten the reversing time. As a result, the cleaning performance was improved and the reliability of the motor and the speed reduction mechanism was improved.
[0004]
[Problems to be solved by the invention]
  However, in such a conventional configuration, since the brake torque cannot be controlled, when the load is light, the agitating blade stops rapidly, and a rattling noise is generated due to play between the drive shaft and the agitating blade. As a result, a large torque is generated in the speed reduction mechanism and the motor shaft, resulting in a problem that mechanical wear increases.
[0005]
  The present invention solves the above-described conventional problems, and makes it easy to control the deceleration braking torque (negative torque) for forward reversal control of the agitating blade to obtain the optimum brake torque, and to prevent the rattling noise generated by the play of the mechanism. In addition, the regenerative energy generated during braking can be controlled easily, and the generated energy is consumed by the motor's internal resistance to prevent an abnormal increase in the DC voltage of the inverter circuit. Furthermore, the motor is stopped during deceleration braking. The purpose is to ensure.
[0006]
[Means for Solving the Problems]
  In order to achieve the above object, the present invention converts the DC power of a rectifier circuit connected to an AC power source into AC power by an inverter circuit, adds it to the motor, and agitates the motor.WingsDriving, detecting the rotor position of the motor by the rotor position detection means, and controlling the inverter circuit by the control means, the control means,Control the motor forward and reverse to execute the stirring stroke, andStirring bladeForward / reverse controlDuring deceleration braking, the motorVoltageWhether phase is motor induced voltageEt 180 degreesless thanAdvanceLet power not regenerateIt is what I did.
[0007]
  As a result, the braking torque can be easily controlled, so that the abnormal torque to the motor shaft and the speed reduction mechanism can be reduced, the forward reversing time of the stirring blade can be shortened, and the rattling sound generated by the play of the mechanism can be reduced. Therefore, the regenerative power can be easily controlled, and the generated energy can be consumed by the internal resistance of the motor to prevent an abnormal increase in the DC voltage of the inverter circuit.
[0008]
DETAILED DESCRIPTION OF THE INVENTION
  The invention according to claim 1 of the present invention is an AC power source, a rectifier circuit connected to the AC power source, an inverter circuit that converts DC power of the rectifier circuit into AC power, and stirring driven by the inverter circuit.WingsA motor for driving, rotor position detecting means for detecting a rotor position of the motor, and control means for controlling the inverter circuit, the control means comprising:While performing a stirring process by controlling the motor to reverse in the reverse direction, in the stirring processOf the stirring bladeForward / reverse controlDuring deceleration braking, the motorVoltageWhether phase is motor induced voltageEt 180 degreesless thanAdvanceLet power not regenerateSince the braking torque can be easily controlled by controlling the motor current or motor current phase, optimal braking torque control during deceleration braking is possible, and the driving torque during agitation driving is increased. The forward inversion time can be shortened.
[0009]
  According to a second aspect of the present invention, in the first aspect of the present invention, the control means is configured to reduce the braking speed of the motor.When the rotation speed of the motor decreasesmotorVoltagePhase relative to motor induced voltage240 degreesTo the vicinityAdvances the motor current phase by advancing the motor current phase by less than 180 degrees, so that power generation energy is consumed by the internal resistance of the motor, preventing abnormal rise in the DC voltage of the inverter circuit, and low speed rotation Sometimes, advancement of 180 degrees or more becomes a strong field, and rotation stop time can be shortened..
[0010]
Claim3The invention according to claim 1 is an AC power source, a rectifier circuit connected to the AC power source, an inverter circuit that converts DC power of the rectifier circuit into AC power, and an agitator driven by the inverter circuit.WingsA motor for driving; rotor position detecting means for detecting a rotor position of the motor; current detecting means for detecting the motor current; and control means for controlling the inverter circuit;While carrying out a stirring process by controlling the motor in the reverse direction,Decomposing the motor current into a current component corresponding to magnetic flux and a current component corresponding to torque,In the stirring stepForward reversal control of the stirring bladeDeceleration brakingSometimes the current component corresponding to the torqueAnd current components corresponding to magnetic fluxSet to negativeThe current component corresponding to the torque is set according to the load amountSystemControlBy decelerating and braking the motor current into a current component corresponding to the magnetic flux and a current component corresponding to the torque, and controlling the current component corresponding to the braking torque by vector control, The optimum braking torque can be obtained, the driving torque during agitation driving can be increased, and the forward inversion time can be shortened..
[0011]
Claim4The invention described in claim 13In the present invention, the control means sets and controls the current component corresponding to the braking torque during the deceleration braking of the motor according to the current corresponding to the torque during the agitation drive. Since the load amount can be determined by the torque current, by controlling the torque current component during deceleration braking according to the load amount, the braking time can be shortened and the agitation inversion time can be shortened. Since the braking torque current can be reduced, the rattling noise caused by the play of the agitating blade and drive shaft can be reduced..
[0012]
【Example】
  Embodiments of the present invention will be described below with reference to the drawings.
[0013]
  (Example 1)
  As shown in FIG. 1, the AC power source 1 applies AC power to the rectifier circuit 2, and the rectifier circuit 2 converts the DC power into DC power using a rectifier 20 and a capacitor 21, and applies a DC voltage to the inverter circuit 3.
[0014]
  The inverter circuit 3 is constituted by a three-phase full-bridge inverter circuit composed of six power switching semiconductors and an antiparallel diode, and normally includes an insulated gate bipolar transistor (IGBT), an antiparallel diode, its driving circuit, and a protection circuit. It consists of an intelligent power module (hereinafter referred to as IPM). The motor 4 is connected to the output terminal of the inverter circuit 3 to drive a stirring blade (not shown) or a washing and dewatering tub (not shown).
[0015]
  The motor 4 is constituted by a direct current brushless motor, and the rotor position detecting means 4a detects the relative position (rotor position) between the permanent magnet and the stator constituting the rotor. The rotor position detecting means 4a is normally composed of three Hall ICs and detects a position signal for every 60 degrees of electrical angle. The current detection means 5 detects the phase currents Iu, Iv, and Iw of the motor 4, and normally uses a DC current transformer that can be measured from a low frequency including DC current. It can also be detected by an alternating current transformer or a shunt resistor. In the case of a three-phase motor, a general method is to obtain a two-phase current and obtain the remaining one phase from Kirchhoff's law (Iu + Iv + Iw = 0).
[0016]
  The control means 6 controls the rotation speed of the motor 4 by vector-controlling the inverter circuit 3 by the rotor position detection means 4a and the current detection means 5. The control means 6 controls the rotation speed of the motor 4 by the inverter circuit 3, and includes a microcomputer, an inverter control timer (PWM timer) built in the microcomputer, a high-speed A / D conversion circuit, a memory circuit (ROM, RAM). ) And the like, and an electric angle detecting means 60 for detecting an electric angle from an output signal of the rotor position detecting means 4a, and a current component Id corresponding to the magnetic flux from the output signal of the current detecting means 5 and the signal of the electric angle detecting means 60. And three-phase / two-phase dq conversion means 61 that decomposes into a current component Iq corresponding to torque, and sine wave data (sin, cos data) necessary for conversion from a stationary coordinate system to a rotation coordinate system or inverse conversion. The storage means 62 for storing, the voltage component Vd corresponding to the magnetic flux and the voltage component Vq corresponding to the torque into the three-phase motor drive control voltages vu, vv, 2-phase / 3-phase dq inverse conversion means 63 for converting the w, 3-phase motor drive control voltage vu, vv, and a like PWM control means 64 for controlling the IGBT switching of the inverter circuit 3 in accordance with the vw.
[0017]
  Further, setting change means 65 for controlling the start, stop, rotation speed, braking, etc. of the motor 4 according to the stroke, rotation speed detection means 66 for detecting the rotation speed from the output signal of the rotor position detection means 4a, and rotation A rotation speed control means 67 for controlling the rotation speed of the motor 5 according to the output signal of the number detection means 66, and a d-axis (direct-axis) current setting signal Ids, q from the setting change means 65 and the rotation speed control means 67 A voltage component Vd corresponding to the magnetic flux for controlling the motor current by comparing the Id and Iq calculated by the quadrature-axis current setting signal Iqs and the three-phase / 2-phase dq conversion means 61 and a voltage corresponding to the torque. Motor current control means 68 for calculating the component Vq.
[0018]
  Constant torque control is possible by performing feedback control so that the q-axis current Iq corresponding to the torque becomes the set value Iqs. However, since the motor induced voltage rises and the torque current Iq does not increase as the rotational speed increases, the q-axis current can also be increased by so-called weakening magnetic flux control that increases the d-axis current according to the rotational speed. Torque can be increased.
[0019]
  FIG. 2 shows the waveform relationship of each part when the motor is driven, and the edge signals of the output signals H1, H2 and H3 of the rotor position detecting means 4a change every 60 degrees, and the angle obtained by dividing 360 degrees into 6 parts from each part state signal Can be determined. A high edge where the signal H1 changes from low to high is indicated as a reference electrical angle of 0 degree, and the U-phase winding induced voltage Ec of the motor 4 has a waveform delayed by 30 degrees from the reference signal H1. Maximum efficiency is obtained when the phases of the U-phase motor current Iu and the motor-induced voltage Ec are the same. The motor induced voltage Ec is the same axis as the q axis, and the d axis is delayed by 90 degrees. Since the q-axis current is in phase with the motor induced voltage phase, it is called torque current.
[0020]
  FIG. 3 shows a waveform relationship of each part at the time of motor braking, and a negative torque is generated by applying a current substantially in phase opposite to the motor induced voltage Ec to perform deceleration braking. When the agitator blades are rotated forward or reversely, the suspension period can be shortened and the forward reversal time can be shortened by stopping the drive and decelerating braking compared to the case where the motor is stopped and the suspension period is reversed and then reversed. it can. In particular, in order to increase the stirring force, if the motor current is increased to increase the driving torque, it is necessary to increase the pause interval. You can save time.
[0021]
  FIG. 4 shows a vector diagram when the motor is driven, and the motor applied voltage Va is slightly advanced with respect to the motor induced voltage Ec (q axis), and the motor current I is substantially in phase with the motor induced voltage Ec. Is increasing.
[0022]
  FIG. 5 shows a vector diagram at the time of motor deceleration braking. The phase of the motor applied voltage Va is advanced to the vicinity of 180 degrees (π) with respect to the motor induced voltage Ec (q axis), and the phase β of the motor current I with respect to the q axis is set. A braking torque is generated in a phase almost opposite to the motor induced voltage Ec.
[0023]
  In general, the torque T per one phase of the motor, when ωr is an angular velocity, from T = EcI cos β / ωr, when the phase β is 90 degrees (π / 2) or more, the torque T becomes negative and braking torque is generated. I cos β corresponds to the q-axis current, that is, the torque current, and when the torque current becomes negative, it becomes the brake current or the braking torque current. The applied power P per motor phase is P = VaI cos α when the phase of the motor current I and the motor applied voltage Va is α, and becomes negative power when the phase α is 90 degrees (π / 2) or more. The motor back electromotive force is regenerated and the inverter circuit DC power supply voltage rises.
[0024]
  In order to prevent power regeneration, the motor current phase β or the motor applied voltage phase (α + β) is controlled so that the phase α is 90 degrees or less. Normally, when the motor voltage application phase (α + β) is set to 180 degrees or less, there is no power regeneration, and when the motor rotation speed decreases and the back electromotive force decreases, (α + β) is advanced to around 240 degrees (120 degrees delayed). But power regeneration hardly occurs.
[0025]
  FIG. 6 shows a vector diagram at the time of deceleration braking when the motor rotation speed is lowered. Even if the motor applied voltage phase is advanced by approximately 240 degrees, the phase of the motor applied voltage Va and the motor current is reduced and power regeneration is not performed. Is shown. As a result, when the rotational speed further decreases, the voltage drop ωLI due to the reactance component becomes almost zero, and only the voltage drop RI due to the motor winding resistance R becomes present, and the current phase becomes substantially equal to the motor applied voltage phase. At this time, since strong field control is performed, the field magnetic pole and the rotor magnetic pole are attracted to increase the braking torque, and the rotor is likely to stop.
[0026]
  In FIG. 2, the U-phase motor current Iu is slightly advanced from the U-phase winding induced voltage Ec, and the motor applied voltage Vu shows a waveform approximately 30 degrees ahead of the U-phase winding induced voltage Ec. vc is a sawtooth waveform carrier signal generated in the PWM control means 64, and vu is a sinusoidal U-phase control voltage. The PWM signal u, which compares the carrier signal vc and the U-phase control voltage vu, is generated in the PWM control means 64. And is added as a control signal for the U-phase upper arm transistor of the inverter circuit 3. ck is a synchronizing signal of the carrier signal vc, and is an interrupt signal when the carrier counter counts up and overflows.
[0027]
  Coordinate conversion from the stationary coordinate system to the rotating coordinate system, that is, dq conversion, is performed with the electrical angle at which the rotor magnet axis of the motor 4 and the magnetic flux axis of the stator coincided as the d axis, and a reference electrical angle of 0 degrees. 60 detects electrical angles such as 30 degrees, 90 degrees, and 150 degrees from the output signals H1, H2, and H3 of the rotor position detecting means 4a, and obtains the electrical angle θ by estimation except every 60 degrees.
[0028]
  In general, the current component corresponding to the magnetic flux is called a d-axis current Id, and the torque is zero because the permanent magnet magnetic flux and the field magnetic flux are coaxially attracted to the field magnet. Further, the lead angle is 90 degrees in electrical angle from the d-axis, the phase is the same as the induced voltage phase, and the maximum torque is called the q-axis, and the current component corresponding to the torque is called the q-axis current Iq. Further, increasing the d-axis current in the negative direction is equivalent to weakening the field magnetic flux on the d-axis, and is called field weakening control or weakening magnetic flux control (or magnetic flux weakening control). Also, since it is divided into d-axis current and q-axis current and controlled independently, it is called vector control.
[0029]
  The three-phase / two-phase dq conversion means 61 converts the motor currents Iu, Iv, and Iw into the d-axis current Id and the q-axis current Iq according to (Equation 1), and the motor current detected corresponding to the electrical angle θ. Id and Iq are calculated from the instantaneous values.
[0030]
[Expression 1]

[0031]
  Since the storage unit 62 stores the data of sin θ and cos θ, it can be decomposed into the d-axis current Id and the q-axis current Iq by calling the data corresponding to the electrical angle data and performing the product-sum operation. The detection of the electrical angle θ and the detection of the instantaneous motor current value are performed in synchronization with the carrier signal, and will be described in detail according to the flowchart described later.
[0032]
  The rotation speed detection means 66 detects the motor rotation speed from the output reference signal H1 of the rotor position detection means 4a and applies the rotation speed signal to the setting change means 65 and the rotation speed control means 67. The setting changing means 65 performs start-up control of the motor 4 and setting of the rotation speed, switching between rotation speed priority control or current priority control, and calculation of the d axis current Ids according to the rotation speed and the q axis current set value Iqs. A rotation speed setting signal Ns or a current priority control signal Iqo is applied to the rotation speed control means 67 during deceleration braking, and a d-axis setting signal Ids is added to the motor current control means 68.
[0033]
  The rotation speed control means 67 includes a rotation speed comparison means 67a that compares the detected rotation speed n with the set rotation speed Ns, an error signal Δn between the detected rotation speed n and the set rotation speed Ns, and a change rate (acceleration) of the rotation speed. Torque current setting means 67b for controlling the q-axis current set value Iqs in response to the output, and the q-axis current set value Iqs is output as it is from the torque current setting means 67b, or the current priority control is performed by a command from the setting change means 65 The q-axis current switching means 67c is used to switch between the two.
[0034]
  In the case of the rotational speed control, so-called rotational speed control current minor loop control is performed to control the q-axis current set value Iqs according to the error signal Δn, and current control is performed during deceleration braking. The current set value Iqo is set, and Iqs = Iqo.
[0035]
  The motor current control means 68 compares the output signals Iq and Id of the three-phase / two-phase dq conversion means 61 with the setting signals Iqs and Ids and outputs control voltage signals Vq and Vd. 68a, q-axis voltage setting means 68b, d-axis current comparison means 68c, and d-axis voltage setting means 68d, and generate voltage signals Vq and Vd for controlling the q-axis current and the d-axis current, respectively.
[0036]
  The d-axis current set value Ids is a signal applied from the setting changing means 65 to the motor current control means 68. In the case of an embedded magnet motor, the d-axis current set value Ids is increased in accordance with the rotation speed to control the field weakening. I do. In the case of a surface magnet motor, Ids is normally set to zero, and Ids is increased in the case of high rotational speed driving.
[0037]
  The two-phase / three-phase dq reverse conversion means 63 calculates the three-phase motor drive control voltages vu, vv, vw from the voltage signals Vq, Vd from (Equation 2), and detects the electrical angle in synchronization with the carrier signal. A sinusoidal signal corresponding to the electrical angle θ detected by the means 60 is applied to the PWM control means. The method of product-sum calculation of sin θ and cos θ stored in the storage means 62 is almost the same as the calculation of the three-phase / 2-phase dq conversion means 61.
[0038]
[Expression 2]

[0039]
  FIG. 7 is a vector diagram in which the motor current during rotation driving and braking is decomposed into a q-axis current Iq and a d-axis current. I1 is a current vector during forward rotation or reverse driving, and Iq1 and -Id1 are Q-axis current Iq and d-axis current, I2 is a current vector at the time of deceleration braking, q-axis current and d-axis current are respectively set to negative (−Iq2, −Id2), and I3 is a current immediately before braking is stopped. In the vector, the q-axis current is set negative and the d-axis current is set positive (-Iq2, Id3).
[0040]
  When the motor rotation speed is high, the d-axis current during deceleration braking is set to a negative value and field weakening control is performed, so that power regeneration can be suppressed, and the motor back electromotive force is consumed by the motor resistance. When the motor rotation speed decreases, the motor back electromotive force decreases, energy is consumed by the motor winding resistance, and regeneration does not occur. By performing strong field control to make the d-axis current positive, the motor braking torque increases and stops. It becomes easy.
[0041]
  In the case of vector control that breaks down into q-axis current Iq and d-axis current, it is necessary to set q-axis current Iq and d-axis current, respectively. It doesn't matter. In addition, when stirring is performed without water in the washing and dewatering tank, for example, in the drying and stirring process in the drying process of the washing and drying machine, the stirring blades are rapidly stopped and the regenerative power is rapidly decreased. It is only necessary to advance the applied voltage vector by 240 degrees.
[0042]
  The operation of the above configuration will be described with reference to FIGS. 8, 9, 10, 11, 12, 13, and 14. FIG. FIG. 8 is a flowchart of the agitation process, which is used for the washing process of the washing machine or the washing process of the vertical washer / dryer and the agitation of clothes by the agitating blades in the drying process.
[0043]
  FIG. 9 is a time chart of the agitation stroke, showing how the rotation speed changes with time, the forward and reverse motor drive time ton, the deceleration braking period after motor driving tb, and the motor drive stop period Is toff.
[0044]
  In FIG. 8, the agitation process is started from step 100, and various initial settings of the washing agitation process are performed in step 101. Next, the process proceeds to step 102 where forward / reverse control setting is performed. The next time, reverse is set, and if the previous motor drive is reversed, the next time is set as normal reverse. Next, the routine proceeds to step 103, where the motor drive subroutine shown in FIG. 11 is executed, and the motor current is vector-controlled so as to obtain a predetermined rotational speed. Next, the routine proceeds to step 104, where it is determined whether or not the motor drive time is not less than the set value ton.
[0045]
  In step 105, deceleration braking is set, and the process proceeds to step 106 where the q-axis current setting value and the d-axis current setting value are set to be negative. Next, the motor drive subroutine of step 107 is executed. At this time, since the motor drive subroutine is set to deceleration braking in step 105, negative torque current control is performed.
[0046]
  For deceleration control, deceleration braking is possible even if the set speed is set to zero instead of current control instead of current control. In this case, upper and lower limit values are set to the q-axis current and d-axis current set values. Need to be controlled. That is, during deceleration braking, the q-axis current is set from zero to a negative maximum value -Iqmax, the d-axis current is set from a negative maximum value -Idmax to a positive maximum value + Idmax, and the positive maximum value + Idmax is decreased. Otherwise, the inverter DC voltage will rise abnormally due to the power regeneration action. Further, when the braking torque is increased and the load is light, a rattling sound is generated due to the play of the speed reduction mechanism and the stirring blade. Therefore, it is necessary to control the braking torque maximum value according to the load amount, which is the same as the actual current control.
[0047]
  Next, the routine proceeds to step 108, where it is determined whether or not the motor speed n has reached approximately zero. If it is approximately zero, the routine proceeds to step 109, where the motor is stopped for a predetermined time. Next, the process proceeds to step 110 to determine whether or not the agitation process is completed. If the process is completed, the process proceeds to step 111 and the subroutine is returned. If the process is not completed, the process proceeds to step 102 and the agitation process is continued.
[0048]
  FIG. 10 shows the control characteristics of the q-axis current set value (−Iqs) at the time of deceleration braking and the q-axis current at the time of forward / reverse driving. When the number of revolutions of the stirring blade is controlled at the set number of revolutions, the load is large. Since the q-axis current increases when the motor is driven, the load amount can be determined by the q-axis current. Therefore, the negative torque current set value (−Iqs) during deceleration braking is controlled according to the load amount. The larger the load amount, the shorter the braking time by increasing the brake current. If the load amount is small, the brake current can be reduced to reduce the rattling noise caused by the play of the stirring blade and the speed reduction mechanism generated by the rapid stop of the stirring blade. it can.
[0049]
  As another method of determining the load amount, a method of determining based on the inertia rotation speed after stopping the stirring drive or the number of position detection pulses of the rotor position detection means can be considered. Further, it can be determined from the increase in the rotational speed with the torque current at the time of stirring driving being constant. In addition, detection accuracy is further improved by detecting the load amount using two signals, ie, a q-axis current at the time of stirring drive and an inertial rotation speed after stopping the stirring drive. When detecting inertial rotation speed, if deceleration braking is performed, load amount detection accuracy is deteriorated. Therefore, deceleration braking is not provided in load amount detection.
[0050]
  Next, the flowchart of the motor drive subroutine shown in FIG. 11 will be described. From step 200, the motor drive subroutine starts. Step 201 is an initial determination that is determined at the beginning of the subroutine execution. The activation or braking initial state is determined. If the activation or braking is initial, the process proceeds to Step 202, various initial settings are performed, parameter passing from the main routine and various settings are performed. Execute.
[0051]
  Next, the routine proceeds to step 203, where rotation start control or braking initial control is performed. Steps 202 and 203 are executed only once at the beginning. In the start control, at the time of start in which the rotation speed feedback control cannot be performed, a predetermined motor applied voltage is set and energized 120 degrees, and soft start is performed to increase the voltage from the low motor applied voltage to the high voltage with time. In the case of braking operation, the negative d-axis current is increased, the negative q-axis current is decreased, power regeneration at high rotational speed is prevented, and soft braking is performed so that sudden braking torque is not applied.
[0052]
  Next, the routine proceeds to step 204 where it is determined whether or not there is a carrier signal interrupt. The carrier signal interrupt is executed by an interrupt signal ck that is generated when the carrier counter of the PWM control means 65 overflows, and the process proceeds to step 205 to execute a carrier signal interrupt subroutine.
[0053]
  FIG. 12 shows a detailed flowchart of the carrier signal interrupt subroutine. The carrier signal interrupt subroutine is started from step 300, and the interrupt signal ck is counted in step 301. Next, the routine proceeds to step 302 where the rotor position electrical angle θ is calculated. The rotor position signal θ is obtained by multiplying the electrical angle Δθ per cycle of the carrier signal separately obtained by a value k · Δθ obtained by multiplying the count value k of the carrier counter by an electrical angle φ every 60 degrees that can be detected by the rotor position detecting means 4a. Addition to estimate calculation.
[0054]
  If the motor 4 has 8 poles, the carrier frequency is 15.6 kHz, and the rotation speed is 900 r / min, the motor drive frequency is 60 Hz, and the carrier counter count value k within an electrical angle of 60 degrees is about 43. Therefore, Δθ is about 1.4 degrees. As the motor rotational speed is lower, the count value k within an electrical angle of 60 degrees is higher, and the electrical angle detection resolution in calculation is improved. Therefore, it can be understood that there is no problem even when the rotational speed is low and accuracy is required.
[0055]
  Next, the routine proceeds to step 303 where the motor currents Iu and Iv are detected. Since noise may be included in one current detection, the process proceeds to step 304 to detect again, and in step 305, the average value is obtained to remove the noise, and the motor current Iw is calculated from Iw = − (Iu + Iv) To do.
[0056]
  Next, the process proceeds to step 306, and the calculation shown in (Equation 1) is performed from the electrical angle θ and the motor current, and the three-phase / 2-phase dq conversion is performed to obtain the d-axis current Id and the q-axis current Iq. In step 307, Id and Iq are stored in memory and used separately as rotation speed control data.
[0057]
  Next, the process proceeds to step 308 to call up the d-axis control voltage Vd and the q-axis control voltage Vq, and the process proceeds to step 309 to perform 2-phase / 3-phase dq reverse conversion according to (Equation 2) to obtain the 3-phase control voltages vu, vv. , Vw is obtained. This inverse transformation uses sin θ and cos θ data corresponding to the electrical angle of the storage means 62 as in step 306 and performs a product-sum operation at high speed. Next, the process proceeds to step 310 to perform PWM control corresponding to the three-phase control voltages vu, vv, vw, and the process proceeds to step 311 to return the subroutine.
[0058]
  As described in FIG. 2, the PWM control is performed by comparing the sawtooth wave (or triangular wave) carrier signal and the control voltages vu, vv, and vw corresponding to the U phase, V phase, and W phase. An IGBT on / off control signal for the circuit 3 is generated to drive the motor 4 in a sine wave. The signals of the upper arm transistor and the lower arm transistor are inverted waveforms, and the output voltage becomes positive when the conduction ratio of the upper arm transistor is increased. When the voltage is increased and the conduction ratio of the lower arm transistor is increased, the output voltage becomes a negative voltage. When the conduction ratio is 50%, the output voltage becomes zero.
[0059]
  When the control voltage is changed in a sine wave shape corresponding to the electrical angle θ, a sine wave current flows. In the case of sine wave drive, when the transistor conduction ratio is set to the maximum value of 100%, the output voltage is maximum and the modulation degree Am is 100%. When the maximum value of the conduction ratio is 50%, the output voltage is minimum and the modulation is performed. The degree Am is called 0%.
[0060]
  Since 3-phase / 2-phase dq conversion and 2-phase / 3-phase dq reverse conversion for vector control of motor current are executed at high speed for each carrier, high-speed current control becomes possible, and torque current for each carrier. Since Iq is detected, the load amount can be determined instantaneously. Further, the vector control for each carrier can always drive with a constant torque, and if the negative constant torque control is performed at the time of deceleration braking, the brake torque can be kept constant and an overcurrent can be prevented.
[0061]
  Returning to FIG. 11, after executing the carrier signal interrupt subroutine, the routine proceeds to step 206, where it is determined whether or not there is a position signal interrupt. When any of the position signals H1, H2, and H3 changes, an interrupt signal is generated, and the process proceeds to step 207 to execute the position signal interrupt subroutine shown in FIG. As shown in FIG. 2, an interrupt signal is generated every 60 electrical angles.
[0062]
  Here, the position signal interrupt subroutine will be described. The position signal interruption subroutine is started from step 400, the process proceeds to step 401, the position signals H1, H2, and H3 are input to detect the position, and then the process proceeds to step 402 to detect the rotor electrical angle θc from the position signal. . Next, the process proceeds to step 403, where the count value k counted in the carrier signal interrupt subroutine is stored in kc, the process proceeds to step 404, the count value k is cleared, the process proceeds to step 405, and an electrical angle between 60 degrees is obtained. The electrical angle Δθ of one carrier is calculated from the carrier counter count value kc.
[0063]
  Next, the process proceeds to step 406 to determine whether the interrupt signal is based on the reference position signal H1. If the reference position signal interrupts, the process proceeds to step 407 and the count value T of the rotation period measurement timer T is stored as the period To. In step 408, the timer T is cleared, and in step 409, the motor rotational speed n is calculated. Next, the routine proceeds to step 410 to start counting of the rotation period measuring timer, and the routine proceeds to step 411 and the subroutine is returned.
[0064]
  If the detection resolution of the rotation period measurement timer is set to 8-bit accuracy, the clock becomes 64 μs, and the carrier signal can be used as the clock. However, in order to improve the rotation control performance, it is necessary to improve the rotation period detection resolution. Needs to be set to 1 to 10 μs. In this case, the microcomputer system clock is divided and used as the clock.
[0065]
  Although the rotation speed detection method described above has shown the method of calculating | requiring from the period of the position signal H1, you may use all the position signals H1, H2, and H3. If the carrier signal is a triangular wave, the carrier counter timer has a period twice that of the sawtooth wave. Therefore, if the triangular wave overflow signal is used as a clock, the resolution is improved. Therefore, the triangular wave timer overflow signal may be used as a clock.
[0066]
  Next, in FIG. 11, after the position signal interruption subroutine 207 is executed, the routine proceeds to step 208, where it is determined whether or not there is a rotational speed control flag, and if rotational speed control is set, the routine proceeds to step 209 and the rotational speed control subroutine is executed. If the deceleration braking setting is set, the routine proceeds to step 210, where the deceleration braking subroutine included in the rotation speed control subroutine is executed. Details of the rotation speed control subroutine are shown in FIG.
[0067]
  In FIG. 14, the rotational speed control subroutine is started from step 500, the motor rotational speed n is called at step 501, and the routine proceeds to step 502, where it is determined whether normal driving or deceleration braking is performed.
[0068]
  If it is normal driving, the process proceeds to step 503, the torque control is performed by controlling the q-axis current set value Iqs based on the error between the set rotational speed and the detected rotational speed, and then the process proceeds to step 504 to set the q-axis current set value Iqs to the upper limit If it is greater than Iqmax, the process proceeds to step 505 where Iqs is set to Iqmax so that the q-axis current Iq does not exceed the upper limit value Iqmax.
[0069]
  If it is deceleration braking, the routine proceeds to step 506, where the q-axis current set value is set to -Iqs for negative torque control, that is, brake torque control. Next, the routine proceeds to step 507, where the d-axis current set value Ids is controlled according to the rotational speed. When the rotational speed is high, the d-axis current set value Ids is made negative to make the field weak so as not to regenerate power, and when the rotational speed is lowered, Ids is made positive to make the field strong.
[0070]
  Next, in step 508, the d-axis current Id is called, and in step 509, Id and Ids are compared in magnitude. If the d-axis current Id is larger than the set value Ids, the process proceeds to step 510 and the d-axis control voltage Vd is reached. If the d-axis current Id is smaller than the set value Ids, the process proceeds to step 511 to increase the d-axis control voltage Vd.
[0071]
  Next, the routine proceeds to step 512, where the q-axis current Iq obtained from the three-phase / two-phase dq conversion means 61 is called, and Iq and Iqs are compared in magnitude at step 513, and the q-axis current Iq is determined from the set value Iqs. If so, the process proceeds to step 514 to decrease the q-axis control pressure Vq, and if the q-axis current Iq is smaller than the set value Iqs, the process proceeds to step 515 to increase the q-axis control voltage Vq.
[0072]
  Next, the process proceeds to step 516 to store the calculated d-axis control voltage Vd and q-axis control voltage Vq, respectively, and the process proceeds to step 517 to return the subroutine.
[0073]
  Since the d-axis current Id and the q-axis current Iq are basically converted for each carrier signal, fluctuations including torque ripple are large. If the converted d-axis current Id and q-axis current Iq and the set values Ids and Iqs are compared and determined for each carrier, the fluctuation element is large and the control is not stable. Therefore, it is necessary to add an integration element such as averaging. Therefore, as shown in FIG. 11, the rotation speed control subroutine is not executed in the carrier signal interruption subroutine or the position signal interruption subroutine, but is executed independently in the motor drive control. However, in order to increase the response speed of the rotation control, a method performed in the position signal interruption subroutine is also conceivable, but there is a drawback that the response is delayed when the rotation speed is low.
[0074]
  (Example 2)
  In the first embodiment, the motor current is vector-controlled at the time of deceleration braking. However, since fine control of the deceleration braking torque is unnecessary in the agitation stroke, it may be sufficient only to control the motor applied voltage vector. In this embodiment, the motor applied voltage vector is controlled and will be described with reference to FIGS.
[0075]
  As shown in FIG. 15, the control means 6 ′ is obtained by adding a slight change to the two-phase / three-phase dq inverse conversion means 63 and the setting change means 65 constituting the control means 6 of the first embodiment. The two-phase / three-phase dq reverse conversion means 63 ′ not only converts the output signals Vq and Vd of the motor current control means 68 into a three-phase motor voltage according to (Equation 2) but also the q-axis from the setting change means 65 ′. The voltage setting signal Vqs and the d-axis voltage setting signal Vds are directly converted into a motor applied voltage. That is, during normal motor driving, the motor current is vector-controlled, but during deceleration, the q-axis voltage control signal Vq and the d-axis voltage control signal Vd are controlled by the setting signal from the setting change means 65 'to control the motor applied voltage. To do. Other configurations are the same as those of the first embodiment.
[0076]
  The control flowchart is almost the same as that of the first embodiment, and can be realized by partially changing the flowchart of the rotation speed control subroutine as shown in FIG.
[0077]
  In FIG. 16, the rotational speed subroutine starts from step 500 '. In step 501, the detected rotational speed n of the motor 4 is called, and then the routine proceeds to step 502, where it is determined whether normal stirring drive or deceleration control. The flowchart in the case of normal driving is the same as FIG. In the case of the deceleration drive, the process proceeds to step 518 to set the q-axis voltage setting signal Vqs to negative, and then proceeds to step 519 to control the d-axis voltage setting signal Vds according to the motor rotation speed n.
[0078]
  In order not to regenerate the power, it is desirable to set the d-axis voltage setting signal Vds to be negative to make the field weak, and to make Vds positive and make the field strong after the rotation speed is lowered. However, when the moment of inertia at the time of stirring is small, Vds may be made positive at the start of braking and Vds may not be changed according to the rotational speed.
[0079]
  In particular, in the drying and stirring process of the vertical drying washing machine, since clothes are stirred and dried without water, the moment of inertia is small, and power regeneration does not occur even if the voltage vector is constant. However, since the braking torque increases when the voltage vector is increased, it is necessary to reduce the voltage vector and set the braking torque to be small in advance.
[0080]
  As shown in the first embodiment, it is obvious that more optimal deceleration braking control can be performed by controlling the voltage vector according to the load amount.
[0081]
  After executing step 519, the process proceeds to step 516, the q-axis voltage signal Vq and the d-axis voltage signal Vq are stored, and the subroutine is returned. Since the rotation speed control subroutine returns to the motor drive subroutine shown in FIG. 11, the 2-phase / 3-phase dq reverse conversion is executed in the carrier signal interrupt subroutine shown in FIG. 12 in the motor drive subroutine to control the motor applied voltage. The motor current is controlled.
[0082]
  As described above, the feature of the present invention is that the motor current of the opposite phase to the motor induced voltage is controlled to perform deceleration braking before stopping the motor for forward inversion control of the agitating blade. The torque can be easily controlled to obtain the optimum braking torque, and the rattling noise generated by the play of the speed reduction mechanism can be reduced, and the pause time for reversing after the motor 4 stops can be shortened. It can be shortened and the washing performance at the time of washing and stirring can be improved.
[0083]
  Further, in the drying process of the vertical washer / dryer, it is necessary to perform stirring without water so that warm air can be blown uniformly on the clothing, and optimal torque control is required as the clothing dries. According to the above, since the agitation driving and the deceleration braking torque can be optimally controlled, waterless agitation with less fabric pain is possible.
[0084]
  Furthermore, the regenerative energy generated at the time of braking can be easily controlled, the generated energy can be consumed by the internal resistance of the motor 4, and an abnormal increase in the DC voltage of the inverter circuit 3 can be prevented. Can be ensured.
[0085]
【The invention's effect】
  As described above, according to the first aspect of the present invention, an AC power source, a rectifier circuit connected to the AC power source, an inverter circuit that converts DC power of the rectifier circuit into AC power, and the Agitated driven by inverter circuitWingsA motor for driving, rotor position detecting means for detecting a rotor position of the motor, and control means for controlling the inverter circuit, the control means comprising:While performing a stirring process by controlling the motor to reverse in the reverse direction, in the stirring processOf the stirring bladeForward / reverse controlDuring deceleration braking, the motorVoltageWhether phase is motor induced voltageEt 180 degreesless thanAdvanceLet power not regenerateAs a result, the braking torque can be easily controlled by controlling the motor current or the motor current phase, so that the optimum braking torque control during deceleration braking can be performed, and the driving torque during agitation driving can be increased to increase the braking torque. The inversion time can be shortened.
[0086]
  According to the invention of claim 2, the control means is configured to reduce the braking speed of the motor.When the rotation speed of the motor decreasesmotorVoltagePhase relative to motor induced voltage240 degreesTo the vicinitySince the angle is advanced, the motor current phase is advanced by less than 180 degrees, so that the generated energy is consumed by the internal resistance of the motor, and an abnormal increase in the DC voltage of the inverter circuit can be prevented. Advancing at an angle of more than 50 degrees results in a stronger field and shortens the rotation stop time.
[0087]
Claims3According to the invention, the AC power source, the rectifier circuit connected to the AC power source, the inverter circuit that converts the DC power of the rectifier circuit into AC power, and the stirring driven by the inverter circuitWingsA motor for driving; rotor position detecting means for detecting a rotor position of the motor; current detecting means for detecting the motor current; and control means for controlling the inverter circuit;While carrying out a stirring process by controlling the motor in the reverse direction,Decomposing the motor current into a current component corresponding to magnetic flux and a current component corresponding to torque,In the stirring stepForward reversal control of the stirring bladeDeceleration brakingSometimes the current component corresponding to the torqueAnd current components corresponding to magnetic fluxSet to negativeThe current component corresponding to the torque is set according to the load amountSystemControlTherefore, the motor current is decomposed into a current component corresponding to the magnetic flux and a current component corresponding to the torque, and the braking is decelerated. By controlling the current component corresponding to the braking torque by vector control, the optimum braking is performed. Torque can be obtained, and the drive torque during agitation drive can be increased to shorten the forward inversion time..
[0088]
Claims4According to the invention described in the above, the control means sets and controls the current component corresponding to the braking torque during the deceleration braking of the motor according to the current corresponding to the torque during the stirring drive. Since the load amount can be determined by the torque current, by controlling the torque current component during deceleration braking according to the load amount, the braking time can be shortened and the agitation inversion time can be shortened. Since the braking torque current can be reduced, the rattling noise caused by the play between the agitating blade and the drive shaft can be reduced..
[Brief Description of Drawings]
FIG. 1 is a partial block circuit diagram of a motor drive device for a laundry machine according to a first embodiment of the present invention.
[Fig. 2] Time chart at the time of stirring drive of the motor drive device of the laundry machine
FIG. 3 is a time chart during deceleration braking of the motor drive device of the laundry machine.
FIG. 4 is a motor voltage / current vector diagram during agitation drive of the motor drive device of the laundry machine.
FIG. 5 is a motor voltage / current vector diagram during deceleration braking of the motor drive device of the laundry machine.
FIG. 6 is a motor voltage / current vector diagram during deceleration braking at a low rotational speed of the motor drive device of the laundry machine.
FIG. 7 is a current vector diagram in which the motor current of the motor driving device of the laundry machine is decomposed into a d-axis current and a q-axis current.
FIG. 8 is a flowchart of the stirring process of the motor drive device of the laundry machine
FIG. 9 is a control time chart of the stirring process of the motor drive device of the laundry machine
FIG. 10 is a control characteristic diagram of a load amount and a negative torque current setting value of the motor drive device of the laundry machine.
FIG. 11 is a flowchart of a motor drive subroutine of the motor drive device of the laundry machine
FIG. 12 is a flowchart of a carrier signal interrupt subroutine of the motor drive device of the laundry machine
FIG. 13 is a flowchart of a position signal interrupt subroutine of the motor drive device of the laundry machine.
FIG. 14 is a flowchart of a rotation speed control subroutine of the motor drive device of the laundry machine
FIG. 15 is a circuit diagram in which a part of a motor driving device of a laundry machine according to a second embodiment of the present invention is formed into a block.
FIG. 16 is a flowchart of a rotation speed control subroutine of the motor drive device of the laundry machine
[Explanation of symbols]
  1 AC power supply
  2 Rectifier circuit
  3 Inverter circuit
  4 Motor
  4a Rotor position detection means
  6 Control means

Claims (4)

AC power source, a rectifier circuit connected to the AC power source, an inverter circuit that converts DC power of the rectifier circuit into AC power, a motor that is driven by the inverter circuit and drives a stirring blade, and a rotor position of the motor a rotor position detecting means for detecting, and control means for controlling said inverter circuit, wherein the control means is configured to perform the stirring step the motor in the forward reverse control, positive the stirring blade in the stirring step when the inversion control deceleration braking, the motor driving apparatus of laundry equipment to avoid power regeneration by advancing the motor voltage phase below or al 1 80-degree motor induced voltage. 2. The motor drive of a laundry machine according to claim 1, wherein the control means is configured to advance the motor voltage phase to about 240 degrees with respect to the motor induced voltage when the rotational speed of the motor decreases during deceleration braking of the motor. apparatus. AC power source, a rectifier circuit connected to the AC power source, an inverter circuit that converts DC power of the rectifier circuit into AC power, a motor that is driven by the inverter circuit and drives a stirring blade, and a rotor position of the motor A rotor position detecting means for detecting the motor current, a current detecting means for detecting the motor current, and a control means for controlling the inverter circuit, wherein the control means controls the motor to perform forward / reverse control to execute a stirring process. At the same time, the motor current is decomposed into a current component corresponding to the magnetic flux and a current component corresponding to the torque, and the current component corresponding to the torque and the magnetic flux corresponding to the torque at the time of deceleration braking in the forward reversal control of the stirring blade in the stirring stroke. the set current component to negative respectively, mode of laundry equipment that the current component corresponding to the torque so that Gyosu setting system in accordance with the load Motor drive unit. 4. The motor drive device for a laundry machine according to claim 3 , wherein the control means sets and controls the current component corresponding to the braking torque during the deceleration braking of the motor according to the current corresponding to the torque during the agitation drive.

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