JP3687590B2 - PM motor control method and control apparatus - Google Patents

Description

【００３３】
これを、図７のように、Δθ＝θ−θ_Cだけずれた制御位相ｄ_C，ｑ_C座標に変換すると次の（２）式となる。
【００３４】
【数２】

【００３５】
これを、電流微分を求める状態方程式に変形すると、次の（３）式になる。
【００３６】
【数３】

【００３７】
上記の（３）式に対して、以下のΔＶ_hが振幅で、ω_hが角周波数である（４）式のようなｄｃ軸にのみ単振動の高周波電圧成分を印加することとする。
【００３８】
【数４】

【００３９】
このとき、発生する電流は、（４）式を（３）式に代入すればよく、さらに以下の（５）式、（６）式の近似を行う。
【００４０】
【数５】

【００４１】
この近似により、（３）式の右辺の第２項および第３項は零値となり、以下の（７）式のような電流の状態方程式となる。
【００４２】
【数６】

【００４３】
ここで、近似できる根拠は、（５）式はモータの速度ωが零速度または極低速状態に適用することによる。また、（６）式は印加する高周波電圧の周波数が高いため、インダクタンスの誘起起電力に比べて抵抗Ｒの電圧降下成分は小さいことによる。
【００４４】
この（７）式の両辺を積分すると、以下の（８）式のような電流状態方程式が得られる。
【００４５】
【数７】

【００４６】
上記の（８）式では、ｑｃ軸には電圧成分を印加していないにもかかわらず、ｑｃ軸にも電流成分が発生している。これは、インダクタンスの突極性（Ｌ₂成分）によるものであり、以降にはこれを利用して軸ずれを検出していく。
【００４７】
上記の（８）式の電流には、高調波成分と定常成分が含まれているため、帯域フィルタや１周期の平均値を減算する等の操作により、高調波電流成分だけを抽出することができ、以下の（９）式となる。
【００４８】
【数８】

【００４９】
この高調波電流成分から、角周波数＋ω_hで回転する正相軸と、−ω_hで回転する逆相軸に対する写像を求める。ここで、正相／逆相軸の定義については、時刻ｔ＝０のときの初期位相をｄｃ軸に一致させる成分ｄｆｃ，ｄｒｃと、初期位相をｑｃ軸に一致させる成分ｑｆｃ，ｑｒｃとの２種類が存在する。これらの関係および以下で求める写像成分の関係を図９に示す。
【００５０】
まず、正相軸成分である２種類の成分ｄｆｃ，ｄｒｃについて計算する。正相軸への写像成分は、ω_hｔで回転する回転座標成分と等しく、これに電流式（９）式を代入すると以下の（１０）式となる。
【００５１】
【数９】

【００５２】
次に、逆相軸成分は、−ω_hｔで回転する回転座標成分と等しく、これに電流式（９）式を代入すると以下の（１１）式となる。
【００５３】
【数１０】

【００５４】
上記の（１０）、（１１）式は、三角関数の倍角の公式を利用すると、２ω_hｔという電圧の２倍の角速度で振動する成分となり、以下の（１２）式のようになる。
【００５５】
【数１１】

【００５６】
次に、単振動の正相と逆相の差を利用する。ここで、（２）式のインダクタンスＬ₂に関する突極性の要素のみにするために、前記（１０）式と（１１）式の合成をとることにする。電流成分である（１２）式のままでは複雑であるが、差をとることにより簡単な式になってくる。
【００５７】
ここで、Ｌ₁の項を消去するように合成することとし、ｄ軸を初期位相とする成分については差分を、ｑ軸を初期位相とする成分については加算をとることとする。この差分と加算を取ったものが以下の（１３）式と（１４）式である。
【００５８】
【数１２】

【００５９】
上記の（１３）式は前記の（１０）、（１１）式より、ｄ軸を初期値とする正相軸と逆相軸の写像の差分になる。また、（１４）式は前記の（１０）、（１１）式より、ｑ軸を初期値とする正相軸と逆相軸の写像の加算（合成）になる。
【００６０】
これら正相軸と逆相軸の差分と合成成分をタイムチャートで表すと図１０のようになる。同図より、次のことがわかる。
【００６１】
（差分成分：ｉ_dfc−ｉ_drc）
・０〜π／２の期間を最小単位とする線対称波形（実線）になる。
【００６２】
・Δθ＝０の場合には、振幅は零となる。
【００６３】
・ｃｏｓ（２ω_hｔ）−１の波形は、Ｌ₂・ｓｉｎ（２Δθ）が正極性のとき負のみの波形になるし、逆に、Ｌ₂・ｓｉｎ（２Δθ）が負極性のとき正のみの波形になる。
【００６４】
（合成成分：ｉ_qfc＋ｉ_qrc）
・０〜π／４の期間を最小単位とする点対称波形（破線）になる。
【００６５】
・Δθ＝０の場合には、振幅は零となる。
【００６６】
・ｓｉｎ（２ω_hｔ）の波形は、零を中心とする正弦波波形であるため、π／２毎に符号が反転する。したがって、０〜π／４の期間に区切って、面積とその符号をまとめると、位相ずれ量とその極性が判別可能である。
【００６７】
前記の（１４）式を最小単位である０〜π／４期間に亙って積分すると、以下の（１５)式になる。
【００６８】
【数１３】

【００６９】
この（１５）式より、電流の検出面積からｄｃ軸と磁極軸との位相ずれ量Δθを逆に求めることができ、以下の（１６）式になる。また、同様に、他の７つの期間についても位相ずれ量Δθを求めることができる。
【００７０】
【数１４】

【００７１】
上記の位相ずれ量Δθを求めることができれば、そのままｄｃ軸を修正してもよいし、またノイズなどの要因がある場合には以下の（１７）式のように緩和ゲインｋを乗じて積分動作をさせて収束させてもよい。（１７）式ではπ／２周期で極性が反転するため、符号補正関数Ｓ（ω_hｔ）を乗じている。
【００７２】
【数１５】

【００７３】
上記の（１７）式において、逆三角関数については、｜Δθ｜≪πであると近似すれば、ｓｉｎΔθ≒Δθと近似できる。また、係数部分も緩和ゲインの一部とみなすと、以下の（１８）式のように簡略化することもできる。
【００７４】
【数１６】

【００７５】
また、最小期間毎に限定せずに、位相Δθを逐次修正してもよい。この場合、ω_pを遮断周波数とする低域通過フィルタなどを通して、ゆっくりと修正動作を行わせる必要がある。そこで、以下の（１９）式のように推定を行うこともできる。
【００７６】
【数１７】

【００７７】
このようにして、ｄｃ軸を（１５）式の成分が零となるまで収束させると、ｄｃ軸は磁極軸と一致させることができる。
【００７８】
以上までの説明が、正相軸と逆相軸の写像を利用した本発明による磁極軸の位相推定方式である。
【００７９】
（発明の基本構成）
上記までの原理的な説明から、前記の文献１ではＦＦＴにより電流ベクトル成分を抽出してから計算しているが、本発明では新たに単振動の正相分と逆相分で取り扱うようにしたもので、以下の特徴事項になる。
【００８０】
（１）ｄｃ軸上に正弦波状の高周波成分の電圧（または電流）を重畳する点は文献１と同じである。この入力した単振動状の電圧が磁極軸またはその直交軸と一致していると、電流も同一位相上の単振動となる。磁極軸またはその直交軸に一致していない場合には、高周波電流成分はΔφだけずれた軸上の単振動となる。
【００８１】
（２）本来、単振動成分は、正回転する成分（正相分）と逆回転する成分（逆相分）という振幅は等しいが回転方向の異なる２つの回転ベクトル成分に分離することができる。もし、この入力した単振動成分と磁極軸またはその直交軸とが一致していれば、正相分と逆相分の時間波形は等しい正弦波となる。もし、磁極軸の位相がずれている場合には、正相分と逆相分の時間波形に位相ずれが発生する。したがって、正相／逆相成分の差を利用し、これが零となるように制御すれば位相推定が可能となる。
【００８２】
（３）そこで、正相分と逆相分を求めるために、正相軸と逆相軸に対する高周波電流成分の瞬時ベクトルの写像を利用する。
【００８３】
（４）この２つの軸の写像成分の位相が一致するように高周波注入位相を修正すると、収束した位相が磁極位相またはその垂直位相となる。
【００８４】
以上のことから、本発明の基本構成は、図１に示すようになり、破線ブロックの部分で図６と異なるものである。
【００８５】
図１において、積分器２１と高周波電圧発生部２２によってｄｃ軸に高周波電圧を印加するもので、高周波角速度指令ω_hの積分で高周波位相指令ω_hｔを得、この位相をもち振幅を調整した正弦波状の単振動高周波電圧ΔＶ_hｃｏｓ（ω_hｔ）をｄｃ軸の電圧指令に注入する。
【００８６】
また、高調波抽出部２３、２４によってｄｃ，ｑｃ軸電流ｉ_dc，ｑ_dcに含まれる高周波電流成分を抽出する。
【００８７】
また、正相軸写像演算部２５と逆相軸写像演算部２６は、抽出された高周波電流成分と正相軸と逆相軸に対する写像を求める。
【００８８】
また、正／逆相非対称性抽出部２７は、２つの写像成分の非対称性を特徴量として求める。
【００８９】
また、位相推定演算部２８は、特徴量から推定位相θｃを求める。
【００９０】
（制御方法の発明）
（１）ＰＭモータに単振動正弦波状の高周波電圧または電流を注入し、この高周波電圧または電流の注入によってＰＭモータに流れる高周波電流又は電圧から磁極の位相を推定をするＰＭモータの制御方法において、
前記位相推定は、
前記高周波電流成分又は電圧から正方向に回転する正相軸と逆回転方向に回転する逆相軸に対する写像成分を求め、
前記正相軸の写像成分と逆相軸の写像成分の非対称性を特徴量として求め、
前記特徴量から磁極の位相を推定することを特徴とするＰＭモータの制御方法。
【００９１】
（制御装置の発明）
（２）ＰＭモータに単振動正弦波状の高周波電圧または電流を注入し、この高周波電圧または電流の注入によってＰＭモータに流れる高周波電流又は電圧から磁極の位相を推定するＰＭモータの制御装置において、
前記高周波電流成分又は電圧から正方向に回転する正相軸と逆回転方向に回転する逆相軸に対する写像成分を求め、
前記正相軸の写像成分と逆相軸の写像成分の非対称性を特徴量として求め、
前記特徴量から磁極の位相を推定する位相推定手段を備えたことを特徴とするＰＭモータの制御装置。
【００９２】
（３）ＰＭモータの磁極位置になるｄｃ軸とそれに直交したｑｃ軸に分離した電流制御系と、前記ｄｃ軸の電圧指令に単振動正弦波状の高周波電圧または電流を注入し、この高周波電圧または電流の注入によってＰＭモータに流れる高周波電流又は電圧から磁極の位相を推定する位相推定装置とを備えたＰＭモータの制御装置において、
前記位相推定装置は、
前記高周波電圧または電流の位相指令ω_hｔを基にして前記単振動の高周波電圧または電流を発生する高周波発生手段と、
ＰＭモータの前記ｄｃ軸電流とｑｃ軸電流に含まれる高周波電流成分ｉ_dc，ｉ_qcを抽出する高調波抽出手段と、
前記高周波電流成分ｉ_dc，ｉ_qcから正方向に回転する正相軸と逆回転方向に回転する逆相軸に対する写像成分ｉ_qfc、ｉ_qrcを求める写像抽出手段と、
前記写像成分ｉ_qfc、ｉ_qrcを合成し、高周波位相指令ω_hｔの０〜π／４期間に亙って積分し、係数を乗じて位相ずれ量Δθを求める積分手段と、
前記位相ずれ量Δθから推定位相θｃを求める位相推定演算手段とを備えたことを特徴とするＰＭモータの制御装置。
【００９３】
（４）ＰＭモータの磁極位置になるｄｃ軸とそれに直交したｑｃ軸に分離した電流制御系と、前記ｄｃ軸の電圧指令に単振動正弦波状の高周波電圧または電流を注入し、この高周波電圧または電流の注入によってＰＭモータに流れる高周波電流又は電圧から磁極の位相を推定する位相推定装置とを備えたＰＭモータの制御装置において、
前記位相推定装置は、
前記高周波電圧または電流の位相指令ω_hｔを基にして前記単振動の高周波電圧または電流を発生する高周波発生手段と、
ＰＭモータの前記ｄｃ軸電流とｑｃ軸電流に含まれる高周波電流成分ｉ_dc，ｉ_qcを抽出する高調波抽出手段と、
前記高周波電流成分ｉ_dc，ｉ_qcから正方向に回転する正相軸と逆回転方向に回転する逆相軸に対する写像成分ｉ_dfc、ｉ_drcを求める写像抽出手段と、
前記写像成分ｉ_dfc、ｉ_drcの差分を求め、高周波位相指令ω_hｔの０〜π／２期間に亙って積分又は連続的な積分をし、係数を乗じて位相ずれ量Δθを求める積分手段と、
前記位相ずれ量Δθから推定位相θｃを求める位相推定演算手段とを備えたことを特徴とするＰＭモータの制御装置。
【００９４】
【発明の実施の形態】
（実施形態１）
図２は、本発明の実施形態１を示す構成図であり、図１と同等の部分は同一符号で示す。また、各部には前記の各式を対応付けて示す。
【００９５】
図２において、正相軸写像演算部２９と逆相軸写像演算部３０は、以下のように定義された座標変換行列を使用して写像演算を行い、前記（１０）式の２行目と（１１）式の２行目のｑｃ軸の正相成分と逆相成分を求める。
【００９６】
【数１８】

【００９７】
合成部３１は、写像成分ｉ_qfc，ｉ_qrcの加算（合成）によってｑ軸を初期位相とする前記（１４）式の合成値（ｉ_qfc＋ｉ_qrc）を求める。
【００９８】
積分器３２は、合成値（ｉ_qfc＋ｉ_qrc）を高周波位相指令ω_hｔの０〜π／４期間に亙って積分し、前記（１５）式の積分結果を求め、これに係数を乗じて位相ずれ量Δθを求める。
【００９９】
位相推定演算部３３は、位相ずれΔθに符号補正関数等を乗じて前記の（１７）式または（１８）式の演算で推定位相θ_Cを求める。
【０１００】
本実施形態において、図１０の特性は、前記の（４）式のようにｄｃ軸に余弦波波形の高周波電圧を印加することにより得ている。仮に、入力軸を変更した余弦波を正弦波に変更すると、正相と逆相の写像の合成波形は発生する軸や極性が異なってくるが、基本的にはｃｏｓ（２ω_hｔ）−１とｓｉｎ（２ω_hｔ）の波形の組み合わせに限定される。そのため、入力位相や電圧形状によって、写像の選択を変更するだけで、同じ制御アルゴリズムを使用できる。
【０１０１】
（実施形態２）
図３は、本発明の実施形態２を示す構成図である。同図が図２と異なる部分を以下に説明する。
【０１０２】
正相軸写像演算部３４と逆相軸写像演算部３５は、前記の（１０）式の１行目と（１１）式の１行目のｄｃ軸の正相成分と逆相成分を求める。
【０１０３】
差分演算部３６は、写像成分ｉ_dfc，ｉ_drcの差分演算によって前記（１３）式の差分（ｉ_dfc−ｉ_drc）を求める。
【０１０４】
積分器３７は、差分（ｉ_dfc−ｉ_drc）を高周波位相指令ω_hｔの０〜π／２期間に亙って積分し、差分成分による積分結果を求め、これに係数を乗じて位相ずれ量Δθを求める。
【０１０５】
位相推定演算部３８は、位相ずれΔθに符号補正関数等を乗じて前記の（１７）式〜（１９）式のいずれかの演算で推定位相θ_Cを求める。
【０１０６】
前記の実施形態１では、正相軸と逆相軸の写像の非対称性を求めるのにｑ軸を初期位相とする（１４）式を利用している。本実施形態では、ｄ軸を初期位相とする（１３）式を利用した方式とする。
【０１０７】
また、積分器３７は、π／２区間の積分器としているが、（ｉ_dfc−ｉ_drc）は脈動するものの、極性は同じであるため、連続的な積分器でもよい。
【０１０８】
【発明の効果】
以上のとおり、本発明によれば、以下の効果がある。
【０１０９】
（１）文献１では、ＦＦＴを適用しているため、入力高周波電圧成分のπ期間の電流検出データが必要であったが、本発明では最小でπ／４期間でよく、４倍高速に位相推定が実行できる。
【０１１０】
（２）文献２では電流微分を使用しているが、本発明は電流微分を使用していない。逆に、電流の写像成分を一定期間積分している。そのため、文献２に比べて、強いノイズの抑制効果が得られる。
【図面の簡単な説明】
【図１】本発明の基本構成図。
【図２】本発明の実施形態１を示すブロック図。
【図３】本発明の実施形態２を示すブロック図。
【図４】従来のＰＭモータの制御方式のブロック図。
【図５】従来のＰＭモータの位置センサレス制御方式のブロック図。
【図６】従来の文献１のブロック図。
【図７】ＰＭモータの高調波電圧と磁極位置関係の図。
【図８】従来の文献２のブロック図。
【図９】本発明に係る正相軸と逆相軸への写像成分の図。
【図１０】本発明に係る高周波成分の電圧成分と、正相軸と逆相軸の差分との関係図。
【符号の説明】
１…速度制御器
４…逆回転座標変換器
５…電力変換器
６…ＰＭモータ
８…回転座標変換器
１０…速度検出演算部
１２Ａ、１２Ｂ…電流制御器
２１…積分器
２２…高周波発生部
２３、２４…高調波抽出部
２５…正相軸写像抽出部
２６…逆相軸写像抽出部
２７…非対称性抽出部
２８、３３、３８…位相推定演算部
２９…正相軸写像演算部
３０…逆相軸写像演算部
３１…合成部
３２、３７…積分器
３６…差分演算部[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a PM motor control device that controls the speed and torque of a PM motor using a permanent magnet as a field source by a variable speed drive device such as an inverter.
[0002]
[Prior art]
A synchronous machine using a permanent magnet as a field source incorporates a powerful damper winding (equivalent to the cage conductor of an inductor) on the field side, and can be started by directly turning on the commercial power supply. There are two types, namely, there is no damper winding, and the generated torque and stabilization are controlled by controlling the voltage and current by a power converter such as an inverter.
[0003]
The present invention controls a type of PM motor that does not have a damper or that has a weak damper function and cannot be started directly.
[0004]
In order to control such a PM motor, it is necessary to detect the position of the magnetic pole and pass a current corresponding to the magnetic pole. Therefore, the position is generally detected by attaching a position sensor to the rotating shaft. FIG. 4 shows an example of the configuration.
[0005]
The figure shows an example in which control is performed on a speed command, and a torque command is obtained from the speed control unit 1 by comparing the speed command with a speed detection value. The current command calculation unit 2 converts the torque command into a current command by using information such as field magnetic flux and inductance. The current control unit 3 obtains a voltage command by comparison with the detected current value, gives a voltage control signal to the power converter (inverter) 5 by coordinate conversion by the reverse rotation coordinate conversion unit 4, and outputs an armature current to the PM motor 6. Supply. The current at this time is detected by the current detector 7, and a detected current signal is given to the current control unit 3 by coordinate conversion by the rotating coordinate conversion unit 8.
[0006]
In this current control system, since the current command is calculated with reference to the magnetic pole phase, the AC current obtained from the current detector 7 is converted to the same magnetic pole phase as the current command by using the position information in the coordinate converter 8. Convert to reference rotation coordinates. After performing current control on this coordinate, the output voltage is reversely rotated again by the coordinate conversion unit 4 to give an AC voltage command to the power converter 5, and finally the PM motor 6 is driven.
[0007]
The position detector 9 detects the magnetic pole position of the PM motor 6, obtains a speed detection signal from the speed detection calculation unit 10, and gives it to the speed control unit 1. Further, the position detection signal of the position detector 9 is taken into the coordinate conversion units 4 and 8 as position information.
[0008]
However, the position detector 9 has a built-in electronic circuit, and has problems such as low environmental resistance and high price.
[0009]
Therefore, in addition to a method using such a position detector, there is a position sensorless control method in which a magnetic flux is estimated from an output voltage or voltage command and current detection information to estimate the phase of the magnetic pole. An example of the configuration is shown in FIG. 5, and the position estimation calculation unit 11 estimates the position from the voltage command and the current detection signal.
[0010]
In this configuration example, voltage information is required. More strictly, the speed electromotive force by the permanent magnet needs to be included therein.
[0011]
However, since the rotational speed is zero at the time of starting, the important speed electromotive force is not generated. Therefore, a method of estimating the position by flowing a high frequency or a pulse current at the time of starting or measuring an inductance change of a synchronous machine having a saliency by applying a high frequency voltage has been proposed.
[0012]
In a synchronous machine using a permanent magnet, a field pole having a high magnetic permeability is composed of a material such as a silicon steel plate and a permanent magnet having a low magnetic permeability. Therefore, a magnetic pole axis (d axis) and an axis orthogonal to it (q axis) ), A difference occurs due to the asymmetry of the shape. The position is estimated using the difference in inductance.
[0013]
Such a method is called a pulse application method, a high-frequency application method, a high-frequency superposition method, or the like. As a method for injecting harmonics, there are the following documents.
[0014]
Reference 1: Takashi Aihara, Akio Toba, Takao Yanase, “Zero-speed torque control of salient-pole synchronous motor by sensorless method”, 1996 Annual Conference of the Institute of Electrical Engineers of Japan, N0.170, and related proposals There is document 2: Japanese Patent Application No. 6-550255 (Japanese Patent Laid-Open No. 7-245981).
[0015]
According to Document 1, this method has a configuration as shown in FIG. However, the terms and symbols are modified as defined in the present invention. Here, the current control system is represented by an example in which two controllers 12A and 12B are configured for a dc axis that is a magnetic pole position estimated by the control system and a qc axis component orthogonal thereto.
[0016]
The details of FIG. 6 are described in Reference 1, but the feature is that the high frequency component is analyzed by FFT to obtain the dc and qc axis components, and the magnetic pole deviation angle Δθ is estimated from that, as shown in FIG. Is briefly described below.
[0017]
(1) A sinusoidal high frequency voltage v _h is superimposed on the output of the current control system on the dc axis that is the magnetic pole estimation axis of the control coordinates.
[0018]
Here, when the magnetic pole axis d of the motor is deviated from the control estimation axis dc by Δθ, the high frequency component i of the current is caused by the difference (saliency) between the inductance components L _d and L _q of the motor d and q axes. _The locus of _h is located on a straight line shifted by Δφ.
[0019]
(2) A high frequency component i _h synchronized with a sinusoidal high frequency voltage is extracted from the detected current by an FFT (Fast Fourier Transform) 13, and Δφ is calculated by the phase error calculation unit 14.
[0020]
(3) The information of the current deviation angle Δφ is integrated by the integrator 15 to correct the dc axis (θ) so that Δφ = 0. Thereby, after convergence, an estimated phase θ _{C in} which the dc axis coincides with the magnetic pole axis can be obtained. This estimated phase θ _C is given as a phase signal to the coordinate conversion units 4 and 8, and the speed detection calculation unit 10 obtains it as a speed detection signal by speed detection by difference calculation and high-frequency component removal by a low-pass filter.
[0021]
The advantages of the above method are as follows.
[0022]
・ Position can be estimated even when stopped (zero speed).
・ High-frequency voltage components are input only in approximately the same phase as the magnet axis, so torque ripple due to high-frequency current does not occur.
[0023]
Next, in Document 2 (Japanese Patent Laid-Open No. 7-245981), instead of FFT calculation, a method using current differentiation after extracting a high-frequency current is adopted. Part of this embodiment is shown in FIG. 8 with the speed control system omitted. The characteristics of this method will be described below.
[0024]
(1) Similar to Document 1, a high-frequency voltage is superimposed on a magnetic pole estimation axis (dc axis) for control.
[0025]
(2) A high frequency current component synchronized with the high frequency voltage is extracted from the current detection using the notch filters 16A and 16B.
[0026]
(3) Differentiate the high-frequency current component by the differential operation unit 17 to estimate the inductance component equivalent, and estimate the magnetic pole phase.
[0027]
(4) As a waveform, a square wave and a triangular wave are extended in addition to a sine wave, and the input is extended to voltage superposition and current command superposition.
[0028]
[Problems to be solved by the invention]
(First issue)
The current detection in FIG. 6 includes a fundamental wave component necessary for driving the motor and a high frequency component necessary for magnetic pole estimation. Of these, in order to separate only the high frequency components necessary for magnetic pole estimation, an FFT algorithm is used. However, in order to execute FFT, data of one period or more is necessary, and the data detection period is limited for each high frequency period.
[0029]
(Second problem)
Document 2 is a method that uses a differential amount of current without using FFT, and at least, position estimation can be executed with two samples. However, in current detection, noise is likely to be mixed into the detector due to switching of the main circuit element generated for performing PWM modulation. Therefore, the method using current differentiation has a problem that it is vulnerable to noise.
[0030]
It is an object of the present invention to provide a PM motor control method and control apparatus that can perform phase estimation at high speed without using FFT and reduce an estimated phase error with respect to noise.
[0031]
[Means for Solving the Problems]
(Principle description of the invention)
In the dq orthogonal biaxial coordinate system rotating in synchronization with the motor magnetic pole axis in FIG. 7, the current-current equation of the permanent magnet type synchronous machine including the case where the inductance of the magnetic pole and that orthogonal axis is different is the following equation (1): Become.
[0032]
[Expression 1]

[0033]
When this is converted into control phases d _C and q _C coordinates shifted by Δθ = θ−θ _C as shown in FIG. 7, the following equation (2) is obtained.
[0034]
[Expression 2]

[0035]
When this is transformed into a state equation for obtaining a current derivative, the following equation (3) is obtained.
[0036]
[Equation 3]

[0037]
For the above equation (3), a high frequency voltage component of simple vibration is applied only to the dc axis as in equation (4) where ΔV _h is amplitude and ω _h is angular frequency.
[0038]
[Expression 4]

[0039]
At this time, the generated current may be obtained by substituting the equation (4) into the equation (3), and the following equations (5) and (6) are approximated.
[0040]
[Equation 5]

[0041]
By this approximation, the second term and the third term on the right side of the equation (3) become zero values, and the current state equation as the following equation (7) is obtained.
[0042]
[Formula 6]

[0043]
Here, the reason for the approximation is that equation (5) is applied when the motor speed ω is zero or extremely low. Further, the expression (6) is because the voltage drop component of the resistor R is smaller than the induced electromotive force of the inductance because the frequency of the applied high frequency voltage is high.
[0044]
By integrating both sides of the equation (7), a current state equation such as the following equation (8) is obtained.
[0045]
[Expression 7]

[0046]
In the above equation (8), although no voltage component is applied to the qc axis, a current component is also generated on the qc axis. This is due to the inductance of the salient pole (L ₂ component), it will detect the axial displacement by use of this later.
[0047]
Since the current of the above formula (8) includes a harmonic component and a steady component, it is possible to extract only the harmonic current component by an operation such as subtracting the average value of a band filter or one period. The following equation (9) can be obtained.
[0048]
[Equation 8]

[0049]
From this harmonic current component, a mapping is obtained for a positive phase axis rotating at an angular frequency + ω _h and a negative phase axis rotating at −ω _h . Here, with respect to the definition of the normal phase / reverse phase axis, the components dfc and drc that make the initial phase coincident with the dc axis at time t = 0 and the components qfc and qrc that make the initial phase coincide with the qc axis are two. There are types. FIG. 9 shows these relationships and the relationship between the mapping components obtained below.
[0050]
First, two types of components dfc and drc that are positive phase axis components are calculated. The mapping component to the positive phase axis is equal to the rotating coordinate component rotating at ω _h t, and the following equation (10) is obtained by substituting the current equation (9) into this.
[0051]
[Equation 9]

[0052]
Next, the negative phase axis component is equal to the rotational coordinate component rotating at −ω _h t, and the following equation (11) is obtained by substituting the current equation (9) into this.
[0053]
[Expression 10]

[0054]
The above equations (10) and (11) are components that oscillate at an angular velocity twice as high as a voltage of 2ω _h t when the double angle formula of the trigonometric function is used, and is represented by the following equation (12).
[0055]
[Expression 11]

[0056]
Next, the difference between the positive and negative phases of simple vibration is used. Here, in order to make only the element of the saliency related to the inductance L _{2 in the} equation (2), the above equations (10) and (11) are combined. Although the equation (12) that is the current component is complicated as it is, it becomes a simple equation by taking the difference.
[0057]
Here, the synthesis is performed so as to eliminate the term of L ₁ , and the difference is taken for the component having the d-axis as the initial phase, and the addition is taken for the component having the q-axis as the initial phase. The following equations (13) and (14) are obtained by taking the difference and addition.
[0058]
[Expression 12]

[0059]
The above equation (13) is the difference between the positive phase axis and the negative phase axis mapping with the d axis as the initial value from the above equations (10) and (11). Further, the equation (14) is an addition (combination) of the mapping of the positive phase axis and the negative phase axis with the q axis as an initial value from the above equations (10) and (11).
[0060]
FIG. 10 shows the difference between the normal phase axis and the negative phase axis and the synthesized component in a time chart. The figure shows the following.
[0061]
(Difference component: i _dfc −i _drc )
A line-symmetric waveform (solid line) having a period of 0 to π / 2 as a minimum unit.
[0062]
• When Δθ = 0, the amplitude is zero.
[0063]
The waveform of cos (2ω _h t) −1 is only a negative waveform when L ₂ · sin (2Δθ) is positive, and conversely, it is only positive when L ₂ · sin (2Δθ) is negative. It becomes the waveform.
[0064]
(Composite component: i _qfc + i _qrc )
A point-symmetric waveform (broken line) having a period of 0 to π / 4 as a minimum unit.
[0065]
• When Δθ = 0, the amplitude is zero.
[0066]
Since the waveform of sin (2ω _h t) is a sine wave waveform centered on zero, the sign is inverted every π / 2. Therefore, the phase shift amount and its polarity can be discriminated by dividing the area and its sign by dividing into periods of 0 to π / 4.
[0067]
When the above equation (14) is integrated over the minimum unit of 0 to π / 4 period, the following equation (15) is obtained.
[0068]
[Formula 13]

[0069]
From this equation (15), the phase shift amount Δθ between the dc axis and the magnetic pole axis can be obtained in reverse from the current detection area, and the following equation (16) is obtained. Similarly, the phase shift amount Δθ can be obtained for the other seven periods.
[0070]
[Expression 14]

[0071]
If the above-mentioned phase shift amount Δθ can be obtained, the dc axis may be corrected as it is, and if there is a factor such as noise, the integration operation is performed by multiplying the relaxation gain k as shown in the following equation (17). May be converged. In the equation (17), since the polarity is inverted every π / 2 period, the sign correction function S (ω _h t) is multiplied.
[0072]
[Expression 15]

[0073]
In the above equation (17), if the inverse trigonometric function is approximated as | Δθ | << π, it can be approximated as sin Δθ≈Δθ. If the coefficient portion is also considered as a part of the relaxation gain, it can be simplified as in the following equation (18).
[0074]
[Expression 16]

[0075]
Further, the phase Δθ may be corrected sequentially without being limited to every minimum period. In this case, the correction operation needs to be performed slowly through a low-pass filter having ω _p as a cutoff frequency. Therefore, estimation can also be performed as in the following equation (19).
[0076]
[Expression 17]

[0077]
In this way, when the dc axis is converged until the component of equation (15) becomes zero, the dc axis can be made to coincide with the magnetic pole axis.
[0078]
The above explanation is the phase estimation method of the magnetic pole axis according to the present invention using the mapping of the positive phase axis and the negative phase axis.
[0079]
(Basic configuration of the invention)
From the above principle description, in the above-mentioned document 1, the calculation is performed after extracting the current vector component by FFT, but in the present invention, it is newly dealt with the normal phase component and the reverse phase component of the single vibration. It becomes the following characteristic matters.
[0080]
(1) The point of superimposing a voltage (or current) of a sinusoidal high frequency component on the dc axis is the same as in Document 1. When the input single vibration voltage coincides with the magnetic pole axis or its orthogonal axis, the current also becomes a single vibration on the same phase. When it does not coincide with the magnetic pole axis or its orthogonal axis, the high-frequency current component is a single vibration on the axis shifted by Δφ.
[0081]
(2) Basically, the simple vibration component can be separated into two rotation vector components having the same amplitude but different rotation directions, ie, a component that rotates in the forward direction (for the positive phase) and a component that rotates in the reverse direction (for the reverse phase). If the input single vibration component matches the magnetic pole axis or its orthogonal axis, the time waveform for the positive phase and the reverse phase are equal sine waves. If the phase of the magnetic pole axis is shifted, a phase shift occurs in the time waveform for the positive phase and the reverse phase. Therefore, the phase can be estimated by using the difference between the positive phase / negative phase components and controlling the difference to be zero.
[0082]
(3) Therefore, in order to obtain the positive phase component and the negative phase component, the instantaneous vector mapping of the high-frequency current component with respect to the positive phase axis and the negative phase axis is used.
[0083]
(4) When the high-frequency injection phase is corrected so that the phases of the mapping components of the two axes coincide, the converged phase becomes the magnetic pole phase or its perpendicular phase.
[0084]
From the above, the basic configuration of the present invention is as shown in FIG. 1, and is different from FIG.
[0085]
In FIG. 1, a high frequency voltage is applied to the dc axis by an integrator 21 and a high frequency voltage generator 22, and a high frequency phase command ω _h t is obtained by integrating the high frequency angular velocity command ω _h , and the amplitude is adjusted using this phase. A sinusoidal single vibration high-frequency voltage ΔV _h cos (ω _h t) is injected into the voltage command of the dc axis.
[0086]
Further, the high frequency current components included in the dc and qc axis currents i _dc and q _dc are extracted by the harmonic extraction units 23 and 24.
[0087]
Further, the positive phase axis mapping calculation unit 25 and the negative phase axis mapping calculation unit 26 obtain a mapping of the extracted high-frequency current component, the positive phase axis, and the negative phase axis.
[0088]
Further, the normal / reverse phase asymmetry extraction unit 27 obtains the asymmetry of the two mapping components as a feature amount.
[0089]
Further, the phase estimation calculation unit 28 obtains the estimated phase θc from the feature amount.
[0090]
(Invention of control method)
(1) In a PM motor control method for injecting a single vibration sinusoidal high-frequency voltage or current into a PM motor and estimating the phase of the magnetic pole from the high-frequency current or voltage flowing in the PM motor by the injection of the high-frequency voltage or current.
The phase estimation is
Determine the mapping component for the positive phase axis rotating in the positive direction and the negative phase axis rotating in the reverse rotation direction from the high-frequency current component or voltage,
Obtain the asymmetry of the mapping component of the normal phase axis and the mapping component of the negative phase axis as a feature amount,
A PM motor control method, wherein the phase of a magnetic pole is estimated from the feature amount.
[0091]
(Invention of control device)
(2) In a PM motor control device that injects a single-vibration sinusoidal high-frequency voltage or current into a PM motor and estimates the phase of a magnetic pole from the high-frequency current or voltage flowing through the PM motor by the injection of the high-frequency voltage or current.
Determine the mapping component for the positive phase axis rotating in the positive direction and the negative phase axis rotating in the reverse rotation direction from the high-frequency current component or voltage,
Obtain the asymmetry of the mapping component of the normal phase axis and the mapping component of the negative phase axis as a feature amount,
A PM motor control apparatus comprising phase estimation means for estimating the phase of a magnetic pole from the feature quantity.
[0092]
(3) A current control system separated into a dc axis which is a magnetic pole position of the PM motor and a qc axis orthogonal thereto, and a single vibration sinusoidal high frequency voltage or current is injected into the voltage command of the dc axis. In a PM motor control device comprising a phase estimation device that estimates the phase of a magnetic pole from a high-frequency current or voltage flowing in the PM motor by current injection,
The phase estimation device includes:
High-frequency generating means for generating the single-vibration high-frequency voltage or current based on the phase command ω _h t of the high-frequency voltage or current;
Harmonic extraction means for extracting high-frequency current components i _dc and i _qc included in the dc-axis current and qc-axis current of the PM motor;
Mapping extraction means for _obtaining mapping components i _qfc and i _qrc from the high-frequency current components i _dc and i _qc to a normal phase axis rotating in the positive direction and a negative phase axis rotating in the reverse rotation direction;
The mapping component i _QFC, to synthesize i _qrc, integrated over the 0~π / 4 period of the high frequency phase command omega _h t, and integrating means for obtaining a phase shift amount Δθ by multiplying the coefficients,
A PM motor control apparatus comprising phase estimation calculation means for obtaining an estimated phase θc from the phase shift amount Δθ.
[0093]
(4) A current control system separated into a dc axis which is a magnetic pole position of the PM motor and a qc axis orthogonal thereto, and a single vibration sinusoidal high frequency voltage or current is injected into the voltage command of the dc axis. In a PM motor control device comprising a phase estimation device that estimates the phase of a magnetic pole from a high-frequency current or voltage flowing in the PM motor by current injection,
The phase estimation device includes:
High-frequency generating means for generating the single-vibration high-frequency voltage or current based on the phase command ω _h t of the high-frequency voltage or current;
Harmonic extraction means for extracting high-frequency current components i _dc and i _qc included in the dc-axis current and qc-axis current of the PM motor;
Mapping extraction means for _obtaining mapping components i _dfc and i _drc for the positive phase axis rotating in the positive direction and the negative phase axis rotating in the reverse rotation direction from the high-frequency current components i _dc and i _qc ;
_Integration to obtain the difference _between the mapping components i _dfc and i _drc , to perform integration or continuous integration over the period of 0 to π / 2 of the high-frequency phase command ω _h t, and to multiply the coefficient to obtain the phase shift amount Δθ Means,
A PM motor control apparatus comprising phase estimation calculation means for obtaining an estimated phase θc from the phase shift amount Δθ.
[0094]
DETAILED DESCRIPTION OF THE INVENTION
(Embodiment 1)
FIG. 2 is a block diagram showing Embodiment 1 of the present invention, and parts equivalent to those in FIG. Also, each part is shown in association with each of the above expressions.
[0095]
In FIG. 2, the normal phase axis mapping calculation unit 29 and the antiphase axis mapping calculation unit 30 perform mapping calculation using a coordinate transformation matrix defined as follows, and the second row of the above equation (10) The positive phase component and the negative phase component of the qc axis in the second row of the equation (11) are obtained.
[0096]
[Expression 18]

[0097]
The synthesizing unit 31 _obtains a synthesized value (i _qfc + i _qrc ) of the equation (14) having the q axis as an initial phase by adding (synthesizing) the mapping components i _qfc and i _qrc .
[0098]
The integrator 32 integrates the composite value (i _qfc + i _qrc ) over the period of 0 to π / 4 of the high-frequency phase command ω _h t, obtains the integration result of the equation (15), and multiplies this by the coefficient. To obtain the phase shift amount Δθ.
[0099]
The phase estimation calculation unit 33 multiplies the phase shift Δθ by a sign correction function or the like to obtain the estimated phase θ _C by the calculation of the above equation (17) or (18).
[0100]
In the present embodiment, the characteristics shown in FIG. 10 are obtained by applying a high-frequency voltage having a cosine wave waveform to the dc axis as shown in the above equation (4). If the cosine wave whose input axis has been changed is changed to a sine wave, the composite waveform of the normal phase and reverse phase maps will have different axes and polarities, but basically cos (2ω _h t) −1. And a combination of sin (2ω _h t) waveforms. Therefore, the same control algorithm can be used simply by changing the mapping selection depending on the input phase and voltage shape.
[0101]
(Embodiment 2)
FIG. 3 is a block diagram showing Embodiment 2 of the present invention. The difference between FIG. 2 and FIG. 2 will be described below.
[0102]
The positive phase axis mapping calculation unit 34 and the negative phase axis mapping calculation unit 35 obtain the positive phase component and the negative phase component of the dc axis in the first row of the equation (10) and the first row of the equation (11).
[0103]
Difference calculation unit 36 calculates the equation (13) of the difference (i _dfc -i _drc) mapping component i _dfc, the difference operation i _drc.
[0104]
The integrator 37 integrates the difference (i _dfc −i _drc ) over the period of 0 to π / 2 of the high-frequency phase command ω _h t, _obtains an integration result by the difference component, and multiplies this by a coefficient to shift the phase. The quantity Δθ is determined.
[0105]
The phase estimation calculation unit 38 multiplies the phase shift Δθ by a sign correction function or the like to obtain the estimated phase θ _C by any one of the expressions (17) to (19).
[0106]
In the first embodiment, the equation (14) with the q axis as the initial phase is used to obtain the asymmetry of the mapping between the positive phase axis and the negative phase axis. In this embodiment, a method using the equation (13) with the d-axis as the initial phase is used.
[0107]
Further, the integrator 37 is an integrator in a π / 2 section, but although (i _dfc -i _drc ) pulsates, the polarity is the same, so a continuous integrator may be used.
[0108]
【The invention's effect】
As described above, the present invention has the following effects.
[0109]
(1) In Reference 1, since FFT is applied, current detection data for the π period of the input high-frequency voltage component is necessary. However, in the present invention, a minimum π / 4 period is sufficient, and the phase is four times faster. Estimation can be performed.
[0110]
(2) Although current differentiation is used in Document 2, the present invention does not use current differentiation. Conversely, the current mapping component is integrated for a certain period. Therefore, compared to Document 2, a stronger noise suppression effect can be obtained.
[Brief description of the drawings]
FIG. 1 is a basic configuration diagram of the present invention.
FIG. 2 is a block diagram showing Embodiment 1 of the present invention.
FIG. 3 is a block diagram showing Embodiment 2 of the present invention.
FIG. 4 is a block diagram of a conventional PM motor control system.
FIG. 5 is a block diagram of a conventional PM motor position sensorless control system.
FIG. 6 is a block diagram of Reference 1 in the related art.
FIG. 7 is a diagram showing the relationship between the harmonic voltage of the PM motor and the magnetic pole position.
FIG. 8 is a block diagram of the conventional document 2.
FIG. 9 is a diagram of mapping components on the positive phase axis and the negative phase axis according to the present invention.
FIG. 10 is a diagram showing the relationship between the voltage component of the high frequency component and the difference between the positive phase axis and the negative phase axis according to the present invention.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 ... Speed controller 4 ... Reverse rotation coordinate converter 5 ... Power converter 6 ... PM motor 8 ... Rotation coordinate converter 10 ... Speed detection calculating part 12A, 12B ... Current controller 21 ... Integrator 22 ... High frequency generation part 23 , 24 ... harmonic extraction unit 25 ... normal phase axis mapping extraction unit 26 ... reverse phase axis mapping extraction unit 27 ... asymmetry extraction units 28, 33, 38 ... phase estimation calculation unit 29 ... normal phase axis mapping calculation unit 30 ... reverse Phase axis mapping calculation unit 31... Synthesis unit 32, 37.

Claims (4)

In a PM motor control method for injecting a single vibration sinusoidal high-frequency voltage or current into a PM motor and estimating the phase of the magnetic pole from the high-frequency current or voltage flowing in the PM motor by the injection of the high-frequency voltage or current.
The phase estimation is
Determine the mapping component for the positive phase axis rotating in the positive direction and the negative phase axis rotating in the reverse rotation direction from the high-frequency current component or voltage,
Obtain the asymmetry of the mapping component of the normal phase axis and the mapping component of the negative phase axis as a feature amount,
A PM motor control method, wherein the phase of a magnetic pole is estimated from the feature amount. In a PM motor control device that injects a single-oscillation sinusoidal high-frequency voltage or current into a PM motor and estimates the phase of a magnetic pole from the high-frequency current or voltage flowing through the PM motor by the injection of the high-frequency voltage or current.
Determine the mapping component for the positive phase axis rotating in the positive direction and the negative phase axis rotating in the reverse rotation direction from the high-frequency current component or voltage,
Obtain the asymmetry of the mapping component of the normal phase axis and the mapping component of the negative phase axis as a feature amount,
A PM motor control apparatus comprising phase estimation means for estimating the phase of a magnetic pole from the feature quantity. A current control system separated into a dc axis that is a magnetic pole position of the PM motor and a qc axis that is orthogonal to the dc axis, and a high-frequency voltage or current in the form of a single oscillating sinusoidal wave is injected into the voltage command of the dc axis. In a PM motor control device comprising a phase estimation device for estimating the phase of a magnetic pole from a high-frequency current or voltage flowing in the PM motor by
The phase estimation device includes:
High-frequency generating means for generating the single-vibration high-frequency voltage or current based on the phase command ω _h t of the high-frequency voltage or current;
Harmonic extraction means for extracting high-frequency current components i _dc and i _qc included in the dc-axis current and qc-axis current of the PM motor;
Mapping extraction means for _obtaining mapping components i _qfc and i _qrc from the high-frequency current components i _dc and i _qc to a normal phase axis rotating in the positive direction and a negative phase axis rotating in the reverse rotation direction;
The mapping component i _QFC, to synthesize i _qrc, integrated over the 0~π / 4 period of the high frequency phase command omega _h t, and integrating means for obtaining a phase shift amount Δθ by multiplying the coefficients,
A PM motor control apparatus comprising phase estimation calculation means for obtaining an estimated phase θc from the phase shift amount Δθ. A current control system separated into a dc axis that is a magnetic pole position of the PM motor and a qc axis that is orthogonal to the dc axis, and a high-frequency voltage or current in the form of a single oscillating sinusoidal wave is injected into the voltage command of the dc axis. In a PM motor control device comprising a phase estimation device for estimating the phase of a magnetic pole from a high-frequency current or voltage flowing in the PM motor by
The phase estimation device includes:
High-frequency generating means for generating the single-vibration high-frequency voltage or current based on the phase command ω _h t of the high-frequency voltage or current;
Harmonic extraction means for extracting high-frequency current components i _dc and i _qc included in the dc-axis current and qc-axis current of the PM motor;
Mapping extraction means for _obtaining mapping components i _dfc and i _drc for the positive phase axis rotating in the positive direction and the negative phase axis rotating in the reverse rotation direction from the high-frequency current components i _dc and i _qc ;
The difference _between the mapping components i _dfc and i _drc is obtained, integrated over a period of 0 to π / 2 of the high-frequency phase command ω _h t, or continuously integrated, and the phase shift amount Δθ is obtained by multiplying the coefficient. Integration means;
A PM motor control apparatus comprising phase estimation calculation means for obtaining an estimated phase θc from the phase shift amount Δθ.

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