JP3633715B2 - Digital wireless transmission system - Google Patents

Description

【００２９】
の自乗の和の平方を包絡信号とし、一の情報シンボル期間とそれに続く情報シンボル期間の時間差または、該一の情報シンボル期間を分割した複数のサブシンボル期間の時間差に対応する包絡信号の差分を振幅差信号として復調することを特徴とする。
【００３０】
更に、本発明の目的を達成するデジタル無線伝送方式の第五の構成は、キャリア信号を所定位相に一定期間保ち、情報を伝送するデジタル無線伝送方式において、時刻ｔにおける受信信号ｆ（ｔ） を実部
【００３１】
【数１７】

【００３２】
とし、該受信信号ｆ（ｔ） のヒルベルト変換信号
【００３３】
【数１８】

【００３４】
を虚部とする解析信号
【００３５】
【数１９】

【００３６】
と、時刻ｔ−Ｔ における解析信号
【００３７】
【数２０】

【００３８】
の複素共役
【００３９】
【数２１】

【００４０】
との積ｇ（ｔ） ｇ’（ｔ−Ｔ）の実部
【００４１】
【数２２】

【００４２】
を復調信号Ｉ（ｔ） とし、虚部
【００４３】
【数２３】

【００４４】
をＱ（ｔ） とすることを特徴とする。
【００４５】
また、本発明の目的を達成するデジタル無線伝送方式の第六の構成は、キャリア信号を所定位相に一定期間保ち、情報を伝送するデジタル無線伝送方式において、時刻ｔにおける受信信号ｆ（ｔ） を実部
【００４６】
【数２４】

【００４７】
とし、該受信信号ｆ（ｔ） のヒルベルト変換信号
【００４８】
【数２５】

【００４９】
を虚部とする解析信号
【００５０】
【数２６】

【００５１】
と、時刻ｔ−Ｔ における解析信号
【００５２】
【数２７】

【００５３】
の複素共役
【００５４】
【数２８】

【００５５】
との積ｇ（ｔ） ｇ’（ｔ−Ｔ）を、時刻ｔにおける包絡Ａ（ｔ） ならびにｔ−Ｔ における包絡Ａ（ｔ−Ｔ） で正規化した信号［ｇ（ｔ） ｇ’（ｔ−Ｔ）／Ａ（ｔ） Ａ（ｔ−Ｔ） ］の実部
【００５６】
【数２９】

【００５７】
を復調信号Ｉ（ｔ） とし虚部
【００５８】
【数３０】

【００５９】
をＱ（ｔ）とすることを特徴とする。
【００６０】
【発明の実施の形態】
以下本発明の実施の形態を図面にしたがって説明する。尚、図において同一または、類似のものには、同一の参照番号または、参照記号を付して説明する。
【００６１】
本発明のデジタル無線伝送方式は、ＰＳＫ伝送において、シンボル区間の位相差と振幅差に情報を載せるようにしたことを特徴とする。
【００６２】
さらに本発明の実施の形態として、振幅に情報を載せる方法として、シンボル内の差分を用いる方法と、シンボル間の差分を用いる方法を提案している。以下それぞれの方法について、本発明の特徴を説明する。
【００６３】
［シンボル内差分法］
先に説明したように、フェーディング等で受ける振幅歪みがシンボル内で準定常であるので、一部あるいは全てのシンボルに図１に示す様に２個以上に分割されたサブシンボルを定義する時、それぞれの振幅変動が略等しくなる。
【００６４】
即ち、図１は、１シンボル区間に２つのサブシンボルｓｓ１、ｓｓ２を設定した例であり、少なくとも隣接するサブシンボルｓｓ１とｓｓ２の相対的な振幅値はフェーディングを受けても変化しない。
【００６５】
例えば、フェーディングにより受信波のシンボル振幅が変動し、基準値より大きくなった場合、サブシンボル振幅もそれぞれ変動し、相対的な振幅関係は保全される。
【００６６】
したがって、サブシンボルｓｓ１とｓｓ２の振幅差に図２に示すように情報を与え即ち、変調して送信して、対応して現れる振幅値の大きさの相対的な変動を受信波の包絡線から測定する事で受信側では、送信情報を受信することが検出できる。
【００６７】
図２では、第一のシンボル区間のサブシンボルｓｓ１の振幅に対し、サブシンボルｓｓ２の振幅を相対的に大きくし、この時データ「１」を表わす。また第二のシンボル区間では、サブシンボルｓｓ１の振幅に対し、サブシンボルｓｓ２の振幅を相対的に小さくし、データ「０」を表わす。このようにサブシンボルの相対的な変化により情報を送ることが可能である。
【００６８】
尚、振幅値を変化させている時、同時に位相に情報を載せて、通信することができる。この場合は、位相検出に高度な受信技術が要求させるが、たとえば後に簡単に説明するように解析受信の技術により対応が可能である。
【００６９】
さらに上記のサブシンボルの相対的な変化としては、第１のサブシンボルｓｓ１に比較して第２のサブシンボルｓｓ２の振幅値が、大きい場合（状態１）、同じ場合（状態２）、及び小さい場合（状態３）の３状態が取り得る。
【００７０】
この３状態を情報に割り付ける方法は、種々考えられる。例えば、上記のように状態１をデータ「１」、状態３をデータ「０」と対応させる第１の割付方法がある。
【００７１】
また、状態１を直前の送信データが「０」であり、該当サブシンボルがデータ「１」に変化、状態２を直前の送信データと該当シンボルで送信されるデータが同じ、状態３を直前の送信データが「１」であり該当シンボルがデータ「０」に変化する場合に対応させる第２の割付方法がある。
【００７２】
更に、状態１をデータ「０」、状態２をデータ「１」、状態３をデータ「２」と３進データに対応させる第３の割付方法がある。
【００７３】
［シンボル間差分法］
フェーディング伝送環境で受ける振幅歪みはシンボル内でほとんど定常状態で一定値を保ち、互いに連続する数シンボルの振幅もほぼ一定を保つ。これは、シンボル期間の逆数であるシンボル周波数に対して最大ドップラー周波数ｆ_Ｄ が十分に小さい事からも理解できる。
【００７４】
現シンボルの振幅値として２値と３値を採る２方法がある。
【００７５】
今、第１の方法であるシンボル間差分が２値を採る方法を考える。現シンボルの振幅が大なる値をデータ「１」に、現シンボルの振幅が小さくなる値をデータ「０」に対応させる時、前シンボルと現シンボルとの振幅差は、前シンボルの送信データが「０」でかつ現シンボルの送信データが「１」の場合に十分大きな正の値となる。
【００７６】
前シンボルの送信データが「１」でかつ現シンボルの送信データが「０」の場合に十分大きな負なる値となり、前シンボルの送信データと現シンボルの送信データが等しい場合にほとんど零なる値を示す。
【００７７】
第２の方法としてシンボル間差分が３値を採る方法を考える。３値のうち最大の値をデータ「１」に、中央の値をデータ「０」に、最小の値をデータ「−１」に対応させる。
【００７８】
この時、前シンボルと現シンボルの振幅差は、前シンボルの送信データが「０」でかつ現シンボルの送信データが「１」の場合と、前シンボルの送信データが「−１」でかつ現シンボルの送信データが「０」の場合には、正なる値δとなる。
【００７９】
前シンボルの送信データが「１」でかつ現シンボルの送信データが「０」の場合と、前シンボルの送信データが「０」でかつ現シンボルの送信データが「−１」の場合には負なる値−δとなる。
【００８０】
また、前シンボルの送信データと現シンボルの送信データが等しい場合にほとんど零となる。前シンボルの送信データが「１」でかつ現シンボルの送信データが「０」の場合と、前シンボルの送信データが「−１」でかつ現シンボルの送信データが「１」の場合には十分大きな正なる値２δとなる。
【００８１】
さらに、前シンボルの送信データが「１」でかつ現シンボルの送信データが「−１」の場合には十分大きな負なる値−２δとなる。このようにして、５種類の値が得られる。これらにトレリス符号化を適用すればより多くの情報を通信できる。
【００８２】
なお、上記説明では、簡単のために振幅の変化幅を中央値に値δを加算して得られる振幅値を最大値とし、振幅の変化幅を中央値に値δを減算して得られる振幅値を最小値としたが、閾値判定に十分な変化を与えれば良い。必ずしも、加算量と減算量を等しくする必要は無い。
【００８３】
また、シンボル間差分法として、振幅差分を前シンボルと現シンボル間で考えたが、対応するサブシンボル間でも同様であり容易に類推できるので説明を省略する。
【００８４】
本発明は、上記の通り従来技術に対し、位相変調波の振幅の差分に情報を載せる点が新しい技術であり、振幅を正確に検出するためには厳密な包絡検出回路を適用することが望ましい。
【００８５】
かかる包絡検出回路（ＳＳＥｄ：Ｓｔｒｉｃｔ Ｓｅｎｓｅ Ｅｎｖｅｌｏｐｅ Ｄｅｔｅｃｔｉｏｎ）は、既に、１９９４年年６月スエーデンのストックホルムで開催された国際会議ＩＥＥＥＶＴＣ９４にて発表されている。その内容は、会議議事録（“Ｅｎｖｅｌｏｐｅ Ｄｅｔｅｃｔｉｏｎ ｉｎ Ｓｔｒｉｃｔ Ｓｅｎｓｅ ａｎｄ ｉｔｓ Ａｐｐｌｉｃａｔｉｏｎ ｔｏ Ｓｙｌｌａｂｉｃ Ｃｏｍｐａｎｄｅｒｓ”， Ｐｒｏｃｅｅｄｉｎｇ Ｖｏｌ．３， ＰＰ．１７０４−１７０８） に記載がある。
【００８６】
本発明においては、包絡検出回路ＳＳＥｄで求まる前シンボルと現シンボルの時間差Ｔあるいはサブシンボル時間差Ｔ／２を有する２時点の各々の瞬時エンビロープ値を求め，各々の瞬時エンビロープ値の差を振幅差として用いる事に新たな特徴を有している。
【００８７】
［解析受信］
振幅に情報を載せた位相変調波（従来のＱＡＭ変調、あるいはＰＳＫ−ＡＳＫ変調に相当）を、正確に復調するためには、上記の解析信号の概念に基づく解析受信が必要になる。
【００８８】
受信信号あるいは中間周波数に変換した受信信号とする時、
【００８９】
【数３１】

【００９０】
を実部に、ｆ（ｔ） のヒルベルト変換信号である
【００９１】
【数３２】

【００９２】
を虚部とする概念上の信号である解析信号ｇ（ｔ） を考える。
【００９３】
したがって、解析信号ｇ（ｔ） は、
【００９４】
【数３３】

【００９５】
の式（１）のように与えられる。
【００９６】
解析信号ｇ（ｔ） は、複素表示すれば
【００９７】
【数３４】

【００９８】
の式（２）のようにも表現できる。
【００９９】
式（２）における解析信号の絶対値Ａ（ｔ） は、いわゆる包絡信号であり時刻ｔにおける瞬時振幅値を表す。Ａ（ｔ） は、受信信号ｆ（ｔ） を実部
【０１００】
【数３５】

【０１０１】
と、ｆ（ｔ） のヒルベルト変換信号
【０１０２】
【数３６】

【０１０３】
を用いて、
【０１０４】
【数３７】

【０１０５】
の式（３）のように与えられる。
【０１０６】
式（３）が厳密な意味での包絡信号ＳＳＥｄである。したがって、時刻ｔにおける振幅差分Ｂ（ｔ） は、時刻ｔの瞬時振幅値Ａ（ｔ） と時刻ｔよりτだけ事前の時刻ｔ−τの瞬時振幅値Ａ（ｔ− τ） の差分として、
【０１０７】
【数３８】

【０１０８】
の式（４）に与えられる。
【０１０９】
時間差τをＴと設定すれば、Ｂ（ｔ） はシンボル間差分を、時間差τをＴ／２と設定すればＢ（ｔ） はサブシンボル差分を、それぞれ表す。
【０１１０】
また、式（２）における解析信号の位相成分θ（ｔ） は、いわゆる瞬時位相であり、その実数成分ｃｏｓ｛θ（ｔ） ｝は時刻ｔにおける復調信号Ｉ（ｔ） を、その虚数成分ｓｉｎ｛θ（ｔ） ｝は時刻ｔにおける復調信号Ｑ（ｔ） を表す。
【０１１１】
瞬時位相θ（ｔ） は、時刻ｔ における瞬時方位を表しているので、解析信号ｇ（ｔ） をその絶対値で正規化した正規化解析信号β（ｔ） は、複素平面上での瞬時方位を表す事になる。正規化解析信号β（ｔ） は、受信信号ｆ（ｔ）
【０１１２】
【数３９】

【０１１３】
と、ｆ（ｔ） のヒルベルト変換信号
【０１１４】
【数４０】

【０１１５】
を用いて、
【０１１６】
【数４１】

【０１１７】
の式（５）のように与えられる。
【０１１８】
ここに、θ（ｔ） は、
【０１１９】
【数４２】

【０１２０】
の式（６）の関係にある。
【０１２１】
したがって、時刻ｔにおける位相差φ（ｔ） は、時刻ｔの瞬時位相θ（ｔ） と時刻ｔよりτだけ事前の時刻ｔ−τの瞬時位相θ（ｔ−τ） の差分として、
【０１２２】
【数４３】

【０１２３】
の式（７）のように与えられる。
【０１２４】
ところで、位相差φ（ｔ） は、式（５）で与えられる時刻ｔにおける正規化解析信号β（ｔ） と時刻ｔ−τにおける正規化解析信号β（ｔ−τ） の複素共役の積として与える。すなわち、
【０１２５】
【数４４】

【０１２６】
の式（８）の関係がある。
【０１２７】
復調信号Ｉ（ｔ） は式（８）の実数部として、Ｑ（ｔ） は式（８）の虚数部として与えられる。正規化解析信号β（ｔ） は、受信信号ｆ（ｔ） を
【０１２８】
【数４５】

【０１２９】
と、ｆ（ｔ） のヒルベルト変換信号
【０１３０】
【数４６】

【０１３１】
を用いて
【０１３２】
【数４７】

【０１３３】
の式（９）のように与えられる。
【０１３４】
また、式（８）を、受信信号ｆ（ｔ） を
【０１３５】
【数４８】

【０１３６】
と、ｆ（ｔ） のヒルベルト変換信号
【０１３７】
【数４９】

【０１３８】
を用いて
【０１３９】
【数５０】

【０１４０】
の式（１０）のように表す事ができる。
【０１４１】
式（１０）の位相角は、確かに式（７）に一致している事からも、解析受信の正当性が立証できる。
【０１４２】
さらに式（１０）が明らかに示すように、振幅の大きさに係わらず、その実部、虚部は定まる。これは、フェーディングに拠る通信信頼度が阻害される事を防止する事を示すものである。
【０１４３】
受信信号ｆ（ｔ） の振幅を一定に保つ様にリミッタで振幅を制限する場合は、分母が一定値となり、分子が支配項となる。リミッタ制限する場合は、正規化解析信号の代わりに受信信号ｆ（ｔ） を直接用いる事ができる。式（１０）の実部を復調信号Ｉ（ｔ） 、虚部を復調信号Ｑ（ｔ） とすれば、それぞれ
【０１４４】
【数５１】

【０１４５】
式（１１）、（１２）のように与える事ができる。
【０１４６】
図３は、本発明の実施例回路構成図である。図において、ヒルベルト変換器１は、上記の式（１）で与えられるように、受信信号ｆ（ｔ） を入力し、解析信号の実部と虚部を求めるためのものである。かかるヒルベルト変換器１の構成例については、本出願の発明者により先に特願平５−２８８３８７（特開平７−１４１３２４号公報）でも提案している。
【０１４７】
一例として図５に示されるヒルベルト変換器１は、シフトレジスタ１０とＲＯＭ及び累算器１２で構成される。シフトレジスタ１０には、受信信号ｆ（ｔ） のｎ時点のサンプル値ｆ（ｎ） が、入力される。
【０１４８】
シフトレジスタ１０の段数２Ｔが、ＲＯＭ１１に記憶された応答長ｕ（ｎ）と同じ数にセットされる場合、ヒルベルト変換信号
【０１４９】
【数５２】

【０１５０】
が、シフトレジスタ１０の値とＲＯＭ１１の値との内積により求められる。
【０１５１】
図３に戻り説明すると、他の構成要素の機能が図４に示される。即ち、図３において、３０乃至３６は、乗算器［図４（ａ）］であり、入力ａとｂの積を出力する。２０乃至２２は、遅延器［図４（ｂ）］であり、入力をＴ時間遅延して出力する。
【０１５２】
４０乃至４３は、加算器［図４（ｃ）］であり、入力ａとｂに対し、出力ａ±ｂ（入力に対し、複合同順）を求める。更に、５０は平方回路［図４（ｄ）］であり、入力ａに対し、出力√ａを出力する。６０、６１は、除算回路［図４（ｅ）］であり、入力ａとｂに対し、ａ÷ｂを求め出力する。
【０１５３】
かかる図４の回路構成の機能により、除算器６０、６１のそれぞれから、式（１１）、式（１２）により表わされる復調信号Ｉ（ｔ） 、Ｑ（ｔ） が出力される。更に、加算器４３から式（３）により表わされる包絡信号ＳＳＥｄであるＡ（ｔ） の時刻ｔおける、式（３）により表わされる振幅差分Ｂ（ｔ） が出力される。
【０１５４】
尚上記ヒルベルト変換器１は、ベースバンド復調の場合に必要であるが、中間周波復調では中間周波の周波数の１／４周期の遅延器で代用する事ができる。
【０１５５】
図６は本発明の上記図３の構成を適用する例として、従来のπ／４−ＤＱＰＳＫ（Ｄｉｆｆｅｒｅｎｔｉａｌ Ｑｕａｄｉ−Ｐｈａｓｅ Ｓｈｉｆｔ ｋｅｙｉｎｇ） 変調波に振幅１ビットの情報を付加し、周波数帯域を従来と同じ周波数帯域に制限した場合の、図２に示すようなサブシンボルに分割した時のＲＺ（Ｒｅｔｕｒｎ ｔｏ Ｚｅｒｏ ）変調に対する、振幅信号のアイパターンを示す。
【０１５６】
シンボルの中央時点で振幅値が立ち上がる場合と、逆に立ち下がる場合が明確に観測できる。
【０１５７】
更に、図７は上記図６と同様の適用であり、従来のπ／４−ＤＱＰＳＫ変調波に振幅１ビットの情報を付加し、周波数帯域を従来と同じ周波数帯域に制限した場合でシンボル差分のＮＲＺ（Ｎｏｎ Ｒｅｔｕｒｎ ｔｏ Ｚｅｒｏ）変調の振幅信号のアイパターンを示す。シンボルの中央時点で振幅値が大と小の２値を示している状態が同様に明確に観測できる。
【０１５８】
図８は、本発明と従来例とのビット誤り率（ＢＥＲ：Ｂｉｔ Ｅｒｒｏｒ Ｒａｔｅ） の比較を示す図である。サーマル雑音が支配的な静的伝送路の場合の特性Ａと、動的なフェーディング伝送路における特性Ｂを示す。
【０１５９】
同図において、実線ａは、π／４−ＤＱＰＳＫ変調波の振幅に１ ビットの情報をＮＲＺ方式で付加した場合の受信電界Ｅｂ／Ｎｏ 対ビット誤り率ＢＥＲ特性を示す。点線ｂは、π／４−ＤＱＰＳＫ変調波の振幅に１ビットの情報をＲＺ方式で付加した場合の特性である。また、一点鎖線ｃは、従来のπ／４−ＤＱＰＳＫ変調波の特性である。
【０１６０】
ビット誤り率ＢＥＲを受信電界Ｅｂ／Ｎｏ で評価する限り、本発明に適用するＮＲＺ方式は、１周波数あたり３ビットの高効率伝送状態を、従来のπ／４−ＤＱＰＳＫ変調方式より優れた通信品質（低いビット誤り率）で実現できる事が理解できる。
【０１６１】
また、本発明に従うＲＺ方式は、１周波数あたり３ビットの高効率伝送状態を、従来のπ／４−ＤＱＰＳＫ変調方式とほとんど同じ通信品質（低いビット誤り率）で実現できる事も理解できる。
【０１６２】
図９は、更に本発明を位相差８値と振幅差２値を用いた１６値伝送に適用した時のビットコンステレーションを示す。従来のビットコンステレーションとは異なり、位相差分と振幅差分を表している。
【０１６３】
この図９のビットコンステレーションを用い、本発明に開示されている伝送技術を用いて１周波数当たり４ビットの高効率伝送を実施した場合に得られる受信電界Ｅｂ／Ｎｏ 対ビット誤り特性を図１０に示す。
【０１６４】
図１０において、実線ａはＮＲＺの場合の１６ＤＱＡＭのＥｂ／Ｎｏ−ＢＥＲ特性を、点線ｂはＲＺの場合の１６ＤＱＡＭのＥｂ／Ｎｏ−ＢＥＲ（ビット誤りレート）特性を示す。さらに、同図１０における１点鎖線ｃは、実線ａと点線ｂで示す伝送実験例と同じ周波数帯域を用いて同じシンボル伝送速度で得られる従来のπ／４−ＤＱＰＳＫ変調の受信電界Ｅｂ／Ｎｏ−ＢＥＲ特性を示している。
【０１６５】
周波数利用効率が４ビット／Ｈｚと高い従来の１６ＱＡＭでは、フェーディング下では、殆ど伝送出来ない事は周知である。しかし、本発明で開示する１６ＤＱＡＭを用いることにより、π／４−ＤＱＰＳＫとほぼ同程度の誤り率であることが図１０に示すＥｂ／Ｎｏ−ＢＥＲ特性から明らかになる。このπ／４−ＤＱＰＳＫの周波数利用効率は、２ビット／Ｈｚと低いが、フェーディング耐力に優れ誤り率が低い事から、一般的に広く使用されている。
【０１６６】
すなわち、本発明の１６ＤＱＡＭは、フェーディング下でもπ／４−ＤＱＰＳＫと同程度の誤り率で、４ビット／Ｈｚと２倍の情報を同じ周波数帯域幅で伝送でき、電波資源の枯渇対策に強力な手段を提供する。
【０１６７】
さらに、本発明は、ＰＳＫ変調波の振幅差分に情報を付与する特徴を有するものであり、回線状況が良くなくビット誤り率が悪い場合には、１６ＤＱＡＭの移送差を８値から４値に切り換えて、本発明で開示しているＰＳＫ−ＤＯＥ技術を用いる事ができる。この場合、図８に示すように、誤り率を表すＥｂ／Ｎｏ−ＢＥＲ特性は、従来のπ／４−ＤＱＰＳＫの誤り率をさらに改善する事になる。同図において、同じ誤り率のＢＥＲ値を得るために必要なＰＳＫ−ＤＯＥの受信電界は、フェーディング下でπ／４−ＤＱＰＳＫのそれに比較し約５ｄＢ低くなり、それだけ悪い受信環境に耐える事を意味している。しかし、ＰＳＫ−ＤＯＥでは３ビット送信しているので、π／４−ＤＱＰＳＫの２ビット送信に対して約１．８ｄＢ（１．５倍＝３／２倍）だけ多くの電力を放射していることになり、差引、３ｄＢ強の改善が観られる。しかも、この時、周波数利用効率が３ビット／Ｈｚになっており、従来のπ／４−ＤＱＰＳＫに比較し１．５倍に周波数利用効率が改善されている。
【０１６８】
さらに、回線状況が悪化した場合には、振幅差に情報を付与する事を止め、移送差だけ用いた伝送に切り換え、誤り率の低下を防止する事ができる。この時は、従来のπ／４−ＤＱＰＳＫと周波数利用効率は同じになるが、復調方式に本発明で開示する解析受信を用いる事で、従来の遅延検波回路では必須であったローパスフィルタを不要とし受信波の位相歪みを与える事無く誤り率を改善し、かつ回路構造が簡単で経済的に実現できる利点を有する。
【０１６９】
解析受信回路は、図３に示すように、中間周波数の１／４周期時間の遅延器あるいはヒルベルト変換器１個と、シンボル期間Ｔ時間の遅延器２個と、乗算器４個と加算器２個だけ用いた簡単な回路で実現され、従来の遅延検波に必要なローパスフィルタを不要とする特徴を有する。ＰＳＫ−ＤＯＥあるいは１６ＤＱＡＭを対象にする場合でも、前記の中間周波数の１／４周期時間の遅延器あるいはヒルベルト変換器１個と、シンボル期間Ｔ時間の遅延器２個と、乗算器４個と加算器２個に、シンボル期間Ｔ時間の遅延器１個と、乗算器３個と加算器２個と除算器２個と平方根回路１個だけ加えた簡単な回路で実現できる。遅延検波で必須なローパスフィルタを必要としない利点は、フェーディング伝播による位相歪みが多々ある所へフィルタの遮断帯域近傍に存在する位相回転は致命的な欠陥になっている。ローパスフィルタの遮断域近傍の位相歪みを避けるためには、情報が存在する帯域を十分に越えた帯域までを通過帯域として広げたローパスフィルタを使用する必要がある。帯域幅を広げなければならない事は、通信している周波数帯域に隣接帯域を使用する事が出来ない事であり引いては周波数利用効率の低下が著しく生じる事になる。さらにローパスフィルタの遮断域近傍の位相歪みを避けるために通過帯域幅を広げる事は、不要雑音をそれだけ受信する事になり、誤り率の低下も同時に誘起する欠点がある。
【０１７０】
従来の遅延検波では、受信波とその遅延の積を求めるという所謂ＡＭ変調と同じ操作が含まれ、復調波に両側側帯波を含み、不要な側帯波を除去するために、どうしてもローパスフィルタを必要としていた。このため、片側の側帯波に含まれる情報エネルギーが損失される大きな問題が存在していた。
【０１７１】
さらに、ローパスフィルタは遮断周波数近傍で大きな位相歪みを与える性質を持ち、復調信号の位相成分に大きな歪みを与えＰＳＫ検波の誤り率を悪化させていた。さらに、ローパスフィルタの遮断特性を急峻にすると、遮断周波数近傍での位相歪みは増大し誤り率の劣化は増大する。逆に、ローパスフィルタの遮断特性を緩慢にして遮断周波数近傍での位相歪みを少なくすると、フィルタの通過域周波数幅が増大し、そのため伝送帯域以外の雑音の混入が増大し、やはり誤り率の劣化を増大させていた。両側帯波から片側の側帯波を除去するためにフィルタ（ローパスフィルタ）を使用する限り、誤り率が劣化するという避けえない欠点が存在していた。
【０１７２】
一方、解析受信では、クレームに述べているように解析信号理論に基づき信号処理するため所謂ＳＳＢ変調形式となり、復調信号に単一側帯波成分のみ含む事になる。このため、解析受信の復調信号には情報エネルギーが全て含まれている利点が得られる。
【０１７３】
加えるに、側帯波の片側を除去する必要は無いため、フィルタ操作で増大した位相歪みが原因となる誤り率の劣化や雑音の混入が原因となる誤り率の劣化を完全に防止する利点が得られる。このため、解析受信を用い復調信号に全情報エネルギーを含ませ、かつ情報エネルギー以外の雑音エネルギーは殆ど含ませない事で、従来と同程度の誤り率で、３ビット／Ｈｚないし４ビット／Ｈｚの高い周波数利用効率を達成する利点を得る。
【０１７４】
無論、従来の遅延検波が有する特徴の１つであるキャリア捕捉を不要とする利点を、本発明が開示する解析受信も有する。このため、移動通信のようなマルチパス伝搬は前提となる通信システムにおいては、最大電力を示す受信波を復調する事で常にＤＵＲを０ｄＢ以上に保つ事が可能となっている。
【０１７５】
最大電力を有する受信波を捜し出す必要は無く、単純にマルチパス伝搬後の受信波ならびに中間周波あるいはベースバンドへ周波数変換した受信信号をそのまま本発明で開示する受信器へ入力すれば、キャリア捕捉を不要とする性質から勝手に最大電力を有する信号が支配項となりＤＵＲが０ｄＢ以上となり、位相ジッタの最大幅は±９０度となる。
【０１７６】
従来のキャリア捕捉が必要とする場合は、目的波がゼロ振幅値で非目的波が全振幅となる場合まで想定する必要があり、最大位相ジッタは±πラジアン＝±１８０度となっていた。
【０１７７】
このため、本発明の伝送方式では位相ジッタが従来の±１８０度＝±１／２シンボル期間から、±９０度＝±１／４シンボル期間と縮退し、ビット誤り率が大幅に低下し通信品質が大幅に改善される。ひいては同じ通信エネルギーでも１．５倍から２倍の情報量を通信できる利点を有する。
【０１７８】
尚、本発明の適用にあたっては、本発明のＰＳＫ−ＤＯＥ、Ｅ−１６値ＤＱＡＭを固定的に用いるのではなく、回線状態に応じて適応的に用いることが望ましい。
【０１７９】
例えば、回線状態が最良の時に、本発明に従う位相８値×振幅２値変調による１６値ＤＱＡＭを適用し、回線状態が悪化してきた時に、本発明に従うＰＳＫ信号に振幅変調を行なうＰＳＫ−ＤＯＥ方式を採用する。
【０１８０】
更に回線状態が悪化した場合は、従来のＰＳＫ方式を採用するように送受信間で変調方式を切り替えるようにすればよい。また、回線状態が回復してきた場合は、上記と反対方向に変調方式を戻すことが出来る。更に加えるに、回線状態が更に良くなる場合には、位相を８値から多値化し、又振幅も２値から多値化し、周波数効率を増大させ通信することが出来る。
【０１８１】
図１１は、かかる本発明のデジタル無線伝送方式において、上記回線の伝搬状態に応じて適応的に、変調方式を変えることができる送信側に置かれる変調器の構成例である。
【０１８２】
シリアルーパラレルビット変換器１００は、入力のシリアルビット列を、伝搬状態に応じて２ビット、３ビット、４ビットにグループ化する機能を有する。差動位相変調器１０１は、伝搬状態に応じて４値、または８値の位相変調を行なう機能を有する。エンビロープ発生器１０２は、伝搬状態に応じて振幅変調を行なう。以下にかかる変調器の動作例を説明する。
［伝搬状態が良好な場合▲１▼］
この場合に、差動位相変調器１０１は４値をとるものとする。シリアルーパラレルビット変換器１００は、入力ビット列を３ビットずつにグループ化し、２ビットを差動位相変調器１０１に、１ビットをエンビロープ発生器１０２にそれぞれ送出する。
【０１８３】
差動位相変調器１０１は２ビットの信号列を９０度毎に異なる４値のシンボルに割り当てる位相変調を行ない、乗算器１０３に出力する。エンビロープ発生器１０２では１ビットの信号に基ずき２値のレベルを作成し、乗算器１０３に出力する。乗算器１０３では、４値の位相変調出力に２値のレベルを合成して、８値の多値化信号を出力する。これを、ＰＳＫーＤＯＥ（ＰＳＫ−Ｄａｔａ ｏｎ Ｅｎｖｅｌｏｐｅ） と定義する。
［伝搬状態が▲１▼より更に良好な場合］
この場合は、差動位相変調器１０１は８値をとるものとする。シリアルーパラレルビット変換器１００は、入力ビット列を４ビットずつにグループ化し、３ビットを差動位相変調器１０１に、１ビットをエンビロープ発生器１０２にそれぞれ送出する。
【０１８４】
差動位相変調器１０１は３ビットの信号列を４５度毎に異なる８値のシンボルに割り当てる位相変調を行い、乗算器１０３に出力する。エンビロープ発生器１０２では、１ビットの信号に基ずき２値のレベルを作りだし乗算器１０３に出力する。乗算器１０３では、８値の位相変調出力に２値にレベルを合成して、１６値の多値化信号を出力する。これを１６ＤＱＡＭ（Ｄｉｆｆｅｒｅｎｔｉａｌ Ｑｕａｄｉ−Ａｍｐｌｉｔｕｄｅ Ｍｏｄｕｌａｔｉｏｎ）と定義する。
［伝搬状態が▲１▼よりも良好でない場合］
この場合、差動襲う変調器１０１は４値をとるものとする。シリアルーパラレルビット変換器１００は、入力ビット列を２ビットずつにグループ化し、そのすべてを差動位相変調器１０１に送出する。
【０１８５】
差動位相変調器１０１は、２ビットの信号列９０度毎に異なる４値のシンボルに割り当てる位相変調を行い、乗算器１０３に出力する。エンビロープ発生器１０２には何も入力されておらず、単一のレベルを保持する。結果として、乗算器１０３では、４値の位相変調出力をそのまま４値の多値化信号として、出力する。
【０１８６】
上記のように、３種類の変調方法について説明したが、その選択方法は、上記のように伝搬状態（ビット誤り率、電解強度等から求められる）、システムの固有の要求等を勘案して適宜選択されるべきものである。
【０１８７】
【発明の効果】
以上実施例にしたがい説明したように、本発明によりデジタル無線伝送方式において、ＰＳＫ伝送のフェーディング耐力を損なうこと無く、かつ占有周波数帯域を増大すること無く、同じ周波数帯域幅を使用して伝送する情報量を増大できる。
【図面の簡単な説明】
【図１】１シンボル区間を２つのサブシンボルｓｓ１，ｓｓ２を設定した例を説明する図である。
【図２】サブシンボル間の振幅差に情報を与えることを説明する図である。
【図３】本発明の実施例回路の構成図である。
【図４】図３の各構成要素を説明する図である。
【図５】ヒルベルト変換器の構成例を説明する図である。
【図６】π／４−ＤＱＰＳＫ変調波に１ビット情報付加の例（ＲＺ符号の例）を説明する図である。
【図７】π／４−ＤＱＰＳＫ変調波に１ビット情報付加の例（ＮＲＺ符号の例）を説明する図である。
【図８】本発明のＰＳＫ−ＤＯＥと従来のπ／４−ＤＱＰＳＫ変調方式についてのＢＥＲの比較を説明する図である。
【図９】本発明の位相８値ー振幅２値の１６値ＤＱＡＭのビットコンステレーションを説明する図である。
【図１０】図９のビットコンステレーションを用いた時のＢＥＲの比較を説明する図である。
【図１１】送信側の変調器の構成例を示す図である。
【図１２】従来のＰＳＫ信号のフェーィングにおける振幅ひずみを説明する図である。
【図１３】従来のＰＳＫ信号のフェーィングにおける位相ジッタを説明する図である。
【符号の説明】
１ ヒルベルト変換器
２０〜２２ 遅延器
３０〜３６ 乗算器
４０〜４３ 加算器
６０、６１ 除算器
５０ 平方器
１００ シリアル−パラレルビット変換器
１０１ 差動位相変調器
１０２ エンビロープ発生器
１０３ 乗算器[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a digital radio transmission system for efficiently transmitting digital information under intense fading such as mobile communication.
[0002]
[Prior art]
In principle, a carrier signal having a constant amplitude is transmitted at a phase corresponding to a symbol value in a symbol interval, and the phase in the symbol interval is detected (synchronous detection) on the receiving side, or the immediately preceding symbol interval and Phase modulation transmission systems such as PSK modulation or differential PSK modulation for detecting information (delay detection) and transmitting information have been put into practical use.
[0003]
In general, transmitted radio waves do not arrive at the receiving side via a single radio wave propagation path, but are received as a composite wave of a plurality of reflected waves from a wall surface of a building. In mobile communication such as a car phone and a mobile phone, reception is often performed through a plurality of propagation paths, and the distance and direction of the propagation path changes drastically with time because the transceiver itself moves.
[0004]
Here, DUR which is the ratio of the target wave (= D wave) and the non-target wave (= U wave) is considered. That is, when it is possible to assume that the received wave is two waves, when the received wave with the maximum power is the target wave (= D wave) and the remaining received wave is the non-target wave (= U wave), the D wave power and the U wave The worst case where the power ratio is 1 is DUR = 10 log (D / U) = 0 dB.
[0005]
On the other hand, in synchronous detection where it is necessary to faithfully capture a carrier signal, the signal power to be captured often falls below the remaining signal power, and the DUR may be less than 0 dB, indicating a negative DUR value.
[0006]
Also, in asynchronous detection that does not assume carrier capture, an incoming radio wave with a strong received electric field can be a D wave, and the remaining received waves can be U waves. In this case, the DUR never falls below 0 dB.
[0007]
Now, the maximum Doppler frequency f _D FIG. 12A shows a time response of amplitude fluctuation of a general received wave of mobile communication when = 40 Hz and DUR = 0 dB, and FIG. 12B shows a time average power spectrum of amplitude fluctuation of the received wave.
[0008]
As is clear from FIG. 12B, the power distribution of the amplitude fluctuation is almost normal, and the power component is the maximum Doppler frequency f. _D The amplitude fluctuation component that concentrates below the frequency of 2 times and exceeds the frequency of 2 times is less than -20 dB.
[0009]
On the other hand, in the multipath fading channel, the instantaneous value of the amplitude of the radio wave changes slowly and greatly. In the time response of the 500 msec period shown in FIG. 12A, for example, in the case of 4-valued PSK of 32 kbps, there are 8000 symbol periods, and the amplitude variation difference between symbols is considerably small.
[0010]
The maximum Doppler frequency f, which is the same condition as when the characteristics of FIG. _D FIG. 12A shows the phase jitter time response of the PSK wave that has passed through the fading propagation path of = 40 Hz and DUR = 0 dB, and FIG. 13B shows the time average power spectrum of the phase jitter.
[0011]
As can be clearly seen in FIG. 13 (b), the phase jitter is the maximum Doppler frequency f. _D A uniform distribution is shown with the following approximately equal frequency components. On the other hand, the amount of instantaneous phase jitter is large, and it frequently appears that the judgment critical value ± π / 4 in the case of quaternary PSK is exceeded.
[0012]
In the example of FIG. 13A, sections exceeding π / 4 appear at 110-121 msec and 471-478 msec. In the case of 4-valued PSK of 32 kbps, an error occurs continuously in 176 symbol intervals at 11 msec in the 110-121 msec interval, and an error occurs in 112 symbol intervals in 7 msec in the 471-478 msec interval.
[0013]
Therefore, 288 / 8000≈4 × 10 due to fading ^-2 Symbol error rate, ie 2 × 10 ^-2 The bit error rate (BER) is obtained, and the communication quality is considerably deteriorated. Although fading alone is not the cause of error occurrence, it also causes thermal noise errors in receivers and the like, but fading is the main cause of error occurrence when the received electric field is strong.
[0014]
Thus, a large phase displacement appears in fading. However, the phase jitter difference between adjacent symbol sections can be suppressed to be quite small. This means that the frequency component of the phase jitter fluctuation is the maximum Doppler frequency f. _D It can be understood from the following.
[0015]
In conventional PSK transmission, information is placed on the carrier phase or phase difference. Therefore, for demodulation on the receiving side, carrier acquisition is essential for phase identification of the received carrier, and in a multipath propagation environment, it is not always possible to set a received wave showing the maximum power as a target wave.
[0016]
This is because it is necessary to continue capturing the received wave once captured, but there is no guarantee that the reception level of the captured received wave indicates the maximum power of the multipath propagation wave.
[0017]
If another received wave with a higher received wave level is used as the D wave among the multipath propagation waves, it is necessary to perform another synchronous pull-in process for carrier acquisition, and is correct until carrier acquisition is completed. You cannot receive.
[0018]
For this reason, errors occur for a long time continuously after switching to the maximum received wave until the carrier acquisition is completed.
[0019]
Rather than abandoning a received wave that has been acquired once because the reception level has dropped, tracking the received wave whose reception level has decreased and continuing carrier acquisition results in fewer errors. it can.
[0020]
For this reason, the conventional reception method premised on carrier acquisition has a drawback that a negative state occurs in the DUR which is a reception comparison value with the U wave, and the phase jitter becomes large.
[0021]
Information is transmitted only on the phase component and cannot be used to put information on the amplitude component that fluctuates greatly due to fading. The conventional transmission system has drawbacks.
[0022]
For this reason, the conventional method has a drawback that the frequency utilization efficiency is low and the transmission capacity cannot be increased even if the same frequency bandwidth is used.
[0023]
[Problems to be solved by the invention]
Accordingly, an object of the present invention is to solve the problems in the conventional system described above, and to reduce the amount of information transmitted using the same frequency bandwidth without deteriorating the fading resistance of PSK transmission and without increasing the occupied frequency band. It is an object of the present invention to provide a highly efficient digital wireless transmission system that can increase the frequency.
[0024]
[Means for Solving the Problems]
The first configuration of the digital radio transmission system that achieves the object of the present invention is the digital radio transmission system that keeps the carrier signal in a predetermined phase for a certain period and transmits information, and in the digital radio transmission system that transmits information, A second period following the first period is set, and the amplitude value of the carrier signal in the second period is increased or decreased using the amplitude value of the carrier signal in the first period as a reference value. Thus, information of 1 bit or more is modulated on the difference between the amplitude values of the first period and the second period.
[0025]
Furthermore, a second configuration of the digital radio transmission system that achieves the object of the present invention is characterized in that, in the first configuration, the certain period is an information symbol period.
[0026]
A third configuration of the digital radio transmission system that achieves the object of the present invention is a digital radio transmission system in which a carrier signal is kept in a predetermined phase for a certain period and information is transmitted. By using the amplitude value as a reference value, the amplitude value of the carrier signal in the information symbol period following the one information symbol period is increased or decreased, whereby the amplitude value of the first period and the second period It is characterized in that information of 1 bit or more is modulated on the difference between the two.
[0027]
Furthermore, the fourth configuration of the digital radio transmission system that achieves the object of the present invention is a digital radio transmission system that transmits information by keeping a carrier signal in a predetermined phase for a certain period of time, and receives a signal f (t) at time t. And a signal obtained by Hilbert transform of the received signal f (t)
[0028]
[Expression 16]

[0029]
The square of the sum of the squares of the two is used as an envelope signal, and the difference in the envelope signal corresponding to the time difference between one information symbol period and the subsequent information symbol period or the time differences between a plurality of subsymbol periods obtained by dividing the one information symbol period Demodulated as an amplitude difference signal.
[0030]
Furthermore, the fifth configuration of the digital radio transmission system that achieves the object of the present invention is the digital radio transmission system that keeps the carrier signal in a predetermined phase for a certain period and transmits information, and the received signal f (t) at time t is Real part
[0031]
[Expression 17]

[0032]
And the Hilbert transform signal of the received signal f (t)
[0033]
[Expression 18]

[0034]
Analysis signal with imaginary part
[0035]
[Equation 19]

[0036]
And an analysis signal at time t-T
[0037]
[Expression 20]

[0038]
Complex conjugate of
[0039]
[Expression 21]

[0040]
Product g (t) and real part of g ′ (t−T)
[0041]
[Expression 22]

[0042]
Is the demodulated signal I (t) and the imaginary part
[0043]
[Expression 23]

[0044]
Is Q (t).
[0045]
The sixth configuration of the digital radio transmission system that achieves the object of the present invention is the digital radio transmission system that keeps the carrier signal in a predetermined phase for a certain period and transmits information, and the received signal f (t) Real part
[0046]
[Expression 24]

[0047]
And the Hilbert transform signal of the received signal f (t)
[0048]
[Expression 25]

[0049]
Analysis signal with imaginary part
[0050]
[Equation 26]

[0051]
And an analysis signal at time t-T
[0052]
[Expression 27]

[0053]
Complex conjugate of
[0054]
[Expression 28]

[0055]
Signal g (t) g ′ (t−T) normalized with the envelope A (t) at time t and the envelope A (t−T) at time tT. −T) / A (t) A (t−T)]
[0056]
[Expression 29]

[0057]
Is the demodulated signal I (t) and the imaginary part
[0058]
[30]

[0059]
Is Q (t).
[0060]
DETAILED DESCRIPTION OF THE INVENTION
Embodiments of the present invention will be described below with reference to the drawings. In the drawings, the same or similar elements will be described with the same reference numerals or reference symbols.
[0061]
The digital wireless transmission system of the present invention is characterized in that information is placed on the phase difference and the amplitude difference in the symbol section in PSK transmission.
[0062]
Further, as an embodiment of the present invention, as a method for placing information on the amplitude, a method using a difference in a symbol and a method using a difference between symbols are proposed. The features of the present invention will be described below for each method.
[0063]
[Intra-symbol difference method]
As described above, since the amplitude distortion caused by fading or the like is quasi-stationary within a symbol, when sub-symbols divided into two or more as shown in FIG. 1 are defined for some or all symbols. The amplitude fluctuations are substantially equal.
[0064]
That is, FIG. 1 is an example in which two subsymbols ss1 and ss2 are set in one symbol period, and at least the relative amplitude values of the adjacent subsymbols ss1 and ss2 do not change even when fading is applied.
[0065]
For example, when the symbol amplitude of the received wave varies due to fading and becomes larger than the reference value, the sub-symbol amplitude also varies, and the relative amplitude relationship is maintained.
[0066]
Therefore, information is given to the amplitude difference between the sub-symbols ss1 and ss2 as shown in FIG. 2, that is, the signal is modulated and transmitted, and the relative variation in the magnitude of the corresponding amplitude value appears from the envelope of the received wave. By measuring, the receiving side can detect that transmission information is received.
[0067]
In FIG. 2, the amplitude of the subsymbol ss2 is relatively increased with respect to the amplitude of the subsymbol ss1 in the first symbol period, and data “1” is represented at this time. In the second symbol period, the amplitude of the sub symbol ss2 is relatively small with respect to the amplitude of the sub symbol ss1 to represent data “0”. In this way, it is possible to send information by relative changes of sub-symbols.
[0068]
When the amplitude value is changed, it is possible to communicate by placing information on the phase at the same time. In this case, an advanced reception technique is required for phase detection, but it can be dealt with by an analysis reception technique as will be described later.
[0069]
Further, as the relative change of the sub-symbol, the amplitude value of the second sub-symbol ss2 is larger (state 1), the same (state 2), and smaller than the first sub-symbol ss1. There are three possible cases (state 3).
[0070]
There are various methods for assigning these three states to information. For example, as described above, there is a first allocation method in which state 1 is associated with data “1” and state 3 is associated with data “0”.
[0071]
Also, in state 1, the previous transmission data is “0”, the corresponding sub-symbol changes to data “1”, state 2 is the same as the previous transmission data and the data transmitted in the corresponding symbol, and state 3 is the previous transmission data. There is a second allocation method that corresponds to the case where the transmission data is “1” and the corresponding symbol changes to data “0”.
[0072]
Further, there is a third allocation method in which state 1 corresponds to data “0”, state 2 corresponds to data “1”, state 3 corresponds to data “2”, and ternary data.
[0073]
[Difference method between symbols]
The amplitude distortion that is received in the fading transmission environment is almost constant in the steady state within the symbol, and the amplitude of several consecutive symbols is also almost constant. This is the maximum Doppler frequency f for the symbol frequency which is the reciprocal of the symbol period. _D Can be understood from the fact that is sufficiently small.
[0074]
There are two methods of taking the binary value and the ternary value as the amplitude value of the current symbol.
[0075]
Now, consider a first method in which the inter-symbol difference takes a binary value. When a value that increases the amplitude of the current symbol corresponds to data “1” and a value that decreases the amplitude of the current symbol corresponds to data “0”, the amplitude difference between the previous symbol and the current symbol is the transmission data of the previous symbol. When “0” and the transmission data of the current symbol is “1”, a sufficiently large positive value is obtained.
[0076]
When the transmission data of the previous symbol is “1” and the transmission data of the current symbol is “0”, a sufficiently large negative value is obtained, and when the transmission data of the previous symbol is equal to the transmission data of the current symbol, the value is almost zero. Show.
[0077]
As a second method, consider a method in which the inter-symbol difference takes a ternary value. Of the three values, the maximum value corresponds to data “1”, the center value corresponds to data “0”, and the minimum value corresponds to data “−1”.
[0078]
At this time, the amplitude difference between the previous symbol and the current symbol is as follows: the transmission data of the previous symbol is “0” and the transmission data of the current symbol is “1”, and the transmission data of the previous symbol is “−1”. When the symbol transmission data is “0”, the value is a positive value δ.
[0079]
Negative when the previous symbol transmission data is “1” and the current symbol transmission data is “0”, and when the previous symbol transmission data is “0” and the current symbol transmission data is “−1”. The value −δ.
[0080]
Further, when the transmission data of the previous symbol is equal to the transmission data of the current symbol, it becomes almost zero. It is sufficient when the transmission data of the previous symbol is “1” and the transmission data of the current symbol is “0” and when the transmission data of the previous symbol is “−1” and the transmission data of the current symbol is “1”. A large positive value 2δ is obtained.
[0081]
Further, when the transmission data of the previous symbol is “1” and the transmission data of the current symbol is “−1”, a sufficiently large negative value −2δ is obtained. In this way, five types of values are obtained. If trellis coding is applied to these, more information can be communicated.
[0082]
In the above description, for the sake of simplicity, the amplitude value obtained by adding the value δ to the median value of the amplitude change width is set to the maximum value, and the amplitude obtained by subtracting the value δ from the median value of the amplitude change range. Although the value is set to the minimum value, it suffices to give a sufficient change to the threshold determination. It is not always necessary to make the addition amount and the subtraction amount equal.
[0083]
Further, as the inter-symbol difference method, the amplitude difference is considered between the previous symbol and the current symbol. However, the same applies to the corresponding sub-symbols and can be easily analogized.
[0084]
As described above, the present invention is a new technique in which information is put on the difference in amplitude of the phase modulation wave as compared with the conventional technique, and it is desirable to apply a strict envelope detection circuit in order to accurately detect the amplitude. .
[0085]
Such an envelope detection circuit (SSEd) has already been presented at the international conference IEEE VTC94 held in Stockholm, June 1994. The contents are described in the meeting minutes (“Envelopment Detection in Strate Sense and Applications to Syllabic Companders”, Proceeding Vol. 3, PP. 1704-1708).
[0086]
In the present invention, the instantaneous envelope values at two time points having the time difference T or sub-symbol time difference T / 2 between the previous symbol and the current symbol obtained by the envelope detection circuit SSEd are obtained, and the difference between the instantaneous envelope values is used as the amplitude difference. It has a new feature to use.
[0087]
[Analysis reception]
In order to accurately demodulate a phase-modulated wave with information on amplitude (equivalent to conventional QAM modulation or PSK-ASK modulation), analysis reception based on the concept of the analysis signal described above is required.
[0088]
When it is a received signal or a received signal converted to an intermediate frequency,
[0089]
[31]

[0090]
Is the Hilbert transform signal of f (t)
[0091]
[Expression 32]

[0092]
Let us consider an analytic signal g (t) that is a conceptual signal with imaginary part as.
[0093]
Therefore, the analytic signal g (t) is
[0094]
[Expression 33]

[0095]
(1).
[0096]
The analytic signal g (t) can be expressed in complex form.
[0097]
[Expression 34]

[0098]
It can also be expressed as equation (2).
[0099]
The absolute value A (t) of the analytic signal in equation (2) is a so-called envelope signal and represents the instantaneous amplitude value at time t. A (t) is the real part of the received signal f (t)
[0100]
[Expression 35]

[0101]
And the Hilbert transform signal of f (t)
[0102]
[Expression 36]

[0103]
Using,
[0104]
[Expression 37]

[0105]
(3).
[0106]
Equation (3) is the envelope signal SSEd in a strict sense. Therefore, the amplitude difference B (t) at the time t is the difference between the instantaneous amplitude value A (t) at the time t and the instantaneous amplitude value A (t−τ) at the time t−τ prior to the time t by τ.
[0107]
[Formula 38]

[0108]
Is given in equation (4).
[0109]
If the time difference τ is set to T, B (t) represents the inter-symbol difference, and if the time difference τ is set to T / 2, B (t) represents the sub-symbol difference.
[0110]
In addition, the phase component θ (t) of the analytic signal in the expression (2) is a so-called instantaneous phase, and the real component cos {θ (t)} represents the demodulated signal I (t) at time t and the imaginary component sin. {Θ (t)} represents the demodulated signal Q (t) at time t.
[0111]
Since the instantaneous phase θ (t) represents the instantaneous azimuth at time t 1, the normalized analysis signal β (t) obtained by normalizing the analytic signal g (t) with its absolute value is the instantaneous azimuth on the complex plane. Will be expressed. The normalized analysis signal β (t) is the received signal f (t)
[0112]
[39]

[0113]
And the Hilbert transform signal of f (t)
[0114]
[Formula 40]

[0115]
Using,
[0116]
[Expression 41]

[0117]
(5).
[0118]
Where θ (t) is
[0119]
[Expression 42]

[0120]
(6).
[0121]
Therefore, the phase difference φ (t) at time t is the difference between the instantaneous phase θ (t) at time t and the instantaneous phase θ (t−τ) at time t−τ prior to time t by τ,
[0122]
[Expression 43]

[0123]
(7).
[0124]
By the way, the phase difference φ (t) is a product of the complex conjugate of the normalized analysis signal β (t) at time t given by the equation (5) and the normalized analysis signal β (t−τ) at time t−τ. give. That is,
[0125]
(44)

[0126]
(8)
[0127]
The demodulated signal I (t) is given as the real part of equation (8), and Q (t) is given as the imaginary part of equation (8). The normalized analysis signal β (t) is obtained by converting the received signal f (t) to
[0128]
[Equation 45]

[0129]
And the Hilbert transform signal of f (t)
[0130]
[Equation 46]

[0131]
Using
[0132]
[Equation 47]

[0133]
(9).
[0134]
Further, the equation (8) is expressed as the received signal f (t)
[0135]
[Formula 48]

[0136]
And the Hilbert transform signal of f (t)
[0137]
[Formula 49]

[0138]
Using
[0139]
[Equation 50]

[0140]
(10) can be expressed as follows.
[0141]
Since the phase angle of Expression (10) certainly matches Expression (7), the validity of analysis reception can be verified.
[0142]
Furthermore, as the equation (10) clearly shows, the real part and the imaginary part are determined regardless of the magnitude of the amplitude. This indicates that communication reliability due to fading is prevented from being hindered.
[0143]
When limiting the amplitude with a limiter so as to keep the amplitude of the received signal f (t) constant, the denominator becomes a constant value and the numerator becomes the dominant term. When limiting the limiter, the received signal f (t) can be directly used instead of the normalized analysis signal. If the real part of Equation (10) is the demodulated signal I (t) and the imaginary part is the demodulated signal Q (t), respectively,
[0144]
[Equation 51]

[0145]
It can be given as in equations (11) and (12).
[0146]
FIG. 3 is a circuit diagram of an embodiment of the present invention. In the figure, a Hilbert transformer 1 is for receiving a received signal f (t) and obtaining a real part and an imaginary part of an analysis signal, as given by the above equation (1). An example of the configuration of the Hilbert converter 1 has been previously proposed in Japanese Patent Application No. 5-288387 (Japanese Patent Laid-Open No. 7-141324) by the inventor of the present application.
[0147]
As an example, the Hilbert transformer 1 shown in FIG. 5 includes a shift register 10, a ROM, and an accumulator 12. The shift register 10 receives the sample value f (n) at the time n of the received signal f (t).
[0148]
When the number of stages 2T of the shift register 10 is set to the same number as the response length u (n) stored in the ROM 11, the Hilbert transform signal
[0149]
[Formula 52]

[0150]
Is obtained from the inner product of the value of the shift register 10 and the value of the ROM 11.
[0151]
Returning to FIG. 3, the functions of the other components are shown in FIG. That is, in FIG. 3, reference numerals 30 to 36 denote multipliers (FIG. 4A), which output the product of inputs a and b. Reference numerals 20 to 22 denote delay units [FIG. 4B], which delay the input by T time and output.
[0152]
Reference numerals 40 to 43 denote adders [FIG. 4 (c)], which obtain output a ± b (combined order with respect to input) for inputs a and b. Reference numeral 50 denotes a square circuit (FIG. 4D), which outputs an output √a with respect to the input a. Reference numerals 60 and 61 denote a dividing circuit [FIG. 4 (e)], which calculates a ÷ b for the inputs a and b and outputs them.
[0153]
The demodulated signals I (t) and Q (t) represented by the equations (11) and (12) are output from the dividers 60 and 61 by the function of the circuit configuration of FIG. Further, the amplitude difference B (t) represented by the equation (3) is output from the adder 43 at the time t of A (t) which is the envelope signal SSEd represented by the equation (3).
[0154]
The Hilbert transformer 1 is necessary in the case of baseband demodulation, but in the intermediate frequency demodulation, a delay device having a quarter cycle of the frequency of the intermediate frequency can be used instead.
[0155]
FIG. 6 shows, as an example of applying the configuration of FIG. 3 of the present invention, information of 1-bit amplitude is added to a conventional π / 4-DQPSK (Differential Quadrature-Phase Shift keying) modulation wave, and the frequency band is the same as the conventional one. 3 shows an eye pattern of an amplitude signal for RZ (Return to Zero) modulation when divided into sub-symbols as shown in FIG. 2 when limited to a frequency band.
[0156]
A case where the amplitude value rises at the central point of the symbol and a case where the amplitude value falls on the contrary can be clearly observed.
[0157]
Further, FIG. 7 is the same application as FIG. 6 described above. In the case where information of 1-bit amplitude is added to the conventional π / 4-DQPSK modulated wave and the frequency band is limited to the same frequency band as in the conventional case, An eye pattern of an amplitude signal of NRZ (Non Return to Zero) modulation is shown. Similarly, a state in which the amplitude value shows a binary value of large and small at the central point of the symbol can be clearly observed.
[0158]
FIG. 8 is a diagram showing a comparison of bit error rate (BER) between the present invention and the conventional example. A characteristic A in the case of a static transmission line in which thermal noise is dominant and a characteristic B in a dynamic fading transmission line are shown.
[0159]
In the figure, a solid line a represents the received electric field Eb / No versus bit error rate BER characteristic when 1-bit information is added to the amplitude of the π / 4-DQPSK modulated wave by the NRZ method. A dotted line b represents a characteristic when 1-bit information is added to the amplitude of the π / 4-DQPSK modulated wave by the RZ method. A one-dot chain line c is a characteristic of a conventional π / 4-DQPSK modulated wave.
[0160]
As long as the bit error rate BER is evaluated by the received electric field Eb / No, the NRZ system applied to the present invention has a communication quality superior to that of the conventional π / 4-DQPSK modulation system in a highly efficient transmission state of 3 bits per frequency. It can be understood that it can be realized with (low bit error rate).
[0161]
It can also be understood that the RZ system according to the present invention can realize a high-efficiency transmission state of 3 bits per frequency with almost the same communication quality (low bit error rate) as the conventional π / 4-DQPSK modulation system.
[0162]
FIG. 9 shows a bit constellation when the present invention is further applied to 16-value transmission using 8 phase differences and 2 amplitude differences. Unlike the conventional bit constellation, it represents a phase difference and an amplitude difference.
[0163]
The received electric field Eb / No versus bit error characteristics obtained when high-efficiency transmission of 4 bits per frequency is performed using the transmission technique disclosed in the present invention using the bit constellation of FIG. Shown in
[0164]
In FIG. 10, the solid line a indicates the Eb / No-BER characteristic of 16DQAM in the case of NRZ, and the dotted line b indicates the Eb / No-BER (bit error rate) characteristic of 16DQAM in the case of RZ. Further, a dashed-dotted line c in FIG. 10 indicates a received electric field Eb / No of conventional π / 4-DQPSK modulation obtained at the same symbol transmission rate using the same frequency band as the transmission experiment example indicated by the solid line a and the dotted line b. -The BER characteristic is shown.
[0165]
It is well known that conventional 16QAM, which has a high frequency utilization efficiency of 4 bits / Hz, can hardly transmit under fading. However, by using 16DQAM disclosed in the present invention, it becomes clear from the Eb / No-BER characteristic shown in FIG. 10 that the error rate is almost the same as that of π / 4-DQPSK. The frequency utilization efficiency of π / 4-DQPSK is as low as 2 bits / Hz, but is generally widely used because of its excellent fading resistance and low error rate.
[0166]
That is, the 16DQAM of the present invention can transmit information twice as high as 4 bits / Hz at the same frequency bandwidth with an error rate similar to that of π / 4-DQPSK even under fading, and is a powerful countermeasure for depletion of radio resources. Provide a simple means.
[0167]
Furthermore, the present invention has a feature of adding information to the amplitude difference of the PSK modulated wave. When the line condition is not good and the bit error rate is bad, the transport difference of 16DQAM is switched from 8 values to 4 values. Thus, the PSK-DOE technology disclosed in the present invention can be used. In this case, as shown in FIG. 8, the Eb / No-BER characteristic representing the error rate further improves the error rate of the conventional π / 4-DQPSK. In the figure, the received electric field of PSK-DOE necessary to obtain the BER value of the same error rate is about 5 dB lower than that of π / 4-DQPSK under fading, so that it can withstand a bad reception environment. I mean. However, since 3 bits are transmitted in PSK-DOE, more power is radiated by about 1.8 dB (1.5 times = 3/2 times) than 2 bits transmission of π / 4-DQPSK. As a result, subtraction, an improvement of slightly over 3 dB can be seen. Moreover, at this time, the frequency utilization efficiency is 3 bits / Hz, and the frequency utilization efficiency is improved by 1.5 times compared to the conventional π / 4-DQPSK.
[0168]
Furthermore, when the line condition deteriorates, it is possible to stop adding information to the amplitude difference and switch to transmission using only the transfer difference, thereby preventing a reduction in error rate. At this time, the frequency use efficiency is the same as that of the conventional π / 4-DQPSK, but the analysis reception disclosed in the present invention is used for the demodulation method, so that the low-pass filter that is essential in the conventional delay detection circuit is unnecessary. As a result, the error rate is improved without giving phase distortion of the received wave, and the circuit structure is simple and economical.
[0169]
As shown in FIG. 3, the analysis receiving circuit includes a delay unit or a Hilbert transformer having a quarter period time of an intermediate frequency, two delay units having a symbol period T time, four multipliers, and an adder 2. This is realized by a simple circuit using only one circuit, and has a feature that a low-pass filter necessary for conventional delay detection is not required. Even when PSK-DOE or 16DQAM is targeted, add one delay unit or Hilbert transformer of ¼ period time of the intermediate frequency, two delay units of symbol period T time, and four multipliers. This can be realized by a simple circuit including two delay units, one symbol period delay unit, three multipliers, two adders, two dividers and one square root circuit. The advantage of not requiring a low-pass filter that is essential for delay detection is that a phase rotation existing in the vicinity of the cutoff band of the filter is a fatal defect where there are many phase distortions due to fading propagation. In order to avoid phase distortion in the vicinity of the cut-off region of the low-pass filter, it is necessary to use a low-pass filter that extends the pass band to a band that sufficiently exceeds the band in which information exists. The fact that the bandwidth has to be widened is that the adjacent band cannot be used for the frequency band being communicated, and the frequency utilization efficiency is significantly reduced. Furthermore, widening the passband width to avoid phase distortion in the vicinity of the cut-off region of the low-pass filter has a drawback that unnecessary noise is received as much, and a decrease in error rate is induced at the same time.
[0170]
Conventional delay detection includes the same operation as so-called AM modulation in which the product of the received wave and its delay is obtained. The demodulated wave includes both sidebands, and a low-pass filter is inevitably required to remove unnecessary sidebands. I was trying. For this reason, there has been a serious problem that information energy contained in one sideband wave is lost.
[0171]
Further, the low-pass filter has a property of giving a large phase distortion in the vicinity of the cutoff frequency, and gives a large distortion to the phase component of the demodulated signal, thereby deteriorating the error rate of PSK detection. Furthermore, when the cutoff characteristic of the low-pass filter is made steep, the phase distortion near the cutoff frequency increases and the error rate deteriorates. Conversely, if the low-pass filter's cutoff characteristics are slowed down to reduce the phase distortion near the cutoff frequency, the filter's passband frequency width increases, which increases the mixing of noise outside the transmission band and also degrades the error rate. Was increasing. As long as a filter (low-pass filter) is used to remove one sideband from both sidebands, there is an unavoidable disadvantage that the error rate deteriorates.
[0172]
On the other hand, in the analysis reception, as described in the claims, since the signal processing is performed based on the analysis signal theory, it becomes a so-called SSB modulation format, and the demodulated signal includes only a single sideband component. For this reason, the advantage that the demodulated signal for analysis reception contains all the information energy is obtained.
[0173]
In addition, since it is not necessary to remove one side of the sideband, there is an advantage of completely preventing the error rate deterioration caused by the phase distortion increased by the filter operation and the error rate deterioration caused by noise contamination. It is done. For this reason, all information energy is included in the demodulated signal using analysis reception, and noise energy other than information energy is hardly included, so that the error rate is about 3 bits / Hz to 4 bits / Hz with the same error rate as before. The advantage of achieving high frequency utilization efficiency is obtained.
[0174]
Of course, the analysis reception disclosed by the present invention also has the advantage of eliminating the need for carrier acquisition, which is one of the features of conventional delay detection. For this reason, in a communication system that assumes multipath propagation such as mobile communication, it is possible to always keep the DUR at 0 dB or more by demodulating the received wave indicating the maximum power.
[0175]
There is no need to search for a received wave having the maximum power, and the carrier wave can be acquired by simply inputting the received wave after multipath propagation and the received signal frequency-converted to the intermediate frequency or baseband to the receiver disclosed in the present invention. Due to the unnecessary properties, the signal having the maximum power is the dominant term, the DUR is 0 dB or more, and the maximum width of the phase jitter is ± 90 degrees.
[0176]
When conventional carrier acquisition is required, it is necessary to assume that the target wave has a zero amplitude value and the non-target wave has the full amplitude, and the maximum phase jitter is ± π radians = ± 180 degrees.
[0177]
Therefore, in the transmission system of the present invention, the phase jitter is degenerated from the conventional ± 180 degrees = ± 1/2 symbol period to ± 90 degrees = ± 1/4 symbol period, and the bit error rate is greatly reduced, resulting in communication quality. Is greatly improved. As a result, there is an advantage that the amount of information can be communicated 1.5 to 2 times with the same communication energy.
[0178]
In applying the present invention, it is desirable to use the PSK-DOE and E-16 value DQAM of the present invention adaptively according to the line state, rather than fixedly.
[0179]
For example, a PSK-DOE system that applies 16-value DQAM by phase 8-value × amplitude binary modulation according to the present invention when the line condition is the best and performs amplitude modulation on the PSK signal according to the present invention when the line condition deteriorates. Is adopted.
[0180]
If the line condition further deteriorates, the modulation method may be switched between transmission and reception so that the conventional PSK method is adopted. Also, when the line state has recovered, the modulation scheme can be returned in the opposite direction. In addition, when the line condition is further improved, the phase is changed from 8 values to multi-value, and the amplitude is also changed from 2 values to increase the frequency efficiency for communication.
[0181]
FIG. 11 shows an example of the configuration of a modulator placed on the transmission side that can adaptively change the modulation method in accordance with the propagation state of the line in the digital wireless transmission method of the present invention.
[0182]
The serial-parallel bit converter 100 has a function of grouping an input serial bit string into 2 bits, 3 bits, and 4 bits according to the propagation state. The differential phase modulator 101 has a function of performing 4-level or 8-level phase modulation according to the propagation state. The envelope generator 102 performs amplitude modulation according to the propagation state. An example of the operation of the modulator will be described below.
[When propagation is good (1)]
In this case, the differential phase modulator 101 assumes four values. The serial-parallel bit converter 100 groups the input bit string into 3 bits, and sends 2 bits to the differential phase modulator 101 and 1 bit to the envelope generator 102, respectively.
[0183]
The differential phase modulator 101 performs phase modulation that assigns a 2-bit signal sequence to different quaternary symbols every 90 degrees, and outputs the result to the multiplier 103. The envelope generator 102 creates a binary level based on the 1-bit signal and outputs it to the multiplier 103. The multiplier 103 combines the binary level with the quaternary phase modulation output and outputs an 8-level multilevel signal. This is defined as PSK-DOE (PSK-Data on Envelope).
[When the propagation state is better than (1)]
In this case, the differential phase modulator 101 assumes eight values. The serial-to-parallel bit converter 100 groups the input bit string into 4 bits, and sends 3 bits to the differential phase modulator 101 and 1 bit to the envelope generator 102, respectively.
[0184]
The differential phase modulator 101 performs phase modulation that assigns a 3-bit signal sequence to different 8-valued symbols every 45 degrees, and outputs the result to the multiplier 103. The envelope generator 102 creates a binary level based on the 1-bit signal and outputs it to the multiplier 103. Multiplier 103 combines the binary level with the 8-level phase modulation output and outputs a 16-level multilevel signal. This is defined as 16DQAM (Differential Quadrature-Amplitude Modulation).
[When the propagation state is not better than (1)]
In this case, the differentially attacking modulator 101 assumes four values. The serial-parallel bit converter 100 groups the input bit strings into two bits and sends all of them to the differential phase modulator 101.
[0185]
The differential phase modulator 101 performs phase modulation that is assigned to different quaternary symbols every 90 degrees of the 2-bit signal sequence, and outputs the result to the multiplier 103. Nothing is input to the envelope generator 102, and a single level is maintained. As a result, the multiplier 103 outputs the quaternary phase modulation output as it is as a quaternary multilevel signal.
[0186]
As described above, the three types of modulation methods have been described. The selection method is appropriately determined in consideration of the propagation state (determined from the bit error rate, electrolytic strength, etc.) and the specific requirements of the system as described above. To be selected.
[0187]
【The invention's effect】
As described above according to the embodiments, in the digital wireless transmission system according to the present invention, transmission is performed using the same frequency bandwidth without impairing the fading resistance of PSK transmission and without increasing the occupied frequency band. The amount of information can be increased.
[Brief description of the drawings]
FIG. 1 is a diagram illustrating an example in which two sub-symbols ss1 and ss2 are set in one symbol section.
FIG. 2 is a diagram for explaining giving information to an amplitude difference between sub-symbols;
FIG. 3 is a block diagram of an embodiment circuit of the present invention.
4 is a diagram for explaining each component of FIG. 3;
FIG. 5 is a diagram illustrating a configuration example of a Hilbert transformer.
FIG. 6 is a diagram for explaining an example of adding 1-bit information to an π / 4-DQPSK modulated wave (an example of an RZ code).
FIG. 7 is a diagram illustrating an example of adding 1-bit information to an π / 4-DQPSK modulated wave (an example of an NRZ code).
FIG. 8 is a diagram for explaining a BER comparison between the PSK-DOE of the present invention and the conventional π / 4-DQPSK modulation method.
FIG. 9 is a diagram illustrating a bit constellation of 16-value DQAM of phase 8 value-amplitude value 2 according to the present invention.
10 is a diagram for explaining a comparison of BER when the bit constellation of FIG. 9 is used.
FIG. 11 is a diagram illustrating a configuration example of a modulator on the transmission side.
FIG. 12 is a diagram for explaining amplitude distortion in fading of a conventional PSK signal.
FIG. 13 is a diagram for explaining phase jitter in fading of a conventional PSK signal.
[Explanation of symbols]
1 Hilbert converter
20-22 delay device
30-36 multiplier
40-43 adder
60, 61 Divider
50 square meters
100 Serial-parallel bit converter
101 Differential phase modulator
102 Envelope generator
103 multiplier

Claims (4)

In a digital wireless transmission system that keeps a carrier signal in a predetermined phase for a certain period and transmits information,
A first period and a second period following the first period are set in the certain period, and the carrier value in the second period is set using the amplitude value of the carrier signal in the first period as a reference value. A digital wireless transmission system characterized by modulating information of 1 bit or more to a difference between amplitude values of the first period and the second period by increasing or decreasing an amplitude value of a signal. In claim 1,
The digital wireless transmission system characterized in that the predetermined period is an information symbol period. In a digital wireless transmission system that keeps a carrier signal in a predetermined phase for a certain period and transmits information,
One information symbol period and an information symbol period following the one information symbol period are set in the certain period,
As a reference value an amplitude value of the carrier signal of the one information symbol period, and increases the amplitude of the carrier signal of the subsequent information symbol period to the one information symbol period, or by decreasing, the one of information A digital wireless transmission system characterized by modulating information of 1 bit or more to a difference between amplitude values of a symbol period and an information symbol period following the one information symbol period . In a digital wireless transmission system that keeps a carrier signal in a predetermined phase for a certain period and transmits information,
Received signal f (t) at time t and signal obtained by Hilbert transform of the received signal f (t)

The square of the sum of the squares of
One information symbol period and an information symbol period following the one information symbol period are set in the certain period,
A difference between envelope signals corresponding to a time difference between the one information symbol period and a subsequent information symbol period or a plurality of sub-symbol periods obtained by dividing the one information symbol period is demodulated as an amplitude difference signal, Digital wireless transmission method.

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