JP3569726B2 - Apparatus and method for measuring geometric thickness and refractive index of sample - Google Patents

Description

ここで、ＮＡは対物レンズの開口数を表す。
【００１２】
この原理では、試料２０の移動距離Δｚ と、光路長の増加Δｌ の２量を測定する必要がある。試料２０の移動距離Δｚ の測定を行うためには、焦点の検出を行う必要がある。これには次に述べる波長走査型ヘテロダイン干渉法を用いる。
【００１３】
（共焦点顕微鏡の原理）
共焦点顕微鏡は、図１３に示すように点光源１１から発した光を対物レンズ１９を通して試料２０に照射し、試料２０から反射した光を偏光ビームスプリッタ１７によって反射させ、レンズ２３によってピンホール２４上に集光し、その透過光を検出する。ここで、焦点位置が試料表面に一致したときには、ピンホールを通して検出される信号は極大値をとる（図１（ｃ））。逆に、信号がピークをとる位置が、試料表面に焦点があっているときである。
【００１４】
（波長走査型ヘテロダイン干渉法の原理）
波長走査型干渉法は光路差を持たせた干渉計において、光源の波長を一定量だけ連続的に変化させたときに生ずる干渉信号の位相変化を測定し、それより干渉計の光路差を可動部なしで測定する手法である。光源に半導体レーザー（あるいはレーザーダイオード，ＬＤ）を用いた場合、連続的に波長を掃引できる幅が小さいため、位相の変化は小さい。そこで、高分解能で光路差測定を行うためには、位相の測定精度を高くする必要がある。そのため、光ヘテロダイン干渉法を併用した手法（波長走査型ヘテロダイン干渉法と呼ぶ）を用いる。この波長走査型ヘテロダイン干渉法を図２をもとに説明する。
【００１５】
まず、はじめに注入電流変調をかけたＬＤ１１の出射光の電場は、ＬＤ１１からの距離をｚ、時刻をｔとすると次式で表される。
【数２】

ここで、ｃは光速、Ｅ_０ （ｔ）およびＶ（ｔ）は時刻ｔにおけるＬＤ端での電場と光周波数であり、正弦変調の場合には次式を用いて表される。
【００１６】
Ｅ_０ （ｔ）＝Ａ ……（４）
Ｖ（ｔ）＝Ｖ_０ ＋ΔＶｃｏｓ（２πｆ_ｍ ｔ＋φ） ……（５）
ここで、Ａは電場の振幅で簡単のために一定としておく。Ｖ_０ は変調をかけないときの光周波数、ΔＶは周波数変調幅、φは注入電流の変調信号を基準としたときのＶ（ｔ）の初期位相である。
【００１７】
図２を参照すると、ＬＤ１１から出た光は、光周波数シフタ１２によって、たがいに周波数がＶ_ｂ だけ異なる２つの直交直線偏光となる。このうち一方の分離光を参照鏡１５へ向ける参照光として用い、もう一方の分離光を試料２０へ向ける物体光として用いる。検出器２５における参照光および物体光の電場Ｅ_ｒｅｆ ，Ｅ_ｏｂｊ は次のように表される。
【数３】

ここで、α、βは偏光ビームスプリッタ１７の振幅分割比、Ｚ_ｒｅｆ 、Ｚ_ｏｂｊ は参照光および物体光の光路長、Ｖ_ｒｅｆ 、Ｖ_ｏｂｊ は参照光および物体光の周波数で、
Ｖ_ｒｅｆ （ｔ）＝Ｖ（ｔ）＋Ｖ_１ ……（８）
Ｖ_ｏｂｊ （ｔ）＝Ｖ（ｔ）＋Ｖ_１ ＋Ｖ_ｂ ……（９）
で表される。Ｖ_１ およびＶ_１＋Ｖ_ｂは光周波数シフタ１２による光周波数シフト量をあらわし、その差Ｖ_ｂ が観測されるビート周波数に相当する。
【００１８】
これより、検出器２５で得られるビート信号は、
【数４】

となる。
【００１９】
ここで両光の光路差をＬ＝２（Ｚ_ｒｅｆ−Ｚ_ｏｂｊ）としている。これよりビート信号の位相が変調周波数ｆ_ｍ で変調され、その振幅を測定することにより光路差Ｌが求められる。
【００２０】
（共焦点信号と光路差の同時測定）
屈折率・厚さの分離測定では、これら共焦点顕微鏡および波長走査型ヘテロダイン干渉法を用いて、焦点検出と光路長測定を行う。従来の測定手法では、これら２つの光学系が１つの光学系に組み込まれており、それぞれを切り替えて測定していた。すなわち、参照光路に入れたシャッターを閉じることによって、まず、光学系を共焦点系にし、試料を光軸方向に移動させ、その共焦点系の信号からピーク位置、すなわち界面に焦点が合う位置を計算する。次に、そのピーク位置に試料を移動させ、シャッターを開放することにより光学系を波長走査型ヘテロダイン干渉計に切り替えて、光路長測定を行う。これをそれぞれのピーク位置に対して順に行っていた。
【００２１】
この方法では、一度取得した共焦点信号ピーク位置に試料を再移動させた後、それぞれの位置で光路差を測定するため時間がかかる。さらに、機械的なステージを用いるので、バックラッシュの影響がさけられず、その影響を低減するために、一度、ステージを機械原点に戻してからピーク位置に移動させていた。これにより、さらに時間がかかることになる。
【００２２】
本発明は試料２０の光軸方向の移動時に、共焦点系の情報と、波長走査型ヘテロダイン干渉計による光路長情報とを同時に取得するものである。これにより、試料の光軸方向の走査は１回で済むので測定時間の短縮につながり、またバックラッシュの問題も解決される。
【００２３】
本発明においては、検出器２５で検出されるビート信号から共焦点プロファイルに相当する強度情報と、波長走査型ヘテロダイン干渉計に相当する位相変調率とを分離することが必要となる。
共焦点信号および光路差信号の分離
（検出される干渉信号の解析式）
まず、はじめに検出されるビート信号を解析的に求め、次に分離手法を検討する。解析のモデルは図２に示す光学系である。
【００２４】
図２の説明中、簡単のために光強度は一定であるとしたが、実際には周波数変調とともに強度変調もかかる。そこでＬＤの強度変調を表現するには、（４）式の電場Ｅ_０ （ｔ）を
【数５】

で置き換える。ただし、ｍは強度の変調率である。さらに、試料２０の反射率はその位置によらずつねに一定としたが、実際には共焦点原理により、試料位置によって検出光の強度が変化する。そこで、この効果を取り入れるために、等価的に試料の反射率ｐ（ｚ）が位置によって変化するとする。すなわち、βをβｐ（ｚ）で置き換えればよい。これらを考慮すると検出されるビート信号は、
【数６】

となる。
【００２５】
図３にこのビート信号の周波数成分を図示する。図３において、それぞれの振幅は試料位置の関数になっている。これより、ビート信号は周波数がゼロの成分、変調周波数と同じ周波数成分、ビート周波数の成分、およびその両側帯波成分からなっている。ただし、位相変調もかかっているのでこの側帯波は広がる（図には示していない）。ここで、全ての信号成分の振幅がピークとなる試料位置Ｚ_０ が、試料の表面に焦点があっているときに相当する。これらの信号から位相変調率と振幅を独立に抽出する。
【００２６】
（位相変調率（光路差信号）の測定法）
まず、位相計（Ｓｔａｎｆｏｒｄ Ｒｅｓｅａｒｃｈ．ＳＲ８５０）２８（図９参照）に振幅変調がかかった信号を入力したときの影響を調べる。任意波形発生器（Ｈｅｗｌｅｔｔ−Ｐａｃｋａｒｄ，ＨＰ８９０４Ａ）から２つの疑似信号を発生させる。１つは周波数１００ｋＨｚ、振幅１Ｖ_ｐ−ｐ の正弦波でこれを参照信号として用いる。また、もう１つは周波数１００ｋＨｚ、振幅１Ｖ_ｐ−ｐ の正弦波に振幅変調（変調周波数０．１Ｈｚ）をかける。これらの信号を位相計２８に入力し、出力信号のうち変調周波数と同じ周波数で変化する信号の振幅をオシロスコープで読み取る。振幅変調率を０から１００％まで１０％ずつ変化させたときの結果を図４に示す。
【００２７】
図４において、縦軸の１０Ｖが位相の１８０°に対応する。これより、振幅変調率が大きくなるにつれて位相計の出力も大きくなるが線形な関係にはない。したがって、位相計において正確な位相変調率の測定を行うためには、振幅変調がかかっていない波形を用いる必要である。
【００２８】
そこで、振幅変調および位相変調がかかった信号から、振幅変調を除去するために、ゼロクロスコンパレータ２７（図９参照）を用いて波形を方形波に整形する。
【００２９】
図５（ａ）（ｂ）に振幅変調（周波数１００ｋＨｚ、振幅１Ｖ_ｐ−ｐ 、振幅変調率３０％、変調周波数８０Ｈｚ）がかかった信号をゼロクロスコンパレータに通した前後の波形を示す。これより、ゼロクロスコンパレータにより振幅変調が除去され、振幅が一定の方形波に整形されていることがわかる。同様に位相変調（周波数１００ｋＨｚ、振幅１Ｖ_ｐ−ｐ 、位相変調率３０°、変調周波数８０Ｈｚ）の信号の様子も図５（ｃ）（ｄ）に示す。これより、整形後の波形は、振幅が一定の方形波でかつ位相変調が保存されていることがわかる。
【００３０】
ここで図５（ａ）−（ｄ）において参照信号を各々の上部に示す。また図５（ａ）の下部は振幅変調がかかった信号を示し、図５（ｂ）の下部はこれを整形した信号を示す。また図５（ｃ）の下部は位相変調がかかった信号を示し、図５（ｄ）の下部はこれを整形した信号を示す。
【００３１】
位相変調率を変え、ゼロクロスコンパレータ通過後の位相変調の幅をオシロスコープから直接読み取った測定結果を図６に示す。これより、ゼロクロスコンパレータ前後で位相変調率は比例関係にあることがわかる。
【００３２】
（振幅（共焦点信号）の測定法）
図３のいずれかの周波数の信号から、試料２０の各位置に対する振幅の大きさを求める必要がある。そこで、まずバイアスおよび変調周波数の信号から振幅を得ることを考える。これらはローパスフィルタあるいはバンドパスフィルタを用いることにより抽出することができる。また、この信号には位相変調がかかっていないという利点もある。しかし、試料を動かしたときの振幅の変化は大きくない。これと比較して、ビート周波数およびその側帯波の信号は、変化が大きくより効率的に検出できるものと考えられる。そこで、ビート周波数のみを抽出することを考えられる。しかし、実際の変調周波数は８０Ｈｚで、ビート周波数は１００ｋＨｚであるので、側帯波は１００ｋＨｚに対して８０Ｈｚしか離れていない。さらに、位相変調もかかっているので、周波数差はより小さくなるものと考えられる。したがって、側帯波からキャリア周波数成分を抽出するのは非常に難しくなる。したがって、側帯波を含めた信号から、振幅変調および位相変調の影響を受けずに、干渉信号の振幅のみを抽出することが必要である。
【００３３】
（ＲＭＳ−ＤＣ変換器による振幅の抽出）
干渉信号の振幅情報の抽出には、変調周期より十分長い時間で信号を積分すれば、振幅変調および位相変調は、ともに平均化され、その影響は抑圧されると考え、信号の積分にＲＭＳ−ＤＣ変換器３２（図９参照）を用いる。
【００３４】
まず、ＲＭＳ−ＤＣ変換器（ＡＤ７３６、アナログ・デバイセズ）３２に振幅変調および位相変調がかかった信号を入力し、その特性を調べる。周波数１００ｋＨｚ、振幅１．０Ｖ_ｐ−ｐ の正弦波をＲＭＳ−ＤＣ変換器に入力し、出力電圧を２０ｄＢに増幅して測定する。入力信号は、振幅を０．１Ｖ_ｐ−ｐ から０．５Ｖ_ｐ−ｐ まで０．１Ｖ_ｐ−ｐ ずつ変化させる。また、振幅変調率は、０％から１００％まで５％ずつ変化させ、さらに、位相変調率は０°から１８０°まで５°ずつ変化させる。測定例として振幅が０．１Ｖ_ｐ−ｐ の時の結果を図７に示す。これより、ＲＭＳ−ＤＣ変換器３２の出力電圧は入力信号の位相変調率には依存しないことがわかる。しかし、振幅変調率を大きくすると、出力電圧も若干大きくなっていくことがわかる。
【００３５】
そこで、入力信号の振幅の大きさに対するＲＭＳ−ＤＣ変換器３２の出力特性を詳しく調べるために、振幅を１０ｍＶ_ｐ−ｐ から５００ｍＶ_ｐ−ｐ まで、１０ｍＶ_ｐ−ｐ ずつ変化させて同様な測定を行なう。ここで位相変調はかけていないことにする。
【００３６】
図８に測定結果を示す。図８において振幅は５０ｍＶ_ｐ−ｐ おきに示す。図８に示すように、振幅が４００ｍＶ_ｐ−ｐ までは、振幅変調率の増加にともなって、出力電圧も増加していくが、振幅が４００ｍＶ_ｐ−ｐ を過ぎると、この傾向は逆転する。しかし、同じ振幅変調率に関して、振幅が大きいほど出力も大きく、この関係が崩れることはないことがわかる。したがって、ＲＭＳ−ＤＣ変換器３２を用いて、干渉信号の振幅を抽出する場合、同じ振幅でも、得られる値は振幅変調率によって異なる。しかし、実際の測定では振幅変調率は一定であり、また、共焦点信号のピーク位置が特定できればよいので、ＲＭＳ−ＤＣ変換器３２を干渉信号の振幅の抽出に用いることができる。
具体的実施の形態
次に図９により上記基本原理を用いた具体的実施の形態について説明する。
【００３７】
図９に示すように、本発明による測定装置は試料２０に対して周波数可変光を投光する光源としての半導体レーザダイオード（ＬＤ）１１と、ＬＤ１１からの光を偏光の相違により一対の分離光に分けるとともに各分離光の周波数を変化させる周波数シフタ１２と、周波数シフタ１２と試料２０との間に配置された偏光ビームスプリッタ１７とを備えている。また、偏光ビームスプリッタ１７の試料２０側には１／４波長板（ｑｕａｔｅｒ−ｗａｖｅ ｐｌａｔｅ）１８と対物レンズ１９が設けられている。
【００３８】
また周波数シフタ１２と試料２０との間の光路と交差するとともに偏光ビームスプリッタ１７を通る直線上において、偏光ビームスプリッタ１７の一側に１／４波長板（ｑｕａｒｔｅｒ−ｗａｖｅ ｐｌａｔｅ）１６および参照鏡１５が順次配置され、偏光ビームスプリッタ１７の他側に偏光子２２およびレンズ２３を介してピンホール２４と光電子増倍管（検出器）２５が順次配置されている。
【００３９】
さらにこの光電子増倍管２５には、ハイパスフィルタ２６、ゼロクロスコンパレータ２７および位相計２８が順次接続され、また位相計２８にはロックインアンプ２９が接続されている。またハイパスフィルタ２６にはアンプ３１、ＲＭＳ−ＤＣ変換器（平方自乗平均−直流変換器）３２、およびアンプ３３が順次接続され、このアンプ３３とロックインアンプ２９は演算部３５に接続されている。
【００４０】
また演算部３５はステージ制御部３６に接続され、このステージ制御部３６は試料２０を保持する移動ステージ２１を駆動制御するようになっている。
【００４１】
なお、上記構成部分のうち、ＲＭＳ−ＤＣ変換器３２により振幅検出部が構成され、ゼロクロスコンパレータ２７、位相計２８およびロックインアンプ２９により位相変調率信号検出部が構成される。
【００４２】
また位相計２８と周波数シフタ１２との間には、ローパスフィルタ４０、二重平衡変調器４１および高周波信号発生器４２ａ，４２ｂが配設されている。さらに周波数シフタ１２は、偏光ビームスプリッタ１２ａ，１２ｂと、音響光学素子１２ｃ，１２ｄとを有している。
【００４３】
またロックインアンプ２９には、ファンクションジェネレータ３７が接続され、このファンクションジェネレータ３７はＬＤ駆動部３８を介してＬＤ１１を駆動制御するようになっている。
【００４４】
次に図９により本発明による測定方法について説明する。
【００４５】
図９に示すように、半導体レーザー１１（ＴＯＳＨＩＢＡ，ＴＯＬＤ−９１４０，λ＝６８８ｎｍ＠４０ｍＡ、１０ｍＷ）は、ファンクションジェネレータ３８からの信号により変調周波数ｆ_ｍ ＝８０Ｈｚの正弦波で注入電流変調される。これによりＬＤ１１の発振周波数ｖ（ｔ）および電場Ｅ（ｔ）は、それぞれ式（５）（１１）で示すように変調される。ＬＤ素子の温度はペルチェ素子により１８．００±０．０１℃に安定化されている。
【００４６】
ＬＤ１１からの出射光はレンズ５０，５１により平行光（ビーム径１ｍｍ）にされ、直交偏波型光周波数シフタ（ＨＯＹＡ、Ｓ−２１０−６３３）１２に入射する。周波数シフタ１２内で入射光は偏光の違いによって２つに分けられ、異なる周波数（８０ＭＨｚと８０．１ＭＨｚ）により駆動される２つの音響光学素子１２ｃ，１２ｄにより、各分離光は周波数シフト（周波数の変化）を受け、再び合波される。これにより出射光は互いに偏光方向が直交し、かつ周波数差Ｖ_ｂ が１００ｋＨｚの同軸の一対の分離光となる。この光をレンズ１３，１４により光束を拡大し偏光ビームスプリッタ１７に導入する。
【００４７】
偏光ビームスプリッタ１７で直交する直線偏光に二分された光のうち、一方は対物レンズ１９を通して試料２０を照明し、他方は参照鏡１５を照明する参照光として用いる。
【００４８】
ここで、もし他方の光の焦点が試料２０内のある界面に合っているとすると、試料２０の界面および参照鏡１５からの反射光は１／４波長板１６，１８を往復して通過することによりそれぞれ偏光方向は９０°回転し、偏光偏光ビームスプリッタ１７で再び合波され、偏光子２２により偏光干渉される。さらにレンズ２３でピンホール２４上に集光し、光電子増倍管２５（浜松ホトニクス、光センサモジュールＨ５７８３−０１）により検出される。
【００４９】
ところで、光周波数シフタ１２を駆動する高周波信号発生器４２ａ，４２ｂからの信号が分波され、この信号により２重平衡変調器４１およびローパスフィルタ４０を用いて、周波数１００ｋＨｚの信号を発生させている。これを位相計２８の基準ビート信号として用いることにより、従来必要となっていたＬＤ光およびＨｅ−Ｎｅ光とを混合・分離するための光学素子が不要となり、光学系の簡素化とともにＬＤ光の利用効率も向上する。
【００５０】
光電子増倍管２５により検出したビート信号は、ハイパスフィルタ２６により、バイアス成分が取り除かれ、位相変調率測定用の回路と振幅測定用の回路とに分岐される。
【００５１】
まず、位相変調率測定用の回路では、光電子増倍管２５により検出した信号はゼロクロスコンパレータ２７によって方形波に変換され、振幅変調成分が取り除かれる。この信号は位相計（Ｓｔａｎｆｏｒｄ Ｒｅｓｅａｒｃｈ Ｓｙｓｔｅｍｓ，Ｄｉｇｉｔａｌ Ｌｏｃｋ−Ｉｎ Ａｍｐｌｉｆｉｅｒ，ＳＲ８５０）２８に入力される。位相計２８では二重平衡変換器４１を用いて発生した基準ビート信号と、ゼロクロスコンパレータ２７からの信号の位相を比較して、その差に比例した電圧を出力する。さらにこの出力信号は、後続のロックインアンプ（エヌエフ回路設計ブロック、Ｌｏｃｋ−Ｉｎ Ｖｏｌｔｍｅｔｅｒ、５５６０）２９に入力される。ロックインアンプ２９の参照信号にはファンクションジェネレーター３７からの信号を用いているので、位相計２８の出力のうちＬＤ１１の変調周波数と同一の周波数成分の信号の振幅と位相とを測定することができる。ロックインアンプ２９からの測定値は演算部３５に取り込まれる。
【００５２】
次に、振幅測定用の回路では、光電子増倍管２５により検出した信号はアンプ３１によって適当に減衰され、ＲＭＳ−ＤＣ変換器３２に入力される。次にこの直流出力信号はアンプ３３で増幅され、演算部３５に取り込まれる。
【００５３】
試料２０を載せたステージ２１は演算部３５およびステージ制御部３６によって制御されており、各試料２０の位置において位相変調率と振幅を記録していく。さらに取得したデータの解析はオフラインで行なわれる。
【００５４】
次に演算部３５における演算方法について述べる。波長走査型ヘテロダイン干渉法による光路長差測定では、可変波長幅が既知でなければならない。しかし、波長計を用いて動作状態の波長を測定は困難であるため、ここでは予め実際に既知の光路長を測定し、それより位相変調率と光路差との関係を求める。
【００５５】
すなわち予めマイケルソン干渉計で手動ステージにのせた基準となる鏡を光軸方向に移動させ位相変調率を測定しておく。測定結果を図１０に示す。図１０に示すように鏡の移動に伴って、測定値（○印）は線形に変化している。この傾きを求めると、０．５８９ｍＶ／ｍｍとなる。これは、１６．０１°／ｍｍに相当する。また、ゼロ光路差（４．３ｍｍ）の前後で位相（□印）が反転することにより光路差の符号の判定を行なう。そして図１０に示す光路差と位相変調率との関係に基づいて、位相変調率から光路長の増加分を求めることができる。
【００５６】
ところで平面鏡を、光軸方向に走査することによって検出器の深度応答を測定する。まず図１１（ａ）（ｂ）により、本発明による干渉信号の振幅から求めた結果（図１１（ａ））と、比較例としての参照光を遮ることによって強度から求めた結果（図１１（ｂ））をそれぞれに示す。図１１（ａ）に示す本発明のほうが、幅が広がっているが、この原因は、（１２）式から予想されるとおり、干渉信号の振幅から求めた深度応答は、強度から求めた深度応答の平方根を取った形となるからであり、また、ＲＭＳ−ＤＣ変換器３２の入力信号の振幅に対する非線形性の影響も含まれていると考えられる。これらの結果から半値半幅を求めると、それぞれ１７μｍ、１３μｍとなり、また、双方のピーク位置の違いは１μｍ程度である。
【００５７】
次に、演算部３５において、屈折率・厚さの分離測定を行なう。試料２０として例えば平行平面基板（ＢＫ７ガラス、厚さ１０８４μｍ、屈折率１．５１３）を用いた場合におけるＲＭＳ−ＤＣ変換器３２で測定した干渉信号の振幅を図１２（ａ）に示し、また位相計２８の出力信号を図１２（ｂ）に示す。
【００５８】
図１２（ａ）に示すピークは試料２０の表面・裏面に焦点が合ったときに相当している。また、図１２（ｂ）では、図１２（ａ）のピーク位置の付近しか値が求まっていない。これは、検出された干渉信号がゼロクロスコンパレータ２７のしきい値より下回るので、入力波形が正確に方形波に整形されず、位相計２８での測定が不能となるためである。
【００５９】
図１２（ａ）から、まずピーク間隔に基づいて光の焦点が試料２０の一面から他面へ移るのに必要な試料の移動距離を求める。次に位相計２８およびロックインアンプ２９により得られた図１２（ｂ）に示す位相変調率に基づいて、図１０に示す光路長と位相変調率との関係により光路の増加分を求める。
【００６０】
その後、式（１）−（２）を用いて試料２０の屈折率と幾何学的厚さを算出すると１．５２±０．０２、１１７０±１３μｍとなる。
【００６１】
ところで位相計２８で測定できる範囲は−１８０°から１８０°までの範囲である。位相変調率（位相変調の位相の振幅）がこの範囲に収まっているときは、測定は問題ない。しかし、位相変調率が±１８０°の外に出ると測定は不能となる。これが位相変調率（すなわち光路差）の最大測定レンジとなる。ただし、位相変調率がこれより小さくても位相のバイアス分（初期位相）の値によっては、途中で、１８０°を上回り、あるいは、−１８０°を下回ってしまうことも考えられる。これを回避するためにこの初期位相を例えば０、９０、１８０、２７０°と４回位相をシフトさせ、位相変調率の測定を行なうことにより測定不能状態を回避することができる。
【００６２】
【発明の効果】
以上のように本発明によれば、試料の幾何学厚さおよび屈折率を迅速かつ確実に求めることができる。
【図面の簡単な説明】
【図１】本発明の測定原理を示す図。
【図２】波長走査型ヘテロダイン干渉法の測定原理を示す基本構成図。
【図３】ビート信号の周波数成分を示す図。
【図４】振幅変調率と位相計出力との関係を示す図。
【図５】ゼロクロスコンパレータを通過前後の信号波形を示す図。
【図６】ゼロクロスコンパレータ通過後の位相変調の幅を示す図。
【図７】ＲＭＳ−ＤＣ変換器の出力電圧を示す図。
【図８】ＲＭＳ−ＤＣ変換器の出力電圧を示す図。
【図９】本発明による試料の幾何学的厚さおよび屈折率測定装置を示す概略図。
【図１０】鏡の変位と位相変調率の関係を示す図。
【図１１】試料の変位と検出器の出力の関係を示す図。
【図１２】ＲＭＳ−ＤＣ変換器と位相計の出力を示す図。
【図１３】共焦点顕微鏡の基本原理を示す図。
【符号の説明】
１１ ＬＤ
１２ 周波数シフタ
１５ 参照鏡
１７ 偏光ビームスプリッタ
１９ 対物レンズ
２０ 試料
２１ ステージ
２４ ピンホール
２５ 検出器
２７ ゼロクロスコンパレータ
２８ 位相計
２９ ロックインアンプ
３２ ＲＭＣ−ＤＣ変換器
３５ 演算部
３６ ステージ制御部[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to an apparatus and a method for measuring the geometric thickness and refractive index of a sample for determining the geometric thickness and the refractive index of the sample.
[0002]
[Prior art]
Conventionally, low coherence interferometry has been used to measure the optical thickness of a transparent sample such as a thin film or optical glass. Low coherence interferometry utilizes the fact that an interference fringe appears only when the optical path difference between both arms of the interferometer is near zero by using a low-coherence light source such as white light or a light emitting diode in the interferometer. In this method, the absolute position of the measurement object is known from the position of the reference mirror at that time. This is applied to absolute measurement of a block gauge, calibration of a base line, surface shape measurement, and the like. Further, in recent years, an extended method has been actively studied in the fields of ophthalmology and biological science.
[0003]
On the other hand, there is known a confocal laser microscope in which a laser beam is spot-irradiated onto a sample and reflected light or fluorescence from the sample is re-imaged on a point detector.
[0004]
This confocal microscope can not only obtain high-contrast images compared to conventional optical microscopes, but also has a high resolution in the optical axis direction and can construct three-dimensional images. Has established.
[0005]
The optical thickness of the transparent sample can be measured by utilizing the resolution of the confocal microscope in the optical axis direction.
[0006]
[Problems to be solved by the invention]
However, the quantity directly obtained by the confocal principle is the optical thickness, and the refractive index of each layer must be obtained by another method in order to obtain the geometric thickness. Such a problem similarly occurs in the thickness measurement in the low coherence interferometry described above. However, the fact is that a method for quickly and surely measuring the refractive index of a molded sample has not yet been developed.
[0007]
The present invention has been made in view of such a point, and a sample geometric thickness and refractive index measuring device capable of quickly and reliably determining the sample geometric thickness and refractive index, and a device therefor. It is intended to provide a measuring method.
[0008]
[Means for Solving the Problems]
The present invention provides a light source that emits frequency-variable light to a sample, and separates the light from the light source into a pair of separated lights according to a difference in polarization, and changes the frequency of each separated light by a predetermined shift amount. A frequency shifter, a polarizing beam splitter disposed in an optical path between the frequency shifter and the sample, and one of the polarizing beam splitters intersecting the optical path between the frequency shifter and the sample and on a straight line passing through the polarizing beam splitter. The reference beam arranged on the other side of the beam splitter, the separated light reflected from the reference mirror passing through the polarizing beam splitter on the other side of the beam splitter, and the other separated light reflected from the sample passing through the polarizing beam splitter and reflected from the sample A detector for detecting the generated interference signal, an amplitude detector for obtaining the amplitude based on the interference signal from the detector, and an amplitude detector for detecting the interference signal from the detector. A phase modulation rate signal detector for obtaining a phase modulation rate signal, and based on the amplitude from the amplitude detector, determine a moving distance of the sample necessary for the focus of the other separated light to move from one surface of the sample to the other surface. The amount of shift of the frequency of each separated light and the amount of increase in the optical path length of the other separated light when the focus shifts from one surface of the sample to the other surface based on the phase modulation rate signal from the phase modulation signal detection unit, An arithmetic unit for calculating the geometric thickness and the refractive index of the sample based on the increase of the moving distance and the optical path length, and a measuring device for measuring the geometric thickness and the refractive index of the sample. .
In the measuring method using the geometric thickness and refractive index measuring device described above, a step of moving the sample relative to the frequency shifter, and one of the light from the light source separated by the frequency shifter The light is reflected to the reference mirror via the polarizing beam splitter, and the other separated light separated by the frequency shifter is reflected to the sample via the polarizing beam splitter, and the pair of separated lights interfere with each other as an interference signal by a detector. Detecting, detecting an amplitude by an amplitude detector based on an interference signal from the detector, obtaining a phase modulation rate signal by a phase modulation rate signal detector based on the interference signal from the detector, and calculating In the unit, based on the amplitude from the amplitude detection unit, determine the moving distance of the sample necessary for the focus of the other separated light to move from one surface of the sample to the other surface The amount of shift of the frequency of each separated light and the increase in the optical path length of the other separated light when the focal point shifts from one surface of the sample to the other surface are determined based on the phase modulation ratio signal from the phase modulation ratio detection unit. Obtaining the geometric thickness and the refractive index of the sample based on the increment of the moving distance and the optical path length.
[0009]
According to the present invention, the amplitude detection section obtains the amplitude based on the interference signal from the detector, and the phase modulation rate signal detection section obtains the phase modulation rate signal based on the interference signal from the detector. In the calculation unit, based on the amplitude from the amplitude detection unit, while determining the moving distance of the sample required to shift the focus of the other separated light from one surface of the sample to the other surface, and the amount of frequency shift of each separated light, Based on the phase modulation rate signal from the phase modulation rate signal detection unit, an increase in the optical path of the other separated light when the focus shifts from one surface to the other surface of the sample is obtained. The geometric thickness and the refractive index of the sample are determined based on the moving distance and the increase in the optical path.
[0010]
BEST MODE FOR CARRYING OUT THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
[0011]
First, the basic principle of the present invention will be described.
Measurement principle
(Separation measurement of refractive index and thickness)
FIG. 1 shows the principle of separation and measurement of refractive index and thickness. Here, a single-layer sample is described for simplicity, but application to a multilayer is easy. The sample 20 is irradiated with parallel light through the objective lens 19. When the sample 20 is moved closer to the objective lens 19, first, one surface of the sample 20 (a surface close to the objective lens 19) is focused (FIG. 1A). Further, when the distance is brought closer, the optical path length from the objective lens 19 to the focal position increases (FIG. 1 (d)), and then the other surface of the sample 20 is focused (FIG. 1 (b)). Here, assuming that the moving distance of the sample 20 required to shift the focal position from one surface of the sample 20 to the other surface is Δz, and the increase in the optical path length at that time is Δl, the refractive index n of the sample 20 and The geometric thickness d is calculated using the equations (1) and (2).
(Equation 1)

Here, NA represents the numerical aperture of the objective lens.
[0012]
According to this principle, it is necessary to measure two amounts of the moving distance Δz of the sample 20 and the increase Δl of the optical path length. In order to measure the moving distance Δz of the sample 20, it is necessary to detect the focus. For this, a wavelength scanning type heterodyne interferometry described below is used.
[0013]
(Principle of confocal microscope)
As shown in FIG. 13, the confocal microscope irradiates the sample 20 with light emitted from the point light source 11 through the objective lens 19, reflects the light reflected from the sample 20 by the polarization beam splitter 17, The light is collected on the upper side and the transmitted light is detected. Here, when the focal position coincides with the sample surface, the signal detected through the pinhole takes a maximum value (FIG. 1C). Conversely, the position where the signal peaks is when the sample surface is focused.
[0014]
(Principle of wavelength scanning heterodyne interferometry)
Wavelength scanning interferometry measures the phase change of an interference signal that occurs when the wavelength of a light source is continuously changed by a fixed amount in an interferometer with an optical path difference, and the optical path difference of the interferometer can be moved. This is a method of measuring without a part. When a semiconductor laser (or laser diode, LD) is used as the light source, the change in phase is small because the width over which the wavelength can be continuously swept is small. Therefore, in order to perform optical path difference measurement with high resolution, it is necessary to increase the phase measurement accuracy. Therefore, a technique that uses optical heterodyne interferometry in combination (referred to as wavelength-scanning heterodyne interferometry) is used. This wavelength scanning type heterodyne interferometry will be described with reference to FIG.
[0015]
First, the electric field of the light emitted from the LD 11 that has been subjected to the injection current modulation is represented by the following equation, where z is the distance from the LD 11 and t is the time.
(Equation 2)

Where c is the speed of light, E₀(T) and V (t) are the electric field and the optical frequency at the LD end at time t, and are expressed by the following equations in the case of sine modulation.
[0016]
E₀(T) = A (4)
V (t) = V₀+ ΔVcos (2πf_mt + φ) (5)
Here, A is constant at the amplitude of the electric field for simplicity. V₀Is the optical frequency when no modulation is applied, ΔV is the frequency modulation width, and φ is the initial phase of V (t) with reference to the modulation signal of the injection current.
[0017]
Referring to FIG. 2, the light emitted from the LD 11 has a frequency of V_bAnd two orthogonal linearly polarized lights that differ only by One of the separated lights is used as reference light directed to the reference mirror 15, and the other is used as object light directed to the sample 20. Electric field E of reference light and object light in detector 25_ref, E_objIs expressed as follows.
(Equation 3)

Here, α and β are the amplitude division ratios of the polarization beam splitter 17, and Z_ref, Z_objIs the optical path length of the reference light and the object light, V_ref, V_objIs the frequency of the reference light and the object light,
V_ref(T) = V (t) + V₁              …… (8)
V_obj(T) = V (t) + V₁+ V_b        …… (9)
Is represented by V₁And V₁+ V_bRepresents the optical frequency shift amount by the optical frequency shifter 12, and the difference V_bCorresponds to the observed beat frequency.
[0018]
Thus, the beat signal obtained by the detector 25 is
(Equation 4)

It becomes.
[0019]
Here, the optical path difference between the two lights is L = 2 (Z_ref-Z_obj) To be. From this, the phase of the beat signal becomes the modulation frequency f_mThe optical path difference L is obtained by measuring the amplitude.
[0020]
(Simultaneous measurement of confocal signal and optical path difference)
In the separation measurement of refractive index and thickness, focus detection and optical path length measurement are performed using these confocal microscopes and wavelength scanning heterodyne interferometry. In a conventional measurement method, these two optical systems are incorporated in one optical system, and each is switched to perform measurement. That is, by closing the shutter placed in the reference optical path, first, the optical system is made a confocal system, the sample is moved in the optical axis direction, and the peak position, that is, the position where the interface is focused on is determined from the signal of the confocal system. calculate. Next, the sample is moved to the peak position, the shutter is opened, and the optical system is switched to a wavelength scanning heterodyne interferometer, and the optical path length is measured. This was performed sequentially for each peak position.
[0021]
In this method, it takes a long time to move the sample to the confocal signal peak position once obtained and then measure the optical path difference at each position. Furthermore, since a mechanical stage is used, the effects of backlash cannot be avoided, and in order to reduce the effects, the stage is once returned to the mechanical origin and then moved to the peak position. This requires more time.
[0022]
According to the present invention, when the sample 20 moves in the optical axis direction, information of the confocal system and optical path length information by the wavelength scanning heterodyne interferometer are simultaneously obtained. As a result, only one scan of the sample in the optical axis direction is required, which leads to a reduction in measurement time, and the problem of backlash is also solved.
[0023]
In the present invention, it is necessary to separate the intensity information corresponding to the confocal profile and the phase modulation rate corresponding to the wavelength scanning type heterodyne interferometer from the beat signal detected by the detector 25.
Separation of confocal signal and optical path difference signal
(Analysis formula of detected interference signal)
First, the detected beat signal is analytically obtained, and then a separation method is examined. The analysis model is the optical system shown in FIG.
[0024]
In the description of FIG. 2, the light intensity is assumed to be constant for the sake of simplicity, but in practice, intensity modulation is applied together with frequency modulation. Therefore, to express the intensity modulation of the LD, the electric field E of the equation (4) is used.₀(T)
(Equation 5)

Replace with Here, m is the intensity modulation rate. Further, the reflectance of the sample 20 is always constant irrespective of its position, but actually, the intensity of the detection light changes according to the sample position according to the confocal principle. Therefore, in order to incorporate this effect, it is assumed that the reflectance p (z) of the sample equivalently changes depending on the position. That is, β may be replaced with βp (z). Considering these, the detected beat signal is
(Equation 6)

It becomes.
[0025]
FIG. 3 shows the frequency components of the beat signal. In FIG. 3, each amplitude is a function of the sample position. Thus, the beat signal is composed of a component having a frequency of zero, a frequency component equal to the modulation frequency, a component of the beat frequency, and a band component on both sides thereof. However, since side modulation is also applied, the sideband spreads (not shown). Here, the sample position Z where the amplitudes of all the signal components peak.₀Corresponds to when the surface of the sample is in focus. The phase modulation rate and the amplitude are independently extracted from these signals.
[0026]
(Method of measuring phase modulation rate (optical path difference signal))
First, the effect of inputting a signal subjected to amplitude modulation to a phase meter (Stanford Research. SR850) 28 (see FIG. 9) will be examined. Two pseudo signals are generated from an arbitrary waveform generator (Hewlett-Packard, HP8904A). One is 100kHz frequency, 1V amplitude_ppAnd this is used as a reference signal. The other one has a frequency of 100 kHz and an amplitude of 1 V_ppIs subjected to amplitude modulation (modulation frequency 0.1 Hz). These signals are input to the phase meter 28, and the amplitude of the output signal that changes at the same frequency as the modulation frequency is read by an oscilloscope. FIG. 4 shows the results when the amplitude modulation rate is changed from 0 to 100% in steps of 10%.
[0027]
In FIG. 4, 10 V on the vertical axis corresponds to 180 ° of the phase. As a result, the output of the phase meter increases as the amplitude modulation rate increases, but is not in a linear relationship. Therefore, in order to accurately measure the phase modulation rate in the phase meter, it is necessary to use a waveform that is not subjected to amplitude modulation.
[0028]
Therefore, in order to remove the amplitude modulation from the signal subjected to the amplitude modulation and the phase modulation, the waveform is shaped into a square wave using the zero cross comparator 27 (see FIG. 9).
[0029]
FIGS. 5A and 5B show amplitude modulation (frequency 100 kHz, amplitude 1 V)._pp, Amplitude modulation rate 30%, modulation frequency 80 Hz) before and after passing through a zero-cross comparator. From this, it can be seen that the amplitude modulation is removed by the zero-cross comparator, and the amplitude is shaped into a constant square wave. Similarly, phase modulation (frequency 100 kHz, amplitude 1 V_pp5 (c) and 5 (d) also show the state of a signal having a phase modulation rate of 30 ° and a modulation frequency of 80 Hz. This indicates that the shaped waveform is a square wave having a constant amplitude and preserves the phase modulation.
[0030]
Here, reference signals are shown above each of FIGS. The lower part of FIG. 5A shows a signal subjected to amplitude modulation, and the lower part of FIG. 5B shows a signal obtained by shaping the signal. The lower part of FIG. 5 (c) shows a signal subjected to phase modulation, and the lower part of FIG. 5 (d) shows a signal obtained by shaping the signal.
[0031]
FIG. 6 shows a measurement result obtained by directly reading from the oscilloscope the width of the phase modulation after passing through the zero cross comparator while changing the phase modulation rate. This indicates that the phase modulation ratio is in a proportional relationship before and after the zero cross comparator.
[0032]
(Method of measuring amplitude (confocal signal))
It is necessary to determine the magnitude of the amplitude for each position of the sample 20 from the signal of any frequency in FIG. Therefore, first, consider obtaining the amplitude from the bias and modulation frequency signals. These can be extracted by using a low-pass filter or a band-pass filter. Another advantage is that the signal is not phase modulated. However, the change in amplitude when the sample is moved is not large. Compared to this, it is considered that the beat frequency and its sideband signal have a large change and can be detected more efficiently. Therefore, it is conceivable to extract only the beat frequency. However, since the actual modulation frequency is 80 Hz and the beat frequency is 100 kHz, the sideband is only 80 Hz apart from 100 kHz. Further, since the phase modulation is applied, the frequency difference is considered to be smaller. Therefore, it is very difficult to extract the carrier frequency component from the sideband. Therefore, it is necessary to extract only the amplitude of the interference signal from the signal including the sideband without being affected by the amplitude modulation and the phase modulation.
[0033]
(Extraction of amplitude by RMS-DC converter)
In order to extract the amplitude information of the interference signal, if the signal is integrated for a time sufficiently longer than the modulation period, both the amplitude modulation and the phase modulation will be averaged, and the effects thereof will be suppressed. A DC converter 32 (see FIG. 9) is used.
[0034]
First, a signal subjected to amplitude modulation and phase modulation is input to an RMS-DC converter (AD736, Analog Devices) 32, and its characteristics are examined. Frequency 100kHz, amplitude 1.0V_ppIs input to the RMS-DC converter, and the output voltage is amplified to 20 dB for measurement. The input signal has an amplitude of 0.1 V_ppFrom 0.5V_ppUp to 0.1V_ppChange by one. Further, the amplitude modulation rate is changed by 5% from 0% to 100%, and the phase modulation rate is changed by 5 ° from 0 ° to 180 °. As an example of measurement, amplitude is 0.1V_ppFIG. 7 shows the results at the time. This indicates that the output voltage of the RMS-DC converter 32 does not depend on the phase modulation rate of the input signal. However, when the amplitude modulation rate is increased, the output voltage is slightly increased.
[0035]
Therefore, in order to examine in detail the output characteristics of the RMS-DC converter 32 with respect to the amplitude of the input signal, the amplitude is set to 10 mV._ppFrom 500mV_ppUp to 10mV_ppThe same measurement is performed while changing each time. Here, it is assumed that no phase modulation is applied.
[0036]
FIG. 8 shows the measurement results. In FIG. 8, the amplitude is 50 mV._ppShown every other day. As shown in FIG. 8, the amplitude is 400 mV._ppUntil, the output voltage also increases as the amplitude modulation rate increases, but the amplitude is 400 mV._ppAfter, this trend reverses. However, for the same amplitude modulation rate, the larger the amplitude, the larger the output, and it can be seen that this relationship is not broken. Therefore, when the amplitude of the interference signal is extracted using the RMS-DC converter 32, the obtained value differs depending on the amplitude modulation rate even with the same amplitude. However, in the actual measurement, the amplitude modulation rate is constant and the peak position of the confocal signal need only be specified, so that the RMS-DC converter 32 can be used for extracting the amplitude of the interference signal.
Specific embodiments
Next, a specific embodiment using the above basic principle will be described with reference to FIG.
[0037]
As shown in FIG. 9, a measuring apparatus according to the present invention includes a semiconductor laser diode (LD) 11 as a light source for projecting frequency-variable light to a sample 20, and a pair of separated light beams based on a difference in polarization. A frequency shifter 12 for changing the frequency of each separated light, and a polarization beam splitter 17 disposed between the frequency shifter 12 and the sample 20. A quarter-wave plate 18 and an objective lens 19 are provided on the sample 20 side of the polarization beam splitter 17.
[0038]
Further, on a straight line passing through the polarizing beam splitter 17 while intersecting the optical path between the frequency shifter 12 and the sample 20, a quarter-wave plate 16 and a reference mirror 15 are provided on one side of the polarizing beam splitter 17. Are sequentially arranged, and a pinhole 24 and a photomultiplier tube (detector) 25 are sequentially arranged on the other side of the polarization beam splitter 17 via a polarizer 22 and a lens 23.
[0039]
Further, a high-pass filter 26, a zero-cross comparator 27, and a phase meter 28 are sequentially connected to the photomultiplier tube 25, and a lock-in amplifier 29 is connected to the phase meter 28. An amplifier 31, an RMS-DC converter (root mean square-DC converter) 32, and an amplifier 33 are sequentially connected to the high-pass filter 26, and the amplifier 33 and the lock-in amplifier 29 are connected to an arithmetic unit 35. .
[0040]
The computing unit 35 is connected to a stage control unit 36, which controls the drive of the moving stage 21 that holds the sample 20.
[0041]
Note that, of the above components, the RMS-DC converter 32 forms an amplitude detection unit, and the zero-cross comparator 27, the phase meter 28, and the lock-in amplifier 29 form a phase modulation rate signal detection unit.
[0042]
A low-pass filter 40, a double balanced modulator 41, and high-frequency signal generators 42a and 42b are provided between the phase meter 28 and the frequency shifter 12. Further, the frequency shifter 12 has polarization beam splitters 12a and 12b and acousto-optic elements 12c and 12d.
[0043]
Further, a function generator 37 is connected to the lock-in amplifier 29, and the function generator 37 drives and controls the LD 11 via an LD driver 38.
[0044]
Next, the measuring method according to the present invention will be described with reference to FIG.
[0045]
As shown in FIG. 9, the semiconductor laser 11 (TOSHIBA, TOLD-9140, λ = 688 nm ＠ 40 mA, 10 mW) modulates the modulation frequency f by a signal from the function generator 38._m= 80 Hz sine wave for injection current modulation. Thereby, the oscillation frequency v (t) and the electric field E (t) of the LD 11 are modulated as shown by the equations (5) and (11). The temperature of the LD element is stabilized at 18.00 ± 0.01 ° C. by the Peltier element.
[0046]
Light emitted from the LD 11 is converted into parallel light (beam diameter: 1 mm) by the lenses 50 and 51, and enters the orthogonal polarization type optical frequency shifter (HOYA, S-210-633) 12. In the frequency shifter 12, the incident light is divided into two by the difference in polarization, and the two separated acousto-optic elements 12c and 12d driven at different frequencies (80 MHz and 80.1 MHz) cause each separated light to undergo a frequency shift (frequency shift). Change) and are multiplexed again. As a result, the emitted lights have polarization directions orthogonal to each other and the frequency difference V_bIs a pair of separated light beams of 100 kHz coaxial. The light is expanded by the lenses 13 and 14 and introduced into the polarization beam splitter 17.
[0047]
Of the light split into two linearly polarized lights orthogonal to each other by the polarizing beam splitter 17, one illuminates the sample 20 through the objective lens 19, and the other is used as reference light for illuminating the reference mirror 15.
[0048]
Here, if the other light is focused on an interface in the sample 20, the reflected light from the interface of the sample 20 and the reflected light from the reference mirror 15 reciprocate through the quarter-wave plates 16 and 18. As a result, the polarization directions are rotated by 90 °, respectively, are multiplexed again by the polarization beam splitter 17, and are polarized by the polarizer 22. Further, the light is condensed on the pinhole 24 by the lens 23 and is detected by the photomultiplier tube 25 (Hamamatsu Photonics, photosensor module H5783-01).
[0049]
By the way, the signals from the high-frequency signal generators 42a and 42b for driving the optical frequency shifter 12 are demultiplexed, and the double-balanced modulator 41 and the low-pass filter 40 are used to generate a signal having a frequency of 100 kHz. . By using this as a reference beat signal of the phase meter 28, an optical element for mixing and separating LD light and He-Ne light, which has been conventionally required, becomes unnecessary. Usage efficiency also improves.
[0050]
The beat signal detected by the photomultiplier tube 25 has its bias component removed by a high-pass filter 26, and is branched into a circuit for measuring a phase modulation factor and a circuit for measuring an amplitude.
[0051]
First, in the circuit for measuring the phase modulation rate, the signal detected by the photomultiplier tube 25 is converted into a square wave by the zero cross comparator 27, and the amplitude modulation component is removed. This signal is input to a phase meter (Stanford Research Systems, Digital Lock-In Amplifier, SR850) 28. The phase meter 28 compares the phase of the reference beat signal generated by using the double balance converter 41 with the phase of the signal from the zero cross comparator 27, and outputs a voltage proportional to the difference. Further, this output signal is input to a subsequent lock-in amplifier (NF circuit design block, Lock-In Voltmeter, 5560) 29. Since the signal from the function generator 37 is used as the reference signal of the lock-in amplifier 29, the amplitude and phase of the signal of the same frequency component as the modulation frequency of the LD 11 in the output of the phase meter 28 can be measured. . The measurement value from the lock-in amplifier 29 is taken into the calculation unit 35.
[0052]
Next, in the circuit for amplitude measurement, the signal detected by the photomultiplier tube 25 is appropriately attenuated by the amplifier 31 and input to the RMS-DC converter 32. Next, this DC output signal is amplified by the amplifier 33 and taken into the arithmetic unit 35.
[0053]
The stage 21 on which the samples 20 are placed is controlled by the calculation unit 35 and the stage control unit 36, and records the phase modulation rate and the amplitude at the position of each sample 20. Further analysis of the acquired data is performed off-line.
[0054]
Next, a calculation method in the calculation unit 35 will be described. In the measurement of the optical path length difference by the wavelength scanning heterodyne interferometry, the variable wavelength width must be known. However, since it is difficult to measure the operating wavelength using a wavelength meter, a known optical path length is actually measured in advance, and the relationship between the phase modulation factor and the optical path difference is determined therefrom.
[0055]
That is, a reference mirror placed on a manual stage by a Michelson interferometer is moved in the optical axis direction in advance to measure the phase modulation rate. FIG. 10 shows the measurement results. As shown in FIG. 10, the measured value (circle) changes linearly with the movement of the mirror. When this inclination is obtained, it becomes 0.589 mV / mm. This corresponds to 16.01 ° / mm. Further, the sign of the optical path difference is determined by reversing the phase (marked by squares) before and after the zero optical path difference (4.3 mm). Then, based on the relationship between the optical path difference and the phase modulation rate shown in FIG. 10, an increase in the optical path length can be obtained from the phase modulation rate.
[0056]
By the way, the depth response of the detector is measured by scanning the plane mirror in the optical axis direction. First, referring to FIGS. 11A and 11B, a result obtained from the amplitude of the interference signal according to the present invention (FIG. 11A) and a result obtained from the intensity by blocking the reference light as a comparative example (FIG. b)) are shown respectively. The reason for this is that the depth response obtained from the amplitude of the interference signal is different from the depth response obtained from the intensity, as expected from equation (12). It is considered that the square root of RMS-DC converter 32 is included, and the influence of nonlinearity on the amplitude of the input signal of the RMS-DC converter 32 is included. When the half width at half maximum is obtained from these results, they are 17 μm and 13 μm, respectively, and the difference between both peak positions is about 1 μm.
[0057]
Next, the calculation unit 35 performs separation measurement of the refractive index and the thickness. FIG. 12A shows the amplitude of the interference signal measured by the RMS-DC converter 32 when a parallel plane substrate (BK7 glass, thickness 1084 μm, refractive index 1.513) is used as the sample 20, for example. The output signals of the total 28 are shown in FIG.
[0058]
The peak shown in FIG. 12A corresponds to when the front and back surfaces of the sample 20 are focused. Further, in FIG. 12B, values are obtained only near the peak position in FIG. 12A. This is because the detected interference signal is lower than the threshold value of the zero-cross comparator 27, so that the input waveform is not accurately shaped into a square wave, and the measurement by the phase meter 28 becomes impossible.
[0059]
From FIG. 12A, first, the moving distance of the sample necessary for the focal point of light to move from one surface of the sample 20 to the other surface is obtained based on the peak interval. Next, based on the phase modulation rate shown in FIG. 12B obtained by the phase meter 28 and the lock-in amplifier 29, the increment of the optical path is obtained from the relationship between the optical path length and the phase modulation rate shown in FIG.
[0060]
Thereafter, when the refractive index and the geometric thickness of the sample 20 are calculated using the equations (1) and (2), they are 1.52 ± 0.02 and 1170 ± 13 μm.
[0061]
The range that can be measured by the phase meter 28 is a range from -180 ° to 180 °. When the phase modulation rate (phase modulation phase amplitude) is within this range, there is no problem in measurement. However, when the phase modulation rate goes out of ± 180 °, the measurement becomes impossible. This is the maximum measurement range of the phase modulation rate (that is, optical path difference). However, even if the phase modulation rate is smaller than this, depending on the value of the phase bias (initial phase), it is conceivable that the phase modulation rate may exceed 180 ° or fall below -180 ° on the way. In order to avoid this, the initial phase is shifted four times, for example, 0, 90, 180, and 270 °, and the phase modulation rate is measured, whereby the unmeasurable state can be avoided.
[0062]
【The invention's effect】
As described above, according to the present invention, the geometric thickness and the refractive index of a sample can be quickly and reliably obtained.
[Brief description of the drawings]
FIG. 1 is a diagram showing a measurement principle of the present invention.
FIG. 2 is a basic configuration diagram showing a measurement principle of the wavelength scanning type heterodyne interferometry.
FIG. 3 is a diagram showing a frequency component of a beat signal.
FIG. 4 is a diagram showing a relationship between an amplitude modulation rate and a phase meter output.
FIG. 5 is a diagram showing signal waveforms before and after passing through a zero cross comparator.
FIG. 6 is a diagram showing the width of phase modulation after passing through a zero cross comparator.
FIG. 7 is a diagram showing an output voltage of an RMS-DC converter.
FIG. 8 is a diagram showing an output voltage of the RMS-DC converter.
FIG. 9 is a schematic diagram illustrating a device for measuring the geometric thickness and refractive index of a sample according to the present invention.
FIG. 10 is a diagram showing a relationship between a displacement of a mirror and a phase modulation factor.
FIG. 11 is a diagram showing a relationship between a displacement of a sample and an output of a detector.
FIG. 12 is a diagram showing outputs of an RMS-DC converter and a phase meter.
FIG. 13 is a diagram showing the basic principle of a confocal microscope.
[Explanation of symbols]
11 LD
12 Frequency shifter
15 Reference mirror
17 Polarizing beam splitter
19 Objective lens
20 samples
21 stages
24 pinhole
25 Detector
27 Zero Cross Comparator
28 Phase meter
29 Lock-in amplifier
32 RMC-DC converter
35 Calculation unit
36 Stage control unit

Claims (6)

A light source for projecting a variable frequency light to the sample,
A frequency shifter that divides the light from the light source into a pair of separated lights by a difference in polarization, and changes the frequency of each separated light by a predetermined shift amount,
A polarizing beam splitter disposed in an optical path between the frequency shifter and the sample;
On a straight line that crosses the optical path between the frequency shifter and the sample and passes through the polarizing beam splitter, the reference mirror disposed on one side of the polarizing beam splitter passes through the polarizing beam splitter disposed on the other side of the polarizing beam splitter. One of the separated lights reflected from the reference mirror, and a detector that detects an interference signal generated by interference between the other separated light reflected from the sample through the polarizing beam splitter,
An amplitude detection unit for obtaining an amplitude based on an interference signal from the detector,
A phase modulation rate signal detection unit that obtains a phase modulation rate signal based on an interference signal from the detector,
Based on the amplitude from the amplitude detector, the moving distance of the sample required for the focal point of the other separated light to move from one surface of the sample to the other surface is determined, the shift amount of the frequency of each separated light, and the phase modulation signal. The amount of increase in the optical path length of the other separated light when the focal point shifts from one surface to the other surface of the sample is obtained based on the phase modulation rate signal from the detection unit, and the amount of the sample An arithmetic unit for determining a geometric thickness and a refractive index;
An apparatus for measuring the geometric thickness and refractive index of a sample, comprising: 2. The sample geometry according to claim 1, wherein the calculation unit obtains the increase in the optical path from the phase modulation rate signal based on a preset relational expression between the phase modulation rate signal and the increase in the optical path length. Thickness and refractive index measuring device. 2. The apparatus for measuring the geometric thickness and refractive index of a sample according to claim 1, wherein the sample is held by a stage which can be freely moved toward and away from the frequency shifter. In a measuring method using the geometric thickness and refractive index measuring device according to claim 1,
Moving the sample relative to the frequency shifter;
Of the light from the light source, one of the separated lights separated by the frequency shifter is reflected to the reference mirror via the polarizing beam splitter, and the other separated light separated by the frequency shifter is reflected to the sample via the polarizing beam splitter, A step of causing the detector to detect the interference light by causing the pair of separated lights to interfere with each other;
Obtaining an amplitude by an amplitude detector based on an interference signal from the detector;
Obtaining a phase modulation rate signal by a phase modulation rate signal detection unit based on an interference signal from the detector,
In the calculation unit, based on the amplitude from the amplitude detection unit, while determining the moving distance of the sample required to shift the focus of the other separated light from one surface of the sample to the other surface, and the amount of frequency shift of each separated light, Based on the phase modulation rate signal from the phase modulation rate detection unit, an increase in the optical path length of the other separated light when the focus shifts from one surface of the sample to the other surface is determined, and based on this movement distance and the increase in the optical path length Determining the geometric thickness and refractive index of the sample by
A measurement method, comprising: 4. The measuring method according to claim 3, wherein an increase in the optical path length is obtained from the phase modulation rate signal based on a preset relational expression of the phase modulation rate signal and the increase in the optical path. 4. The measuring method according to claim 3, wherein the sample is held by a stage which moves freely relative to the frequency shifter, and the stage moves the sample with respect to the frequency shifter.

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