JP3307155B2 - High frequency filter design method and high frequency filter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION The present invention mainly relates to a VHF band,
The present invention relates to widening the bandwidth of a high-frequency filter used in the UHF band, the microwave band, and the millimeter wave band.

[0002]

2. Description of the Related Art FIGS. 34 to 35 show S. Herbert, "Microstr.
ip Interdigital Filter Design, ”Microwave Journal
, pp. 187-188, Nov. 1990, or JAGMalherbe,
“Microwave Transmission Line Filters”, ARTECH H
OUSE, INC, pp.111-130, 1979. is a diagram showing the design procedure of a conventional high-frequency filter. FIG. 34 is a design flow, FIG. 35 is a circuit of a low-pass filter, and FIG. It is an equivalent circuit of a bandpass filter.
In FIG. 35, Ci '(i = 1,3) and Li' (i = 2,4) are the elements of the original low-pass filter, G0 'is the power supply impedance, G5' is the load impedance, and gi (i = 0,3). 1,…,
5) is a coefficient determined by the transfer function of the filter.
In FIG. 36, Li (i = 1,..., 4) is a parallel inductance, Ji, i + 1 (i, = 1,2,3) is a J inverter, and δ0
And δ5 are the impedance transformation ratios. Li and J
i and i + 1 are near the frequency where θ = π / 2 and are represented by equivalent circuits of the distributed constant lines shown in FIGS. 36 (b) and (c).

Next, the design procedure will be described. First, the element value of the original low-pass filter is determined from the desired pass band width, the ripple in the pass band, the number of resonator stages, and the coefficient of the Chebyshev series used as the transfer function.
Next, frequency conversion is performed on the original low-pass filter, and a J-inverter is introduced to convert it into an equivalent circuit of the band-pass filter shown in FIG. At this time J
The inverter and the parallel inductance are shown in FIG.
As shown in (c), the filter is replaced with a distributed constant type equivalent circuit using a tip short-circuit stub having an electrical length θ approximately near the center frequency of the filter. A conventional interdigital band-pass filter is designed using this distributed constant type equivalent circuit.

A high-frequency filter is generally constituted by a resonator having a distributed constant line, and the frequency characteristics can be calculated relatively accurately by an equivalent circuit shown in FIG. The reflection characteristic of the conventional high-frequency filter shown in FIG. 36 has a large error due to the approximation of FIGS. 36 (b) and (c) at frequencies away from the center frequency. As shown, the reflection loss increases at the pass band edge.

FIGS. 38 to 40 show RECollin, “F
oundations for Microwave Engineering ”, McGraw-Hi
ll, Inc., pp. 639-642, 1979. or another conventional high-frequency filter design procedure disclosed in Japanese Patent Application Laid-Open No. 6-216608. FIG. 38 is a design flow, and FIG. FIGS. 40 and 41 are equivalent circuits of a waveguide band-pass filter. In FIG. 39, Ci '(i = 1,3, ..., 11) and Li', (i = 2,4, ..., 10)
Is the element of the original low-pass filter, G0 'is the source impedance, G5' is the load impedance, gi, (i = 0,1,
.., 12) are coefficients determined by the transfer function of the filter. In FIG. 40A, Li (i = 1, 2,..., 1
1) is a series inductance, Ci (i = 1,2, ..., 11) is a series capacitance, and Ki, i + 1 (i, = 1,2,3) is a K inverter. In the vicinity of the frequency where θ = π, Ki, j
Is represented by a parallel susceptance Bi, j shown in FIG. Bi, j is realized, for example, by an inductive iris. A series resonance circuit composed of Li and Ci is represented by a waveguide circuit having an electrical length θi.

Next, the design procedure will be described. First, the element value of the original low-pass filter is determined from a desired pass bandwidth, ripple in the pass band, the number of resonator stages, and a coefficient obtained by series expansion of a Chebyshev function used as a transfer function. Next, frequency conversion is performed on the original low-pass filter, and further, a K inverter is introduced to convert to an equivalent circuit of the band-pass filter shown in FIG. At this time, near the center frequency of the filter,
As shown in FIGS. 40 (a) and (b), Ki, j is derived by approximating the parallel susceptance Bi, j, and approximating the series resonant circuit composed of Li and Ci by the waveguide circuit having the electrical length θi. The waveguide type bandpass filter is represented by a distributed constant type equivalent circuit shown in FIG. A conventional waveguide bandpass filter is designed using this distributed constant type equivalent circuit.

The reflection characteristic of the conventional waveguide band-pass filter shown in FIG. 41 shows that the error due to the approximation shown in FIGS. In particular, when the pass band is wide, the reflection loss increases at the end of the pass band as shown by the solid line in FIG. In particular, in a waveguide band-pass filter, the dispersion of the wavelength in the waveguide with respect to the frequency change is large, and the ratio of the electrical length change to the frequency change is large. Deterioration of characteristics becomes remarkable.

FIGS. 43 and 44 show RECollin,
“Foundations for Microwave Engineering”, McGra
FIG. 43 is a diagram showing a design procedure of a conventional impedance transformer shown in w-Hill, Inc., pp. 343-360, 1979.
Is a design flow, and FIG. 44 is an equivalent circuit. In FIG. 44, Zi (i = 1, 2,..., 4) is the characteristic impedance of the distributed constant line, Z0 is the characteristic impedance of the input line, ZL 'is the characteristic impedance of the output line, and θ is the electrical length of the distributed constant line. is there.

Next, the design procedure will be described. First, the number of stages of the impedance transformer is defined, and the expression of the reflection coefficient of the impedance transformer is obtained using the equivalent circuit of FIG. Assuming that the electric length of each distributed constant line is set equal and that the reflection at the connection surface between adjacent distributed constant lines is sufficiently small, the reflection coefficient is expanded into a polynomial using a trigonometric function. At this time, the reflection coefficient in the pass band is accurately expressed using the Chebyshev function if the pass band width and the number and magnitude of the ripples in the pass band are defined. The Chebyshev function can be expanded by a trigonometric function, and the characteristic impedance of each distributed constant line of the impedance transformer can be obtained by comparing the Chebyshev function with a polynomial of the reflection coefficient obtained from the equivalent circuit of FIG. At this time, the line length of the distributed constant line is set so that the electrical length θ is π / 2 at the center frequency.

In an actual impedance transformer,
Since physical steps such as conductor width occur at the connection between adjacent distributed constant lines with different characteristic impedances,
This discontinuity causes a susceptance. This susceptance has the function of equivalently changing the electric length of the distributed constant line. In a normal design, the distributed constant line is so controlled that the total electric length including this electric length change becomes π / 2 at the design center frequency. Determine the physical length of However, since the electrical length due to this susceptance differs in frequency from the electrical length of the distributed constant line, the reflection coefficient of the impedance transformer based on the equivalent circuit including the discontinuous susceptance component is based on the Chebyshev function at frequencies away from the center frequency. Shift. Accordingly, the reflection characteristic of the conventional impedance transformer shown in FIG. 44 shows that the reflection loss is large at the pass band edge as shown by the solid line in FIG. 45 at a frequency away from the center frequency as in the case of the filters of FIGS. Become.

[0011]

Since the conventional high-frequency filter design method and high-frequency filter are as described above, at a frequency distant from the center frequency of the design, the reflection characteristic of the equivalent circuit corresponding to the physical shape is reduced by the transfer function. However, there has been a problem that the characteristic of reflection is deteriorated from the ideal reflection characteristic obtained from the above, and therefore, a characteristic of low reflection over a wide band cannot be obtained.

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and it is an object of the present invention to provide a design method capable of reducing reflection over a wide band and to obtain a high-frequency filter having small reflection over a wide band by using this design method. Aim.

[0013]

According to a first aspect of the present invention, there is provided a method for designing a high-frequency filter, comprising the steps of:
Inter-resonator coupling means for coupling the resonators to each other;
In a method for designing a high-frequency filter including input / output coupling means for mutually coupling the resonator and the input / output line,
Resonator length of the resonator, dimensions of the inter-resonator coupling means,
And selecting frequencies equal to or more than the number of parameters for determining the size of the input / output coupling means, and adjusting the resonance of the resonator so that the pass characteristic or the reflection characteristic of the high frequency filter approaches a desired value at the selected frequency. The length of the device, the dimensions of the inter-resonator coupling means, and the dimensions of the input / output coupling means are determined.

A method of designing a high-frequency filter according to a second aspect of the present invention is a method of designing a high-frequency filter, comprising: a plurality of transmission line resonators; inter-resonator coupling means for mutually coupling the resonators; And a input / output coupling means for mutually coupling the filter and the input / output coupling means, wherein a plurality of slope zeros of a Chebyshev function as a function of a frequency of a radio wave in the resonator are set in a pass band of the high frequency filter. The frequency corresponding to the plurality of slope zeros is defined as a reflection zero frequency or a reflection maximum frequency, and the resonator length of the resonator so that the reflection becomes a minimum or a maximum at the reflection zero frequency or the reflection maximum frequency. The dimensions of the resonator coupling means and the dimensions of the input / output coupling means are determined.

In a third aspect of the present invention, there is provided a method for designing a high-frequency filter, comprising: a plurality of transmission line resonators; inter-resonator coupling means for mutually coupling the resonators; In the method of designing a high-frequency filter comprising input and output coupling means for mutually coupling the Chebyshev function as a function of the frequency of the radio wave in the resonator in the pass band of the high-frequency filter, a plurality of zeros, A frequency corresponding to the plurality of zeros is defined as a reflection zero frequency, and the length of the resonator of the resonator, the dimensions of the coupling means between the resonators, and the input / output between the input and output are set so that the reflection is minimized at the reflection zero frequency. The dimensions of the coupling means are determined.

A method of designing a high-frequency filter according to a fourth aspect of the present invention is a method of designing a high-frequency filter, comprising: a plurality of transmission line resonators; inter-resonator coupling means for mutually coupling the resonators; In the method of designing a high-frequency filter comprising input and output coupling means for mutually coupling the Chebyshev function as a function of the frequency of the radio wave in the resonator in the pass band of the high-frequency filter, a plurality of zeros, A frequency corresponding to the plurality of zeros is defined as a reflection zero frequency, and the length of the resonator of the resonator, the dimensions of the coupling means between the resonators, and the input / output between the input and output are set so that the reflection is minimized at the reflection zero frequency. The size of the coupling means is determined, and among the reflection characteristics of the high-frequency filter obtained using the above-mentioned dimensions, a frequency having a larger reflection than that of an ideal Chebyshev filter is selected. The resonator length of the resonator, the dimensions of the inter-resonator coupling means, and the dimensions of the input / output coupling means such that the reflection is less than or equal to a predetermined magnitude at the large reflection frequency and the zero reflection frequency. It has been decided.

In a fifth aspect of the present invention, there is provided a method for designing a high-frequency filter, comprising: a plurality of transmission line resonators; inter-resonator coupling means for mutually coupling the resonators; In the method for designing a high-frequency filter including input / output coupling means for mutually coupling the maximum and the minimum points of the Chebyshev function as a function of the frequency of the radio wave in the resonator, within a pass band of the high-frequency filter. ,
The frequency corresponding to the plurality of local maximum points is defined as a reflective local maximum frequency, and the resonator length of the resonator, the dimension of the resonator-to-resonator coupling means, such that the reflection at the reflective local maximum frequency has a predetermined local maximum value, and The dimensions of the input / output coupling means are determined.

A method for designing a high-frequency filter according to a sixth aspect of the present invention is a method of designing a high-frequency filter, comprising: a plurality of transmission line resonators; inter-resonator coupling means for mutually coupling the resonators; In the method for designing a high-frequency filter including input / output coupling means for mutually coupling the maximum and the minimum points of the Chebyshev function as a function of the frequency of the radio wave in the resonator, within a pass band of the high-frequency filter. ,
The frequency corresponding to the plurality of local maximum points is defined as a reflective local maximum frequency, and the resonator length of the resonator, the dimension of the resonator-to-resonator coupling means, such that the reflection at the reflective local maximum frequency has a predetermined local maximum value, and Determine the size of the input / output coupling means, and select a frequency having a large reflection from the characteristics of the ideal Chebyshev filter among the reflection characteristics of the high-frequency filter obtained using the size, The resonator length of the resonator, the dimensions of the inter-resonator coupling means, and the dimensions of the input / output coupling means are determined so that the reflection is equal to or less than a predetermined magnitude at the reflection maximum frequency.

A method for designing a high-frequency filter according to a seventh aspect of the present invention is a method of designing a high-frequency filter comprising: a plurality of cavity resonators each comprising a waveguide; and a susceptance element serving as a coupling means for the cavity resonators and an input / output coupling means. In the method for designing a waveguide type high frequency filter provided, a plurality of zeros of a Chebyshev function as a function of a guide wavelength of the waveguide are set in a pass band of the waveguide type high frequency filter, and the plurality of zeros are set. The frequency corresponding to the guide wavelength at the zero point is defined as the reflection zero frequency, and the resonator length of the resonator, the susceptance element value, and the dimensions of the susceptance element are determined so that the reflection is minimized at the reflection zero frequency. It was done.

Further, a method of designing a high-frequency filter according to an eighth aspect of the present invention is a method of designing a plurality of cavity resonators each comprising a waveguide and a susceptance element as a means for coupling the cavity resonators to each other and an input / output coupling means. In the method for designing a waveguide type high frequency filter provided, a plurality of zeros of a Chebyshev function as a function of a guide wavelength of the waveguide are set in a pass band of the waveguide type high frequency filter, and the plurality of zeros are set. The frequency corresponding to the guide wavelength at the zero point is defined as the reflection zero frequency, and the resonator length of the resonator, the susceptance element value, and the dimensions of the susceptance element are determined so that the reflection is minimized at the reflection zero frequency. Then, among the reflection characteristics of the waveguide filter obtained using the above dimensions, a frequency having a larger reflection than that of an ideal Chebyshev filter is selected. Resonator length of the resonator as reflected in large frequency and the reflection zero frequency of the reflection is less than or equal to a predetermined size, the susceptance element value, and is obtained by determining the dimension of the susceptance elements.

A ninth aspect of the present invention provides a method of designing a high-frequency filter, comprising: a plurality of cavity resonators each comprising a waveguide; and a susceptance element serving as a coupling means between the cavity resonators and an input / output coupling means. In the method for designing a waveguide-type high-frequency filter provided, a plurality of local maximum points of Chebyshev function as a function of the guide wavelength of the waveguide are set in a pass band of the waveguide-type high-frequency filter. The frequency corresponding to the guide wavelength of the maximum point of the tube is defined as the reflection maximum frequency, the resonator length of the resonator, the susceptance element value, and the susceptance so that the reflection has a predetermined maximum value at the reflection maximum frequency. The dimensions of the element are determined.

A tenth aspect of the present invention provides a method for designing a high-frequency filter, comprising: a plurality of cavity resonators each comprising a waveguide; and a susceptance element serving as a coupling means for the cavity resonators and an input / output coupling means. In the method for designing a waveguide-type high-frequency filter provided, a plurality of local maximum points of Chebyshev function as a function of the guide wavelength of the waveguide are set in a pass band of the waveguide-type high-frequency filter. The frequency corresponding to the guide wavelength of the maximum point of the tube is defined as the reflection maximum frequency, the resonator length of the resonator, the susceptance element value, and the susceptance so that the reflection has a predetermined maximum value at the reflection maximum frequency. The dimensions of the element are determined, and among the reflection characteristics of the waveguide type filter obtained using the above dimensions, the reflection characteristics are more than those of the ideal Chebyshev type filter. The resonator length of the resonator, the susceptance element value, and the dimensions of the susceptance element were determined so that the reflection was not more than a predetermined magnitude at the large reflection frequency and the maximum reflection frequency. Things.

A high frequency filter designing method according to an eleventh aspect of the present invention is the high frequency filter designing method according to any one of claims 7 to 10, wherein the number of resonators is N and the center frequency of the filter is N. When the susceptance value of the i-th susceptance element determined from the equivalent circuit element value is B0i, the susceptance value Bi of the i-th susceptance element is set to a range given by the following equation. 1 <Bi / B0i <1.2 (1 ≦ i ≦ 0.2N, N ≦ i ≦ N + 1) 0.8 <Bi / B0i <1 (0.2N <i ≦ 0.4N, 0.8N ≦ i < N) 0.9 <Bi / B0i ≦ 1 (0.4N <i <0.8N)

According to a twelfth aspect of the present invention, there is provided a high frequency filter configured by the method for designing a high frequency filter according to any one of claims 7 to 11, wherein the susceptance element is an inductive iris or It is an inductive post.

[0025]

According to the first aspect of the present invention, the frequencies are selected by the number of parameters which are equal to or greater than the number of parameters for determining the resonator length of the resonator, the dimension of the inter-resonator coupling means, and the dimension of the input / output coupling means as design parameters. The resonator length of the resonator, the dimensions of the inter-resonator coupling means, and the dimensions of the input / output coupling means are determined so that the pass characteristic or the reflection characteristic of the high frequency filter approaches a desired value at the selected frequency. Therefore, not only the center frequency but also a plurality of frequencies within the selected frequency range are used to set a conditional expression equal to or more than the number of the above parameters. The value of the obtained design parameter can be determined.

In the second invention, a plurality of slope zeros of the Chebyshev function as a function of the frequency of the radio wave in the resonator are set in the pass band of the high frequency filter, and the frequencies corresponding to the plurality of slope zeros are reflected. A resonator length, dimensions of coupling means between resonators, and coupling between input and output, which are defined as a zero frequency or a reflection maximum frequency, and are design parameters so that reflection is minimum or maximum at the reflection zero frequency or the reflection maximum frequency. Since the size of the means is determined, the increase in reflection at the end of the pass band and the deviation of the reflection zero frequency are small, and the desired reflection characteristic close to the ideal characteristic by the Chebyshev function at all the frequencies in the pass band by an optimization method or the like. Can be determined.

In the third invention, a plurality of zeros of a Chebyshev function as a function of the frequency of the radio wave in the resonator are set in a pass band of the high frequency filter, and a frequency corresponding to the plurality of zeros is set to a reflection zero frequency. And the dimensions of the resonator length, dimensions of the coupling means between resonators, and dimensions of the coupling means between the input and output are determined so that the reflection is minimized at the reflection zero frequency. The value of the design parameter that can obtain the desired reflection characteristic close to the ideal characteristic by the Chebyshev function at all frequencies in the passband can be determined by the used optimization technique or the like.

In the fourth invention, a plurality of zeros of the Chebyshev function as a function of the frequency of the radio wave in the resonator are set in a pass band of the high-frequency filter, and a frequency corresponding to the plurality of zeros is defined as a reflection zero frequency. The length of the resonator, the dimensions of the coupling means between the resonators, and the dimensions of the coupling means between the input and output are determined so that the reflection becomes a minimum value at the reflection zero frequency. Among the reflection characteristics of the high-frequency filter, a frequency having a large reflection is selected from the characteristics of an ideal Chebyshev-type filter, and the resonance is performed so that the reflection is equal to or less than a predetermined magnitude at the large reflection frequency and the zero reflection frequency. Since the resonator length of the resonator, the dimensions of the inter-resonator coupling means, and the dimensions of the input-output coupling means were determined, optimization was performed a plurality of times, The optimization method or the like using a frequency of the small limit can determine the value of the design parameters obtained of the desired reflection characteristics very close to the ideal characteristics of Chebyshev function at all frequencies within the passband.

In the fifth invention, a plurality of maximum points of the Chebyshev function as a function of the frequency of the radio wave in the resonator are set in the pass band of the high-frequency filter, and the frequencies corresponding to the plurality of maximum points are reflected by the reflection maximum. Since the length of the resonator, the dimensions of the resonator-to-resonator coupling means, and the dimensions of the input-output coupling means were determined as design parameters so that the reflection would have a predetermined maximum value at the reflection maximum frequency, the minimum The value of the design parameter that can obtain a desired reflection characteristic close to the ideal characteristic by the Chebyshev function at all frequencies within the passband can be determined by an optimization method using a limited frequency or the like.

In the sixth invention, a plurality of maximum points of the Chebyshev function as a function of the frequency of the radio wave in the resonator are set in the pass band of the high-frequency filter, and the frequencies corresponding to the plurality of maximum points are reflected by the reflection maximum. The resonator length, the dimension of the resonator coupling means, and the dimension of the input / output coupling means are determined as design parameters so that the reflection has a predetermined maximum value at the reflection maximum frequency. From among the reflection characteristics of the high-frequency filter obtained by using the above, a frequency having a large reflection is selected from the characteristics of an ideal Chebyshev-type filter, and the reflection is equal to or less than a predetermined magnitude at the large reflection frequency and the maximum reflection frequency. Since the length of the resonator, the dimensions of the inter-resonator coupling means, and the dimensions of the input / output coupling means are determined as described above, The optimization is performed, and the value of the above-mentioned design parameter which can obtain the desired reflection characteristic very close to the ideal characteristic by the Chebyshev function can be determined at all the frequencies in the above-mentioned passband by an optimization technique using the minimum frequency. .

In the seventh invention, a plurality of zeros of the Chebyshev function as a function of the guide wavelength of the waveguide are set in the pass band of the waveguide filter, and the zeros correspond to the guide wavelengths of the plurality of zeros. Is defined as the zero reflection frequency, and the resonator length, the susceptance element value, and the dimensions of the susceptance element as design parameters are determined so that the reflection is minimized at the zero reflection frequency. Especially for a large waveguide type filter, the above-mentioned design parameters for obtaining a desired reflection characteristic close to the ideal characteristic by the Chebyshev function at all frequencies within the above passband by an optimization method using the minimum frequency and the like. Can be determined.

In the eighth invention, a plurality of zeros of the Chebyshev function as a function of the guide wavelength of the waveguide are set in the pass band of the waveguide filter, and the zeros correspond to the guide wavelengths of the plurality of zeros. Is defined as the zero reflection frequency, the resonator length as a design parameter, the susceptance element value, and the dimensions of the susceptance element are determined so that the reflection is minimized at the zero reflection frequency, and is obtained using the above dimensions. Of the reflection characteristics of the waveguide type filter, a frequency having a large reflection is selected from the characteristics of an ideal Chebyshev type filter, and the reflection is set to a predetermined value or less at the large reflection frequency and the zero reflection frequency. Since the resonator length of the resonator, the value of the susceptance element, and the dimensions of the susceptance element were determined, optimization was performed multiple times. Therefore, even for a waveguide type filter having a particularly large change in frequency of the electrical length, it is very close to the ideal characteristic by the Chebyshev function at all the frequencies in the passband by an optimization method using the minimum frequency or the like. It is possible to determine the value of the above-mentioned design parameter that obtains a desired reflection characteristic.

In the ninth aspect, a plurality of maximum points of the Chebyshev function as a function of the guide wavelength of the waveguide are set in the pass band of the waveguide filter, and the guide wavelengths of the plurality of maximum points are set. Is defined as the reflection maximum frequency, the resonator length as a design parameter so that the reflection at the reflection maximum frequency has a predetermined maximum value,
Since the value of the susceptance element and the dimensions of the susceptance element were determined, even in the case of a waveguide type filter with a particularly large change in the frequency of the electrical length, all of the values within the above passband were determined by the optimization method using the minimum frequency. It is possible to determine the value of the above-mentioned design parameter at which a desired reflection characteristic close to the ideal characteristic by the Chebyshev function is obtained at the frequency of.

In the tenth aspect, a plurality of maximum points of the Chebyshev function as a function of the guide wavelength of the waveguide are set in the pass band of the waveguide filter, and the guide wavelengths of the plurality of maximum points are set. Is defined as the reflection maximum frequency, the resonator length as a design parameter, the susceptance element value, and the dimensions of the susceptance element are determined so that the reflection at the reflection maximum frequency becomes a predetermined maximum value, and the dimensions of the susceptance element are determined. Of the reflection characteristics of the waveguide filter obtained by using the above, a frequency having a large reflection is selected from the characteristics of the ideal Chebyshev type filter, and the reflection has a predetermined magnitude at the large reflection frequency and the maximum reflection frequency. The resonator length of the resonator as follows,
Since the susceptance element value and the dimensions of the susceptance element are determined, optimization is performed a plurality of times. It is possible to determine the values of the design parameters that can obtain the desired reflection characteristics very close to the ideal characteristics based on the Chebyshev function at all the frequencies in the passband by a conversion technique or the like.

In the eleventh aspect, the number of resonators in any of the seventh to tenth aspects is N, and the susceptance value of the i-th susceptance element determined from the equivalent circuit element value at the center frequency of the filter is B0i, the susceptance value Bi of the i-th susceptance element
1 <Bi / B0i <1.2 (1 ≦ i ≦ 0.2N, N ≦ i ≦ N + 1) 0.8 <Bi / B0i <1 (0.2N <i ≦ 0.4N, 0.8N ≦ i <N) 0.9 <Bi / B0i ≦ 1 (0.4N <i <0.8N) Therefore, even if the passband is wide, the Chebyshev function is performed at all frequencies within the passband. The desired reflection characteristics close to the ideal characteristics are obtained.

According to the twelfth aspect of the invention, the high frequency filter is designed by the method of any of the seventh to eleventh aspects and the inductive iris or the inductive post is used as the susceptance element. By adjusting the iris width or the number, diameter, or interval of the inductive posts, a desired susceptance value can be easily obtained, and an excellent power resistance can be obtained.

[0037]

[Embodiment 1] FIG. 1 is a schematic diagram showing an embodiment of the present invention, and FIG. 2 is a diagram showing the shape of an inner conductor pattern.
a and 1b are dielectric substrates, and 2a is an outer conductor formed by closely attaching a conductive film to one entire surface of the dielectric substrate 1a.
Reference numeral 7 denotes an inner conductor formed by closely attaching a conductive film to the other surface of the dielectric substrate 1a. Reference numeral 9 denotes a through-hole passing through the dielectric substrates 1a and 1b, the outer conductors 2a and 2b, and the inner conductor 3. A through-hole 30 is formed by closely adhering a conductor film continuous with the outer conductors 2a and 2b and the inner conductor 3 on the inner peripheral surface of the inner conductor 3. Numerals 30 denote the dielectric substrates 1a and 1b, the outer conductors 2a and 2b, and the inner conductor 3. And 60 is a dielectric substrate 1
a, 1b, the outer conductors 2a, 2b and the inner conductor 6 are input lines of a strip line, 70 is a dielectric substrate 1a, 1
b, outer conductors 2a and 2b, and an inner conductor 7, an output line of a strip line, P1 is an input terminal, and P2 is an output terminal. The dielectric substrates 1a and 1b include inner conductors 3, 6, and
7 are sandwiched therebetween. Each of the plurality of inner conductors 3 has a length set to approximately １ ／ wavelength, and one end is connected to the outer conductors 2 a and 2 b by a through hole 9 and short-circuited. For this reason, the resonator 30 is a quarter-wavelength resonator in which one end is short-circuited and the other end is open. Multiple resonators 3
0 are all arranged in parallel, and adjacent ones are alternately short-circuited at opposite ends by through holes 9.

Next, the operation will be described. Since the adjacent resonators 30 are arranged so that the short-circuit end and the open end face each other, they are mutually coupled by an electric field. The amount of coupling is adjusted by the distance between the inner conductors 3 or the width of the inner conductors 3.

Now, assuming that the length of each of the inner conductors 3 is set to a predetermined length in the vicinity of 1/4 wavelength, and that all the resonators 30 resonate at the same frequency, for example, f0, At f0, the resonators 30 in the resonance state are strongly coupled to each other, are guided to the resonator 30 at the first stage of the incident wave to the input line 60, and are repeatedly coupled to the adjacent resonator by the electric field, thereby repeating the output line. 70. However, at frequencies other than f0, the coupling between the resonators 30 is very weak, and most of the power of the incident wave on the input line 60 is reflected. Thus, the stripline filters shown in FIGS. 1 and 2 have a function as a bandpass filter.

Next, the flow of the design procedure will be described with reference to FIG. First, in step 1, the equivalent of a band-pass filter having ideal Chebyshev characteristics, which is formed from a lumped-constant element and is deformed from the original low-pass filter, and a distributed constant type determined according to the physical shape of the filter of FIG. Compare the circuit at the center frequency of the passband of the filter,
An initial design of the resonator interval, resonator width, and resonator length as design parameters is performed. Step 1 is shown in FIG.
This is the same as the conventional design procedure shown in FIG. At this time,
A pair resonator obtained by taking out only two adjacent resonators in the filter of FIG. 1 is represented by an equivalent circuit including distributed constant lines as shown in FIG. In FIG. 4, reference numeral 10 denotes an equivalent circuit of a short-circuit stub having an electrical length θ, and reference numeral 11 denotes an equivalent circuit of a line having an electrical length θ. By using the equivalent circuit of FIG. 4 in combination, the filter of FIG. 1 is represented by a distributed constant line equivalent circuit similar to that shown in FIG.

Next, in step 2, frequencies higher than the number of design parameters are selected, and insertion loss or reflection loss at these frequencies is defined as shown in FIG.
FIG. 5 is a diagram showing desired pass and reflection characteristics of the filter of FIG. 1, wherein f1 to f13 are selected frequencies. In order to make the specified levels of the pass loss and the reflection loss the same, the reflection loss is specified at frequencies f2 to f10 in the pass band, and the pass loss is specified at frequencies f1 and f11 to f13 outside the pass band.

Finally, in step 3, the characteristics of the filter are calculated using the distributed constant line type equivalent circuit, and the values of the reflection loss at the frequencies f2 to f10 and the insertion loss at the frequencies f1 and f11 to f13 are set to the desired values. Optimize the parameters of the equivalent circuit. The resonator interval, resonator width, and resonator length are obtained from the parameters obtained by the optimization.

As described above, in the embodiment according to the design procedure of FIG. 3, the reflection loss is specified to a desired value at a plurality of frequencies in the pass band. The increase in reflection is small, and good characteristics are obtained over the entire pass band.

In addition, since the frequency is selected more than the number of parameters, the relational expression can be obtained more than the number of parameters. At the time of optimization, there is a high possibility that the characteristic calculation result by the equivalent circuit approaches the desired characteristic.

Embodiment 2 FIG. FIG. 6 is a flowchart showing a design procedure of another embodiment of the present invention. In step 2, reflection zero and a maximum frequency of Chebyshev characteristics are selected in a pass band, and as shown in FIG. This is a case where the reflection loss in the above is defined. FIG. 7 is a diagram showing desired reflection characteristics of the filter of FIG. 1, where f1, f3, f5, and f7 are shown.
Is the reflection zero frequency and is given by the following second equation.

[0046]

(Equation 1)

Further, f2, f4 and f6 are reflection maximum frequencies and are given by the following third equation.

[0048]

(Equation 2)

As described above, in the embodiment according to the design procedure of FIG. 6, since all the reflection zero frequencies within the pass band are defined, even if the pass band is wide, the increase of the reflection at the end of the pass band and the reflection zero are prevented. The frequency shift is small, and an ideal reflection characteristic by the Chebyshev function can be obtained over the entire pass band.

Since the number of reflection zeros and maximum frequencies is smaller than the total number of parameters, not all parameters can be specified as variables for optimization. For example, only the resonator interval is specified as a variable and other parameters are specified. As for, good reflection characteristics can be realized by correcting the resonator length and the resonator width so that the resonance frequency and the characteristic impedance of each resonator do not change due to the change in the resonator spacing.

Therefore, the embodiment according to the design procedure of FIG.
As in the case of FIG. 3, in addition to being able to obtain a broadband stripline filter, there is an advantage that a filter having good reflection characteristics can be obtained only by defining the reflection loss for a small number of frequencies.

Embodiment 3 FIG. FIG. 8 is a flowchart showing a design procedure of another embodiment of the present invention. In step 2, reflection zero frequencies having Chebyshev characteristics are selected in a pass band, and reflections at these frequencies are selected as shown in FIG. This is the case where the loss is specified. FIG. 9 shows the desired reflection characteristics of the filter of FIG. 1, where f1, f3, f5 and f7 are the zero reflection frequencies and are given by the second equation as in the case of FIG.

As described above, in the embodiment according to the design procedure of FIG. 8, since all the reflection zero frequencies within the pass band are defined, even if the pass band is wide, the increase in the reflection at the end of the pass band and the reflection zero are prevented. The frequency shift is small, and an ideal reflection characteristic by the Chebyshev function can be obtained over the entire pass band.

Since the number of reflected zero frequencies is the same as the number of resonators, not all parameters can be specified as variables for optimization, but only the resonator spacing is specified as a variable.
As for other parameters, good reflection characteristics can be realized by correcting the resonator length and the resonator width so that the resonance frequency and the characteristic impedance of each resonator do not change due to the change in the resonator interval.

Therefore, the embodiment according to the design procedure of FIG.
As in the case of FIG. 3, in addition to being able to obtain a broadband stripline filter, there is an advantage that a filter having good reflection characteristics can be obtained only by defining the reflection loss for a smaller number of frequencies.

Embodiment 4 FIG. FIG. 10 is a flowchart showing a design procedure of another embodiment of the present invention. In step 2, reflection maximum frequencies having Chebyshev characteristics are selected in a pass band, and reflection frequencies at these frequencies are selected as shown in FIG. This is the case where the loss is specified. FIG. 11 shows the desired reflection characteristics of the filter of FIG.
In the figure, f2, f4, and f6 are reflection maximum frequencies, and are given by equation (2) as in the case of FIG.

As described above, in the embodiment according to the design procedure of FIG. 10, since all the reflection maximum frequencies within the pass band are defined, even if the pass band is wide, the reflection at the end of the pass band is increased or the reflection maximum is increased. The frequency shift is small, and characteristics close to ideal reflection characteristics by the Chebyshev function can be obtained over the entire pass band. However, in this case, the reflection characteristics using the equivalent circuit parameters obtained by the optimization are as shown in FIG.
Note that the maximum value may be larger than the specified maximum value as shown in FIG.

Since the number of reflection maximum frequencies is one less than the number of resonators, not all parameters can be specified as variables for optimization. However, only the resonator spacing is specified as a variable, and other parameters are specified. By correcting the resonator length and the resonator width so that the resonance frequency and the characteristic impedance of each resonator do not change due to the change in the resonator spacing, good reflection characteristics can be realized.

Therefore, in the embodiment according to the design procedure of FIG. 10, as in the case of FIG. 3, a broadband stripline filter can be obtained, and the reflection loss is defined for a minimum number of frequencies. Alone has the advantage that good reflection characteristics can be obtained.

Embodiment 5 FIG. FIG. 12 is a flow chart showing the design procedure of another embodiment of the present invention. In the embodiment of FIG. 10, in step 4, the reflection characteristics obtained up to step 3 are replaced with ideal Chebyshev characteristics. In this case, a frequency having a higher reflection than the ideal characteristic is selected as compared with the reflection characteristic, and the reflection loss at this frequency is set as a new constraint condition. The new frequency is given, for example, as f8 and f9 in FIG.

In step 5, f2, f4, f6, f
Using the reflection loss of 8 and f9 as a constraint condition, optimization is performed again, and the resonator spacing, resonator width, and resonator length are obtained from the obtained parameters.

As described above, in the embodiment according to the design procedure of FIG. 12, since all the reflection maximum frequencies within the pass band are defined, even if the pass band is wide, the reflection at the end of the pass band is increased or the reflection maximum is reduced. Since the frequency shift is small and the optimization is performed a plurality of times, an ideal reflection characteristic by the Chebyshev function can be obtained over the entire pass band.

Since the number of reflection maximum frequencies is one less than the number of resonators, not all parameters can be specified as variables for optimization. However, only the resonator spacing is specified as a variable, and other parameters are specified. By correcting the resonator length and the resonator width so that the resonance frequency and the characteristic impedance of each resonator do not change due to the change in the resonator spacing, good reflection characteristics can be realized.

Therefore, the embodiment according to the design procedure of FIG. 12 can obtain a wide band strip line filter as well as the case of FIG. 3, and only defines the reflection loss for a small number of frequencies. Has an advantage that a material having good reflection characteristics can be obtained.

In the description of the above embodiment, the case where the reflection maximum frequency of the Chebyshev characteristic is selected in the pass band in step 2 has been described. In step 2, the reflection zero frequency of the Chebyshev characteristic is selected in the pass band. Even if is selected, an ideal reflection characteristic by the Chebyshev function can be similarly obtained over the entire pass band.

Embodiment 6 FIG. FIG. 14 is a schematic structural view showing another embodiment of the present invention, in which 12 is a rectangular waveguide, 13 is an inductive iris, and 14 is a cavity resonator having both ends partitioned by an inductive iris 13. , P1 are input terminals, and P2 is an output terminal.
The cavity resonator 14 is a half-wavelength resonator with both ends substantially short-circuited.

Next, the operation will be described. Adjacent cavity resonators 14 are mutually coupled via the inductive iris 13, and the amount of coupling is adjusted by the size of the inductive iris 13.

Now, the length of each of the cavity resonators 14 is 1 /
A predetermined length is set near two wavelengths, and all cavity resonators 1 are set.
Assuming that the resonators 4 resonate at the same frequency, for example, f0, the cavity resonators 14 in resonance at the frequency f0 are strongly coupled to each other, and the incident wave to the input terminal P1 is in the first stage. Of the output terminal P2.
Output. However, at frequencies other than f0, the coupling between the cavity resonators 14 is very weak, and the input end P
The incident wave to 1 reflects most of its power. Thus, the waveguide filter shown in FIG. 14 has a function as a band-pass filter.

Next, the flow of the design procedure will be described with reference to FIG. First, in step 1, the equivalent of a band-pass filter having ideal Chebyshev characteristics, which is formed from a lumped-constant element and is deformed from the original low-pass filter, and a distributed constant form determined according to the physical shape of the filter in FIG. The circuit is compared at the center frequency of the pass band of the filter, and the initial design of the resonator length as a design parameter and the susceptance value of the inductive iris is performed. Step 1 is the same as the conventional design procedure shown in FIGS. At this time, the equivalent circuit of the inductive iris 13 in the filter of FIG. 14 is represented by the parallel susceptance Ba as shown in FIG. Here, the parallel susceptance Ba is a function of the spacing d of the inductive iris and the wavelength in the waveguide. The equivalent circuit of the inductive post is as shown in FIG. 17 and can be used in place of the inductive iris. Using the relationship of FIG. 16, the filter of FIG. 14 is represented by a distributed constant type equivalent circuit of FIG.

Next, in step 2, as shown in FIG. 18, zero points of the Chebyshev function as a function of the wavelength in the waveguide are set in the pass band, and the frequencies of the plurality of zeros corresponding to the wavelengths in the waveguide are set. Defined as the reflected zero frequency,
The reflection loss at these frequencies is defined. The characteristic shown by the solid line in FIG. 18 is a desired reflection characteristic of the filter of FIG. In the figure, f1 to f11 are zero reflection frequencies, and the corresponding wavelength λgp in the waveguide is given by the following fourth equation.

[0071]

(Equation 3)

Here, ω _p ′ / ω ₁ ′ is the reflection zero frequency of the original low-pass filter, λg0 is the guide wavelength at the center frequency, and λg1 and λg2 are the guide wavelengths at the lower and upper limits of the pass band, respectively.

Finally, in step 3, the characteristics of the filter are calculated using the distributed constant line type equivalent circuit, and the parameters of the equivalent circuit are optimized so that the value of the reflection loss at the frequencies f1 to f11 becomes a desired value. I do. From the parameters obtained by the optimization, the cavity length and the size of the inductive iris are determined. At this time, the susceptance value Bi of the i-th inductive iris obtained by the optimization is such that the susceptance value in the initial design is B0i and the number of resonator stages is N.
Then, it is obtained within the range of the following first equation.

1 <Bi / B0i <1.2 (1 ≦ i ≦ 0.2N, N ≦ i ≦ N + 1) 0.8 <Bi / B0i <1 (0.2N <i ≦ 0.4N, 0.8N ≦ i <N) 0.9 <Bi / B0i ≦ 1 (0.4N <i <0.8N)

As described above, in the embodiment according to the design procedure of FIG. 15, since all the reflection zero frequencies in the pass band are defined, even if the pass band is wide, the increase of the reflection at the end of the pass band or the reflection zero is not required. The frequency shift is small, and an ideal reflection characteristic by the Chebyshev function can be obtained over the entire pass band.

Since the number of reflected zero frequencies is the same as the number of resonators, not all parameters can be specified as variables for optimization. For example, only the susceptance value of the inductive iris 13 is specified as a variable. As for other parameters, good reflection characteristics can be realized by correcting the resonator length so that the resonance frequency of each resonator does not change due to a change in the susceptance value of the inductive iris 13.

Therefore, in the embodiment according to the design procedure of FIG. 15, a wide band waveguide filter can be obtained, and a filter having good reflection characteristics can be obtained only by defining the reflection loss for a small number of frequencies. Has the advantage of being

Further, the sixth embodiment can be applied to the fifth embodiment to perform the optimization again. The Chebyshev function can be performed at all the frequencies in the pass band by an optimization method using the minimum frequency or the like. , The value of the above-mentioned design parameter can be determined so as to obtain a desired reflection characteristic very close to the ideal characteristic.

Embodiment 7 FIG. FIG. 19 is a flowchart showing the design procedure of another embodiment of the present invention. In step 2, the maximum reflection frequency is selected from the characteristics of the initial design in the pass band, and as shown in FIG. This is a case where the reflection loss in the above is defined. 20, f1 to f6 are the selected reflection maximum frequencies, and the solid line is the reflection characteristic after optimizing the reflection loss at f1 to f6. The reflection characteristics are greatly improved as compared with the characteristics of the wavy line obtained from the initial design.

As described above, in the embodiment according to the design procedure of FIG. 19, since the reflection maximum value is defined at a plurality of frequencies in the pass band, the reflection at the end of the pass band does not increase even if the pass band is wide. A desired reflection loss can be obtained over the entire pass band. However, in this case, the reflection characteristic using the equivalent circuit parameter obtained by the optimization sometimes has a maximum value larger than the specified maximum value. In this case, the optimization similar to that shown in the fifth embodiment is performed. Need to do.

Since the number of reflection maximum frequencies is at most one less than the number of resonators, not all parameters can be specified as variables for optimization. For example, only the susceptance value of the inductive iris 13 is used as a variable. By specifying the other parameters and correcting the resonator length so that the resonance frequency of each resonator does not change due to a change in the susceptance value of the inductive iris 13, good reflection characteristics can be realized.

Therefore, in the embodiment according to the design procedure of FIG. 19, as in the case of FIG. 15, a broadband waveguide filter can be obtained, and in addition to the reflection loss for a minimum number of frequencies. There is an advantage that good reflection characteristics can be obtained only by specifying.

Embodiment 8 FIG. FIG. 21 is a schematic diagram showing another embodiment of the present invention. In the drawing, reference numeral 1 denotes a dielectric substrate, and 2 denotes an outer conductor formed by closely attaching a conductive film to one entire surface of the dielectric substrate 1. , 1
5 is a strip conductor formed by closely attaching a conductive film to the other surface of the dielectric substrate 1, 16 is the dielectric substrate 1, the outer conductor 2,
And a microstrip line composed of the inner conductor 15,
Reference numeral 17 denotes a part of the microstrip line 16.
/ 4 wavelength line, P1 is an input terminal, and P2 is an output terminal. Here, the four quarter wavelength lines 17 are set to have wider line widths closer to the output end P2 side.

Next, the operation will be described. Since the line width of the quarter wavelength line 17 is set wider as it is closer to the output end P2 side, there is a discontinuity in the line width on the connection surface between adjacent lines. In this discontinuity, reflection occurs according to the magnitude of the discontinuity.

Now, assuming that the line lengths of all quarter-wavelength lines 17 are set to be 1/4 wavelength at a predetermined frequency f0, the radio wave of frequency f0 incident from the input terminal P1 will Partly reflected at the discontinuity of. The radio wave passing through the first discontinuity passes through the second quarter wavelength line 17 and is partially reflected again at the second discontinuity. However, 1
Since the line length of the 波長 wavelength line 17 is set to １ ／ wavelength at f 0, the reflected wave reflected at the second discontinuity returns to the first discontinuity and then returns to the first discontinuity at the first discontinuity. Phase is 180 degrees, that is, 1
/ 2 wavelengths. Therefore, these two reflected waves cancel each other out, and if the relative magnitude of the discontinuity is adjusted, it is possible to prevent the reflected wave from returning to the input terminal P1 at all. For the third and subsequent discontinuities, mutual reflection can be canceled by the same principle. Therefore, the incidence from the input terminal P1 having a narrow line width and high impedance is hardly reflected when the frequency is f0. , From the output terminal P2 having a wide line width and a low impedance. Thus, FIG.
1 has a function as an impedance transformer.

Next, the flow of the design procedure will be described with reference to FIG. First, in step 1, according to a conventional design procedure that does not consider the frequency characteristics of susceptance due to discontinuity between quarter-wavelength lines, 1
Initially design the line length and characteristic impedance of the 波長 wavelength line. Step 1 is the same as the conventional design procedure shown in FIGS. At this time, in the impedance transformer shown in FIG.
The discontinuity between them is represented by a parallel susceptance as shown in FIG. In FIG. 23, B12 is a parallel susceptance due to discontinuity, and Z1 and Z2 are characteristic impedances of the quarter wavelength lines 17 at both ends. By using the equivalent circuit of FIG. 23 in combination, the filter of FIG. 21 is represented by the same equivalent circuit as that shown in FIG.

Next, in step 2, frequencies equal to or more than the number of design parameters are selected, and insertion loss or reflection loss at these frequencies is defined as shown in FIG.
FIG. 24 is a diagram showing desired transmission and reflection characteristics of the impedance transformer shown in FIG. 21, where f1 to f1 are shown.
3 is the selected frequency. Here, a reflection loss having a larger frequency characteristic than the insertion loss is defined.

Finally, in step 3, the filter characteristics are calculated by the equivalent circuit, and the frequencies f1 to f13 are calculated.
The electrical length and characteristic impedance of the 波長 wavelength line, which are the parameters of the equivalent circuit, are optimized so that the value of the reflection loss at becomes the desired value. From the parameters obtained by the optimization, the line length and line width of the 波長 wavelength line are obtained.

As described above, in the embodiment of FIG. 21 according to the design procedure of FIG. 22, the reflection loss is specified to a desired value at a plurality of frequencies within the pass band. The increase in reflection at the edges is small, and good characteristics are obtained over the entire pass band.

Further, since the frequency is selected more than the number of parameters, the relational expression can be obtained more than the number of parameters. At the time of optimization, there is a high possibility that the characteristic calculation result by the equivalent circuit approaches the desired characteristic.

Embodiment 9 FIG. FIG. 25 is a flow chart showing a design procedure of another embodiment of the present invention. In step 2, reflection zero and maximum frequency of Chebyshev characteristic are selected in a pass band, and these frequencies are selected as shown in FIG. This is a case where the reflection loss in the above is defined. FIG. 26 is a diagram showing desired reflection characteristics of the impedance transformer of FIG. 21 and prescribed frequencies and reflection losses. In the figure, f1, f3, f5 and f7
Is a reflection zero frequency, and f2, f4, and f6 are reflection maximum frequencies, which can be obtained as in the conventional case.

As described above, in the embodiment according to the design procedure of FIG. 25, since all the reflection zeros and the maximum frequencies in the pass band are defined, even if the pass band is wide, the reflection at the end of the pass band does not increase. At least, an ideal reflection characteristic by the Chebyshev function can be obtained over the entire pass band.

Since the number of reflection zeros and the maximum frequency is smaller than the total number of parameters, not all parameters can be designated as variables for optimization. For example, only the characteristic impedance of the 波長 wavelength line 17 is used as a variable. Specify,
In response to the change in the characteristic impedance, the discontinuity, that is, the phase difference between the parallel susceptances does not change.
By correcting the electrical length of the four-wavelength line 17 subordinately, good reflection characteristics can be realized.

Therefore, in the embodiment according to the design procedure of FIG. 25, as in the case of FIG. 22, a broadband stripline type impedance transformer can be obtained, and the return loss is defined for a small number of frequencies. Alone has the advantage that good reflection characteristics can be obtained.

Embodiment 10 FIG. FIG. 27 is a flowchart showing a design procedure of another embodiment of the present invention. In step 2, reflection zero frequencies having Chebyshev characteristics are selected in a pass band, and reflections at these frequencies are selected as shown in FIG. This is the case where the loss is specified. FIG. 28 is a diagram showing desired reflection characteristics of the impedance transformer of FIG. 21 and prescribed frequencies and reflection losses. In the figure, f1, f3, f5 and f7 are reflection zero frequencies, which can be obtained in the same manner as in the conventional case.

As described above, in the embodiment based on the design procedure of FIG. 27, all the reflection zero frequencies within the pass band are defined, so that even if the pass band is wide, the increase of the reflection at the end of the pass band and the reflection zero are prevented. The frequency shift is small, and an ideal reflection characteristic by the Chebyshev function can be obtained over the entire pass band.

Since the number of reflected zero frequencies is the same as the number of quarter-wave lines, not all parameters can be designated as variables for optimization. Only the variable is designated as a variable, and the electrical length of the 1/4 wavelength line 17 is subordinately corrected so that the discontinuity, that is, the phase difference between the parallel susceptances does not change in response to the change in the characteristic impedance. Reflection characteristics can be realized.

Therefore, in the embodiment according to the design procedure of FIG. 27, as in the case of FIG. 25, a wideband stripline type impedance transformer can be obtained, and the reflection loss is defined for a smaller number of frequencies. There is an advantage that a reflection characteristic can be obtained simply by performing the above operation.

Embodiment 11 FIG. FIG. 29 is a flowchart showing a design procedure of another embodiment of the present invention. In step 2, reflection maximum frequencies having Chebyshev characteristics are selected in a pass band, and reflection frequencies at these frequencies are selected as shown in FIG. This is the case where the loss is specified. FIG. 29 is a diagram showing desired reflection characteristics of the impedance transformer of FIG. 21 and prescribed frequencies and reflection losses. In the figure, f2, f4, and f6 are the zero reflection frequencies, which are obtained in the same manner as in the conventional case.

As described above, in the embodiment based on the design procedure of FIG. 29, since all the reflection maximum frequencies within the pass band are defined, even if the pass band is wide, the increase of the reflection at the end of the pass band or the reflection maximum. The frequency shift is small, and characteristics close to ideal reflection characteristics by the Chebyshev function can be obtained over the entire pass band. However, in this case, the reflection characteristics using the equivalent circuit parameters obtained by the optimization are as shown in FIG.
Note that the maximum value may be larger than the specified maximum value as shown in FIG.

Since the number of reflected maximum frequencies is one less than the number of quarter-wave lines, not all parameters can be specified as variables for optimization. Is designated as a variable, and the electrical length of the ４ wavelength line 17 is subordinately corrected so that the discontinuity, that is, the phase difference between the parallel susceptances does not change in response to the change in the characteristic impedance. Reflection characteristics can be realized.

Therefore, in the embodiment according to the design procedure of FIG. 29, as in the case of FIG. 25, a broadband stripline impedance transformer can be obtained, and the return loss for a minimum number of frequencies can be obtained. Satisfactorily has the advantage that a reflection characteristic is good.

Embodiment 12 FIG. FIG. 31 is a flow chart showing the design procedure of another embodiment of the present invention. In the embodiment of FIG. 29, further, in step 4, the reflection characteristics obtained up to step 3 are converted to ideal Chebyshev characteristics. In this case, a frequency having higher reflection than the ideal characteristic is selected as compared with the reflection characteristic, and the reflection loss at this frequency is set as a new constraint condition. The new frequency is given, for example, as f8 and f9 in FIG.

In step 5, f2, f4, f6, f
With the reflection loss of 8 and f9 as a constraint, optimization is performed again, and the line length and line width of the 1/4 wavelength line are obtained from the obtained parameters.

As described above, in the embodiment based on the design procedure shown in FIG. 31, all the reflection maximum frequencies within the pass band are defined, so that even if the pass band is wide, the reflection at the end of the pass band is increased or the reflection maximum is increased. Since the frequency shift is small and the optimization is performed a plurality of times, an ideal reflection characteristic by the Chebyshev function can be obtained over the entire pass band.

Since the number of reflected maximum frequencies is one less than the number of quarter wavelength lines, not all parameters can be specified as variables for optimization. For example, only the characteristic impedance of the quarter wavelength line 17 Is designated as a variable, and the electrical length of the ４ wavelength line 17 is subordinately corrected so that the discontinuity, that is, the phase difference between the parallel susceptances does not change in response to the change in the characteristic impedance. Reflection characteristics can be realized.

Therefore, in the embodiment according to the design procedure of FIG. 31, as in the case of FIG. 25, a wideband stripline impedance transformer can be obtained, and the reflection loss is defined for a small number of frequencies. There is an advantage that a reflection characteristic can be obtained simply by performing the above operation.

Embodiment 13 FIG. FIG. 33 is a schematic configuration diagram of another embodiment of the present invention,
It is a figure which shows the impedance transformer by a square waveguide. In the figure, 18 is a rectangular waveguide, 19 is a 1/4 wavelength waveguide, P1 is an input terminal, and P2 is an output terminal. Four one
The height of the 波長 wavelength waveguide 19 is set higher as it is closer to the output end P2 side.

Next, the operation will be described. The waveguide height of the 1/4 wavelength waveguide 19 is set higher as it is closer to the output end P2 side, so that there is a discontinuity in the waveguide height at the connection surface between adjacent waveguides. . In this discontinuity, reflection occurs according to the magnitude of the discontinuity.

Now, assuming that the lengths of all the 1/4 wavelength waveguides 19 are set to be 1/4 of the guide wavelength at a predetermined frequency f0, the frequency incident from the input terminal P1 is assumed. The radio wave of f0 is partially reflected at the first discontinuity. The radio wave passing through the first discontinuity passes through the second quarter wavelength waveguide 19 and is partially reflected again at the second discontinuity. However, since the line length of the 1/4 wavelength waveguide 19 is set to 1/4 wavelength at f0, the reflected wave reflected at the second discontinuity returns to 1 at the time of returning to the first discontinuity.
1st phase compared to the first reflected wave at the second discontinuity
It is delayed by 80 degrees, that is, 波長 wavelength. Therefore, these two
The two reflected waves cancel each other out, and if the relative magnitude of the discontinuity is adjusted, the reflected waves can never return to the input terminal P1. For the third and subsequent discontinuities, mutual reflection can be canceled by the same principle. Therefore, incidence from the input terminal P1 having a low waveguide height and low impedance is almost impossible when the frequency is f0. The light is not reflected and is taken out from the output end P2 where the waveguide height is high and the impedance is high. As described above, the waveguide circuit shown in FIG. 33 has a function as an impedance transformer.

In general, since the guide wavelength of a waveguide has dispersibility with respect to frequency, the electrical length of the waveguide is proportional to the reciprocal of the guide wavelength, but not proportional to the frequency. Also in the impedance transformer of FIG. 33, it is necessary to consider the Chebyshev function used as the ideal reflection characteristic as a function of the guide wavelength.

By using the guide wavelength of the Chebyshev function with a zero gradient instead of the frequency with a zero gradient of the Chebyshev function in the eighth to twelfth embodiments according to FIG.
Is applicable to the same design procedure as that shown in Embodiments 8 to 12, and can realize a wide band.

Although the high-frequency filters shown in the first to seventh embodiments have the case where the number of resonators is four or eleven, the number is one to three, five to ten, or twelve or more. The same advantages and effects as those of the embodiment can be obtained.

Although the impedance transformers of the eighth to thirteenth embodiments have been described in connection with the case where the number of quarter wavelength lines or waveguides is four, even if the number of stages is one to three or five or more. Often, the same advantages and effects as those of the embodiment can be obtained.

Further, in the above-described embodiment, the case where the Chebyshev function is used as the transfer function has been described. However, any function such as a Bessel function or an elliptic function that can set a plurality of zeros, local maximums, and local minimums in the passband is used. For example, it goes without saying that the design procedure used for the high-frequency filter or the impedance transformer of the present invention can be similarly applied.

[0116]

As described above, according to the first aspect of the present invention, not only the center frequency but also a plurality of frequencies within the range of the selected frequency are used to set conditional expressions equal to or more than the number of design parameters, and It is possible to determine the value of the above-mentioned design parameter that can obtain a desired reflection characteristic even when the pass band is wide by a conversion method or the like, and it is possible to obtain a high-frequency filter with small reflection over a wide band.

According to the second aspect of the present invention, it is possible to determine the value of the design parameter by which the desired reflection characteristic close to the ideal characteristic by the Chebyshev function is obtained at all the frequencies in the pass band by the optimization method or the like. Thus, there is an effect that a high-frequency filter with small reflection can be obtained.

According to the third aspect of the present invention, a design parameter for obtaining a desired reflection characteristic close to the ideal characteristic by the Chebyshev function at all frequencies in the pass band by an optimization method using the minimum frequency or the like. Can be determined, and there is an effect that a high-frequency filter with small reflection over a wide band can be obtained.

According to the fourth aspect of the present invention, the optimization is performed a plurality of times by the optimization method using the minimum frequency and the like, and the ideal characteristic by the Chebyshev function at all the frequencies in the pass band is very low. It is possible to determine the value of the design parameter that can obtain a close desired reflection characteristic, and there is an effect that a high-frequency filter with small reflection can be obtained over a wide band.

According to the fifth aspect of the present invention, a desired reflection characteristic very close to the ideal characteristic by the Chebyshev function can be obtained at all the frequencies in the pass band by an optimization method using the minimum frequency. The value of the design parameter can be determined, and there is an effect that a high-frequency filter with small reflection over a wide band can be obtained.

According to the sixth aspect of the present invention, a desired reflection characteristic very close to the ideal characteristic by the Chebyshev function can be obtained at all frequencies in the pass band by an optimization method using the minimum frequency or the like. The value of the design parameter can be determined, and there is an effect that a high-frequency filter with small reflection over a wide band can be obtained.

According to the seventh aspect of the present invention, even for a waveguide type filter having a particularly large change in the frequency of the electric length,
The optimization method using the minimum frequency can determine the value of the design parameter that obtains the desired reflection characteristic close to the ideal characteristic by the Chebyshev function at all the frequencies in the pass band, so that the reflection There is an effect that a small high-frequency filter can be obtained.

According to the eighth aspect of the present invention, even for a waveguide type filter having a particularly large change in the frequency of the electric length,
Optimization is performed several times by the optimization method using the minimum frequency, etc., and the design parameter value that obtains the desired reflection characteristic very close to the ideal characteristic by the Chebyshev function is determined at all the frequencies in the passband. Thus, there is an effect that a high-frequency filter with small reflection can be obtained over a wide band.

Further, according to the ninth aspect of the present invention, a waveguide type filter having a particularly large frequency change in electric length is also applicable.
By the optimization method using the minimum frequency, etc., it is possible to determine the value of the design parameter that obtains the desired reflection characteristic close to the ideal characteristic by the Chebyshev function at all the frequencies in the pass band, and the reflection is small over a wide band. There is an effect that a high frequency filter can be obtained.

According to the tenth aspect of the present invention, even for a waveguide type filter having a particularly large change in the frequency of the electrical length, optimization can be performed a plurality of times by an optimization method using the minimum frequency. Then, at all frequencies within the pass band, it is possible to determine the value of the design parameter that obtains the desired reflection characteristic very close to the ideal characteristic by the Chebyshev function, so that the effect of obtaining a high-frequency filter with small reflection over a wide band can be obtained. is there.

According to the eleventh aspect of the present invention, even if the pass band is wide, a desired reflection characteristic close to the ideal characteristic by the Chebyshev function can be obtained at all the frequencies within the pass band, and therefore, over a wide band. There is an effect that a high-frequency filter with small reflection can be obtained.

According to the twelfth aspect of the present invention, since the inductive iris or the inductive post is used as the susceptance element, the iris width of the inductive iris or the number, the diameter, or the interval of the inductive post is reduced. The desired susceptance value can be easily obtained by adjusting, and
Since a filter excellent in power durability can be obtained, there is an effect that a high-frequency filter with small reflection over a wide band can be obtained.

[Brief description of the drawings]

FIG. 1 is a schematic configuration diagram showing a high-frequency filter according to Embodiment 1 of the present invention.

FIG. 2 is a diagram showing an inner conductor pattern of the high-frequency filter according to Embodiment 1 of the present invention.

FIG. 3 is a diagram showing a design procedure of a high-frequency filter according to Embodiment 1 of the present invention.

FIG. 4 is a diagram showing an equivalent circuit of a resonator of the high-frequency filter according to Embodiment 1 of the present invention.

FIG. 5 is a diagram illustrating characteristics of the high-frequency filter according to the first embodiment of the present invention.

FIG. 6 is a diagram showing a design procedure of a high-frequency filter according to Embodiment 2 of the present invention.

FIG. 7 is a diagram showing characteristics of a high-frequency filter according to Embodiment 2 of the present invention.

FIG. 8 is a diagram showing a design procedure of a high-frequency filter according to Embodiment 3 of the present invention.

FIG. 9 is a diagram illustrating characteristics of a high-frequency filter according to Embodiment 3 of the present invention.

FIG. 10 is a diagram showing a design procedure of a high-frequency filter according to Embodiment 4 of the present invention.

FIG. 11 is a diagram showing characteristics of a high-frequency filter according to Embodiment 4 of the present invention.

FIG. 12 is a diagram showing a design procedure of a high frequency filter according to Embodiment 5 of the present invention.

FIG. 13 is a diagram showing characteristics of a high frequency filter according to Embodiment 5 of the present invention.

FIG. 14 is a configuration diagram of a high-frequency filter according to Embodiment 6 of the present invention.

FIG. 15 is a diagram showing a design procedure of a high-frequency filter according to Embodiment 6 of the present invention.

FIG. 16 is a diagram illustrating a susceptance element of a high-frequency filter according to Embodiment 6 of the present invention.

FIG. 17 is a diagram illustrating a susceptance element of a high-frequency filter according to Embodiment 6 of the present invention.

FIG. 18 is a diagram showing reflection characteristics of a high frequency filter according to Embodiment 6 of the present invention.

FIG. 19 is a diagram showing a design procedure of a high-frequency filter according to Embodiment 7 of the present invention.

FIG. 20 shows characteristics of the high-frequency filter according to Embodiment 7 of the present invention.

FIG. 21 is a diagram showing a schematic configuration of an impedance transformer according to Embodiment 8 of the present invention.

FIG. 22 is a diagram showing a design procedure of the impedance transformer according to the eighth embodiment of the present invention.

FIG. 23 is a diagram showing discontinuity of an impedance transformer according to Embodiment 8 of the present invention.

FIG. 24 is a diagram showing characteristics of the impedance transformer according to the eighth embodiment of the present invention.

FIG. 25 is a diagram showing a design procedure of the impedance transformer according to the ninth embodiment of the present invention.

FIG. 26 is a diagram showing characteristics of the impedance transformer according to the ninth embodiment of the present invention.

FIG. 27 is a diagram showing a design procedure of the impedance transformer according to the tenth embodiment of the present invention.

FIG. 28 is a diagram showing characteristics of the impedance transformer according to the tenth embodiment of the present invention.

FIG. 29 is a diagram showing a design procedure of the impedance transformer according to the eleventh embodiment of the present invention.

FIG. 30 is a diagram showing characteristics of the impedance transformer according to Embodiment 11 of the present invention.

FIG. 31 is a diagram showing a design procedure of the impedance transformer according to the twelfth embodiment of the present invention.

FIG. 32 is a diagram showing characteristics of the impedance transformer according to the twelfth embodiment of the present invention.

FIG. 33 is a perspective view showing a configuration of an impedance transformer according to Embodiment 13 of the present invention.

FIG. 34 is a diagram showing a design procedure of a conventional high-frequency filter.

FIG. 35 is a circuit diagram of an original low-pass filter for describing a design procedure of a conventional high-frequency filter.

FIG. 36 is a diagram showing an equivalent circuit of a conventional high-frequency filter.

FIG. 37 is a diagram illustrating characteristics of a conventional high-frequency filter.

FIG. 38 is a diagram showing a procedure for designing a conventional high-frequency filter.

FIG. 39 is a circuit diagram of an original low-pass filter for describing a design procedure of a conventional high-frequency filter.

FIG. 40 is a diagram showing an equivalent circuit of a conventional high-frequency filter.

FIG. 41 is a diagram showing an equivalent circuit of a conventional high-frequency filter.

FIG. 42 is a diagram illustrating characteristics of a conventional high-frequency filter.

FIG. 43 is a diagram showing a design procedure of a conventional impedance transformer.

FIG. 44 is a diagram showing an equivalent circuit of a conventional impedance transformer.

FIG. 45 is a diagram showing characteristics of a conventional impedance transformer.

[Explanation of symbols]

1 dielectric substrate, 2 outer conductor, 3, 6, 7 inner conductor, 9
Through hole, 12 rectangular waveguide, 13 inductive iris, 14 cavity, 15 strip conductor, 16
Microstrip line, 17 1/4 wavelength line, 1
8 rectangular waveguide, 19 1/4 wavelength waveguide, 30 stripline resonator, 60 input lines, 70 output lines.

Continued on the front page (72) Inventor Hidenori Yukawa 5-1-1, Ofuna, Kamakura-shi Mitsubishi Electric Corporation In-house Electronic Systems Research Laboratory (72) Inventor Hideki Asao 5-1-1, Ofuna, Kamakura-shi Mitsubishi Electric Corporation Inside the Electronic Systems Laboratory (72) Inventor Shuji Urasaki 5-1-1, Ofuna, Kamakura City Mitsubishi Electric Corporation Inside the Electronic Systems Laboratory (56) References Yoshihiro Konishi, Fundamentals of microwave circuits and their applications, General Electronic Publishing Co., Ltd., 1990, p.p. 367-373 R.C. E. FIG. Collon, FOUNDA TIONS FOR MICROWAVE ENGINEERING, USA, McGraw-Hill, 1992, pp. 343-360, 587-642 (58) Fields investigated (Int. Cl. ⁷ , DB name) H01P 1/203 H01P 1 / 207

Claims (12)

(57) [Claims] 1. A plurality of transmission line resonators and said resonator
Resonator-to-resonator coupling means for coupling between the resonators, and the resonator
And an input / output coupling means for mutually coupling
A method for designing a high-frequency filter comprising:
Resonator length, dimensions of the inter-resonator coupling means, and
More than the number of parameters that determine the dimensions of the
Number of frequencies, and select
The pass or reflection characteristics of the high frequency filter
The length of the resonators and the distance between the resonators so as to approach the value
Determine the dimensions of the coupling means and the dimensions of the input / output coupling means.
A method for designing a high-frequency filter, characterized in that: 2. A plurality of transmission line resonators and said resonator
Resonator-to-resonator coupling means for coupling between the resonators, and the resonator
And an input / output coupling means for mutually coupling
The method for designing a high-frequency filter provided
Within the pass band of the filter,
Set multiple slope zeros of Chebyshev function as function
And the frequencies corresponding to the plurality of slope zeros are reflected zero frequencies.
Number or the reflection maximum frequency
The reflection is extremely small at the wave number or the above reflection maximum frequency.
Or the maximum length of the resonator so that
The dimensions of the coupling means between the vibrators and the
Design of high-frequency filter characterized by its dimensions
Method. 3. A plurality of transmission line resonators, and said resonator
Resonator-to-resonator coupling means for coupling between the resonators, and the resonator
And an input / output coupling means for mutually coupling
The method for designing a high-frequency filter provided
Within the pass band of the filter,
Set multiple Chebyshev function zeros as a function
The frequencies corresponding to multiple zeros are defined as reflected zero frequencies.
So that the reflection is minimized at the reflection zero frequency.
The length of the resonator and the dimensions of the coupling means between the resonators
And the dimensions of the input / output coupling means are determined.
A method for designing a high-frequency filter, characterized in that: 4. A plurality of transmission line resonators and said resonator
A coupling between the resonators means Ru bound to each other between said cavity
And an input / output coupling means for mutually coupling
The method for designing a high-frequency filter provided
Within the pass band of the filter,
Set multiple Chebyshev function zeros as a function
The frequencies corresponding to multiple zeros are defined as reflected zero frequencies.
So that the reflection is minimized at the reflection zero frequency.
The length of the resonator and the dimensions of the coupling means between the resonators
And the dimensions of the input / output coupling means are determined.
The reflection characteristics of the high-frequency filter obtained using the dimensions
Of which, reflection is more than ideal Chebyshev type filter characteristics
The frequency of the large reflection
At the zero reflection frequency, the reflection is less than a predetermined magnitude.
The resonator length of the resonator and the resonator coupling means
And the dimensions of the input / output coupling means were determined.
A method for designing a high-frequency filter, characterized in that: 5. A plurality of transmission line resonators and said resonator
Resonator-to-resonator coupling means for coupling between the resonators, and the resonator
And an input / output coupling means for mutually coupling
The method for designing a high-frequency filter provided
Within the pass band of the filter,
Set multiple Chebyshev function maxima as functions,
The frequencies corresponding to the plurality of maximum points are defined as the reflection maximum frequency.
At the reflection maximum frequency,
The resonator length of the resonator so that the maximum value is obtained,
Dimensions of the coupling means and the dimensions of the input / output coupling means
Of designing a high frequency filter characterized by determining
Law. 6. A plurality of transmission line resonators and said resonator
Resonator-to-resonator coupling means for coupling between the resonators, and the resonator
And an input / output coupling means for mutually coupling
The method for designing a high-frequency filter provided
Within the pass band of the filter,
Set multiple Chebyshev function maxima as functions,
The frequencies corresponding to the plurality of maximum points are defined as the reflection maximum frequency.
At the reflection maximum frequency,
The resonator length of the resonator so that the maximum value is obtained,
Dimensions of the coupling means and the dimensions of the input / output coupling means
And the high frequency filter obtained using the above dimensions
Ideal Chebyshev filter among the reflection characteristics
Select a frequency with a large reflection from the characteristics of
Reflection at specified frequency and reflection maximum frequency
The resonator length of the size follows Do the resonator so that the above
Size of coupling means between resonators and coupling means between input and output
The size of the high-frequency filter is determined.
Metering method. 7. A plurality of cavity resonators each comprising a waveguide and said cavity resonator.
As coupling means for cavity cavities and input / output coupling means
Waveguide type high frequency filter with susceptance element
In the method of designing a filter, the waveguide type high frequency filter
Within the passband of the waveguide as a function of the guide wavelength of the waveguide.
A plurality of zeros of the Ebyshev function are set, and the
The frequency corresponding to the above tube wavelength is defined as the reflection zero frequency.
So that the reflection is minimized at the reflection zero frequency.
The resonator length of the resonator, the susceptance element value, and
And the dimensions of the susceptance element are determined
Method of designing a high-frequency filter. 8. A plurality of cavity resonators each comprising a waveguide,
As coupling means for cavity cavities and input / output coupling means
Waveguide type high frequency filter with susceptance element
In the method of designing a filter, the waveguide type high frequency filter
Within the passband of the waveguide as a function of the guide wavelength of the waveguide.
A plurality of zeros of the Ebyshev function are set, and the
The frequency corresponding to the above tube wavelength is defined as the reflection zero frequency.
So that the reflection is minimized at the reflection zero frequency.
The resonator length of the resonator, the susceptance element value, and
And the dimensions of the susceptance element, and
Reflection characteristics of the above waveguide type high frequency filter obtained by using
Of the ideal Chebyshev filters
Select a frequency with a large reflection,
And the reflection is equal to or less than the specified magnitude at the above-mentioned zero reflection frequency.
So that the resonator length of the resonator and the susceptance
The element value and the dimensions of the susceptance element were determined
A method for designing a high-frequency filter, characterized in that: 9. A plurality of cavity resonators each comprising a waveguide, and
As coupling means for cavity cavities and input / output coupling means
Waveguide type high frequency filter with susceptance element
In the method of designing a filter, the waveguide type high frequency filter
Within the passband of the waveguide as a function of the guide wavelength of the waveguide.
Set multiple maximum points of the Ebyshev function, and
The frequency corresponding to the above-mentioned guide wavelength of the point is defined as the reflection maximum frequency.
At the reflection maximum frequency,
The resonator length of the resonator and the susceptor so that the maximum value is obtained.
Of the susceptance element and the dimensions of the susceptance element.
A method for designing a high-frequency filter, characterized in that: 10. A plurality of cavity resonators comprising a waveguide and an upper cavity resonator.
As means for coupling the cavity resonators to each other and for input / output coupling
Waveguide-type high-frequency filter with all susceptance elements
In the design method of the filter, the waveguide type high frequency
Within the passband of the waveguide as a function of the guide wavelength of the waveguide.
Set multiple maximum points of the Chebyshev function, and
The frequency corresponding to the above-mentioned guide wavelength is the maximum reflection frequency.
The reflection is specified at the maximum reflection frequency.
The resonator length of the resonator and the susceptor
The value of the susceptance element and the dimensions of the susceptance element
Determine and obtain the high circumference of the waveguide shape obtained using the above dimensions
Ideal Chebyshev type of reflection characteristics of wave filter
Select a frequency with a large reflection from the characteristics of the filter.
At the high frequency of
The resonator of the above-described resonator so that the radiation is less than a predetermined size.
Length, susceptance element value, and susceptance
A high-frequency filter characterized in that the dimensions of the element are determined.
Design method. 11. The number of resonators is N, the center frequency of the filter
I-th subset determined from the equivalent circuit element value in the number
When the susceptance value of the punctance element is B0i,
The susceptance value Bi of the eye susceptance element is given by the following equation.
2. The method according to claim 1, wherein the predetermined range is set.
0. The method for designing a high-frequency filter according to any one of 0 to 0. 1 <Bi / B0i <1.2 (1 ≦ i ≦ 0.2N, N ≦ i ≦ N + 1) 0.8 <Bi / B0i <1 (0.2N <i ≦ 0.4N, 0.8N ≦ i < N) 0.9 <Bi / B0i ≦ 1 (0.4N <i <0.8N) 12. The method according to claim 7, wherein :
High-frequency filter configured by the high-frequency filter design method of
The susceptance element is an inductive iris
Or high-frequency filters characterized by inductive posts.
Ruta.