JP3291127B2 - Control device for synchronous reluctance motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for a synchronous motor. In particular, the present invention relates to a control device for a synchronous motor in which a magnetic resistance differs according to a change in a rotational position of a rotor of the motor, and a rotational force is obtained by utilizing the difference in the magnetic resistance.

[0002]

2. Description of the Related Art In a conventional control apparatus for a reluctance motor, a three-phase AC voltage / current whose frequency is controlled without detecting the position of a rotor is applied to the reluctance motor to control the motor. In the reluctance motor control device shown in FIG. 15, the rotor position and the rotor speed are detected, and precise speed control is performed.

The structure of the synchronous motor to be controlled in the control device shown in FIG. 15 is shown in FIG. 16 as a cross-sectional structure perpendicular to the rotation axis. As shown in FIG. 16, the synchronous motor has a salient pole type rotor 6 and a slot 1 containing a three-phase winding.
4. It is composed of a stator core 15 and a motor case 16. As shown in FIG. 15, a position detector 5 is mechanically coupled to the rotor 6, and a position signal DS is output.

The synchronous motor shown in FIG. 16 is modeled and simplified and expressed as shown in FIG. The three-phase winding is a concentrated winding of 2 slots for each phase, and the U-phase winding is U
P, UN, V phase windings are VP, VN, W phase windings are W
It is represented by P and WN. FIG. 18 is a developed plan view showing the relationship between the windings of each phase, with the horizontal axis representing the rotation direction and the vertical axis representing the axial direction of the motor.

FIG. 19 shows a vector relation of each current component. The direction of the rotor magnetic pole with respect to the U-phase windings UP and UN is AR, and the vector synthesis direction of the three-phase current is AA. The amplitude of the three-phase motor current is DIO, and the amplitude of the field current component is F
I, the amplitude of the armature current component is AI. These relationships are those of an ideal synchronous motor in which the field magnetic flux is not disturbed by the armature current component.

In the control device shown in FIG. 15, the adder 1 adds the speed command SI and the speed signal DAR detected from the position detector 5 via the speed detecting means 3;
The speed deviation ES is obtained. Based on the speed deviation ES, the torque control means 2 performs compensation calculations such as proportional, integral, and derivative, and a torque command T is obtained.

In the field current command circuit 24, the rotor rotational position AR detected by the rotor position detecting means 4 and the speed signal DAR are input, and the field current commands SFIU, SFIV, SFIW represented by the following equation (1). Is created.

SFIU = FI · sin (AR) (1-1) SFIV = FI · sin (AR + 120 °) (1-2) SFIW = FI · sin (AR + 240 °) (1-3) In the above formula (1) FI is the amplitude of the field current. The amplitude FI depends on the speed signal DAR, and the amplitude FI is a constant value below the base rotation speed NB of the electric motor. Above the base rotation speed NB, the amplitude FI decreases as the speed signal DAR increases, and the base rotation speed NB and the amplitude FI
There is a characteristic that a value obtained by multiplying by the constant becomes a constant value.

In the armature current command circuit 21, a torque command T and a rotor rotational position AR are input, and if an amplitude of an armature current component corresponding to the torque command T is AI, it is expressed by the following equation (2). Armature current command SAIU, SAI
V and SAIW are output. SAIU = AI · sin (AR + 90 °) (2-1) SAIV = AI · sin (AR + 90 ° + 120 °) (2-2) SAIW = AI · sin (AR + 90 ° + 240 °) (2-3) In the adder 25 Is a current command SI to which an armature current command SAIU and a field current command SFIU are input and added.
U is obtained. In the adder 26, the armature current command S
The AIV and the field current command SFIV are input, and the added current command SIV is obtained. In the adder 27, the armature current command SAIW and the field current command SFIW are input, and the added current command SIW is obtained. Each current command SIU, SIV, SIW is represented by the following equation (3).

SIU = SAIU + SFIU (3-1) SIV = SAIV + SFIV (3-2) SIW = SAIW + SFIW (3-3) Angle AA indicating the direction of the combined vector of these phase currents
Is the direction of the angle (AR-AA) with respect to the direction of the magnetic pole in each of FIGS.

In the current control circuit 7, a current command SI
U, SIV, and SIW are input and amplified, and the three-phase current I
U, IV, and IW are output. This three-phase current IU, I
V and IW are respectively supplied to the windings 8, 9, and 10 of the three-phase motor.

In the control device shown in FIG. 15, the control of the field magnetic flux and the armature current component for generating the torque are appropriately controlled in accordance with the direction AR of the magnetic pole of the rotor 6 and the rotation speed DAR of the rotor 6. You. Accordingly, a force according to Fleming's law is generated between the stator core 15 and the rotor 6, and a clockwise or counterclockwise torque can be arbitrarily generated, so that the speed control of the electric motor is favorably performed.

[0013]

In the reluctance motor control device shown in FIG. 15 described above, good control performance can be obtained, but the following points are not taken into consideration.

First, the position detector 5 used as a control reference is expensive.

Second, wiring is required for the position detector 5, and the cost for providing the wiring is required.

Third, the position detector 5 is attached to the motor. The attachment of the position detector 5 increases the total size of the motor.

Fourth, since the position detector 5 generally uses a large number of components and includes many electronic components,
The reliability of the entire control device is reduced.

SUMMARY OF THE INVENTION The present invention has been made to solve the above problems, and the present invention is directed to a synchronous motor control device capable of realizing control of a synchronous motor without using a position detector mechanically attached to the synchronous motor. For the purpose of providing. That is,
SUMMARY OF THE INVENTION An object of the present invention is to provide a synchronous motor control device which can reduce costs and has high reliability.

[0019]

According to the present invention, there is provided a synchronous motor control device which obtains a rotational force by utilizing a difference in magnetic resistance according to a change in the rotational position of a rotor of the motor. A motor current for detecting a current command value of each phase winding of the motor and a time change of the current command value, or detecting an actual current flowing in each phase winding of the motor and a time change of the actual current. Detecting means, motor voltage detecting means for detecting a voltage command value of each phase winding of the motor, or detecting voltage of each phase winding of the motor, output of the motor current detecting means and motor voltage detecting means And a rotor position detecting means for receiving the rotor position signal or the rotor position signal and the rotor speed signal.

Each of the detection signals is input to the rotor position detecting means, and the rotational position and the rotational speed of the rotor are obtained by calculation. Further, the rotor position detecting means is provided with a storage means for storing and storing a calculation result according to each input condition instead of the calculation, and reads out a rotation position and a rotation speed of the rotor according to the input condition. Further, a neural network is used in the storage means.

[0021]

In the control device for a synchronous motor according to the present invention, the current command value of each phase winding of the motor and the time change of the current command value are detected by the motor current detecting means, or each phase of the motor is detected. An actual current flowing through the winding and a time change of the actual current are detected. In the motor voltage detecting means, a voltage command value of each phase winding of the motor is detected, or a voltage of each phase winding of the motor is detected. The rotor position detecting means detects a rotor position signal or a rotor position signal and a rotor speed signal based on the output of the motor current detecting means and the output of the motor voltage detecting means. Therefore, speed control and position control of the synchronous motor can be realized without using a position detector.

[0022]

An embodiment according to the present invention will be described below. In the description of the embodiments, the same components as those in the conventional example are denoted by the same reference numerals, and the description will be omitted because the description will be repeated.

FIG. 1 is a block circuit diagram showing a control device for a synchronous motor according to one embodiment of the present invention, and FIG. 2 is a specific block circuit diagram of a main part (current control circuit 7) of the control device. is there.

In the synchronous motor control device shown in FIG. 1, the current control circuit 7 includes current detectors 55, 5 and 5, as shown in FIG.
6, 57, adders 28, 29, 30, current error compensation circuits 31, 32, 33, voltage command adders 34, 35, 36,
It includes comparators 37, 38, 39, drive circuits 40, 41, 42 and a three-phase inverter 43.

In the current detector 55, the U-phase current I
U is detected, and a detection current DIU is generated. The current detector 56 detects the V-phase current IV and generates a detection current DIV. The current detector 57 detects the W-phase current IW and generates a detection current DIW.

In adder 28, U-phase current command SIU
Is subtracted from the U-phase current detection value DIU, and the subtracted current value is output to the current error compensation circuit 31. In the adder 29, the V-phase current detection value DIV is obtained from the V-phase current command SIV.
Is subtracted, and the subtracted current value is used as the current error compensation circuit 32.
Is output to In the adder 30, the W-phase current command SI
The W-phase current detection value DIW is subtracted from W, and the subtracted current value is output to the current error compensation circuit 33.

The current error compensating circuits 31, 32, and 33 perform processes such as proportional and integral control on the input current error signal. The signals subjected to this processing are output to the voltage command adders 34, 35, 36 of each phase as voltage command components based on the current error of each phase. The voltage command adder 34 adds the output voltage command component and the voltage feedforward signal VFU to generate a voltage command signal SVUP. Similarly, the voltage command adder 3
5, the output voltage command component and the voltage feedforward signal VFV are added, and the voltage command signal SVV
P is generated. The voltage command adder 36 outputs the output voltage command component and the voltage feedforward signal VFW.
Are added to generate a voltage command signal SVWP. Although detailed description is omitted, the voltage feedforward signals VFU, VFV, and VFW of each phase are values obtained by summing the impedance drop of the motor, the predicted value of the induced voltage, and the like.

Next, the voltage command signals SVUP, SV of each phase
The PWM processing for pulse width modulation of VP and SVWP will be described.

The signal SAW input to each of the comparators 37, 38 and 39 is a triangular wave signal shown in FIGS. 11 (j), (l) and (n). The comparator 37 compares the signal SAW with the voltage command signal SVUP, and obtains a PWM command signal PWMU shown in FIG. The comparator 38 compares the signal SAW with the voltage command signal SVVP to obtain a PWM command signal PWMV shown in FIG. In the comparator 39, the signal SAW is compared with the voltage command signal SVWP, and a PWM command signal PWMW shown in FIG.

The three-phase inverter 43 is composed of six power transistors as shown in FIG. Drive circuit 40
Is a three-phase inverter 43 based on the PWM command signal PWMU.
Signal B1 for driving the two power transistors
And B2. The drive circuit 41 outputs the PWM command signal P
Based on WMV, base signals B3 and B4 for driving two power transistors of the three-phase inverter 43 are generated. The drive circuit 42 generates base signals B5 and B6 for driving two power transistors of the three-phase inverter 43 based on the PWM command signal PWMW. Normally, each of the base signals B1 to B6 is electrically insulated from the control circuit.

The U-phase will be described.
Is in phase with the PWM command signal PWMU and the base signal B2 is O
The relationship between N and OFF is set substantially opposite to the relationship with the base signal B1. If the upper and lower power transistors are energized simultaneously, an excessive short-circuit current will flow from the power supply,
When the signals of the base signals B1 and B2 are inverted, a time during which the base signals B1 and B2 are both turned off, which is generally called a dead time, is set in order to prevent a vertical short circuit of the transistor. This dead time is set to a value having a margin in which the operation delay time of the power transistor is sufficiently considered.

The PWM control is usually performed at about 1 to 20 KHz, and is controlled by a three-phase inverter 43 composed of six power transistors. As a result, the output current of the three-phase inverter 43, that is, the three-phase currents IU and IV of the motor are obtained. , IW have waveforms that increase and decrease substantially in synchronization with the PWM signal as shown in FIGS. 11 (g), (h), and (i), for example.

In the case where the operation described above is performed almost ideally, the current detection values DUI, DIV, DIW of each phase are obtained.
FIG. 10 shows characteristic examples of the voltages VU-NA, VV-NA, and VW-NA of each phase. Voltages VU, VV, VW are terminal voltages of each phase of the motor, and voltage NA is a neutral point potential of the star-connected motor winding.

The control device shown in FIG. 1 is provided with motor current detecting means 17 to which motor current information DTI is input. Motor current information DT
I is specifically the current detection values DUI, DIV and D of each phase.
This is a general term for IW. Further, in the current control circuit 7 shown in FIG. 2, if it is assumed that precise feedback control of each current is performed, the current commands SIU, S
IV, SIW and current detection value DIU, DIV, DI of each phase
Since it is almost equal to W, current commands SIU, SIV, and SIW of each phase can be used as motor current information DTI.

FIG. 3 shows a specific configuration of the motor current detecting means 17. The motor current detecting means 17 includes a measurement timing setting means 74, sample and hold circuits 60 to 65, analog switches 66 to 71, and an AD converter 72.

The measurement timing setting means 74 generates measurement timing signals PTM1 and PTM2 at timings TM1 and TM2 shown in FIG. 10, for example. The generated measurement timing signals PTM1 and PTM2 are output to each of the sample and hold circuits 60 to 65 and the voltage detecting means 75.

In the sample and hold circuits 60, 62 and 64, the current detection value D of each phase is determined according to the measurement timing signal PTM1 generated at the timing TM1.
IU, DIV, and DIW are held. The sample and hold circuits 61, 63, and 65 hold the current detection values DUI, DIV, and DIW of each phase in accordance with the measurement timing signal PTM2 generated at the timing TM2.

In the AD converter 72, the output values of the sample hold circuits 60 to 65 sequentially output through the analog switches 66 to 71 are AD converted.

The timings TM1 and TM2 have a time interval of a fixed time Δt. When the current value is detected at each phase timing, the current value and the current change at the timings TM1 and TM2 can be detected. Note that the timing TM1
Is determined in consideration of the control cycle of the entire motor. As another method of detecting a change in the current value of each phase, the differential value of the current value of each phase is directly calculated in addition to the method of measuring the current value at a time interval of a fixed time Δt as described above. A method of measuring may be used.

In the voltage detecting means 75, between the timing TM1 and the timing TM2, the output voltage of the inverter bridge of each phase, that is, the voltage VU between the terminals of the motor.
The average value of -VV, VV-VW, and VW-VU is detected. Based on the detected average value, the voltage detection means 75 outputs the terminal voltage detection values DUV, DVW, DWU to the rotor position detection means 11. The specific voltage detection performed by the voltage detection means 75 includes a method of directly detecting the output voltage of the inverter bridge, and a method of detecting the PWM to the inverter bridge.
A method of logically calculating and estimating from the command signal and the inverter DC power supply voltage, a method of using the voltage command signals SVUP, SVVP, SVWP of each phase instead of the voltage between the terminals of the electric motor are used.

The size of the time interval Δt to be measured can be selected from the time interval equal to or less than the PWM cycle to several tens of the PWM cycle in accordance with the required specifications of the control device. For example, FIG. 11 shows an example in which the measured time interval Δt is short. The current detection of each phase is shown in FIG.
Part of the current that increases and decreases in synchronization with the PWM control shown in (h) and (i) is detected at timings TM1 and TM2. In the voltage detection of each phase, FIG. 11 shows the PWM command signal PW of each phase.
As shown by MU, PWMV, PWMW, the timing TM
1, between TM2, the potential VU is equal to the DC voltage VP of the inverter.
M and potentials VV and VW are zero.

When the time interval Δt to be measured is short, it is characterized in that relatively accurate voltage detection can be performed from the PWM command signal without directly measuring each phase output voltage of the inverter bridge of the three-phase inverter 43. Another feature is that high-speed detection of the rotor rotational position is possible. On the other hand, the time interval Δ
When t is long, the amount of change in the motor current during the time interval Δt is large, so that there is a feature that the detection resolution of the time change rate of the current can be increased. Therefore, the measurement accuracy of the rotor rotation position AR can be improved, and each command value signal in the control device can be substituted without directly measuring the values of the motor voltage and the motor current.

The output of the motor current detecting means 17 and the output of the motor voltage detecting means 18 are input to the rotor position detecting means 11 shown in FIGS. 1 and 3, and a rotor position signal or a rotor position signal and a rotor speed signal are inputted. Is detected. The rotor position detecting means 11 has several realizing means described below.

First, an algorithm for detecting the rotor position signal AR and the rotor speed signal DAR from each detection signal will be described. The motor constants, definitions of the control variables, and their correlations are as follows.

Field magnetic flux magnitude TB Rotor rotation position AR Motor current phase AA Motor current phase AR-AA with respect to rotor rotation position Motor current amplitude DIO, field current amplitude FI, armature current amplitude AI and TAN (AR-AA) ) Is represented by the following equation (4).

The square of the electric motor current amplitude DIO DIO ² = FI ² + AI ² = 2/3 (IU ² + IV ² + IW ² ) (4-1) Field current amplitude FI = DIO · sin (AR-AA) (4) -2) Armature current amplitude AI = DIO · sin (AR-AA + 90 °) (4-3) TAN (AR-AA) = FI / AI (4-4) Currents IU, IV, IW actually flowing in each phase Is represented by the following equation (5). The current component of current IU is represented by the sum of field current component FIU and armature current component AIU.
Similarly, the current component of the current IV is represented by the sum of the field current component FIV and the armature current component AIV. The current component of current IW is represented by the sum of field current component FIW and armature current component AIW.

U-phase current IU = DIO · sin (AA + 90 °) = FIU + AIU (5-1) V-phase current IV = DIO · sin (AA + 90 ° + 120 °) = FIV + AIV (5-2) W-phase current IW = DIO · sin (AA + 90 ° + 240 °) = FIW + AIW (5-3) In relation to the field current component flowing through each phase and the armature current component described with reference to FIG. 15, the following equations (6) and (7) are used. Can be represented.

U-phase field current FIU = FI · sin (AR) = DIO · sin (AR-AA) · sin (AR) (6-1) V-phase field current FIV = DIO · sin (AR-AA) · Sin (AR + 120 °) (6-2) W-phase field current FIW = DIO · sin (AR-AA) · sin (AR + 240 °) (6-3) However, the relationship between the field current and the magnetic flux is magnetic. There is a typical non-linear element, and it is necessary to accurately calculate it by compensating for this non-linearity.

U-phase armature current AIU = AI · sin (AR + 90 °) = DIO · sin (AR−AA + 90 °) · sin (AR + 90 °) (7-1) V-phase armature current AIV = AI · sin (AR + 90) {+120}) (7-2) W-phase armature current AIW = AI · sin (AR + 90 ° + 240 °) (7-3) Next, the motor is represented by an equivalent circuit modeled, and the terminal voltage of each phase of the motor is expressed. The correlation between VU, VV, and VW will be described. The motor can be represented by an equivalent circuit shown in FIG. In FIG. 6, reference numeral NA denotes a neutral point potential, VPW denotes a DC voltage of the inverter, VUP, VVP, and VWP denote induced voltages related to field magnetic fluxes of three phases, LL denotes a leakage inductance of the motor, and R denotes an internal resistance. . The line voltage VUV is the difference between the terminal voltages VU and VV, and the line voltage VVW is the terminal voltages VV and VV.
The difference between W and the line voltage VWU is the difference between the terminal voltages VW and VU, that is, the difference between the phase voltages, and is expressed by the following equation (8).

VUV = VU−VV = VUP + IU * R + LL * d (IU) / dt− (VVP + IV * R + LL * d (IV) / dt) (8-1) VVW = VV−VW = VVP + IV * R + LL * d (IV) ) / Dt- (VWP + IW * R + LL * d (IW) / dt) (8-2) VWU = VW-VU = VWP + IW * R + LL * d (IW) / dt- (VUP + IU * R + LL * d (IU) / dt) (8-3) Here, in order to theoretically develop only the induced voltages VUP, VVP, and VWP, values obtained by removing the voltage drop due to the leakage inductance LL and the internal resistance R of the motor from the motor terminal voltage are respectively VUVP, VVWP, and VWUP. And This VU
VP, VVWP, and VWUP are represented by the following equation (9).

VUVP = VUP-VVP = VU-IU * R-LL * d (IU) / dt-VV + IV * R + LL * d (IV) / dt (9-1) VVWP = VVP-VWP = VV-IV * R -LL * d (IV) / dt-VW + IW * R + LL * d (IW) / dt (9-2) VWUP = VWP-VUP = VW-IW * R-LL * d (IW) / dt-VU + IU * R + LL * d (IU) / dt (9-3) FIGS. 8 and 9 are vector relationship diagrams of each voltage. FIG.
Is a characteristic diagram in which an AC voltage and an AC current are applied to the one-phase winding of the electric motor shown in FIGS. 16 and 17, the rotor 6 is rotated from 0 to 360 degrees, and the inductance at each rotor rotation position is measured. It is. The maximum inductance LMX occurs at a position where the direction of the magnetic pole of the rotor 6 is orthogonal to the exciting winding, and the minimum inductance LM occurs at a position where the direction of the magnetic pole of the rotor 6 is parallel to the exciting winding.
X occurs.

Next, the magnitude TB of the field magnetic flux is expressed by the following equation (10), where the number of windings of each phase winding is WNG.

TB = 3/2 · FI · WNG / MR = KTB · FI = KTB · DIO · sin (AR-AA) (10) where MR is the magnetic resistance in the direction of the magnetic pole, and KTB is a constant. KTB is as follows.

KTB = 3 / 2.WNG / MR Next, each known variable is clarified, and the other variables are also represented by known variables. First, the currents IU, I
V and IW are measured and known values. Time interval of each phase Δ
The current change rates ΔIU, ΔIV, and ΔIW that change during t are also measured, and the current change rates ΔIU, ΔIV, and ΔIW are represented by the following equation (11).

D (IU) / dt ≒ ΔIU / Δt (11-1) d (IV) / dt ≒ ΔIV / Δt (11-2) d (IW) / dt ≒ ΔIW / Δt (11-3) By substituting the value obtained by (11) into equation (9), VUVP, VVWP, V
The value of WUP is determined.

Although the magnitude DIO of the motor current is known at the command level, a difference from the measured current may adversely affect other estimation calculations. Therefore, a measured value is used for the magnitude DIO of the motor current, and can be obtained by the following equation (12).

[0057]

(Equation 1) Further, a change rate ΔDIO of the magnitude DIO of the motor current during the time interval Δt is obtained by the following equation (13).

[0058]

(Equation 2) The phase angle AA of the motor current with respect to the U phase can also be obtained for the same reason. The phase angle AA can be actually measured by the following equation and calculated.

IU = DIO · sin (AA + 90 °) IV = DIO · sin (AA + 210 °) The rate of change ΔAA of the phase angle AA during the time interval Δt is obtained by (AA + ΔAA) from the following equation. It can be obtained as a difference.

IU + ΔIU = (DIO + ΔDIO) · s
in (AA + ΔAA + 90 °) IV + ΔIV = (DIO + ΔDIO) · sin (AA +
ΔAA + 210 °) The time change rate dTB / dt of the magnitude TB of the field magnetic flux is obtained by the following equation (14).

DTB / dt = d (KTB · DIO · sin (AR-AA)) / dt = KTB · (dDIO / dt · sin (AR-AA) + DIO · cos (AR-AA) · (d (AR) / Dt−d (AA) / dt)) ≒ KTB · (ΔDIO / Δt · sin (AR-AA) + DIO · cos (AR−AA) · (d (AR) / dt−ΔAA / Δt)) = KTB · (ΔDIO / Δt · sin (AR−AA) + DIO · cos (AR−AA) · (d (AR) / dt−ΔAA / Δt)) (14) In the above equation (14), the first term is a change in current amplitude. This is the induced voltage component caused by the change in magnetic flux due to the minute. The second term is A
This is a change of the exciting current with respect to the change of R-AA.

Next, the induced voltage components VUP, VV of each phase
The relationship between P and VWP will be described. Induced voltage component V
UP, VVP, and VWP are obtained by the following equation (15).

VUP = WNG · (d (TB) / dt · sin (AR) + cos (AR) · d (AR) / dt · TB) (15-1) VVP = WNG · (d (TB) / dt · sin (AR + 120 °) + cos (AR + 120 °) · d (AR) / dt · TB) (15-2) VWP = WNG · (d (TB) / dt · sin (AR + 240 °) + cos (AR + 240 °) · d ( AR) / dt · TB) (15-3) In the equation (15), the first term is an induced voltage caused by a change in the magnitude of the field magnetic flux TB and a change in the interlinkage magnetic flux of each phase winding. Component. The second term is a component in which the interlinkage magnetic flux of each winding changes as the field magnetic flux TB rotates.

Incidentally, it is more reasonable to obtain the measured value of each voltage of the electric motor from the average voltage during the time interval (measurement period) Δt, and the measurement accuracy of the rotor rotational position AR can be improved.

Next, a method for obtaining the rotor position signal AR and the rotor speed signal DAR = d (AR) / dt, which are unknown variables, from the above relational expressions relating to the voltage and current of the motor will be described.

Equation (10) and equation (1) are added to equation (15).
4) is substituted, and the phase induced voltages VUP, VVP, VWP of the substituted equation (15) are respectively substituted into the equation (8). As a result, there are only two unknowns: the rotor position signal AR and the rotor speed signal DAR = d (AR) / dt.
This unknown is obtained by solving a system of equations by algebraic calculation. That is, the rotor rotation position AR is given by the following equation (1).
6).

[0067]

[Equation 3] Here, K, L, and M are known values and are represented by the following equations.

[0068]

(Equation 4) Although two solutions are obtained, the correct answer is a value that is satisfied by substituting into the equation (8). Rotor speed signal DAR = d (A
R) / dt is the rotor rotation angle A in the equations (15) and (9).
It is determined by substituting R.

However, there are three equations in equation (9), and three equations are derived for the equation of the rotor speed signal DAR. It is not always possible to solve with any of the three equations derived. As described above, the first term on the right side of Expression (15) is an induced voltage component caused by a change in the interlinkage magnetic flux of each phase winding due to a change in the magnitude of the field magnetic flux TB. The second term on the right side is a component in which the interlinkage magnetic flux of each winding changes due to the rotation of the field magnetic flux TB. Therefore, for example, the field flux T
When B is constant and rotating at a constant rotation speed, the first term on the right side of equation (15) is zero, and the second term on the right side is a value that varies in a trigonometric function. For example, in the selected equation, when the coefficient of the rotor speed signal d (AR) / dt is zero, the rotor speed signal d (AR) / dt becomes indefinite, and when the coefficient is not zero but close to zero, The problem of poor detection accuracy remains.

One method for stably obtaining the rotor speed signal d (AR) / dt is as follows. In each of the three sets of equations, the magnetic flux of the field rotates due to the rotation of the rotor. Are squared, and then three equations are added to add the rotor speed signal d (AR) / d
This is a method for obtaining t. The advantage of this method is that the rotational speed can be obtained stably, but the disadvantage is that the calculation becomes complicated.

Next, the rotor speed signal DAR = d (AR)
Another method for determining / dt will be described. In this other method, two equations having different phases are compared, and the calculation is performed using the equation having the larger absolute value of the coefficient of the unknown rotor speed signal DAR = d (AR) / dt. Specifically, the following equations (17) and (18) are used for calculation.

[0072]

(Equation 5) | Cos (2 * AR-AA) | <| sin (2 * AR-
AA) | DAR = 1 / (DIO * sin (2 * AR-AA)) * ((VUVP-VWUP) / (3 * WNG * KTB)-[Delta] DIO / [Delta] t * sin (AR) * sin (AR- AA) + DIO * ΔAA / Δt * sin (AR) * cos (AR-AA)) (18) Further, two rotor speed signals DAR = d (AR) / obtained by the above equations (17) and (18). dt to | sin (2
* AR-AA) | It is conceivable to use a method of weighting according to the magnitude of | In this method, for example, when one of the detected values is at a rotational position with low detection accuracy, the rotor position signal is not used, and for the other portion, weighting is performed according to the detection reliability of both the detected signals. And then averaging is performed. When this method is used, the detection accuracy is improved, and the fluctuation at the time of switching of the rotor speed signal, which is observed in the case of switching between the two detection signals instantaneously, can be reduced.

Further, as another method for obtaining the rotor rotational speed DAR, there is a method in which the detected rotor position signal AR is used, and a time variation of the rotor position signal AR is calculated to obtain the rotor speed signal DAR. is there.

In the algorithm for detecting the rotor position signal AR and the rotor speed signal DAR described above,
A field magnetic flux TB is formed by field current components FIU, FIV, FIW, and armature current components AIU, AIV, AIW
Has no effect on the field magnetic flux TB at all. In other words, the algorithm is created by a solving method assuming an ideal motor in which the armature reaction acts only to a negligible extent. Therefore, when the effect of the armature reaction of the motor to be controlled is large, it is necessary to correct the armature reaction. Since the armature reaction changes the position and magnitude of the field magnetic flux TB depending on the armature current components AIU, AIV, and AIW, a specific method of correction is to calculate the position and magnitude of the field magnetic flux TB by using Equation (10) and Equation (10). It is corrected in (14) and equation (15). Particularly, since the position of the field magnetic flux TB changes greatly, the following value obtained by adding compensation proportional to the armature current to the value of the rotor position signal AR indicating the field position in the equation (15) is specifically used. A method of solving by replacing the value with the value of the position signal AR is used.

AR + KAM · DIO · cos (AR−AA) Here, KAM is a proportional constant related to the armature reaction. Another simple method is to add the correction value KAM · DIO to the value of the rotor position signal AR, which is the result of the above solution.
A method may be used in which the cos (AR-AA) is simply added and the rotor position signal AR to which the correction value is added is used as the rotor position signal AR.

Next, the entire control device of the synchronous motor shown in FIG. 1 will be described.

In the control device shown in FIG.
2, an amplifier 13, a motor voltage detecting means 18 and a rotor position detecting means 11 are provided.

In the adder 12, the position command PO
The SC and the rotor position signal AR are compared and subtracted to generate a position error signal ES. The amplifier 13 amplifies the position error signal ES and generates a speed command SI. In the control device according to the present embodiment, position control can be realized without using the external position detector 5 shown in FIG. In the motor voltage detecting means 18, for example, the output potential of the inverter, that is, each terminal voltage VU, V
A method for more accurately estimating V and VW from the control signal in the control device will be described.

For example, when detecting the U-phase output potential VU of the inverter bridge in the three-phase inverter 43, the outline of the output potential VU can be determined by the PWM command signal PWMU. However, in the drive signals of the upper and lower power devices of the inverter bridge, as described above, the upper and lower power devices never conduct at the same time, and an excessive short-circuit current does not flow. In other words, there is a so-called dead time during which both the upper and lower power devices are in the OFF state in order to prevent the occurrence of an excessive short-circuit current, and during this dead time, the U-phase output potential VU is controlled by the PWM command signal PWM.
Cannot be estimated from U. The dead time period depends on the switching characteristics of the power device, but usually 10% of the entire period
Set before and after, this period cannot be ignored.

If a U-phase current IU flows during a period in which both the upper and lower power devices are OFF, the current IU always flows through a reverse diode mounted in parallel in the reverse direction to the power device. U-phase current I
If U is positive, U-phase potential VU becomes zero, and if U-phase current is negative, U-phase potential VU becomes inverter DC voltage VPW.

When both the upper and lower power devices are OFF,
If the U-phase current IU is also zero, the U-phase upper and lower power devices are electrically insulated from the U-phase winding 8 of the motor. The U-phase potential VU does not simply become zero or the potential of the inverter DC voltage VPW, but is affected by the V-phase and W-phase.
The phase potential VU becomes a potential related to the rotor rotation position AR. At this time, since the rotor rotation position AR is not known, the U-phase potential VU cannot be estimated from known data.

In such a case, a method of re-measuring at a timing when the current of each phase does not become zero is adopted, and the U-phase potential VU is obtained. Also, the U-phase potential VU can be obtained by using the following method. First, during the period when the U-phase potential VU is unknown, the voltage command signals SVUP, SVVP, SVW
How to substitute with P. Second, the previous rotor rotation position AR
And the current rotor rotation position A from the rotor rotation speed DAR
A method in which R and the rotor rotation speed DAR are estimated, and these values are substituted into Expressions (10), (14), (15), and (8). Third, a method in which a slight measurement error is allowed, and the U-phase potential VU is calculated as a half of the inverter DC voltage VPW.

The rotor position detecting means 11 is, for example, as shown in FIG.
As shown in FIG. In the memory 77, the rotor rotation position AR and the rotor rotation speed D for the combination of data that can be detected
AR is stored, and the corresponding rotor rotation position AR and rotor rotation speed DAR are read out. Address setting means 7
6, the output ADO of the AD converter 72 including the current signal of each phase of the motor at the timings of the timings TM1 and TM2 and the output voltage of the inverter bridge, that is, the inter-terminal voltage detection values DUV, DVW, DWU of each phase of the motor.
Is input to the memory 7 corresponding to the values of these input variables.
7 are set. Input variable is timing TM
1, respectively, and the three-phase current values at timing TM2. The input variable is the timing T
Each of the three-phase current values at M1 is a current change value that has changed before reaching the timing TM2, and an average value of the motor terminal voltages from the timing TM1 to the timing TM2.

Using the above input variables, it is logically possible to set the address of the data. However, when simply calculated, there are 9 input variables. For example, assuming that the resolution of each input variable is 6 bits in binary, it is 5 in binary.
A memory corresponding to a 4-bit address is required. That is, the required memory capacity of the memory 77 increases.

The following method is employed to reduce the memory capacity. First, the three-phase current signal can be reduced to two-phase values because the sum of the three phases is zero, and the same applies to the current change, so that two input variables can be reduced. Second, the time interval Δt to be measured is set to a cycle shorter than the PWM control cycle, and is limited, for example, at a timing when the value of the U-phase potential VU is the inverter DC voltage VPW and the V-phase potential VV and the W-phase potential VW are zero. How to measure. This second
In the method (1), the input variables relating to the voltage become unnecessary, and three input variables can be reduced. Third, a motor current amplitude DI obtained by synthesizing a three-phase current as a synchronous motor control method
A method in which O is set to be constant and output torque is variable only by the relative phase between the rotor position and the current phase. In the third method, the input variables relating to the current are only the current phase AA of the motor and the current change rate of two of the three phases,
One input variable can be reduced. Fourthly, there is provided a method in which the motor current amplitude DIO is not set to be constant but can take several values such as 2, 4, and 8 types in the third method. Also in the fourth method, the memory capacity of the memory 77 can be reduced.

There are several other methods for reducing the input variables described above. If all of the above reduction methods are employed, the number of input variables becomes three, and the memory capacity of the memory 77 can be further reduced. That is, if the resolution of each input variable is 6 bits in binary, the total is 18 bits, and the rotor rotational position AR
If the detection resolution of the rotor rotation speed DAR is, for example, 1 byte, the required memory capacity is 512 Kbytes.
This value is in a sufficiently practical range.

The above-described rotor position detecting means 11 may be constituted by a neural network shown in FIG. In the neural network, the rotor rotation position AR and the rotor rotation speed DAR for the detectable data combination are obtained in the same manner as described above. The neural network includes registers 78, 79, 80, 81, an input layer, a hidden layer, and an output layer. Registers 78-8
At 1, the current values and the current change rates of the two phases of the three phases at the timings TM1 and TM2 are temporarily stored.
Neural networks need not be special ones, but may be very general ones. The example neural network shown in FIG. 5 has a three-layer structure of an input layer, an intermediate layer, and an output layer, and has a rotor rotation position AR and a rotor rotation speed DA.
R is obtained as output.

The learning process of the neural network can be realized by the following method. First, an absolute encoder for obtaining a true value is mechanically connected to and attached to the rotor of the electric motor. In this absolute value encoder, true values of the rotor rotation position AR and the rotor rotation speed DAR are generated. Next, learning by back propagation is repeatedly performed until the difference between the generated true value and the output of the neural network approaches zero. As a result, a neural network learning process can be realized.

The input variables and the method of reducing the input variables are the same as those described above, and can be applied to a neural network as needed.

In the above description, the description has been made on the assumption that the current and voltage necessary for knowing the state of the motor are appropriately applied to the motor. For example, when the control device of the synchronous motor is turned on. At the time, no voltage and current has yet been applied to the motor. In this state, it is impossible to detect the rotational position of the rotor from the voltage and current of the electric motor, and some auxiliary means is required.

There are the following several methods as this auxiliary means. First, in a system in which the operation preparation operation is allowed immediately after the power is turned on, an appropriate current is applied to each phase before the operation is started, the rotor is oriented in a certain direction, the rotor rotation position is specified, and then the operation is started. How to start. Second, a method of measuring the rotor position by applying a current for rotor position measurement to each phase to such an extent that the rotor does not rotate so much or causes no actual harm. Third, a method in which a certain amount of exciting current always flows even when the rotational speed and the output torque are zero, and the rotor rotational position is easily detected when the synchronous motor is operated.

In the synchronous motor control device according to the present invention which does not use the external position detector 5, the rotor rotational position AR and the rotor rotational speed DAR are relatively easily affected by noise and the like. It is generated based on the detection value and may include a noise-like detection error. Therefore, in the present invention, the detected rotor rotational position AR,
Filter processing, averaging processing, and the like are performed on the rotor rotational speed DAR, and the rotor rotational position AR,
The rotor rotation speed DAR is used.

Further, in the present invention, several examples of the detection method have been described. However, the present invention is not limited to the use of one method alone, but a plurality of methods may be used in combination, or a plurality of methods may be averaged. Can be used. In particular, when a plurality of methods are used in combination, or when a plurality of methods are averaged and used, it is desirable to use a detection method for removing abnormal values in combination. When a plurality of detection methods are used, the reliability of the detection value can be improved, and the detection accuracy can be improved.

The electric motor used in the present invention is a synchronous motor that can obtain a rotational force by utilizing the fact that the magnetic resistance varies depending on the rotational position of the rotor. That is, the synchronous motor having the simple structure shown in FIG.
A wide range of synchronous motors can be used, up to synchronous motors having complicated rotor cross-sectional structures as shown in FIG.

Each of the synchronous motors shown in FIGS. 13 and 14 has four poles. Reference numeral 82 denotes a rotor shaft, 83 denotes a part of a silicon steel plate serving as a magnetic path of a magnetic pole, and 84 denotes a part obtained by punching a part of a silicon steel plate by punching a steel plate. This is a part that shows a large resistance. In particular, the synchronous motor shown in FIG. 13 has a structure in which the rotor magnetic poles have a very large magnetic resistance in the rotation direction, and the effect of the armature current components AIU, AIV, and AIW on the field magnetic flux is very small. The characteristics close to are easily obtained.

Reference numeral 85 denotes a permanent magnet.
Are indicated by N pole and S pole. In the synchronous motor shown in FIG. 14, since a part of the field magnetic flux is constituted by the permanent magnet, the operation principle is similar, but the rotor rotation position AR,
When logically calculating the rotor rotation speed DAR, it is necessary to partially modify or modify the above-mentioned relational expression.

The present invention is not limited to the above-described embodiment, but can be variously modified without departing from the scope of the invention.

For example, first, the three-phase synchronous motor has been described in the above embodiment, but the present invention can be applied to other polyphase synchronous motors. Variations can also be made in the number of poles of the synchronous motor used in the present invention.

Second, in the above-described embodiment, a three-phase full bridge is used for the inverter bridge. However, the present invention can be applied to other configurations that can achieve the same function, or to a simplified driving device such as a three-phase half bridge. it can.

Third, in the control device for a synchronous motor according to the present invention, any control theory and control method that can obtain the same technical idea can be adopted. In particular, the control method used in the control device of the synchronous motor according to the embodiment is a method of controlling the field current component and the armature current component separately, but the present invention is not limited to this control method.

Fourth, the present invention can be applied to a linear motor control device having a similar driving principle.

[0102]

According to the present invention, it is possible to provide a control device for a synchronous motor capable of realizing control of the synchronous motor without using a position detector mechanically attached to the synchronous motor. Further, in the present invention, it is possible to provide a synchronous motor control device which can reduce costs, achieve miniaturization, and has high reliability.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a control device for a synchronous motor according to one embodiment of the present invention.

FIG. 2 is a block diagram showing a current control circuit of the control device.

FIG. 3 is a block diagram showing a motor current detecting means of the control device.

FIG. 4 is a block diagram showing a rotor position detecting means of the control device.

FIG. 5 is a block diagram showing another rotor position detecting means of the control device.

FIG. 6 is a circuit configuration diagram used for measuring the inductance of the synchronous motor.

FIG. 7 is an electrical characteristic diagram of the synchronous motor.

FIG. 8 is a voltage vector diagram of the synchronous motor.

FIG. 9 is a voltage vector diagram of the synchronous motor.

FIG. 10 is a diagram showing each current waveform and each voltage waveform of the synchronous motor.

FIG. 11 is a diagram showing control signals of a control device for the synchronous motor.

FIG. 12 is a circuit diagram showing a three-phase inverter of the control device.

FIG. 13 is a sectional view of another example of the synchronous motor.

FIG. 14 is a sectional view of another example of the synchronous motor.

FIG. 15 is a block diagram showing a conventional synchronous motor control system.

FIG. 16 is a sectional view of the synchronous motor.

FIG. 17 is a sectional view showing a modeled synchronous motor.

FIG. 18 is a winding layout diagram of the synchronous motor.

FIG. 19 is a vector diagram of a current of the synchronous motor.

[Explanation of symbols]

 2 Torque control means 7 Current control circuit 11 Rotor position detecting means 17 Motor current detecting means 18 Motor voltage detecting means 21 Armature current command circuit 24 Field current command circuit 34-36 Voltage command adders 37-39 Comparators 40-42 Drive circuit 43 Three-phase inverter 60 to 65 Sample hold circuit 72 AD converter 74 Measurement timing setting means 75 Voltage detection means 76 Address setting means 77 Memory 78 to 81 Register

Claims (5)

(57) [Claims] 1. A control device for a synchronous reluctance motor that obtains a rotational force by utilizing a difference in magnetic resistance according to a change in the rotational position of a rotor of an electric motor. Motor current detecting means for detecting a current command value of the current command and a time change of the current command value, or detecting a real current flowing through each phase winding of the motor and a time change of the real current, and the motor Motor voltage detecting means for detecting the voltage of each phase winding, and calculating the rotor rotational position or the rotor rotational position and the rotor speed based on the output of the motor current detecting means and the output of the motor voltage detecting means. The PWM signal input to the upper and lower power devices of each phase is constituted by a rotor position calculating means for performing the pulse width modulation control and a power device for controlling the pulse width. And an inverter bridge input at an instant. The motor voltage detecting means determines the output potential of the inverter bridge of each phase by the following logic (1), (2) and (3) algorithms. Control device for synchronous reluctance motor. (1) During a period in which one of the PWM signals input to the upper and lower power devices of the inverter bridge of the corresponding phase is ON, the voltage of the motor of the corresponding phase is the inverter output voltage determined by the PWM signal and the DC power supply voltage of the inverter. Is determined by (2) Both the PWM signals input to the upper and lower power devices of the corresponding inverter bridge are OFF,
During the period when the current of the motor of the corresponding phase is not zero,
The voltage of the motor of the corresponding phase is determined by the inverter output voltage determined by the DC voltage of the inverter in the direction in which the current of the motor of the corresponding phase flows in the inverter bridge. (3) If the current of the motor of the corresponding phase is zero and no current flows during the period in which the PWM signals input to the upper and lower power devices of the corresponding inverter bridge are OFF, the voltage of the motor of the corresponding phase Is the potential obtained by estimating the inverter output potential of the corresponding phase from the previously detected rotor position signal and rotor speed signal and the current and voltage of the other phase, or estimated from the voltage command value of each phase. It is determined by the potential. 2. The synchronous device according to claim 1,
In the control apparatus for a reluctance motor, the rotor position calculating means includes a known leakage inductance, an internal resistance of a motor winding,
Based on the simultaneous equations established from the current flowing through each phase winding and the induced voltage of each phase expressed as a function of the unknown rotor rotational position and unknown rotor rotational speed, and the voltage of each phase winding, A control device for a synchronous reluctance motor, which calculates a rotational position and a rotor speed. 3. The control device for a synchronous reluctance motor according to claim 1, wherein the motor current detecting means and the motor voltage detecting means are respectively the motor current detecting means and the motor voltage detecting means. Wherein each measurement period is set within a period corresponding to a pulse width for pulse width modulation control of each power device of the inverter bridge. 4. The control device for a synchronous reluctance motor according to claim 1, wherein the motor current detecting means and the motor voltage detecting means are respectively the motor current detecting means and the motor voltage detecting means. Is set to a period longer than a period corresponding to a pulse width for performing pulse width modulation control on each power device of the inverter bridge, and a measured value of the motor voltage detecting means is obtained as an average value in the measured period. A control device for a synchronous reluctance motor. 5. The control device for a synchronous reluctance motor according to claim 1, wherein the rotor position detecting means obtains an output of the motor current detecting means and an output of the motor voltage detecting means. Storage means for storing a value of a rotor rotational position and a rotor speed corresponding to an input value to be input; address setting means for determining an address for an input value of a value stored in the storage means; An output means for outputting a stored value in accordance with an input value. A control device for a synchronous reluctance motor, comprising: