JP3282160B2 - Spread spectrum transceiver - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a spread spectrum transceiver, and more particularly to a spread spectrum transceiver which transmits digital data using direct spreading in wireless communication or wired communication.

[0002]

2. Description of the Related Art Conventionally, various modulation schemes have been used for wireless data communication. Among them, a spread spectrum communication scheme has attracted attention as a new communication scheme. The modulation system used for general data communication is a narrow band modulation system, which can be realized with a relatively small circuit. However, such as a room (office or factory), it is effective against multipath and narrow band colored noise. Has the disadvantage of being weak.

[0003] On the other hand, the spread spectrum communication system has an advantage that these drawbacks can be eliminated because the spectrum of data is spread by a spread code and transmitted in a wide band. Among such spread spectrum communication systems, the direct spread (DS) system has already been partially put into practical use.

Hereinafter, the configuration will be described with reference to the drawings. FIG. 11 is a block diagram of a conventional DS type transmitter. The data to be transmitted is spread by the spreading unit 2 using the spreading code generated by the spreading code generation unit 1. Assuming that the code length of the spreading code is k, the data is spread k times.
This spread data is called spread data. Data to be transmitted is called a data unit of one bit, whereas spread data is called a chip of data. That is, one bit becomes k chips. Thereafter, the spread data is modulated by the modulator 3 and transmitted from an antenna via a radio circuit (not shown).

FIG. 12 is a block diagram showing a conventional DS type receiver. The spread data transmitted from the transmitter shown in FIG. 11 is received by the receiver shown in FIG. That is, a radio wave is received by an antenna of a wireless circuit (not shown) and converted into an IF signal. The received IF signal is split into two by the splitter 4 and input to the multipliers 5 and 6, respectively. In the multipliers 5 and 6, the IF signal is converted into a baseband I component and a Q component using the cos and sin components of the local signal from the local signal oscillator (VCO) 7. Here, the sin component of the local signal oscillator 7 and c
Although the os component is used to convert to an I component and a Q component, one of a signal component and a local signal may be converted to an I component and a Q component using a phase shifter that rotates the phase by 90 °.

Thereafter, the I and Q components of the baseband are sampled and quantized by A / D converters 8 and 9 and converted into digital data. This data is input to digital correlators 10 and 11, where it is correlated with a PN code, and a correlation output is provided to a timing detector 12 and latch circuits 13 and 14. Timing detector 1
Numeral 2 detects a correlation spike, latches the respective correlation outputs in the latch circuits 13 and 14 at that timing, and obtains data by making a positive / negative determination. The signals latched by the latch circuits 13 and 14 are input to the phase detector 15 and applied to the local signal oscillator 7 via the loop filter 16 to control the oscillated local signal. The data demodulation unit 17 demodulates data according to the output of the latch circuit 13.

[0007]

The receiver shown in FIG. 12 can demodulate a BPSK-modulated spread spectrum communication signal, but has the disadvantage of a large circuit scale. That is, in the case of a circuit-synchronous system, a carrier phase error detection circuit, a PLL or VC
O or the like is required. Therefore, the circuit of the synchronous receiving system becomes large. Further, the circuit requires two systems of digital correlators 10 and 11, which increases the circuit scale.

FIG. 13 is a block diagram showing an example of the digital correlator shown in FIG. The digital correlator shown in FIG. 13 is a 3-bit digital correlator quoted from a Zi340 Z3340 data sheet. Data input in three bits is input to the shift register 101. For example, if the length of the spreading code is 256 chips, a 256-stage shift register is required. Then, the output of each shift register 101 and a spreading code prepared in advance (hereinafter, referred to as a replica)
102 is a multiplier 103, 104, 10 for each chip.
, 10n and then adder 111
Gives the sum. As described above, the digital correlators 10 and 11 include the shift register 101 having only the length of the spreading code, the replica 102, and the multipliers 103, 104, 1
, 10n and an adder 111 are required. Therefore, when the spreading code becomes long, the circuit scale becomes extremely large, which is an obstacle to cost reduction and miniaturization. Further, in the receiver shown in FIG.
11 requires two systems, so that the circuit scale becomes even larger.

[0009] Therefore, a main object of the present invention is to provide a spread spectrum transceiver which is reduced in cost and size.

[0010]

The invention according to claim 1 is
A spread spectrum transmitter for directly spreading transmission data with a code having a large number of chips (n chips) for one bit of data, wherein a primary signal for spreading a data signal using a primary spreading code of k chips is used. Spreading means, differential coding means for differentially coding the output of the primary spreading means for each chip, and m chips (n =
(k * m) secondary spreading means, and a phase modulating means for phase modulating an output signal of the secondary spreading means.

According to a second aspect of the present invention, there is provided a spread spectrum receiver for receiving a signal transmitted from a spread spectrum transmitter, comprising: a distribution unit for dividing a received signal into two;
A multiplying means for multiplying each of the two divided reception signals by each orthogonal frequency component of the carrier frequency of the transmitter, a first correlating means for correlating the output of the multiplying means with a second spreading code, A signal delayed by one chip of the primary spreading code is generated from the output of the correlating means, and differential demodulation means for performing differential demodulation and the demodulated signal are correlated with the spreading code used for the primary spreading means, and data demodulation is performed. And a second correlation means.

According to a third aspect of the present invention, there is provided a spread spectrum receiver for receiving a signal transmitted from a spread spectrum transmitter, comprising: a distribution unit for dividing a received signal into two;
Multiplying means for multiplying each of the received signals divided into two by each orthogonal frequency component of the carrier frequency of the spread spectrum transmitter; first correlating means for correlating the output of the multiplying means with a secondary spreading code; A delay means for generating a signal delayed by one chip of the primary spreading code from an output of the correlation means, a differential phase detection means for detecting a differential phase from a phase before the delay and a phase after the delay, And a second correlation means for taking the correlation of the differential phase and demodulating the data.

According to a fourth aspect of the present invention, there is provided a cosine value calculating means for calculating a cosine value from the differential phase calculated by the differential phase detecting means of the third aspect, and providing the calculated cosine value to the second correlation means.
The second correlating means correlates the output of the cosine value calculating means.

[0014]

[0015]

[0016]

[0017]

[0018]

[0019]

FIG. 1 is a block diagram showing a first embodiment of the present invention. In particular, FIG. 1 (a) shows a transmitter, and FIG. 1 (b) shows a receiver. In FIG. 1A, on the transmission side, data to be transmitted is provided to a primary spreading unit 21 and spread. At this time, if the code length of the spreading code is k, the data is spread k times. The spread data is
The differential encoding unit 22 performs differential encoding for each chip. Each of the differentially encoded data is further spread by the secondary spreading unit 23. At this time, if the code length is m, the differentially encoded spread data is further spread m times. After that, it is phase-modulated by the DBPSK modulator 24 and transmitted as a DBPSK signal.

On the other hand, in the receiver shown in FIG. 1B, the received signal is split into two by a splitter 31 and supplied to multipliers 32 and 33, respectively. Multipliers 32, 3
3, the local signal c from the local signal oscillator 34
Using the os and sin components, the IF signal is converted into baseband I and Q components. Here, the IF signal was converted into the I component and the Q component using the sin component and the cos component of the local signal oscillator. However, the IF signal was rotated by 90 ° using one of the signal component and the local signal. To I
The conversion may be made into a component and a Q component.

Next, the baseband I and Q components are
Secondary correlators 35, 3 that can be transmitted with m-chip spreading codes
6 is input. The secondary correlators 35 and 36 calculate the correlation between the I component and the Q component and the spreading code, and the correlation output is input to the differential demodulation unit 37. In the differential demodulation unit 37,
The baseband signal is obtained by demodulation from the I and Q correlation outputs. This can be realized by a circuit similar to general differential demodulation.

The output signal of the differential demodulation unit 37 is
And the correlation with the correlation code is obtained by the code of k chips used in the first spreading. A correlation spike is detected from the output of the correlator 38 by the timing detection unit 40, and the data is latched by the latch circuit 39 at that timing. By determining whether the data latched by the latch circuit 39 is positive or negative, the data before spreading can be demodulated.

Next, the features of the conventional example and the first embodiment of the present invention will be compared and described. Conventionally, as a transmission system, as shown in FIG. 11, one bit of data is spread by a spreading unit 2 with a spreading code n (k * m) chip generated by a spreading code generation unit 1 and then a modulation unit 3 On the other hand, on the receiving side, the input signal is input to digital correlators 10 and 11 capable of despreading with n chips as shown in FIG. 12, and demodulated based on the result. Therefore, as shown in FIG. 13, the correlator requires n chips.

On the other hand, in the first embodiment of the present invention, as a transmission system, the data generated by the code spreading unit is spread by the primary spreading unit 21 with k bits of spreading code and 1 bit of data is spread. Then, the signal is spread by the secondary spreading unit 23 for each chip described above, and then modulated by the BPSK modulation unit 24 and transmitted. At this time, the relationship is k * m = n. On the other hand, on the receiving side, the input signal is input to the correlators 35 and 36, which can perform despreading with m chips, and thereafter k
A correlator 38 that can obtain a correlation by a chip is used. For this reason, the correlation operation of the transmission system code generator and the reception system becomes somewhat complicated as compared with the conventional example, and the latch circuit 39 is additionally required. However, in such a circuit, the correlator actually has the largest circuit scale. In the conventional example shown in FIG. 12, n stages of digital correlators 10 and 11 are required. For example, when n = 256, each of these 256 becomes a reference for correlating with a shift register. It was necessary to prepare a replica and a multiplier and add each multiplication output. Also, I, Q
Since these two systems are required, 512 circuits are required, and the circuit for this is enormous in size.

On the other hand, in the first embodiment of the present invention, the m-chip correlator and the k-chip correlator are used.
In order to perform the correlation operation independently, each correlator need only prepare k and m replicas, comparators and adders. For example, if k = 16 and m = 16, the required number is 16 * 2 + 16 = 48. As a result, the required circuit scale of the correlator is 48/5 as compared with the conventional example.
12, which makes it possible to significantly reduce the circuit scale.

FIG. 2 is a diagram showing a modeled correlation output showing the operation of the second embodiment of the present invention. As described above, the present invention can be employed in a semi-synchronous system. In this case, since there is a frequency difference between the transmitting side and the receiving side, the I component and the Q component that have become baseband signals include:
Has residual frequency components. If this is δω, if the baseband signal is free of noise or the like, for example, the I-phase signal will be S (p) = A (t) cosσω.

However, the above equation shows only basic terms in two-phase modulation, and when the spread data is 1, A
(T) = 1, and when −1, A (t) = − 1. Here, assuming that the period of the spreading code n = k * m is T, σ
When ωT is smaller than 360 °, the result becomes a0 in FIG. 4, and when σωT cannot be ignored compared to 360 °, it becomes like b0. Here, a comparison when the transmission data is 1 is shown.

If an n-chip correlator is used as in the conventional example, in the case of a0, a correct correlation output can be obtained because the phase is not inverted for a long code period. , B0, the phase is reversed halfway,
Correct correlation cannot be obtained. For this reason, there is a problem that the correlation output becomes small, the operation such as phase detection becomes impossible, and the demodulation cannot be performed. This is known as the frequency error tolerance of a system using spread spectrum. Therefore, when transmitting with a high spreading factor,
Since the period T becomes long, a high-precision oscillator is required, which has been a problem.

Therefore, first, a case where there is no differential between the k-chip spread and the m-chip spread in a system having k * m spread as in the present invention is different from the present invention. In FIG. 2, the correlation output when the correlation is first obtained for m chips under the condition a is as shown by a1. At this time, k chips are 10 chips, and the code is (1100
010001). A1 and b1 in FIG. 2 show m-chip correlator outputs. The output of a1 is (110001000
Since it is 1), the correlation output becomes 10 by inputting to a correlator that can obtain a correlation with k chips, and there is no deterioration.

However, under the condition of b0 in FIG.
The sign becomes 1 and the sign of the k chip is reversed halfway. That is, the output is (110000)
1110). For this reason, when input to a k-chip correlator, the correlation output becomes 0, the correlation output decreases, and it becomes difficult to demodulate data. This is the same problem as the conventional example described above, and is a problem that occurs because the phase is inverted in the middle of k chips.

Therefore, when the present invention is used under the condition of a0, the correlation is first obtained by m chips, and the output when the differential demodulation is performed is as shown by a2.
101). At this time, the code of the k chip is (10
11001101), a correlation output of 10 is obtained, which is the same as above. In the condition of b0,
Use this invention. First, when the correlation is obtained with m chips, the lower 5 bits in the latter half have an inverted phase, so that b
1 becomes (1100001110). When this signal is differentially demodulated, in the case of differential demodulation, since the in-phase is 1 and the anti-phase is 0, the result is (1011101101). The correlator is (1
Since the correlation can be obtained with (011001101), five solutions are wrong, but since the other nine are correctly correlated, 9 is obtained as a correlation output.

At this time, since the phase is inverted on the way, the sign of the correlation output is inverted, but by demodulating the differential data, correct data can be sent out after the sign is inverted. This is because, since the differential is obtained for each chip, if the signal of one chip before the discrimination is also inverted, the correct value can be obtained without any problem. For this reason, although it is erroneous at an inversion point (fifth) in the middle, correct diffusion data can be calculated before and after that. As a result, when this output is input to a k-chip correlator, the correlation can be correctly obtained with some deterioration,
Data can be demodulated. As described above, according to the present invention, demodulation can be performed even when the frequency difference between transmission and reception, which cannot be demodulated in the conventional example, is large.

Although the above description has been made using DBPSK, the present invention is not limited to this, and the same effect can be obtained in DQPSK.

As the second feature, the case of the quasi-synchronous system has been described, but demodulation can be similarly performed by using this in a synchronous demodulator. Also in this case, since the carrier synchronization can be performed as compared with the conventional example, there is an advantage that the accuracy of the local signal oscillator may be low.

Next, a second embodiment of the present invention will be described. The transmitting side is the same as in the first embodiment.

FIG. 3 is a block diagram of a receiver according to the second embodiment of the present invention. In FIG. 3, the distributor 31
The multipliers 32 and 33, the local oscillator 34, and the second-order correlators 35 and 36 are the same as those in the first embodiment shown in FIG. 1B. The outputs of the secondary correlators 35 and 36 are provided to a phase detector 41 to calculate the phase. I component and Q
Since the components are the cos and sin components of the vector to be obtained, the phase can be calculated by a calculation method for obtaining tan-1. Further, it can also be realized by using a method of obtaining a calculation result from a ROM table, and the calculated phase is 0.
From 360 ° to 360 °.

The output of the phase detector 41 is delayed by the delay unit 42 and is supplied to the differential phase detector 43. The differential phase detector 43 detects the phases before and after the delay and gives the phase difference to the phase correlator 44.

FIG. 4 is a block diagram of the phase correlator 44 shown in FIG. 3, and FIG. 5 is a diagram for explaining its operation.

The phase correlator 44 comprises a correlation phase converter 441 and a phase correlation calculator 442, as shown in FIG. In the correlation phase converter 441, the input 0 °
Is converted from −90 ° to 90 °.
This means that, as shown in FIG. 5, the axes of 90 ° and 270 ° are set to 0, the separation between the first quadrant and the fourth quadrant is set to be positive, while the separation between the second and third quadrants is set to be negative. Is calculated as Although this can be easily obtained by calculation, it is easy to use a conversion table such as a ROM.

After the phase conversion, the signal is supplied to the phase correlation calculator 442. Here, the phase correlator 44 sets 90 ° when the spread code is 1 and -9 when the spread code is −1 based on the spread code to be correlated.
The correlation is calculated based on 0 °. For example, when the spreading code to be correlated is 1, and the phase of the received signal is 90 °, it is 1; when it is 80 °, it is 80/90 °; and when it is −60, it is −60/90.

On the other hand, when the spread code to be correlated is -1, the phase is 1 when the phase of the received signal is -90 degrees, 60/90 when the phase of the received signal is -60 degrees, and 30/90 when the phase of the received signal is 30 degrees. In this way, the correlation is obtained for each chip of the spread code, and the sum is output as a correlation output.

Here, when calculating the digital correlation, the phase angle is quantized and input according to the number of bits. Therefore, if the output of the ROM of the correlation phase converter 441 is quantized in advance, the phase correlator 44
It only needs to be multiplied by positive and negative based on the spreading code. For example, if a 3-bit digital correlator is quantized to eight levels, it is divided into eight levels from -90 ° to 90 °, and generally, -7, -5, -3, -1, 1, 1, 3,
It is quantized to eight values of 5,7. A phase correlation operation can be performed by multiplying the value by the spreading code to obtain a sum, thereby obtaining a phase correlation output.

Using the output of the phase correlator 44, the timing detecting section 40 detects the timing of the correlation spike. As a result, the timing of the correlation synchronization can be known. Therefore, after the phase correlation output is latched by the latch circuit 39, the sign is determined, and the signal is demodulated by the signal demodulation unit 45.

As described above, by using the phase correlator 44, the error rate can be improved.

FIG. 6 is a diagram showing characteristics of the phase correlator itself. The characteristics shown in FIG. 6 are not the characteristics in the second embodiment of the present invention as a whole, but show the case where differential demodulation is performed, data is demodulated and input to the correlator, and the case where the phase correlator 44
Is a simulation of the characteristics in the case of using. The horizontal axis is the C / N ratio before differential, and the vertical axis is the error rate. From this result, it is understood that the error rate can be improved when the phase comparator 44 is used in the second embodiment of the present invention.

FIG. 7 is a block diagram showing a receiver according to the third embodiment of the present invention. The receiver shown in FIG.
A cosine transform unit 46 and a correlator 47 are provided instead of the phase correlator 44 of the receiver according to the second embodiment shown in FIG. 3, and the other configuration is the same as that of FIG. The cosine transform unit 46 obtains a cos component from the angle of the differential phase detected by the differential phase detection unit 43. Then, the output of the cosine transform unit 46 is input to the correlator 47, and is correlated with the correlation code. In this embodiment, since the cosine transform unit 46 only finds the cosine component from the input angle, there is an advantage that the circuit configuration can be smaller than that of the phase correlator 64 shown in FIG.

FIG. 8 is a block diagram showing a fourth embodiment of the present invention, in which (a) shows a transmitter and (b)
Indicates a receiver.

First, on the transmitting side shown in FIG. 8A, data is spread by the primary spreading unit 21 with a spreading code having a code length of k chips. Thereafter, the serial / parallel converter 25 collectively collects k bits at a time,
The signal is parallel-converted and supplied to the CSK modulator 26. C
The SK modulator 26 modulates data with a spreading code having an m-chip spreading code by the CSK method. Here, the CSK modulation is, for example, in the case of the 4CSK system, data is bundled every two bits (k = 2), and if the data is (0, 0),
A spreading code a is sent, a spreading code b is sent if the data is (0, 1), a spreading code c is sent if the data is (1, 0), and a spreading code is sent if the data is (1, 1). d is sent.

On the other hand, in the receiver shown in FIG. 8B, the input signal is demodulated by the CSK demodulation unit 51 to data. From the output of the CSK demodulation unit 51, the timing is detected by a correlation timing detector (not shown), latched by a latch circuit, and parallel / serial converted by the P / S conversion unit 52 by the number bundled on the transmission side. As a result, the output of the latch unit becomes the same in the spread set for k chips. Further, the signal is input to a correlator 38 capable of correlating the k chips, and demodulated based on the result. In this case, though not shown, a timing controller for controlling the timing of generating the spreading code in both the transmission system and the reception system, and a correlation timing detector are included.

By using the fourth embodiment of the present invention, it is possible to increase the transmission rate as compared with the first embodiment. In the first embodiment, the spreading rate is k * m times the transmission rate. As a result, on the contrary, if the chip rate after spreading is fixed (because the wireless bandwidth to be transmitted is determined by the chip rate, if the wireless bandwidth to be transmitted is determined, the chip rate becomes constant). Was limited. However, by using the fourth embodiment, it is possible to increase the transmission rate by changing the number of bundled CSKs, even if the spreading factor is k * m. For example, compared to the DBPSK system, the transmission rate is twice as large in the case of the 4CSK system, and three times as large in the case of the 8CSK system.

In the fourth embodiment, the circuit can be simplified in the same manner as in the first embodiment, and the transmission rate can be increased by changing the number of CSKs to be bundled. In addition, by making it variable, it becomes possible to cope with a variable transmission rate system.

FIG. 9 is a block diagram showing a receiver according to a fifth embodiment of the present invention. In the embodiment shown in FIG. 9, the transmitter shown in the first embodiment is used. 9, a PDI demodulation unit 49 is provided instead of the cosine conversion unit 46 shown in FIG. 7, and the other configuration is the same as that of FIG. The differential phase detection unit 43 calculates the phases before and after the delay by the delay unit 42 and supplies the calculated phases to the PDI demodulation unit 49. The PDI demodulator 49 performs path diversity, and the correlator 47 correlates with the correlation code.
Then, the timing detecting section 40 detects the timing at which the correlation can be obtained, and when the timing of the correlation synchronization is known, the data is latched by the latch circuit 39 and demodulated by the signal demodulating section.

In the first embodiment, a second-order correlation is obtained, and a first-order correlation is obtained using the output of the second-order correlation. However, the error rate after the first-order correlation is determined by the input DBPSK.
The error rate of DBPSK depends on the C / N of the signal input to the second-order correlator. Now, considering only the secondary correlation, if the data is determined to be 1, 0 in the demodulation unit after the secondary correlation, the error rate at that time is as shown in FIG. In FIG. 10, a is the error rate when there is fading, and c is the error rate when there is no fading. Considering the case where there is fading, when the result of the graph of FIG. 10 is put into a primary correlator, the error rate according to the error rate characteristic of the result a in the first embodiment is equal to the overall error rate characteristic. Is obtained as On the other hand, when the PDI method is used at the time of secondary correlation at the time of demodulation as in the fifth embodiment, the error rate before the primary correlation becomes the error rate of b. As a result, the overall error rate characteristics can be improved.

As described above, in this embodiment,
By inserting the PDI path diversity method between the second-order and first-order correlators, the characteristics at the time of fading can be greatly improved.

Next, a sixth embodiment will be described.
In the first to fifth embodiments described above, the circuit is simplified by combining the primary spreading code and the secondary spreading code, but the spreading code will be described here.

The spreading code has an auto-correlation characteristic and a cross-correlation characteristic, and both good ones can be used as the spreading code. Therefore, for example, an m-sequence having a code length of 31 has only three types of codes. In reality, various codes other than the m-sequence are used, but the number of each is limited.

In the present invention, the primary spreading code is k1
And k2 secondary spreading codes. In the present invention, the user can be distinguished by this combination. In this case, k1 * k2 codes can be configured,
The number of codes can be greatly increased.

In the present invention, by using the same secondary spreading code, signals can be detected up to that level. Therefore, when secondary spreading codes are used for the same group, they are shared and assigned to different users using primary spreading codes.

For example, when the present invention is used for several networks, a separate secondary spreading code is assigned to each network, and the same network has the same secondary spreading code. Then, separate primary spreading codes are used for individual identification. In this case, a user having the same secondary spreading code can know his / her access status to the network, and can control his / her own communication according to the frequency.

As described above, the embodiment of the present invention has a feature that it is possible to identify a user suitable for an application by properly using the primary and secondary spreading codes.

[0061]

As described above, according to the present invention, by spreading data using the primary spreading unit and the secondary spreading unit, the number of correlators having the largest circuit scale is set to two, and each of the correlators is correlated. To take action, each correlator is k,
It is only necessary to prepare m replicas, comparators, and adders, and it is possible to significantly reduce the circuit scale. Further, by performing the differential operation after the first spreading, demodulation can be performed even when the frequency difference between transmission and reception is large.

The use of a phase correlator can improve the error rate. Further, by using the CSK method, it is possible to increase the transmission rate or to achieve a variable transmission rate system. Furthermore, by inserting the PDI path diversity method between the second-order and first-order correlation sections, the characteristics at the time of fading can be greatly improved. Also, by combining the primary spreading code and the secondary spreading code, k1 * k2 codes can be formed, and the number of codes can be greatly reduced. Further, by properly using the primary and secondary spreading codes, it is possible to identify a user suitable for an application.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a first embodiment of the present invention.

FIG. 2 is a diagram modeling a correlation output showing an operation of the embodiment of the present invention.

FIG. 3 is a block diagram of a receiver according to a second embodiment of the present invention.

FIG. 4 is a block diagram of the phase correlator shown in FIG. 3;

FIG. 5 is a diagram for explaining the operation of the phase correlator of FIG. 4;

FIG. 6 is a diagram showing characteristics of the phase correlator itself shown in FIG. 4;

FIG. 7 is a block diagram showing a receiver according to a third embodiment of the present invention.

FIG. 8 is a block diagram showing a fourth embodiment of the present invention.

FIG. 9 is a block diagram showing a receiver according to a fifth embodiment of the present invention.

FIG. 10 is a diagram showing an error rate of the receiver of the present invention.

FIG. 11 is a block diagram of a conventional DS type transmitter.

FIG. 12 is a block diagram of a conventional DS type receiver.

FIG. 13 is a block diagram of the digital correlator shown in FIG.

[Explanation of symbols]

 Reference Signs List 21 primary spreading section 22 differential encoding section 23 secondary spreading section 24 BPSK modulation section 25 S / P conversion section 26 CSK modulator 31 distributor 32, 33 multiplier 34 local signal oscillator 35, 36 secondary correlator 37 difference Dynamic demodulation units 38, 47 Correlator 39 Latch circuit 40 Timing detection unit 41 Phase detector 42 Delay unit 43 Differential phase detection unit 44 Phase correlator 45 Signal demodulation unit 46 Cosine conversion unit 49 PDI demodulation unit 51 CSK demodulation unit 52P / S converter

──────────────────────────────────────────────────続 き Continuation of the front page (56) References JP-A-8-88587 (JP, A) JP-A-64-18330 (JP, A) JP-A-63-132544 (JP, A) JP-A-7-88 74725 (JP, A) Mitsuo Yokoyama, "Spread Spectrum Communication System", Science and Technology Publishing, p. 149-153, p. 528-530 Tadami Yoshida et al., Code Shift Keying Synchronization Loop for Modulated Spread Spectrum Communication, IEICE Transactions, Vol. J67-B, No. 5, p. 559-565 (58) Fields surveyed (Int.Cl. ⁷ , DB name) H04J 13/00-13/06 H04B 1/69-1/713

Claims (4)

(57) [Claims] 1. A spread-spectrum transmitter for directly spreading transmission data with a code having a large number of chips (n chips) for one bit of data. Primary spreading means for spreading by using; differential coding means for differentially coding the output of the primary spreading means for each chip; m chips (n = k * m) of the output of the differential coding means A spread spectrum transmitter, comprising: a second spreading means for spreading using the second spreading code, and a phase modulation means for phase modulating an output signal of the second spreading means. 2. A spread-spectrum receiver for receiving a signal transmitted from the spread-spectrum transmitter according to claim 1 , wherein said distribution means divides the received signal into two. Multiplying means for multiplying each received signal by each orthogonal frequency component of the carrier frequency of the spread spectrum transmitter; first correlating means for correlating the output of the multiplying means with the secondary spreading code; Differential demodulation means for generating a signal delayed by one chip of the primary spreading code from the correlating means output and performing differential demodulation, and a spreading code using the signal demodulated by the differential demodulating means for the primary spreading means 2. The spread spectrum receiver according to claim 1, further comprising a second correlating means for obtaining a correlation and demodulating data. 3. A spread-spectrum receiver for receiving a signal transmitted from the spread-spectrum transmitter according to claim 1 , wherein the distribution means divides the received signal into two, and the distribution means divides the received signal into two. Multiplying means for multiplying each received signal by each orthogonal frequency component of the carrier frequency of the spread spectrum transmitter; first correlating means for correlating the output of the multiplying means with the secondary spreading code; Delay means for generating a signal delayed by one chip of the primary spreading code from the output of the correlation means; differential phase detection means for detecting a differential phase from a phase before delay and a phase after delay by the delay means; 2. The spread spectrum receiver according to claim 1, further comprising a second correlating means for correlating the differential phases detected by the differential phase detecting means and demodulating the data. 4. A cosine value calculating means for calculating a cosine value from the differential phase calculated by the differential phase detecting means and giving the cosine value to the second correlating means, wherein the second correlating means comprises: 4. The spread spectrum receiver according to claim 3, wherein the correlation between the outputs of the calculating means is obtained.