JP3245954B2 - Power circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type switching inverter type power supply circuit.

[0002]

2. Description of the Related Art A conventional DC power supply circuit configuration generally uses a large-sized power supply transformer, connects a commercial AC power supply to a primary winding thereof, and rectifies and smoothes a secondary-side output.
Apart from this, there was a small and lightweight power supply called a switching power supply. These include half-bridge, full-bridge, and push-pull type switching inverter power supplies using switching elements such as transistors and thyristors. However, the conventional switching inverter type power supply has a square wave for both operating current and operating voltage,
There was much switching noise. In addition, since high-frequency operation is basically performed, power loss during the recovery time of the rectifier diode and the turn-off time of the switching transistor is large, and heat generation has been a problem. In recent years, in such a switching power supply, it has been proposed to reduce the noise and improve the efficiency by making the current or voltage waveform close to a sine wave using resonance (for example, Japanese Patent Application Laid-Open No. 64-43).
062, JP-A-1-91659).

[0003]

However, in the resonance type switching power supply proposed so far, it is difficult to set ON / OFF timing of the switching element, and if this setting is insufficient, the efficiency cannot be improved. In addition, even if it is called a resonance type, most of the current or voltage waveform actually approximates a sine wave by resonance, and the noise generated by these switching circuits is caused by both voltage and current. Considering this, there was a limit to noise reduction. In general, small switching power supplies generally have a high voltage and a small current on the primary side and a low voltage and a large current on the secondary side. In this case, if a current resonance circuit is configured using the leakage inductance of the transformer on the secondary side, the capacitance value of the capacitor becomes large. This is because, depending on the turns ratio, the leakage inductance seen from the secondary side is smaller than that seen from the primary side. In addition, since this is affected by the square of the turns ratio, if a secondary leakage inductance of the transformer is used, the capacitance value of the current resonance capacitor will be, for example, about 10 μF.
As this kind of current resonance capacitor, from the viewpoint of ripple current characteristics, high frequency characteristics, or changes over time,
Although a film capacitor is desirable, a film capacitor having such a capacitance value has an extremely large shape and size. If a chemical (electrolytic) capacitor is used, a capacitor with this level of capacitance has a small shape and size, which is advantageous in that sense. On the other hand, a chemical capacitor is inferior in ripple current characteristics and high-frequency characteristics, and is also poor in aging characteristics. ,I can not use it. Therefore, it is difficult to reduce the size of the entire power supply device due to the current resonance capacitor to be used. In particular, when a current resonance circuit is configured on the secondary side and a multi-output configuration is used, a plurality of large capacitors are required, which makes it more difficult to reduce the size of the power supply circuit.
The present invention has been made in view of the above points, and has as its object to provide a switching inverter type power supply circuit that has high efficiency and low noise and that can be downsized.

[0004]

A power supply circuit according to the present invention is connected to an input terminal via a primary winding of a transformer.
A switching transistor whose on / off is controlled,
An output circuit provided on the secondary side of the transformer to rectify and smooth the AC output to take out a DC output, and a primary side of the transformer and a leakage inductance having a larger number of turns among the secondary sides.
The switching transistor.
A current resonance circuit for resonating the main current when the
Voltage that resonates the voltage across the switching transistor when it is off.
A voltage resonance circuit, a timing control means having a time constant circuit for driving the switching transistor off when the main current thereof is zero, and driving the switching transistor on when the voltage at both ends is zero , a primary side of the transformer, Number of turns on the secondary side
Clamp that suppresses the peak of the resonance voltage
And a loop circuit .

[0005]

In the switching inverter circuit according to the present invention, current resonance and voltage resonance are used, the current waveform and the voltage waveform are both sinusoidal, and the switching element is driven off at zero current, and the voltage at both ends is zero. It is driven ON in the state. Thereby, the loss in the switching element becomes extremely small, and extremely high efficiency is obtained. In addition, switching noise is greatly reduced, and unnecessary radiation such as harmonics is extremely small. According to the invention, the side of the output transformer having a large number of turns, that is, a high voltage,
Since the current resonance circuit is configured on the small current side, the value of the transformer leakage inductance used for the current resonance circuit can be relatively large, and therefore, the capacitance value of the capacitor of the current resonance circuit can be relatively small, Therefore, a small-capacity and small-sized film capacitor can be used.
As a result, the ripple current characteristics, the high frequency characteristics, the aging characteristics, and the like are improved, the resonance Q can be increased, and the entire switching power supply can be reduced in size. In particular, when the primary side of the output transformer has a high voltage and a small current and the secondary side has a multi-output configuration, the number of large current resonance capacitors is greatly reduced, and the size is effectively reduced. Is done.

[0006]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a schematic configuration of a switching power supply circuit according to one embodiment of the present invention. As shown in the figure, the switching power supply circuit includes a switching unit 11 including a switching element S for converting an input DC voltage into an AC, a timing control unit 12 for controlling on / off of the switching element S, , A voltage resonance circuit 13 composed of a capacitor Cp and an inductance element Lp, and a current resonance circuit 1 composed of a capacitor Cs and an inductance element Ls formed in series with an output current.
4. An output circuit 15 coupled to the output of the switching means 11 via a transformer T. In the output transformer T, the number of turns n1 of the primary winding and the number of turns n2 of the secondary winding have a relationship of n1> n2. Under this relationship, the current resonance circuit 14 is arranged on the primary side of the transformer T. Is the feature.

FIG. 2 shows a power supply circuit configuration of a specific embodiment in which a DC-DC converter and a switching inverter of the present invention are combined. The continuous current mode DC-DC converter 21 has a basic circuit in which a capacitor C1 is connected in parallel to a series circuit of an NPN transistor Q1 as a switching element and a diode D1. A first inductance element L1 is inserted into the input terminal N1, and a second inductance element L2 is connected between a connection node between the diode D1 and the capacitor C1 and the reference terminal N2.
Is inserted. In this embodiment, the first and second inductance elements L1 and L2 are wound around one core and integrated. The first starting circuit 22 includes the converter 2
1 is a circuit for activating a transistor Q1 which is a switching element when an input DC voltage (for example, obtained by rectifying a commercial AC power supply by a rectifier circuit) is supplied to the rectifier circuit 1. The first starting circuit 22 has a collector connected to the input terminal N1.
NP whose emitter is connected to the base of transistor Q1 of converter 21 via diode D2
It comprises an N-transistor Q2, a Zener diode ZD1 for applying a bias thereto, and a resistor R1.

The output terminal N3 of the DC-DC converter 21
Is connected to a current-voltage resonance type switching inverter circuit 24 having the output voltage as an input. The switching inverter circuit 24 controls an NPN transistor Q6 as a switching element, a second starting circuit 26 including a resistor R8 connected between the base of the transistor Q6 and a terminal N3, and ON / OFF timing of the transistor Q6. And a timing control circuit 27 including an inductance element L3 and a capacitor C7. Also, the capacitor C4 provided between the terminal N3 and the reference terminal N2 and the leakage inductance of the output transformer T constitute a current resonance circuit (series resonance circuit) that enters in series with the output current of the inverter, and the collector of the transistor Q6 The self-inductance of the capacitor C5 connected between the emitters and the primary winding tap P1 of the transformer T constitutes a voltage resonance circuit (parallel resonance circuit) which is in parallel with the output voltage of the inverter. That is, the transformer T is equivalently represented by an ideal transformer, a leakage inductance and a self-inductance as shown in FIG. 3, and the leakage inductance and the self-inductance are used in a current resonance circuit and a voltage resonance circuit, respectively. Transistor Q6
The capacitor C6 and the diode D7 provided between the base and the emitter are for timing adjustment and for overvoltage protection, respectively. The primary winding of the output transformer T is provided with an additional tap P4, and a clamping diode D6 is connected between the additional tap P4 and the reference terminal N2.

The output terminal N3 of the DC-DC converter 21
Is provided with a pulse width modulation (PWM) circuit 23 for controlling the ON / OFF time of the NPN transistor Q1 as a switching element. PWM circuit 23
Are NPN transistors Q3, Q4, resistors R2, R3,
It is basically based on a drive circuit section composed of R4, diode D4 and the like. This drive circuit section rectifies the AC voltage taken out from the tap P3 portion of the transformer T by the diode D5, smoothes it by the capacitor C2 , and also takes out the DC voltage of the terminal N4 from the primary winding tap P3. Obtained by integrating the obtained voltage with an integrating circuit including a resistor R7 and a capacitor C3.
It is controlled by the triangular wave voltage at the terminal N5. The voltage at the terminal N4 is determined by the resistance R2 and the diode D3.
To the collector of the switching transistor Q1 of the converter 21 and to the base of the switching transistor Q1 via the resistor R2 and the diode D4. These diodes D3 and D4 are connected to the switching transistor Q1 of the converter 21. Is always provided in the unsaturated region. PWM
The circuit 23 also controls an NPN transistor Q5, a Zener diode ZD2, and resistors R5 and R6 to control the transistor Q4 of the drive circuit section according to the voltage of the terminal N3.
Is provided.

The output of the switching inverter circuit 24 is coupled to an output circuit 25 via an output transformer T. In this embodiment, in the output transformer T, the number of turns n1 of the primary winding and the number of turns n2 of the secondary winding satisfy n1> n2. That is, the primary side has a high voltage and a small current, and the secondary side has a low voltage and a large current. In this embodiment, the output circuit is
In addition to the rectifier diode D8, the smoothing capacitor COUT, a zener diode ZD3 for checking that the output voltage has become excessive and performing feedback control, and a photocoupler P
H. The light emitting part of the photocoupler PH is connected in series with a Zener diode ZD3 between two output terminals, and the light receiving transistor part is connected between the emitter of the transistor Q5 of the PWM circuit 23 and the reference terminal N2.
2 is short-circuited. Thus, when the output voltage on the secondary side becomes higher than a certain value, feedback is applied to the PWM circuit 23 to reduce the output voltage.

A current resonance capacitor C4 is directly exposed between the input terminals of the switching inverter circuit 24. When any circuit is connected to the input side, an isolating means is required to prevent other circuits from interfering with the current resonance capacitor C4. In this embodiment, the first inductance element L1 of the DC-DC converter 21 at the preceding stage also serves as an inductance element for preventing interference with the current resonance means of the switching inverter circuit 24. In other words, this inductance element has a sufficiently large impedance in the region of the resonance frequency determined by the current resonance capacitor C4, and prevents the current resonance capacitor C4 from being substantially coupled with other impedance components in the preceding stage. This inductance element is also used because the resonance circuit of the switching inverter circuit 24 of this embodiment is provided on the primary side of the transformer T and D is provided for the purpose of removing ripples.
This is made possible by the fact that the value of the inductance element L1 of the C-DC converter 21 is appropriate at the resonance frequency of the switching inverter 24 to present an impedance large enough to separate the resonance means from the resonance means. .

Next, the schematic operation of the power supply circuit of the embodiment shown in FIG. 2 will be described. When an input voltage is applied between the terminals N1 and N2, first, the first starting circuit 22 operates, and DC
-The DC converter 21 is started. That is, when an input voltage is applied, the resistance R1, the Zener diode ZD1
The transistor Q2 of the first starting circuit 22 is turned on with a bias determined by the following equation, and the voltage determined by the diode D2 is applied to the base of the transistor Q1 of the converter 21. As a result, the potential of the output terminal N3 of the converter 21 increases. When the voltage of the output terminal N3 reaches a certain value,
The base bias is applied to the transistor Q6 by the second starting circuit 26, that is, the resistor R8 in the switching inverter circuit, and the oscillation is started by the tap P3 of the transformer. The oscillation frequency is determined by the time constant of the inductance element L3 and the capacitor C7 that constitute the timing control circuit 27. At this time, the inductance of the transformer T and the values of the capacitors C4 and C5 are optimally set so that the voltage resonance and the current resonance are established. The detailed operation of the switching inverter circuit 24 utilizing this voltage-current resonance will be described later.

When the on / off operation of the transistor Q6 starts, the voltage obtained at the tap P3 of the transformer is rectified by the diode D5 and smoothed by the capacitor C2. That is, the voltage of the terminal N4 is supplied as the power supply voltage of the PWM circuit 23. The voltage at the tap P3 is converted into a triangular wave by an integrating circuit comprising a resistor R7 and a capacitor C3, and the voltage at the terminal N5 is applied to the base of the transistor Q4. The base of the transistor Q4 has a terminal N4
, A base bias is applied via a resistor R4, and the values of the resistors R4 and R7 are set so that the transistor Q4 is always on when the transistor Q5 is off. When the transistor Q4 turns on, the transistor Q3 turns off and the transistor Q1 of the converter 21 turns on. When the output terminal N3 of the converter 21 reaches the specified voltage, the transistor Q5 is turned on by the resistors R5 and R6 and the Zener diode ZD2, and this works in the direction of lowering the base potential of the transistor Q4. Therefore, PWM modulation for controlling the ON period of the transistor Q1 of the converter 21 is performed based on the potential relationship between the terminals N5 and N3, and the potential of the terminal N3 is stabilized.

After the PWM operation is started and the voltage of the output terminal N3 of the converter 21 is stabilized, the first starting circuit 2
2 automatically stops its function because it should not function. That is, in the case of the embodiment, the constant voltage VD of the Zener diode ZD1 that determines the base bias of the transistor Q2 is selected so as to be lower than the stabilizing voltage of the output terminal N3 of the converter 21. The first starting circuit 22 does not participate in the circuit operation.

The output of the switching inverter circuit 24 is coupled to an output circuit 25 by a transformer T,
The output is smoothed. When the output voltage on the secondary side exceeds a certain value, the zener diode ZD3 is turned on, the photocoupler PH operates, and feedback is applied to the PWM circuit 23. That is, the light receiving transistor of the photocoupler is in parallel with the Zener diode ZD2 inserted in the emitter of the transistor Q5 of the PWM circuit,
When this turns on, the transistor Q5 is forward biased. When transistor Q5 turns on, transistor Q4
Is turned off because the base potential is lowered, and the transistor Q
3 is turned on, and the transistor Q1 of the converter 21 is turned on.
Is driven off. P due to such feedback
By the WM control, the voltage of the output terminal N3 of the converter 21 is corrected downward. At this time, the constant voltage of the Zener diode ZD2 is selected to be lower than the voltage of the terminal N3 that is finally operating.

Next, the configuration and operation of each part of the embodiment of FIG. 2 will be described in detail. The converter 21 in the continuous current mode used in the embodiment of FIG. 2 can be represented as shown in FIG. That is, the first inductance element L1 is inserted in series with the input terminal of the converter circuit body, and the second inductance element L1 is inserted between the input terminal and the reference terminal.
Is inserted. FIGS. 4B to 4D specifically show the converter circuit body. FIG. 4B shows a switching circuit S1 connected in parallel to a series circuit of a capacitor C1 and a diode D1, which is shown in the embodiment of FIG. FIG. 4 (c) is a circuit in which a diode D1 is connected in parallel to a series circuit of a switching element S1 and a capacitor C1, and FIG.
1 and a capacitor C1 connected in parallel with each other. These are also equivalent in circuit function to FIG. 4B, and can replace the circuit of the embodiment of FIG. FIGS. 5A to 5D show the first inductance element L1 of FIGS. 4A to 4D brought to the output terminal side. These are also equivalent in circuit function to those in FIG. 4, and the circuit of the embodiment in FIG. 2 can be replaced with them.

The operation of the DC-DC converter in such a continuous current mode is described with reference to the circuit of the embodiment shown in FIG.
Next, the circuit configuration of FIG. 6B will be described with reference to FIG. FIG. 6 shows the voltage and current waveforms of each part of the circuit of FIG. 4 (b). It is assumed that the impedance of the capacitor C1 is sufficiently small at the switching frequency. Under these conditions, when the switching element S1 is turned on and off,
The output current IOUT is obtained alternately through the switching element S1 when the switching element S1 is on and through the diode D1 when the switching element S1 is off. At this time, the current flowing through the diode D1 is supplied from the second inductance element L2 and the capacitor C1, but the DC current is supplied only from the second inductance element L2. This becomes the current IL2 of the element L2.

When the switching element S1 is turned on, a current IL1 (equal to the input current IIN) flowing through the first inductance element L1 flows to the output side. At this time, the diode D1 is in the off state due to reverse bias. The current IL2 flowing through the inductance element L2 of the
And the output through the switching element S. Therefore, the output current IOUT is IOUT = IL1 + IL2. On the other hand, when the switching element S1 is turned off,
The diode D1 turns on and the second inductance element L
2 flows to the output side through the diode D2. At this time, the current IL1 flowing through the first inductance element L1 also flows to the output side through the capacitor C1 and the diode D1 because the switching element S1 is off. Therefore, also at this time, the output current becomes IOUT IOUT = IL1 + IL2.

Thus, in this DC-DC converter, the output current IOUT is in the continuous mode. The current IC1 of the capacitor C1 alternates, and the voltage between both ends is substantially equal to the input voltage VIN. In this circuit, the voltage across the switching element S1 and the diode D1 is the input voltage VIN at the maximum, and the current is equal to the output current IOUT.
This is the same as the conventional BUCK type converter. Since the duty can be used up to 100%, it has all the advantages of the BUCK type.

On the other hand, in the circuit configuration of FIG. 4B, the first inductance element L1 and the second inductance element L2
Are always equal. This is shown in FIG.
FIG. 7 shows a case where 10 V, 1 A is converted into 5 V, 2 A as an example.
1 and the voltage V1, V across the second inductance element L2.
2 vibrates as shown. Therefore, these inductance elements L1 and L2 can be wound around one core and integrated, and FIG. 2 shows a configuration in that case. 4B, the first and second inductance elements L1 and L2 can be shared by the core for the same reason when the configuration of FIG. 5B is used.

Next, the operation of the switching inverter circuit 24 of FIG. 2 will be described in detail with reference to FIG. As described above, when the converter 21 is activated and a voltage is obtained at the terminal N3, the switching inverter circuit 24 starts operating by the second activation circuit 26 and converts the obtained output voltage into an alternating current. When transistor Q6 turns on,
An electric current flows through this, and at this time, a series resonance circuit is operated by the capacitor C4 and the leakage inductance of the transformer T, resulting in a sine wave oscillating current as shown in FIG. When the main current of the transistor Q6 is almost zero, the transistor Q6 is turned off. When the transistor Q6 is turned off, a parallel resonance circuit operates by the self-inductance of the capacitor C5 and the tap P1 of the transformer T, and the terminal voltage of the transistor Q6 oscillates in a sine wave shape as shown in FIG. The transistor Q6 is turned off at a point where the voltage across the transistor Q6 is substantially zero.

In this way, the series resonance circuit (current resonance circuit) and the parallel resonance circuit (voltage resonance circuit) are driven so that the current of the transistor Q6 is turned off at a point where the voltage is substantially zero and turned on at a point where the voltage between both ends is substantially zero. Circuit) and a timing control circuit 2
7 are set. As a result, in the switching inverter circuit 24, both the current waveform and the voltage waveform become sine waves, and an AC output without harmonics is obtained. The base current of the transistor Q6 is similar to the resonance current waveform of the collector due to LC resonance, thereby realizing high efficiency. The capacitor C6 provided between the base and the emitter of the transistor Q6 makes the base voltage negative when the transistor Q6 is turned off to secure a turn-off current, thereby enabling high-speed turn-off.

As described above, according to this embodiment, the DC-DC converter 21 in the preceding stage is in the continuous current mode and has little noise. Since the switching inverter circuit 24 at the subsequent stage is also turned on and off at zero voltage and zero current by current-voltage resonance, noise is reduced. Therefore, an extremely low-noise switching power supply can be obtained as a whole. Further, in the switching inverter circuit 24, the transistor Q6 as a switching element
Are turned on and off at zero voltage and zero current, resulting in a highly efficient power supply with little loss and little heat generation.

In this embodiment, the switching inverter circuit 24 includes a voltage resonance circuit,
It constitutes a current resonance circuit. As described above, the relationship between the number of turns n1 of the primary winding of the transformer T and the number of turns n2 of the secondary winding is n1> n2, and the leakage inductance of the transformer viewed from the primary side is viewed from the secondary side. (N1 / n2) ² times larger than that. Therefore, a small-capacity and small-sized film capacitor can be used as the current resonance capacitor C4. As a result, various characteristics are improved, the Q of resonance can be increased, and the entire power supply does not increase in size.

Further, in this embodiment, the first inductance element L1 provided in the DC-DC converter at the preceding stage is used.
Are used as an interference preventing inductance element for preventing a current resonance circuit of a switching inverter circuit from interfering with another. This is possible as a result of the switching inverter circuit 24 of this embodiment constituting a current resonance circuit on the primary side. This is very advantageous when configuring a multiple output power supply as shown in FIG. If a multi-output power supply circuit as shown in FIG. 9 is configured by bringing a current resonance circuit on the secondary side, each output circuit on the secondary side needs an inductance for preventing interference and a smoothing inductance, which reduces the size of the switching power supply. Because it inhibits

In the embodiment shown in FIG. 2, the switching inverter circuit 24 is of a so-called single type using one transistor, and both the current resonance circuit and the voltage resonance circuit have a transformer T.
Formed on the primary side. However, the present invention is not limited to this, and can be applied to other types of switching inverter circuits. That is, the single-switching inverter circuit may be configured as shown in FIGS. 10 to 13. However, by providing a current resonance circuit on the primary side or the secondary side of the transformer T having the larger number of turns,
The same effects as in the previous embodiment can be obtained. FIG. 10 shows that, when the number of turns of the transformer T is n1 <n2, the voltage resonance circuit is formed on the primary side by the capacitor C5 and the self-inductance of the primary winding of the transformer, and the leakage inductance of the transformer is formed on the secondary side. A current resonance circuit is constituted by the capacitor C4. In this case, an inductance element LF for preventing interference and smoothing is provided on the secondary side. FIG. 11 shows a case where a clamping diode D6 is provided on the primary side based on the configuration of FIG. 10 when the number of turns of the transformer T is n1 <n2. FIG. 12 shows an example in which the number of turns of the transformer T is n1> n2, and both the current resonance circuit and the voltage resonance circuit are configured on the primary side of the transformer T. In this case, the interference preventing inductance element LF is provided on the primary side. FIG. 13 shows a configuration in which a clamping diode D6 is provided on the primary side based on the configuration of FIG. 12 when the number of turns of the transformer T is also in a relationship of n1> n2. The configuration of FIG. 13 is that of the embodiment of FIG. However, in the embodiment of FIG. 2, as described above, the first-stage converter uses the first inductance element as an interference-prevention inductance element because the first-stage converter includes an inductance component in the output impedance. Even when the configuration of FIG. 12 is used, the inductance element can be shared for the same reason.

A clamping diode D6 is provided on the primary side of the transformer of the switching inverter circuit 24 in the embodiment of FIG. 2 in order to suppress a peak voltage determined by voltage resonance. This has the following meaning. First, if there is no device that suppresses the peak voltage, it causes element destruction due to variations in parts and design errors. By providing the clamping diode, such destruction is prevented, and the reliability is improved.
Second, the on-time of the transistor Q6 can be made relatively long. That is, as shown in comparison with FIGS. 14A and 14B, in the case of FIG. 14A in which the peak voltage Vp is not suppressed, it is difficult to increase the ratio of the ON period for the large peak voltage. . On the other hand, when the peak voltage Vp is determined by the clamp as shown in FIG.
The ON period of the transistor Q6 can be lengthened.
This is because the areas of the positive half wave and the negative half wave of the resonance voltage waveform are equal, and in FIG. 14 (a), S1 = S2.
In (b), S1 '= S2'. This is because the area S1 when the peak value is Vp and the area S1 'when the same peak value Vp is clamped is S1'> S1, and therefore S2 '> S2. Since the ON period of the transistor Q6 can be lengthened,
The peak of the resonance current can be reduced.

The present invention can also be applied to a so-called dual-switching inverter circuit using two switching elements. The switching inverter circuit configuration is shown in FIGS. FIG. 15 shows a half-bridge type switching inverter circuit using two switching elements S1 and S2 and arranging both a current resonance capacitor C4 and a voltage resonance capacitor C5 on the primary side of the transformer T. FIG. 16 shows a half-bridge type switching inverter circuit in which the current resonance capacitor C4 is arranged on the secondary side of the transformer T based on the configuration of FIG. In each case, the winding of the transformer T is n
1> n2. FIG. 17 shows a push-pull type in which both a voltage resonance circuit and a current resonance circuit are formed on the primary side of a transformer T. FIG. 18 shows a current resonance circuit formed on the secondary side of the transformer T, also of the push-pull type. These are also transformer T windings
n1> n2. 15 or 17 in which the current resonance circuit is formed on the primary side is shown in FIG.
When combined with the DC-DC converter of the continuous current mode as in the embodiment, the inductance element LF can be used also as the preceding-stage inductance element as described in the previous embodiment.

Next, several modifications of the timing control circuit 27 and the second starting circuit 26 in the switching inverter circuit 24 in the embodiment of FIG. 2 will be described. FIG. 19 shows an example in which an oscillation circuit is configured using a transistor Q10, resistors R10 and R11, capacitors C10 and C11, and a diode D10 as a timing control circuit for driving the transistor Q6 instead of the LC circuit of the embodiment of FIG. is there. FIG. 20 shows a modification of the second starting circuit 26 which is constituted only by the resistor R8 in the embodiment of FIG. Here, the second
, A PNP transistor Q11 is added to the starting circuit 26, and diodes D11 and D
A bias circuit comprising 12, D13, a resistor R12 and a capacitor C12 is provided. When the DC-DC converter in the preceding stage is activated and the potential of the terminal N3 rises to a certain value,
The NP transistor Q11 is turned on, and a large base current is supplied to the transistor Q6, which is a switching element of the switching inverter circuit, to start the transistor. That is, the startup is enhanced. Once in steady state, the capacitor C12
PNP transistor Q11 is turned off by the voltage of
The second activation circuit 26 stops its function. This makes it possible to suppress the heat generated by the starting resistor when the starting is enhanced by increasing the base current. FIG. 21 is a modification of the second activation circuit 26 in the same manner. Here, two PNP transistors Q13, Q14 and resistors R13, R14, R
15 and a Zener diode ZD10 constitute a bias circuit. In this case, when the potential of the terminal N3 rises to some extent, the PNP transistor Q13 is turned on to supply a base current to the transistor Q6. When the potential of the terminal N3 further rises, the PNP transistor Q14 turns on, the forward bias of the transistor Q13 decreases, and the base current supplied to the transistor Q6 is suppressed. That is, the base current for transistor Q6 is controlled in accordance with the input voltage, and the increase in heat generation due to the enhanced start-up can be suppressed as in FIG.

[0030]

As described above in detail, according to the present invention, the current-voltage resonance type realizes extremely low-noise and extremely high-efficiency characteristics, and at the same time, the current is supplied to the side having a large number of turns of the output transformer. By configuring the resonance circuit, a switching power supply that is reduced in size can be obtained.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a schematic configuration of a power supply circuit according to an embodiment.

FIG. 2 is a diagram showing a specific configuration of a power supply circuit according to the embodiment.

FIG. 3 is a diagram showing an equivalent circuit of a transformer.

FIG. 4 is a diagram showing a configuration of a current continuous mode DC-DC converter.

FIG. 5 is a diagram showing a configuration of a continuous current mode DC-DC converter.

FIG. 6 is a diagram showing operation waveforms of a current continuous mode DC-DC converter.

FIG. 7 is a diagram for explaining the sharing of the core of the inductance element.

FIG. 8 is a diagram showing operation waveforms of the switching inverter circuit.

FIG. 9 is a diagram showing a multi-output power supply configuration.

FIG. 10 is a diagram showing a single-switching inverter of primary-side voltage resonance and secondary-side current resonance.

FIG. 11 is a diagram showing a single-switching inverter having primary voltage resonance and secondary current resonance.

FIG. 12 is a diagram showing a single-switching inverter of primary-side voltage resonance and primary-side current resonance.

FIG. 13 is a diagram showing a single-switching inverter of primary-side voltage resonance and primary-side current resonance.

FIG. 14 is a diagram for explaining an effect of a voltage clamp.

FIG. 15 is a diagram showing a dual-switch half-bridge type switching inverter of primary-side voltage resonance and primary-side current resonance.

FIG. 16 is a diagram showing a dual-switch half-bridge type switching inverter of primary-side voltage resonance and secondary-side current resonance.

FIG. 17 is a diagram showing a dual-push push-pull type switching inverter of primary-side voltage resonance and primary-side current resonance.

FIG. 18 is a diagram showing a dual-push push-pull type switching inverter of primary-side voltage resonance and secondary-side current resonance.

FIG. 19 is a diagram showing a modification of the timing control circuit.

FIG. 20 is a diagram showing a modification of the second startup circuit.

FIG. 21 is a diagram showing a modification of the second startup circuit.

[Explanation of symbols]

11 switching means, 12 timing control circuit,
13: voltage resonance circuit, 14: current resonance circuit, 15: output circuit, T: transformer, 21: current continuous mode DC-DC
Converter, 22 first starting circuit, 23 PWM circuit, 24 current-voltage resonant switching inverter circuit, 25 output circuit, 26 second starting circuit, 27 timing control circuit, L1 first inductance element,
L2: second inductance element, Q1: npn transistor (switching element), Q6: npn transistor (switching element), C4: current resonance capacitor, C5: voltage resonance capacitor.

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02M 3/28 H02M 7/06 H02M 7/537

Claims (1)

(57) [Claims] 1. An input terminal via a primary winding of a transformer.
Switching transistor connected and controlled on / off
An output circuit provided on the secondary side of the transformer for rectifying and smoothing the AC output and extracting a DC output; and a leakage circuit having the larger number of turns of the primary and secondary sides of the transformer.
The switch is constructed using an inductance.
Resonance that resonates the main current when the transistor is on
Circuit and the voltage across the switching transistor when the switching transistor is off.
A voltage resonance circuit for driving the switching transistor , a timing control means having a time constant circuit for driving the switching transistor off when the main current is zero, and driving the switching transistor on when the voltage at both ends is zero, and a primary circuit of the transformer. Side or secondary side with the larger number of turns
A power supply circuit, comprising: a clamp circuit that suppresses a peak of an applied resonance voltage .