JP3179164B2 - High frequency power amplifier - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power amplifier for a high-frequency signal, and more particularly to a transmission power amplifier having high power efficiency used for a mobile radio communication device or the like. .

[0002]

2. Description of the Related Art A high frequency power amplifier for outputting a high level of high frequency power is incorporated in a transmitter of a radio apparatus. Most of the DC power consumed by the wireless device is the power required by the high-frequency power amplifier, and it is indispensable to increase the conversion efficiency to the high-frequency power in order to reduce the power consumption of the wireless device. This is particularly important for a battery-operated mobile phone.

Normally, a class A or class AB high-frequency power amplifier achieves a UHF band conversion efficiency of 30 to 50%. However, if the impedance for a high-frequency signal on the load side as viewed from the active element is properly selected, it will be higher. Conversion efficiency can be obtained.

The relationship between the conversion efficiency and the impedance on the load side will be described with reference to FIG. FIG. 10 shows a basic configuration of a power amplifier. A high-frequency signal from an input terminal 121 is applied to a gate G of a field-effect transistor 101 whose source S is grounded, and an amplified high-frequency signal is obtained from its drain D. Thus, high-frequency power is supplied to the load impedance 150 via a fundamental wave matching circuit 134 for impedance matching of the fundamental wave. Here, L6 and L7 are inductances, which connect the DC power supplies V _GS and V _DS to the gate G and the drain D, respectively.
To supply. Here, I _DC is a DC current to the drain D, V _D (t) is a high-frequency drain voltage, I _D (t) is a high-frequency current to the drain D, and I _M (t) is a high-frequency current to the fundamental matching circuit 134. , I _OUT (t) are the high frequency currents to the load impedance 150.

FIG. 11A shows a current waveform 171 of the high-frequency current I _D (t) to the drain D in FIG.
(B) shows a voltage waveform 172 of the high-frequency drain voltage V _D (t) in FIG. High frequency current I _D (t)
I _D (t) = {I _Ck exp (jkωt) = I ₀ + I ₁ + I ₂ + I ₃ + I ₄ +... (1) where, I ₁ = Re {2I _C1 exp (jωt)} I ₂ = Re {2I _C2 exp (j2ωt)} I ₃ = Re {2I _C3 exp (j3ωt)} I ₄ = Re {2I _C4 exp (j4ωt)}. Here, Σ indicates that k is changed from −∞ to + ∞ and added, I _C indicates a complex current including a real part and an imaginary part, and Re indicates the real part.

Similarly, the high-frequency drain voltage V _D (t) is expressed as follows: V _D (t) = ΣV _Ck exp (jkωt) = V ₀ + V ₁ + V ₂ + V ₃ + V ₄ + (2) where V ₁ = Re {2V _C1 exp (jωt)} V ₂ = Re {2V _C2 exp (j2ωt)} V ₃ = Re {2V _C3 exp (j3ωt)} V ₄ = Re {2V _C4 exp (j4ωt)} . Here Σ represents the adding by changing k from -∞ to + ∞, V _C is the real part and shows the complex current containing the imaginary part, Re denotes the real part.
_Assuming that the impedance seen from the point 181 in FIG. 10 on the fundamental wave matching circuit 134 side is Z _M (ω) and ω = 2πf, each current I _C and voltage V in equations (1) and (2) are obtained.
_{Between C} , V _C1 = −Z _M (ω) I _C1 V _C2 = −Z _M (2ω) I _C2 V _C3 = −Z _M (3ω) I _C3 V _C4 = −Z _M (4ω) I _C4 ... (3), the high-frequency current _ID (t) and the high-frequency drain voltage V of the equations (1) and (2) are obtained from the impedance Z _M (kω) for each harmonic viewed from the point 181 to the right. _It can be seen that the waveform of _D (t) changes.

The power loss P of the field effect transistor 101
_F is P _F = T ^-1 ∫ V _D (t) _ID (t) dt (4) where T = 2π / ω, ω = 2πf, and ∫ is to be integrated from 0 to T Is represented.

The output P _OUT (1) from the drain D of the field effect transistor 101 at the frequency f ₀ of the fundamental wave
Is _{Assuming that} ω ₀ = 2πf ₀ , P _OUT (1) = 2Re {Z _M (ω ₀ )} | I _C1 | ² (5) From the drain D of the field-effect transistor 101 at the k-th harmonic frequency kf ₀ Output power P _OUT (k)
Is P _OUT (k) = 2Re {Z _M (kω ₀ )} | I _Ck | ² (6)

The power consumption P _DC P _DC = V _DS I _DC = P _F + ΣP _OUT (k) supplied from the DC power supply V _{DS in} the circuit of FIG. Here, V _DS is the voltage of the DC power supply V _DS , and Σ represents the sum when k = 1 to ∞. The drain efficiency η for obtaining output power at the frequency f ₀ of the fundamental wave at the drain D of the field effect transistor 101 is η = P _OUT (1) / P _DC (8)

In order to increase the drain efficiency η shown in the equation (8) to increase the efficiency, it is necessary to reduce the power consumption P _DC ,
It is necessary to increase the output power P _OUT (1) at the frequency f ₀ of the fundamental wave.

[0011] Therefore in order to allowed to reduce the power consumption P _DC, it is necessary to reduce the power loss P _F of the field effect transistor 101 of the formula (7), to, Formula (4)
The product of the high-frequency drain voltage V _D (t) causing the power loss P _F and the high-frequency current I _D (t) of the drain D, ie, I _D (t) and V _D (t) in FIG. Are temporally overlapped (a falling portion of the current waveform 171a and a rising portion of the voltage waveform 172a, a falling portion of the 172a and a rising portion of the 171b, a falling portion of the 171b and a rising portion of the 172b). Thus, it is necessary to set the impedance as viewed from the point 181 in FIG. 10 to the right (load side). In FIG. 11, if there is no temporal overlap between the current waveforms 171a and 171b representing I _D (t) and the voltage waveforms 172a and 172b representing V _D (t), the power loss P _F in equation (4) is removed. can do.
Further, in order to set the output power P _OUT (k) (where k is 2 or more) of the k-th harmonic except the fundamental wave (k = 1) of the equation (7) to 0, the k-th order of the equation (6) Resistance component Re in harmonics
{Z _M (kω ₀ )} may be set to 0, or the complex current I _Ck which is a high-frequency current at the k-th harmonic may be set to 0.

As a method of realizing such a condition, a class B amplifier is used to make the high-frequency current I _D (t) of the drain D a half-wave rectified waveform having a flow angle of 180 ° and to make the even-order impedance zero. Have been. High frequency current I _D of the drain D during half-wave rectification (t) _{is, I D (t) = 1} / π + (1/2) cos ω 0 t + (2 / 3π) cos (2ω 0 t) - ( 2 / 7π) cos (4ω ₀ t) + (2 / 35π) cos (6ω ₀ t)-(9) Looking at equation (9), this high-frequency current I _D (t)
Is a fundamental wave represented by ω ₀ and, for example, 2ω ₀ , 4ω ₀ ,
.., Etc., the impedance at the even-order harmonic becomes 0, and the voltage at the even-order harmonic becomes 0. Therefore, the high-frequency drain voltage V
It is known that _D (t) has only the fundamental wave component, and the drain efficiency η shown in Expression (8) increases.

In practice, the harmonic power at higher harmonics of k = 3 or higher is relatively small as compared with the case of k = 2. It is intended to obtain a high drain efficiency η by setting the impedance Re {Z _M (2ω ₀ )} of 6) to 0. In reality, the frequency 2f which is twice the fundamental wave as seen from the point 181 of the drain D of the field effect transistor 101 in FIG.
_This is intended to be realized by bringing the impedance to ₀ (second harmonic) close to 0.

FIG. 12A shows a circuit (hereinafter referred to as a first conventional example) which attempts to reduce the impedance to a frequency 2f ₀ (second harmonic) twice the fundamental wave.
(B) shows the standing wave of the second harmonic in the circuit shown in (a). In the circuit of (a), the descriptions of the inductances L6 and L7 and the DC power supplies V _DS and V _GS are omitted.

In FIG. 12A, a high-frequency signal applied to an input terminal 121 is applied to a gate G of a field-effect transistor 101 whose source S is grounded and amplified, and high-frequency power is obtained at a drain D. The high-frequency power passes through the transmission line 114, passes through the point 182 where the open stub 118 is connected, and passes through the fundamental wave matching circuit 134.
, And is applied to the load impedance 150.

Transmission line 114 and open stub 11
The standing wave of the second harmonic at 8 is shown in FIG. 12 (b), and the electrical length of the transmission line 114 connected between the points 191 and 182 is 2 times the wavelength λ of the second harmonic. The electrical length of the open stub 118 connected to the point 182 to the point 192 at the open end is で of λ. At point 192, the amplitude of the second harmonic is at a maximum, and at points 182 and 191 the amplitude is zero, ie, virtual ground. In this circuit, a long transmission line 114 having a half of λ is used to virtually ground the point 191.
Since phase rotation will occur not only at high frequencies but also at low frequencies, the impedance at point 191 will vary greatly with the frequency of the high frequency signal. For this reason, the virtual ground condition for the second harmonic (frequency 2f ₀ ) is realized only in a narrow frequency range. Therefore, for example, when the frequency of the high-frequency signal changes due to the change of the radio channel and the frequency of the second harmonic deviates from 2f ₀ , a loss occurs and the power efficiency deteriorates rapidly.

FIG. 13A shows a circuit for solving such a problem (hereinafter referred to as a second conventional example), and FIG. 13B shows a Smith chart of a phase diagram of the high-frequency signal. In the circuit of FIG.
6, L7 and DC power supplies V _DS and V _GS are omitted. The field-effect transistor 101 includes an amplifying element section 102 and a parasitic line 103 such as a bonding wire, a package lead, and a lead for connection. An open stub 118 having an electrical length of λ / 4 (λ = 1 / (2f ₀ )) of the second harmonic of the frequency f ₀ of the fundamental wave is connected to the drain D of the field effect transistor 101. Is trying to make the impedance for 2f _{0 zero} .

However, the electric length Δ of the parasitic line 103
Due to the presence of θ, the electrical length of the round trip 2Δθ is added at the point 183, and a phase shift of 20 ° to 40 ° occurs at 2f ₀ = 3 GHz (f ₀ = 1.5 GHz). As shown in point A of FIG. 13 (b), 2Δθ (2
0 ° to 40 °) to 140 ° to 160 °.
The impedance for the second harmonic will not be zero. Therefore, in order to make the impedance of the second harmonic at point 183 zero, the impedance seen from the right side of the load from the drain terminal D of the field effect transistor 101 in consideration of the effect of the parasitic line 103 is as shown in FIG.
The point B rotated by 2Δθ (20 ° to 40 °) from the point C in (b) is the optimum impedance condition. This is supported by the experimental results in the following literature showing the optimum value of the phase angle.

Reference 1. Ikeda et al., IEICE Spring Conference, C-64, 1990

The circuit of the first conventional example shown in FIG.
In addition to the above problem, there is another problem in impedance matching with the fundamental wave as follows.

FIG. 14 shows an equivalent circuit of the second conventional circuit shown in FIG. 13A for a fundamental wave of frequency f ₀ . Here, the high-frequency signal source 107 is an equivalent circuit of the field-effect transistor 101 at the fundamental wave. It includes a high-frequency constant current source 108 and a signal source impedance 109, and the open stub 118 in FIG. 13A is equivalent to a capacitance C6 in a fundamental wave.

The load impedance Z _M (ω ₀₎ in the fundamental wave of the high-frequency signal source 107 and the drain efficiency η is a class B power amplifier to the maximum, the voltage V _DS of the DC power supply V _DS
(See FIG. 10) and the output power P _{0UT at} point 184, there is a relationship Z _M (ω ₀ ) = V _DS ² / (2P _OUT ). For example, as operating conditions of a wireless amplifier,
When V _DS = 5 volts and P _0UT = 1 watt, Z _M
(Ω ₀ ) = 12.5 ohms. The value of the load impedance 150 is 50 ohms, and if there is no capacitance C6, the VSWR (voltage standing wave ratio) of 50 ohms to 12.5 ohms is 4.

However, because of the presence of the open stub 118, the capacitance C6 exists in the fundamental wave. Therefore, when the high-frequency signal source side is viewed from the point 185 in FIG. It is connected and its value is about 9.8 ohms. Therefore, the fundamental wave matching circuit 134 must match the load impedance 150 of 50 ohms with the impedance looking at the high-frequency signal source 107 from the point 185 where the VSWR is about 5.1. As the value of the VSWR increases, the bandwidth of the fundamental wave matching circuit 134 decreases. For this reason, the fundamental wave matching circuit 134 has many components, the power loss in the matching circuit 134 increases, and the power efficiency decreases.

FIG. 15 shows a third conventional example (JP-A-62-1).
68404). Field effect transistor 1
In the middle of the transmission line 115 between the drain D of No. 01 and the fundamental matching circuit 134, a capacitance C7 is grounded. The admittance of the capacitance C7 is impedance-converted by the partial transmission line 116 between the capacitance C7 and the drain D in the transmission line 115, and a short circuit is realized at the drain D with respect to the second harmonic.

However, if an attempt is made to match the fundamental wave, the value of the capacitance C7 cannot be increased, and the impedance at the second harmonic of the capacitance C7 is not sufficiently small. As a result, the second harmonic cannot be short-circuited. Therefore, from the point 186 which is the connection point between the transmission line 115 and the capacitance C7, the fundamental wave matching circuit 1
Since the second harmonic is transmitted to the side 34, unnecessary waves (second harmonics) other than the fundamental wave are output.

[0026]

A first conventional example (FIG. 12)
In (a)), the existence of the parasitic line is not a problem because the parasitic line can be regarded as a part of the transmission line 114, but the bandwidth is narrow because the long transmission line 114 is used. There was a problem.

In the second conventional example (FIG. 13 (a)), although the problem of narrowing the bandwidth of the first conventional example has been solved, the second harmonic is shown in FIG. 13 (a). Point 1
Since the virtual ground is established at 83, the impedance for the second harmonic at the point 182 cannot be set to 0 due to the existence of the parasitic line 103, and the second harmonic also flows to the basic matching circuit 134 side, This results in power loss.
Also in the fundamental wave matching circuit 134, it is necessary to pass only the fundamental wave so as to cut off the second harmonic, so that the circuit configuration is complicated, and the power loss is further increased. For the fundamental wave, the open stub 118 is equivalent to the capacitance C6 in FIG. 14, and the impedance when the high frequency signal source 107 side from the point 185 is smaller than when the capacitance C6 does not exist. The value becomes a large value, and the fundamental wave matching circuit 134 for achieving this impedance matching has a narrow band, the impedance matching degree becomes worse than when the capacitance C6 does not exist, and the power efficiency is not sufficiently improved. There were issues to be addressed.

In the third conventional example (FIG. 15), when the transmission line 115 and the capacitance C7 are selected so as to match the fundamental wave, the impedance of the capacitance C7 is sufficiently small with respect to the second harmonic. Since the value does not reach the value, the second harmonic component cannot be short-circuited at the drain D, and the second harmonic is transmitted in the direction of the fundamental wave matching circuit 134, and the second harmonic power loss occurs. Further, there is a problem to be solved in that even unnecessary second harmonics are output.

[0029]

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and it is an object of the present invention to provide a field effect transistor for amplifying electric power, a drain of the field effect transistor, a load impedance, and a matching in a fundamental wave. The electric length including the parasitic line of the field-effect transistor for connecting with the fundamental wave matching circuit for taking
A first coupling line having a one-half wavelength, and a second coupling parallel to the first coupling line and having mutual coupling, a terminal near the field-effect transistor being grounded and the other end being open. It was configured to include a track.

[0030]

With respect to the second harmonic, the electric length of the parasitic line and the coupling line becomes a quarter wavelength, and the second coupling line is coupled with the first coupling line. The phase angle of the reflection coefficient was set to a required value in anticipation of the load side. As a result, high power efficiency can be obtained. 1/4 of the second harmonic between the first coupled line and the drain terminal
By inserting a transmission line slightly shorter than the wavelength, grounding one end of the transmission line and connecting a capacitance to the other end,
The drain terminal can be short-circuited.

[0031]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS One embodiment of the present invention is shown in FIG. Here, components corresponding to those in FIGS. 10 and 13 are denoted by the same reference numerals. Also,
The illustration of the inductances L6 and L7 and the DC power supplies V _GS and V _{DS in} FIG. 10 is omitted.

The high frequency signal applied to the input terminal 121 is applied to the field effect transistor 101 whose source S is grounded.
Is applied to the gate G. The drain D of the field-effect transistor 101 from which high-frequency power is obtained is connected to the coupling line 20a at a point 193, and the other end of the coupling line 20a is connected to a point 194.
Is connected to a fundamental wave matching circuit 134 for impedance matching with the load impedance 150. There is a coupling line 20b in parallel with the coupling line 20a,
One end of the coupling line 20b on the side of the point 193 is grounded.
The other end of 4 is open. The two coupled lines 20a and 20b are integrated to form a coupled line element. The electrical length of both coupling lines 20a and 20b is one-fourth of the wavelength of the second harmonic having twice the frequency of the fundamental wave, and the coupling line 20b is electromagnetically coupled to 20a.
Short the point 193 for the second harmonic (virtual ground)
However, the circuit size can be reduced without requiring any lines other than the coupling lines 20a and 20b.

FIG. 1B shows an equivalent circuit of a fundamental wave of the circuit shown in FIG.
Is an electrical length, which is a quarter of the wavelength λ of the second harmonic, and θ3 = θ4. The embodiment of FIG.
The electrical length of a is as short as ４ of λ. The first shown in FIG.
In the case of the conventional example of FIG.
Since the electric length of one half of the circuit was used, the circuit of FIG.
As compared with the first conventional example (FIG. 12), the change in the impedance of the line with respect to the change in frequency is small, and the bandwidth is slightly less than twice as wide.

FIG. 2 shows another embodiment of the present invention. Here, the difference from the circuit of FIG. 1 is that the field effect transistor 101 is represented by an electrical length Δθ of an ideal amplifying element portion 102 whose source S is grounded and a bonding wire or a lead wire for soldering. Parasitic line 10
3 in that the terminal on the point 194 side of the coupling line 20b is grounded by the capacitor C4. The sum of the electric lengths of the coupling lines 20a and 20b and the electric length Δθ of the parasitic line 103 is set to be equivalent to a quarter of the wavelength λ of the second harmonic. The point 183 is virtually grounded by the effect of the capacitor C4 and the coupling line 20b.

FIG. 3 is an equivalent circuit of the circuit shown in FIG. 2 and corresponds to FIG. The difference from FIG. 1B is that the electrical length θ6 and the electrical length θ5 obtained by the effect thereof are shown.
(= Θ6) is a quarter of the wavelength λ of the second harmonic, and the capacitance C5 is equivalently added as an effect of the capacitance C4. The capacitance C5 is determined by the degree of coupling 0 <k ≦ 1 between the two coupling lines 20a and 20b, and has a relationship of C5 = kC4. In the circuits of FIGS. 2 and 3, a short circuit is completely realized at the point 183 for the second harmonic, and a wide band is obtained for the fundamental wave due to the short coupling lines 20a and 20b. There are features.

FIG. 4 shows still another embodiment of the present invention. Here, the difference from the one shown in FIG. 2 will be described. A transmission line 141 is added between a point 196 of the drain D and a point 195 at one end of the coupling line 20a, and the coupling line 20b is added.
Are grounded at impedances 158 and 159, which are reactive elements composed of short, open, capacitance and inductance.

Here, the electrical length of the transmission line 141 and the parasitic line 1
03 and the electrical length Δθ are set to be equivalent to a quarter of the wavelength λ of the second harmonic.
8 is open, that is, nothing is connected, and the impedance 159 is short-circuited, that is, short-circuited to ground. Then, for the second harmonic, the point 195 is open and the point 183 is short-circuited because the two coupled lines 20a and 20b are tightly coupled. The capacitance is used as the impedance 158, and this capacitance and the coupling line 20b are used.
Even if the resonance frequency of the resonance circuit formed by the above-described inductance is set to be the frequency of the second harmonic (2f ₀ ), the same effect as when the impedance 158 is opened can be obtained. The impedances 158 and 159 of the reactive elements are adjusted so that the circuit formed by the inductance of the coupling line 20b and the impedances 158 and 159 of the reactive elements resonates at the frequency of the second harmonic (2f ₀ ). You can choose.

FIG. 5 shows the structure of the first embodiment of the coupling line elements constituting the coupling lines 20a and 20b which play an important role in FIGS. Where 1
0 is an outer conductor, 20a and 20b are strip-shaped coupled lines, 70
Is between the coupled lines 20a and 20b and between the coupled lines 20a and 20b.
A dielectric material such as, for example, ethylene tetrafluoride or polyethylene, which has a small high-frequency loss and is filled between the a and 20b and the outer conductor 10.

FIG. 6 shows a cross-sectional view of the coupled line device of FIG. 5 and an electric field there. (A) shows electric fields 31 to 34 in even mode (even mode),
(B) shows an electric field 3 in an odd mode (odd mode).
5-37 are shown. In (a), the coupled line 2
Since 0a and 20b are at the same potential, there is no electric field between them, and electric fields 31 to 34 are generated based on the potential difference with the peripheral external conductor 10. On the other hand, since the mode is an odd mode in FIG.
b, a strong electric field 37 is generated, and based on this,
A large coupling capacitance is formed between the two coupling lines 20a and 20b. Both coupled lines 20a, 20b and peripheral outer conductor 1
Weak electric fields 35 and 36 are generated between 0 and 0. FIG.
In the cases (a) and (b), since the coupling lines 20a and 20b are entirely covered by the outer conductor 10, there is no leakage electric field and no power loss due to the leakage electric field.

In the structure shown in FIG. 5, for example, if the fundamental frequency is f ₀ , and the wavelength of the second harmonic frequency 2f ₀ is λ, λ
/ 4 (λ = (2f ₀ ) ⁻¹ ), polyethylene is used as the dielectric 70, the cross section is about 2 × 2 mm when f ₀ = 5 GHz, and the length is 5 mm
It is about. Since this coupled line element can be handled as a chip component, it can be easily mounted on a printed circuit board or a hybrid integrated circuit board, and the mounting process is simplified. Further, since a material different from the material of the substrate such as a printed circuit board can be selected as the dielectric 70, the use of a low-loss, high-dielectric-constant material (for example, ceramics) for a high-frequency signal can reduce the size of the coupled line element. Becomes

FIG. 7 shows an example in which electrodes are provided for facilitating surface mounting of the coupled line device of FIG. 5 on a printed circuit board or a hybrid integrated circuit board. FIG.
(A) is a front view of the coupled line element, (b) is a view showing a BB 'cross section, (c) is a bottom view of the view shown in (a), (d)
FIG. 3 is a partial perspective view showing electrodes 41a and 41b and peripheral portions thereof. Among the components shown in FIG. 7, those corresponding to those shown in FIG. 5 are denoted by the same reference numerals.

Referring to FIG. 7, parts different from the coupled line elements shown in FIG.
0b are provided with electrodes 41a and 41b at both ends.
Accordingly, a notch 51 for escaping the electrodes 41 a and 41 b is provided in the external conductor 11.

FIG. 8 shows a structure of a second embodiment different from the first embodiment shown in FIG. here,
The lower part of the outer conductor 12 is at the same level as the lower part of the coupling lines 20a and 20b, and a dielectric 71 is provided between the two coupling lines 20a and 20b and between the two coupling lines 20a and 20b and the outer conductor 12. Is filled. Reference numeral 72 denotes a dielectric, on the bottom surface of which an external conductor 13 is provided. With this structure, an electric field distribution similar to the electric field distribution shown in FIG. 6 can be obtained. That is, almost no electromagnetic field leaks into the air through the dielectric. In this structure, the printed circuit board or the hybrid integrated circuit board may be used as the dielectric 72 and the external conductor 13 as they are.

FIG. 9 shows an example in which electrodes are provided for facilitating surface mounting of the coupled line device of FIG. 8 on a printed circuit board or a hybrid integrated circuit board. FIG.
(A) is a front view of the coupled line element, (b) is a view showing a cross section taken along the line CC ', (c) is a bottom view of the view shown in (a), (d)
FIG. 2 is a partial perspective view showing an electrode. Components corresponding to those shown in FIG. 8 among the components shown in FIG. 9 are denoted by the same reference numerals. In FIG. 7, a description will be given of a portion different from the coupled line element shown in FIG. 8 in that electrodes 42a and 42b are provided at both ends of the coupled lines 20a and 20b.

In the coupled line elements described with reference to FIGS. 5 to 9, the phase velocities of the even mode and the odd mode can be matched, and since the coupled lines 20a and 20b face each other, the coupling capacitance is large and the ground is grounded. Outer conductor 10-1
There is no leakage of the electromagnetic field of the high-frequency signal from 3 to the outside, so that no power loss occurs. In addition, each of the dielectrics 70 to 72 can be handled as a chip component, and a material different from the material of the printed circuit board or the hybrid integrated circuit board constituting the circuit of FIGS. It is also possible to select. For example, if a material having a large relative permittivity such as ceramic is used, the coupling line element becomes extremely small, which is useful for reducing the size of a high-efficiency high-frequency signal power amplifier.

In the above description, the case where the field effect transistor is used as the amplifying element has been described.
It will be apparent from the above description that a similar effect can be obtained by using a bipolar transistor instead.

[0047]

As is apparent from the above description, according to the present invention, the output terminal of the amplifying element is virtually grounded with respect to the second harmonic of the fundamental wave, which is one factor that deteriorates the efficiency of the high-frequency power amplifier. I can do it. In order to further enhance such an effect, a small-sized, wide-band, high-efficiency, high-frequency power amplifier is manufactured by using two line-shaped coupled lines facing each other with an outer conductor and using a coupled line element filled with a dielectric material. Became possible. Therefore, the effect of the present invention is extremely large.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention and an equivalent circuit diagram thereof.

FIG. 2 is a circuit diagram showing another embodiment of the present invention.

FIG. 3 is a circuit diagram showing an equivalent circuit of FIG. 2;

FIG. 4 is a circuit diagram showing still another embodiment of the present invention.

FIG. 5 is a perspective view showing one embodiment of a coupled line element which is a component of FIG. 1;

6 is an electric field diagram showing an even mode and an odd mode electric field distribution of the coupled line element of FIG. 3;

FIG. 7 is a structural diagram showing a specific structure of the coupled line element of FIG. 3;

FIG. 8 is a perspective view showing another embodiment of the coupled line device of FIG. 1;

FIG. 9 is a structural diagram showing a specific structure of the coupled line element of FIG. 6;

FIG. 10 is a circuit diagram for explaining the operation principle of a conventional amplifier.

11 is a waveform diagram showing a drain current waveform and a drain voltage waveform of the amplifier of FIG.

FIG. 12 is a circuit diagram showing another conventional example and a standing wave diagram of a second harmonic.

FIG. 13 is a circuit diagram of a mounted conventional amplifier and a phase diagram showing a phase angle of a high-frequency signal.

14 is an equivalent circuit diagram of the circuit in FIG. 10 at a fundamental wave.

FIG. 15 is a circuit diagram showing another conventional example.

[Explanation of symbols]

10-13 External conductor 20 Coupling line 31-37 Electric field 41,42 Electrode 51 Notch 70-72 Dielectric 101 Field effect transistor 102 Amplifying element 103 Parasitic line 106,107 High frequency signal source 108 High frequency constant current source 109 Signal source Impedance 110 Main transmission path 111,112 Partial transmission path 113-115 Transmission path 116 Partial transmission path 118 Open stub 121 Input terminal 134 Fundamental wave matching circuit 141 Transmission path 150 Load impedance 156-159 Impedance 171 Current waveform 172 Voltage waveform 181- 196 points C1 to C5, C6, C7 capacitance I _DC direct current _{I D (t), I M} (t), I OUT (t) a high-frequency current L1 to L3, L6, L7 inductance Δθ, θ1~θ6 electrical length V _DS , V _GS DC power supply V _D t) high frequency drain voltage

────────────────────────────────────────────────── ─── Continuation of the front page (56) References JP-A-62-111 (JP, A) JP-A-4-83408 (JP, A) JP-A-1-279612 (JP, A) US Patent 4647878 (US , A) Koizumi, "Design Method of Filter for High Frequency Band", Transistor Technology, February 1988, pp. 403-412 (Figs. 5 and 6) (58) Fields investigated (Int. Cl. ⁷ , DB name) H03F 3/60 H01P 5/18

Claims (2)

(57) [Claims] 1. A power amplifying means (101) for power-amplifying a high-frequency signal in an amplifying element part (102) to obtain high-frequency power at an output terminal (D), and a first amplifying means serving as an output terminal of the power amplifying means. Connection point (19
3) and first coupling line means (20a) from the first connection point to a second connection point (194) having a first electrical length and obtaining an output of a high-frequency signal; The electric length including the parasitic line (103), which has the same length as the electric length and has a mutual coupling and is parasitic between the amplifying element unit and the output terminal, is amplified by the power amplifying means. A second coupling line that has an electrical length such that an output of the amplifying element is virtually grounded with respect to a second harmonic of a fundamental wave of the signal, and acts so that the output of the amplifying element is virtually grounded; Means (20b) , wherein the first and second coupling line means are arranged in parallel with the surfaces of two strip-shaped conductors facing each other.
A substrate dielectric (7) having a line and a substrate outer conductor (13) on the bottom surface.
2) setting the two connection lines on the upper surface of the substrate
Of the strip-shaped conductor to substantially contain the electric field of the path.
Along the two coupled lines whose plane is perpendicular to the substrate
Around the two coupled lines not facing the substrate dielectric
An outer conductor (12) covering the side, between the two coupled lines
And filling between the two coupled lines and the outer conductor
A high-frequency power amplifier comprising: 2. A power amplifying means (101) for power-amplifying a high-frequency signal in an amplifying element section (102) to obtain a high-frequency power at an output terminal (D), and one end (196) at an output terminal of the power amplifying means. ) Is connected, and the electrical length between the amplification element unit and the output terminal is
A transmission path unit (141) that is not longer than a quarter of the wavelength of the second harmonic of the fundamental wave of the high-frequency signal amplified by the power amplification unit; A first coupled line means (20a) for connecting a point (194) for obtaining a fundamental wave output; and the first coupled line means having the same electrical length, having mutual coupling, and providing the second harmonic. A second coupling line means (20b) that acts to open the other end (195) of the transmission path means to a wave to virtually ground the output of the amplifying element unit . , said first and second coupling line means are arranged in parallel to face surfaces of the two strip-shaped conductor coupling
A substrate dielectric (7) having a line and a substrate outer conductor (13) on the bottom surface.
2) setting the two connection lines on the upper surface of the substrate
Of the strip-shaped conductor to substantially contain the electric field of the path.
Along the two coupled lines whose plane is perpendicular to the substrate
Around the two coupled lines not facing the substrate dielectric
An outer conductor (12) covering the side, between the two coupled lines
And filling between the two coupled lines and the outer conductor
A high-frequency power amplifier comprising: