JP3092246B2 - Amplifier circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a non-cutoff (A)
This is an attempt to improve the efficiency of an amplifier circuit capable of obtaining a linear output combining characteristic by a (class) operation and also being used for large-output amplification.

[0002]

BACKGROUND OF THE INVENTION In a conventional class A amplifier, a base bias is selected so that an operating current (collector current when there is no signal) becomes 1/2 of a peak current (that is, a center point of an operating curve), and a positive or negative input signal is selected. The amplifying element is always in an active state over both cycles to perform linear amplification. FIG. 2 shows a class A push-pull amplifier as an example, in which deep base biases E1 and E2 are added to transistors 1 and 2 having a push-pull configuration. FIG. 3 shows input / output characteristics of the class A push-pull amplifier of FIG. According to this, since the base bias is applied to the transistors 1 and 2, the transistors 1 and 2 are not cut off as in the class B amplifier, and the distortion is low.

However, in such a configuration, a large idle current is required, and the range of class A operation is determined.
There is a drawback that one transistor is cut off to cause distortion. Therefore, an amplifier circuit described in Japanese Patent Application Laid-Open No. 62-214707 has been proposed as a solution to such a drawback so as to reduce the idle current and obtain a wide range of class A operation. I have.

This uses a current-current conversion circuit shown in FIG. This current-current conversion circuit is an NPN
The emitters of the transistor Tr1 and the PNP transistor Tr3 are commonly connected to each other, and a signal is input from the common emitter. The base of the transistor Tr1 is NPN
It is connected to the base of transistor Tr2. Also,
The base of the transistor Tr3 is a PNP transistor Tr
4 are connected to the base. Transistors Tr2, T
In r4, its own base and collector are connected in common, and emitters are connected in common. This common emitter is connected to a reference potential (here, a ground potential). A constant current is supplied to each of the transistor Tr2 and the transistor Tr4. In this circuit, the collector currents of the transistors Tr1 to Tr4 are set to Ic1 to Ic.
4 is assumed. The collector currents Ic2 and Ic4 of the transistors Tr2 and Tr4 are constant currents, and their values are equal here and are _ID .

In the circuit of FIG. 4, when I _D = 1, the values of Ic1 and Ic3 with respect to the input current Ix are obtained and plotted as shown in FIG. 5 (for a detailed analysis, see Japanese Patent Application Laid-Open No. Sho 62-214707). Gazette). According to this, it is understood that even if the input current Ix increases and one of Ic1 and Ic3 increases, the other approaches zero and does not cause zero crossing. The value obtained by adding Ic1 and Ic3 is a straight line passing through the origin.

Summarizing the above, it can be seen that the circuit of FIG. 4 has the following characteristics. The input current Ix is converted into output currents Ic1 and Ic3. Regardless of the sign of the input current Ix, the sign of the output currents Ic1 and Ic3 is constant and does not reverse. Ix = Ic1 + Ic3 Ic1 × Ic3 = Constant value Therefore, if these output currents Ic1 and Ic3 are respectively amplified by the output stage elements and the output currents of these output stage elements are combined and supplied to the load, the output stage The device is always active and operates in Class A operation.

[0007]

2. Description of the Related Art FIG. 6 shows a conventional amplifier circuit using the current-to-current conversion circuit shown in FIG. Input signal Vin
Is input to the voltage-current conversion circuit 8. In the voltage-current conversion circuit 8, the transistors Tr5 and Tr6 constitute a differential amplifier. The transistor Tr9 and the resistor r6 constitute a constant current circuit of the differential amplifier, and a constant current defined by a voltage between both ends of the diode d3 and the resistor r3 flows. A current mirror circuit including transistors Tr7 and Tr8 is connected to the differential amplifier. The input signal Vin is input to the base of the transistor Tr5,
The collector output of the transistor Tr5 is input to the base of the transistor Tr10. Therefore, when the input Vin is high, the base potential of the transistor Tr10 decreases, and when the input Vin is low, the transistor Tr10
Base potential rises.

A transistor Tr10, a resistor r4, and transistors Tr11 and Tr1 are connected between the power supplies + B1 and -B1.
2. A resistor r5 is connected in series. Also, the voltage + B
The voltage between 1 and -B1 is divided by a resistor r1, diodes d1, d2, resistor r2, diode d3, and resistor r3. The base of the transistor Tr11 is a diode d
2 and a resistor r2. Also, it is grounded via the capacitor C _0. Therefore, the voltage of the resistor r1 and the voltages of the diodes d1 and d2 are applied between the collector of the transistor Tr10 and the base of the transistor Tr11. Therefore, when the base potential of the transistor Tr10 rises, the transistor Tr10 → the resistance r4 →
The current flowing through the transistor Tr11 increases, and when the base potential of the transistor Tr10 decreases, the current decreases. The collector of the transistor Tr11 is connected via the line 13 to the current-current conversion circuit 3 at the next stage.

The transistor Tr12 and the resistor r5 form a constant current circuit, and a constant current defined by the voltage across the diode d3 and the resistor r3 flows. Therefore, the line 13 includes the collector current Ic1 of the transistor Tr11.
A current Ix, which is the difference between 1 and the collector current Ic12 of the transistor Tr12, flows.

That is, the voltage-current conversion circuit 8 outputs the following current Ix according to the input Vin. When Vin = 0, Ic11 = Ic12, and Ix = Ic11−Ic12.
= 0. When Vin <0, the collector potential of the transistor Tr5 (that is, the base potential of the transistor Tr10) increases, and Ic11> Ic
The current Ic11-Ic12 of the difference flows out to the current-current conversion circuit 3 as the current Ix. When Vin> 0, the collector potential of the transistor Tr5 decreases, and Ic11
<Ic12, and the difference current Ic12−Ic11
Flows from the current-current conversion circuit 3 as the current Ix.

Thus, the input Vin is applied to the line 13.
, A current Ix proportional to the current flows, and voltage-current conversion is performed. The output current Ix of the line 13 is the current-current conversion circuit 3
To control the values of the currents Ic1 and Ic3 in a relation of Ix = Ic1 + Ic3 Ic1 × Ic3 = constant as described above.

The current mirror circuit 6 includes a differential amplifier 15
The output stage transistor Tr13 is controlled so that the voltages of both inputs become equal. Since the current Ic1 flows through the resistor r7 connected to the + input of the differential amplifier 15, r7
A voltage drop of Ic1 occurs. Therefore, the resistance r8 connected to the negative input of the differential amplifier 15 has Ic1 · Ic so that the voltage drop is equal to the voltage drop of the resistance r7.
A current of (r8 / r7) flows. Therefore, the current amplification factor A of the current mirror circuit 6 is A = r8 / r7.

Similarly, the current mirror circuit 7 controls the output transistor Tr14 so that the voltages of both inputs of the differential amplifier 17 become equal. Since the current Ic3 flows through the resistor r9 (= r7) connected to the + input of the differential amplifier 17, a voltage drop of r9 · Ic3 occurs. Therefore, the resistance r connected to the negative input of the differential amplifier 17
At 10 (= r8), a current Ic3 · (r10 / r9) flows so that the voltage drop becomes equal to the voltage drop of the resistor r9. Therefore, the current amplification factor A of the current mirror circuit 7 is A = r10 / r9 = r8 / r7.

As described above, the transistor Tr1
3, Tr14, collector currents A.Ic1 and A.Ic3 flow. Then, the voltage corresponding to the difference is the output V
Out is taken out from the connection point of the transistors Tr13 and Tr14 and supplied to the load 10. Also, the output Vou
t is connected to the transistor Tr6 via the resistors r11 and r12.
Is fed back to the base.

According to the circuit of FIG. 6, the currents Ic1, Ic2
Does not become 0, the currents A.Ic1 and A.Ic3 also do not become 0, and therefore the output transistor Tr1
3, Tr14 does not cut off and operates in class A.

[0016]

In order to effectively use the characteristics of FIG. 5 in a practical circuit, the output transistors Tr13 and Tr14 in the circuit of FIG. It is necessary to pass current. However, in the circuit of FIG. 6, the output transistor Tr13,
Tr14 is always up may take the output magnitude of the power supply voltage + B _3, because it is driven by -B _3, input a large voltage is applied between the collector and emitter of the output stage transistor Tr 13, Tr14 even at zero ing. For this reason, power loss (heat generation amount) in the output stage transistors Tr13 and Tr14 becomes large, and there is a disadvantage that the efficiency is low although the performance is high. SUMMARY OF THE INVENTION An object of the present invention is to provide an amplifier circuit which improves the efficiency by improving the drawbacks of the prior art and reducing the loss in the output stage element due to the idle current.

[0017]

According to the present invention, the current-
In an amplifier circuit for extracting an output current having characteristics as shown in FIG. 5 using a current conversion circuit and a current mirror circuit, a voltage signal corresponding to the output current of the current mirror circuit and the input signal to be amplified are synthesized and output. A voltage synthesizing circuit, an output stage element for amplifying an output voltage of the voltage synthesizing circuit, an output circuit for synthesizing output currents of these output stage elements and supplying the load to a load, and supplying a driving power supply to the output stage element. A main power supply path, a switching element inserted into the main power supply path to switch on and off the main power supply path, and a switching element inserted into the main power supply path to smooth the output of the switching element and supply the load to the load. A switching circuit that variably switches a ratio between an on-period and an off-period according to a level of the input signal, and a voltage required by the load; Switching control means for mainly supplying power from the main power supply path, and a power supply path for supplying drive power to the output stage element separately from the main power supply path, the input signal of the input signal in a faster response than the main power supply path An auxiliary power supply path capable of supplying power following a level change, and an amount of power supplied from the auxiliary power supply path to the load inserted into the auxiliary power supply path in a state of being cascaded to the output stage element And an auxiliary power supply amount adjusting element for adjusting the power supply.

[0018]

According to the present invention, since the drive voltage of the output stage element is reduced to a voltage corresponding to the input signal level by the main power supply path which is driven by switching, the applied voltage of the output stage element when the input is zero is reduced. Therefore, even if the idle current of the output stage element is set to a large value, the loss (heat generation) here is reduced and the efficiency can be improved. Moreover, since the main power supply path itself is driven by switching, the loss in the main power supply path is small. In addition, when the input power suddenly rises such that the main power supply path cannot follow due to the presence of the smoothing circuit, power is supplied from the auxiliary power supply path having better responsiveness than the main power supply path. It can follow the steep rise of the amplitude input and does not clip the output. Even if the auxiliary power supply path is driven by such a power supply voltage, power is mainly supplied from the main power supply path in normal times, so that the average current value flowing through the auxiliary power supply path can be small, and this auxiliary power supply Road losses are small. As a result, an amplifier circuit having high performance, high efficiency, and good follow-up performance even with a rapid rise of a large amplitude input is realized.

[0019]

FIG. 1 shows an embodiment of the present invention. The input signal is input to the voltage-current conversion circuit 8 and the voltage amplifier 10
The voltage is amplified. The output of the voltage amplifier 19 has a resistor r3
And a current Iin corresponding to the output voltage Vin of the voltage amplifier 19 flows through the resistor r3 to perform voltage-current conversion.

The output current Iin of the voltage-current conversion circuit 8 is
It is input to the current-current conversion circuit 3. This circuit 3 has the same configuration as that of FIG. That is, the NPN transistor Tr1 (first transistor) and the PNP transistor Tr3 (third transistor) are commonly connected to each other, and a signal is input from the common emitter. The base of the transistor Tr1 is connected to the base of an NPN transistor Tr2 (second transistor). The base of the transistor Tr3 is connected to the base of a PNP transistor Tr4 (fourth transistor). The transistors Tr2 and Tr4 have their bases and collectors connected in common, and their emitters are connected in common. This common emitter is connected to a reference potential (here, the midpoint potential of the output circuit 21). A constant current _ID is passed from the transistor Tr2 to the transistor Tr4 by the constant current circuits 23 and 25, respectively. As a result, the currents I ₁ (I _c1 ) and I ₂ (I
_c2 ) flows, and the difference current I ₁ −I ₂ becomes the output current I _{in of the} voltage-current conversion circuit 8.

The current mirror circuit 6, 7 passing the current I _1, the same current as I ₂ I _1, I ₂ to the resistor r1, r2. The resistors r1 and r2 convert the currents I ₁ and I ₂ into voltages. That is, the other ends of the resistors r1 and r2 are connected to the output terminal of the voltage amplifier 19. In this case, if r1 = r2 = r3 = R, a _{V in = -V r3 = -I in} · R = (I 2 -I 1) R. Therefore, the output voltages V1, V2 _{are, V1 = I 1 · R-} V in = I 1 · R + (I 2 -I 1) · R = I 2 · R V2 = I 2 · R-V in = I 2 · R− (I ₂ −I ₁ ) · R = I ₁ · R Therefore, the characteristics of the V _in to V1, V2 V1-V2 = I 2 · R-I 1 · R = (I 2 -I 1) R = V in V1 · V2 = I 1 · I 2 · R 2 - _ID ² · R ² = constant, and the output currents I ₁ and I ₁ of the current mirror circuits 6 and 7
₂ will have been converted to voltages V1, V2 corresponding to.
These configurations of the resistors r1 and r2 and the voltage amplifier 19
The voltage synthesizing circuit 9 is formed.

Voltages V1 and V2 are applied to output stage transistors Qa and Qa 'via drive stage transistors Qd and Qd'.
Of the output stage transistors Qa, Qa '
From the output currents I ₁ ,
Current I _{ou t1,} I _out2 that current amplification of I ₂ are output.
The currents I _out1 and I _out2 are combined in an output circuit 21 composed of resistors Ra and Ra ′ and supplied to the load 10 as a load current _IRL . The characteristics of the current I _out1, I _out2, I _RL shown in FIG. According to this, since the currents I _out1 and I _out2 are always currents in a fixed direction, the output stage transistors Qa and Qa ′ do not cut off and can perform class A operation in a wide range.

Next, a power supply path to the output stage transistors Qa and Qa 'will be described. Here, only the + side will be described, and the portion common to the + side will be denoted by “′” and description thereof will be omitted. The main power supply path 42 that mainly supplies power to the load 10 includes a power supply 44 (power supply voltage + B), a switching transistor Qs (switching element), a smoothing coil L1 (smoothing circuit), an output transistor Qa, and a resistor Ra (= Ra ′). The auxiliary power supply path 48, which supplies the shortage of power supply by the main power supply path 42,
4, resistor Rs, cascaded transistor Qb
(Auxiliary power supply amount adjusting element), an output transistor Qa, and a resistor Ra.

The resistance Rs is a current I flowing through the auxiliary power supply path 48.
_This is a small resistor for detecting _Qb . R by the current I _Qb
The voltage V _Rs generated at both ends of s is input to the hysteresis comparator 50. Hysteresis comparator 50
Constitutes switching control means together with the resistor Rs, and when the voltage _VRs becomes higher than a predetermined upper limit value V _HCH , "0"
Turning the transistor Qs and output, once "0" after reaching the until the voltage V _Rs drops to a predetermined lower limit value V _HCL holds "0", the voltage V _Rs than the lower limit value V _HCL and outputs the composed and "1" low to turn off the transistor Qs, once the voltage V _Rs once they become "1" holds "1" until becomes higher than a predetermined upper limit value V _HCH. Thus, the transistor Qs performs a switching operation. Smoothing coil L1
Is to smooth the output waveform of the transistor Qs by this switching operation and supply it to the load. The diode D3 is a flywheel diode that forms a current path for a current flowing out of the smoothing coil L1 when the transistor Qs is turned off.

A DC power supply 52 is connected between the emitter of the output transistor Qa and the base of the transistor Qb. The power supply voltage V1 is set to V1 = V _Qamin + V _QbBE . Here, V _Qamin is the lowest voltage that can ensure linearity even when the output transistor Qa is at the maximum output, and is usually about 2 to 3 V. V _QbBE is the base-emitter voltage of the transistor Qb and is approximately 0.6
V. The power supply 5 is connected to the emitter of the transistor Qa.
A path 51 extending to the base of the transistor Qb via 2 corresponds to an auxiliary power supply control means.

The operation of the power supply path will be described. First,
It is assumed that the output of the hysteresis comparator 50 is “1” and the switching transistor Qs is off. The audio input signal is supplied to the output transistor Qa via the transistor Qd. As a result, the current I _Qb flows through the auxiliary power supply path 48 to the transistor Qa and is supplied to the load 10. This current I _Qb generates a voltage V _Rs across the resistor Rs. This voltage _VRs is a predetermined upper limit value V
_When the signal reaches _HCH , the output of the hysteresis comparator 50 is inverted to "0" to turn on the transistor Qs. Thus, current is supplied from the main power supply path 42.

When the transistor Qs is turned on, the resistance value of the main power supply path 42 is smaller than the resistance value of the auxiliary power supply path 48. Therefore, once the transistor Qs is turned on, the current I _L1 from the main power supply path 42 increases, The current I _Qb from the auxiliary power supply path 48 decreases accordingly. The currents I _L1 and I _Qb are _equal to the summation point 49.
Are supplied to the load 10 via the transistor Qa as the load current I _RL , thereby supplying necessary electric power to the load 10.

_Just before the current I _Qb of the auxiliary power supply path 48 becomes zero, the voltage V _Rs across the resistor Rs reaches a predetermined lower limit value V _HCL , the output of the comparator 50 is inverted to “1”, and the transistor Qs is turned off. Turn off. When the transistor Qs is turned off, the current I _L1 of the smoothing coil L1 flows through the diode D3. As the current I _L1 gradually decreases, the current I _Qb of the auxiliary power supply path 48 increases, and when the voltage V _Rs reaches a predetermined upper limit value V _HCH , the output of the comparator 50 is again inverted to “0” and the transistor Qs is turned on, and the same operation is repeated thereafter. As described above, the transistor Qs is switched by self-excited oscillation with the ratio of the ON period and the OFF period being variably set according to the input signal level.

According to such an operation, the transistor Q
b is always in an active state, and a load 1
The current I _RL flowing to 0 is supplied from both the main power supply path 42 and the auxiliary power supply path 48. Transistor Qs of main power supply path 42
Is a switching drive, so the loss is small. Although the auxiliary power supply path 48 causes a loss in the transistor Qb, the current I _Qb flowing through the transistor _Qb depends on the value of the resistor Rs and the setting of the reference voltages V _HCH and V _HCL of the hysteresis comparator 50. Since it can be reduced to almost the same as the ripple current of the output current _IL1 ,
_L1 >> I _Qb, and the loss in the transistor Qb can be reduced.

Further, since the transistor Qb is always on and V _QbBE is always about 0.6 V, the collector-emitter voltage V _Qa of the output stage transistor _Qa is always V _Qa = V1−V _QbBE = V _Qamin Is kept. Therefore, the loss of the output stage transistor Qa can be minimized. According to this circuit, the effect of keeping the loss of the output stage transistor Qa, which is the largest loss source, at a minimum is great. This is particularly effective when a large amount of idle current flows.

FIG. 8 shows a portion of the transistors Qb and Qs in FIG. The operation of this circuit will be described with reference to FIG. The signal source 54 is connected to the transistor Qb.
Step-like input V _in falls from (t1), initially the transistor Qs is because off, the output current I _RL is supplied from the transistor Qb for all auxiliary power circuit 48. When the current I _{Qb of the} transistor Qb increases at a stretch and the voltage V _Rs across the resistor Rs reaches the upper reference voltage V _HCH of the comparator 50, the output of the comparator 50 is inverted to “0” to turn on the transistor Qs. . Thus, power supply from the main power supply path 42 is started. The current _IL1 from the main power supply path 42 does not increase at a stretch because of the _presence of the smoothing coil L1, but increases gradually. Main current I _L1
Gradually increases, the auxiliary current I _Qb gradually decreases due to I _Qb = I _RL −I _L1 . The auxiliary current I _Qb decreases and the voltage V
_Rs and reaches the reference voltage V _HCL of lower comparator 50 (t2), the transistor Qs is turned off, the output current I _L1 of the smoothing coil L1 decreases gradually. Then, when the auxiliary current I _Qb increases and the voltage V _RS reaches the upper reference voltage V _HCH (t3), the transistor Qs is turned on again, and the same operation is repeated thereafter. The first rising part (time t1 immediately after), since the output current I _L1 of the smoothing coil L1 does not catch up immediately, the auxiliary current I _Qb is mainly the input V _in
Is small and the change in the load current _IRL is small, power is almost supplied from the main current source 42. Since the music signal has low energy of a high-frequency component (a signal having a large change), the average value of the auxiliary current I _Qb is small, and the load current I _RL is low.
Most of the main power supply path 4 has high efficiency by switching drive
2 as current I _L1 .

In the circuit of FIG. 1, the load current _IRL is all supplied from the transistor Qa (Qa '), but the collector-emitter voltage _VQa of the transistor Qa is necessary to turn on the transistor Qa as described above. Since only the minimum voltage V _Qamin is applied, the loss here is small. Therefore, the efficiency of the entire circuit is very high. Moreover, since the power from a fast auxiliary power line 48 response for steep rise of the input V _in is supplied, it is possible to supply the necessary power to the load 10 to follow the steep rise . Moreover, the auxiliary power supply path 48
Is driven at a high voltage + B, as in the main power supply path 42, so that a predetermined maximum output can be supplied to the load following an abrupt rising of a large amplitude input without clipping the output. .

Further, according to the circuit shown in FIG. 1, an effect in terms of acoustic characteristics can be obtained. That is, since the transistor Qb is not cut off, is in some but active state is always small current, operates to absorb the ripples of the main current I _L1 by switching a current I _Qb from the transistor Qb. Therefore, the supply voltage to the output stage transistor Qa does not have a ripple voltage, the influence of the ripple voltage on the output is eliminated, and deterioration of the distortion factor and S / N due to switching driving can be prevented.

In the circuit of FIG. 1, the average value of the auxiliary current I _Qb is reduced by setting the reference voltage V _HCH on the upper side of the hysteresis comparator 50 to a sufficiently small value to reduce the loss in the transistor Qb. However, the switching frequency of the transistor Qs increases, and the switching loss increases. Therefore, it is desirable to set the magnitude of the upper reference voltage V _HCH so that the sum of the two is minimized in consideration of the loss in the transistor Qb and the switching loss in the transistor Qs. In addition, the lower the reference voltage V _HCL is, the smaller the loss is in the range where the transistor Qb is not cut off. When a bipolar transistor is used, the free-running switching frequency of the transistor Qs can be set, for example, to an upper limit of about 100 kHz (the free-running frequency varies depending on the output). When the upper limit is set to 100 kHz or more, a power MOS transistor or the like can be used if switching cannot be performed by a bipolar transistor.

FIG. 10 shows a configuration example of the hysteresis comparator 50 shown in FIG. A series connection circuit of resistors R3 and R4 is connected between the + B power supply line 56 and a connection point 58 between the transistor Qs and the diode D3. Also,
A transistor Qc, a resistor R2,
The series connection circuit of R1 is connected. And the resistance R
A connection point 60 between R3 and R4 is connected to the base of the transistor Qc, and a connection point 62 between the resistors R1 and R2 is connected to the base of the switching transistor Qs.

According to the above configuration, when the transistor Qs is turned off, the transistor Qc is turned on by the voltage generated across the resistor R3. At this time, a voltage obtained by dividing the voltage _VRs by the resistors R1 and R2 is applied to the transistor Qs. In this state, the auxiliary current I _Qb increases and the voltage V _Rs increases, and the base of the transistor Qs
When the voltage between the emitters rises to about 0.6 V, the transistor Qs is turned on, and the main current _IL1 flows. Transistor Qs
Turns on, the potential at point 58 rises to approximately + BV, so the potential at point 60 also rises and transistor Qc turns off. When the transistor Qs is turned on and the main current I _L1 increases, the auxiliary current I _Qb decreases and the voltage V _Rs decreases. In a state where the transistor Qc is turned off because the base-emitter of the transistor Qs voltage V _Rs is applied as it is, if this voltage V _Rs drops below approximately 0.6V, the transistor Qs is turned off. When the transistor Qs is turned off, the potential at the point 58 drops and the transistor Qc
Turns on.

As described above, according to the hysteresis comparator 50 of FIG. 1, when the transistor Qs is off, a voltage obtained by dividing the voltage _VRs by the resistors R1 and R2 is applied between the base and the emitter of the transistor Qs. , The transistor Qs is turned on only when the voltage V _Rs rises to a voltage that generates a voltage of about 0.6 V or more across the resistor R2. In addition, when the transistor Qs is turned on, it takes between the base and emitter of the voltage V _Rs as it is transistor Qs, the transistor Q when the voltage V _Rs becomes equal to or less than about 0.6V
s turns off. Thereby, a comparator operation with hysteresis is realized, and the transistor Qs performs a switching operation by self-excited oscillation.

The lower reference voltage V _HCL is set by the value of the resistor Rs. The upper reference voltage V _HCH is set by the ratio of the values of the resistors R1 and R2, and the larger the value of the resistor R1 is, the larger the value of V _HCH becomes.

The hysteresis comparator 50 is not limited to the configuration shown in FIG. 10, but may be variously configured using an operational amplifier or the like. However, the configuration as shown in FIG. 10 has an advantage that it can be easily realized such that a separate power supply is not required for the hysteresis comparator 50.

The switching of the power supply path to the load 10 is performed by detecting the auxiliary current value and the main power supply path 42.
And the voltage of the auxiliary power supply path 48. That is, power is supplied from the main power supply path 42 with high efficiency so that the output voltage of the main power supply path 42 becomes higher in normal times, and when the input signal rises rapidly, the output voltage of the main power supply path 42 becomes higher. Does not catch up suddenly, so that the output voltage of the auxiliary power path 48 becomes higher,
The transistor Qb is turned on to supply power from the auxiliary power supply path 48 with high speed response.

The output transistors Qa and Q shown in FIG.
The power supply path to a 'can be conceptually captured as shown in FIG. That is, a high-efficiency current buffer circuit (switching transistor 42) that can supply current to main power supply path 42 with high efficiency but has a slow response to a change in input signal.
And a smoothing coil L1), and a high-speed voltage buffer circuit (a path 51 for controlling the transistor Qb and its base voltage) which has a fast response to a change in the input signal but has a large loss is inserted into the auxiliary power supply path 48. The currents I _L1 and I _Qb of the power supply paths 42 and 48 are added at an addition point 49 to form a transistor Q
A load current I _RL (= I _L1 + I _Qb ) is supplied to the load 10 via a. The current detection circuit (resistor Rs) detects the magnitude of the auxiliary current IQb, and responds accordingly to the high-efficiency current buffer circuit 4 via the control circuit (hysteresis comparator 50).
2. The main current I _L1 is adjusted by controlling _L1 . The high-speed voltage buffer circuits Qb and 51 have a shortage of the main current I _L1 (I _RL −
_IL1 ) is supplied as an auxiliary current IQb.

The voltage V at the collector of the output transistor Qa
s is only slightly greater than the output voltage Vout,
This minimizes the loss of the output transistor Qa. The load current I _RL responds to a steep rising of the input first by the auxiliary current I _Qb , and gradually the main current I _R
L1 becomes dominant. Since the auxiliary current _IQb rises at an extremely high speed, no loss occurs in the output waveform. The supply of the auxiliary current I _Qb is an operation for only a short time, and the energy is mainly supplied by the main current I _L1 from the main power supply path 42 with a small loss, so that the overall efficiency is high. Thus, a power amplifier circuit with high efficiency and a high slew rate is realized.

Next, a specific circuit of the circuit of FIG.
5 is shown. 12 to 15 show A1 to A8, B1 to
B5 are interconnected at E1 to E6. The same reference numerals are used for parts corresponding to those in FIG. In addition,-corresponding to the + side
The part on the side is indicated by adding "'". Note that the configuration of the power supply path to the output stage transistors Qa and Qa 'will be described with respect to the + side circuit (FIG. 14), and the-side circuit (FIG. 15) is configured similarly to the + side circuit. Is omitted.

In FIG. 12, an input signal is input to a transistor Q30 constituting a first-stage differential amplifier circuit, and is transmitted to transistors Qg and Qg 'via transistors Q32 and Q32' and transistors Qf and Qf 'in FIG. I will Then, the currents are output from the transistors Qg and Qg 'and output as currents I ₁ and I ₂ from the transistors Tr 1 and Tr 3 of the current-current conversion circuit 3 via the current mirror circuit 6 via the capacitor Cf and the resistor r 3.
These currents I ₁ and I ₂ are converted into voltages by resistors r ₁ and r ₂ , and transistors Q 10, Q 15 and Q 1 constituting the drive stages of FIGS. 14 and 15 via transistors Qh and Qh ′.
0 ', Q15' and output stage transistors Qa, Qa '
To drive the load (speaker) 10. The output signal is negatively fed back to the first transistor Q31 (FIG. 12) and the current-current conversion circuit 3 (FIG. 13).

In FIG. 14, output stage transistor Qa
And the cascade transistor Qb have a parallel configuration of three transistors Q23 and Q20, respectively. The main power supply path 42 supplies a main current to the output stage transistor Qa via a power supply of +90 V → switching transistor Qs → coil L1 → current addition point 49. The auxiliary power supply path 48 supplies an auxiliary current to the output stage transistor Qa via the resistors R10, R11, R12 and the current detection limiter 90 → the transistor Qb. The path 51 in FIG. 1 for controlling the base potential of the transistor Qb is, in FIG. 14, a path from the emitter of the transistor Q10, the zener diode ZD, the transistor Q11, the resistor R13, the diode D10, the transistor Q14, the resistor R14, and the base of the transistor Qb. Equivalent to. As a result, when the input rises rapidly, the base potential of the transistor Qb immediately rises to increase and follow the auxiliary current, and thereafter the main current gradually increases and the auxiliary current decreases. Let me do it.

The hysteresis comparator 50 is configured as follows. A series connection circuit of resistors R33 and R34 is connected between the + 90V power supply line 56 and a connection point 58 between the transistor Qs (Q22) and the diode D3. The auxiliary power supply path 48 includes resistors R10 and R12.
, A transistor Qc, resistors R11, R1
2 are provided in parallel. The connection point 60 between the resistors R33 and R34 is connected to the base of the transistor Qc. The connection point 62 between the resistors R11 and R12 is connected to the base of the transistor Q21.
1 is connected to the base of the switching transistor Qs via the resistor R21.

According to the above configuration, the transistor Q21
Is off (the transistor Qs is off), the transistor Qc is on due to the voltage generated across the resistor R33. At this time, the auxiliary power supply path 48 has a route passing through the resistor R10, a transistor Qc, and a resistor R11.
Are formed in parallel, and a voltage across the resistor R10 is applied between the base and the emitter of the transistor Q21. In this state, the auxiliary power supply path 4 passing through the resistor R10
8 increases the base-emitter voltage of the transistor Q21 to about 0.6V, the transistor 21
Is turned on, the transistor Qs is turned on, and current is supplied from the main power supply path 42. When the transistor Q21 is turned on, the potential at the point 58 rises to almost +90 V,
The potential of 0 also rises, turning off the transistor Qc. When the transistor Q21 is turned on and the current in the main power supply path 42 increases, the current in the auxiliary power supply path 48 decreases. When the transistor Qc is turned off, the auxiliary power supply path 48 is only a route passing through the resistor R10, so that all the current in the auxiliary power supply path 48 flows through the resistor R10, and the voltage across the resistor R10 is about 0.6 V or less. The transistor Q
21 turns off. When the transistor Q21 is turned off, the potential at the point 58 drops and the transistor Qc turns on.

In this manner, the comparator operation having hysteresis is performed by switching whether to flow the current of the auxiliary power supply path 48 to all the resistors R10 or to shunt a part of the current depending on the on / off state of the transistor Qc. When implemented, the transistor Qs performs a switching operation by self-oscillation.

The output stage transistors Qa and Q shown in FIG.
In the power supply path to a ', the following measures have been taken.
The output stage transistor Qa and the cascade transistor Qb each have a parallel configuration. In accordance with this, the smoothing coil L1 is also formed by parallel winding. With this configuration, the output is increased and a stabilizing resistor (resistance 0.22Ω) is connected to the emitter of the cascade transistor Qb.
) Becomes unnecessary, loss due to this can be prevented, and high efficiency can be maintained. In other words, if the smoothing coil L1 is constituted by a single winding, the number of current injection points 49 is one, and the emitters of the cascade transistors Qb are inevitably connected in common. For stabilization on time, each emitter needs a resistance such as 0.22Ω. On the other hand, when the smoothing coil L1 is parallel-wound, the number of current injection points 49 becomes two, and although the resistance of the coil winding exists between them, although small, the operation of each cascade transistor Qb And the problem of stabilization is reduced, so that the emitter resistance is not required for the cascade transistor Qb, and the loss is reduced accordingly.

Between the output of the smoothing coil L 1 and the power supply line 56, the resistors R 15 and R 16 and the capacitor C 10
And a ripple absorption circuit 92 (low-pass filter) composed of As a result, a part of the ripple of the main current due to switching is released to the AC common potential point (power supply line 56), the operating current load on the cascade transistor Qb is reduced even a little, and the loss of the transistor Qb is prevented from increasing. .

The current detection resistors R10, R11, R12
(When the switching transistor Qs is off (the transistor Qc is on), the DC resistance between the parallel resistance of the resistors R10 and R11 and the resistor R12, and when the switching transistor Qs is on (the transistor Qc is off), the resistor R10 , R12) and a diode D11,
A current detection limiter 90 composed of D12 is connected in parallel. As a result, the upper limit of the current detection voltage transmitted to the transistor Qs is limited to 1.2 V, so that overdrive between the base and the emitter of the transistor Qs is prevented, and current can be supplied from the current detection limiter 90 to the cascade transistor Qb. To improve the current supply capability of the cascade transistor Qb (the current path of the transistor Qb is the resistance Rs (FIG. 1) and the resistance R1
If the current is regulated by 0, R11, and R12, a very large current cannot flow. ).

A hold circuit 94 composed of a capacitor C11 and a diode D10 is inserted in the middle of a path for linking the base potential of the cascade transistor Qb to the emitter potential (output potential) of the output stage transistor Qa. As a result, the high-frequency signal is subjected to peak detection, and the voltage is held in the capacitor C11, so that the base potential of the cascade transistor Qb is made constant, and
The emitter output voltage of the cascade transistor Qb is prevented from becoming insufficient for the output stage transistor Qa due to a phase delay or the like.

Here, the collector-emitter voltage of the output stage transistor Qa is set to about 2V. That is, from the upper end potential of the Zener diode ZD to the base-emitter of the transistor Q11, the diode D10, the base-emitter of the transistor Q14,
b, four diode junctions between the base and the emitter (0.b.
(6 × 4 = 2.4 V), the collector potential of the output stage transistor Qa is determined. In addition, the potential between the lower end of the Zener diode ZD and the base-emitter of the transistor Q15,
The emitter potential of the output stage transistor Qa is determined via two diode junctions (0.6 × 2 = 1.2 V) between the base and the emitter of the transistor Qa. That is, (3.
2-2.4)-(0-1.2) = 2.0 V is the collector-emitter voltage of the output stage transistor Qa.

A switching frequency fine adjustment circuit 96 including a capacitor C12 is inserted in parallel in the current value detection system of the auxiliary current. When the switching free-running system composed of the hysteresis comparator 50 and the switching transistor Qs is optimized in design, the switching frequency may become too high (for example, 100 kHz or more). Can be reduced, thereby increasing the degree of design freedom.

14 and 15, for example, three portions surrounded by alternate long and short dash lines 111, 112, and 113 can be constituted by power ICs (hybrid ICs). The power IC 111 includes output transistors Qa, Qa '(Q23, Q23') on the + and-sides, and constitutes an output stage. The power IC 112 includes a cascade transistor Qb (Q20) on the + side and a switching transistor Qs (Q22), and constitutes a cascade stage and a switching stage on the + side. The power IC 113 has a cascade transistor Qb 'on the negative side.
(Q20 ') and the switching transistor Qs' (Q
22 '), and constitutes the cascade stage and the switching stage on the negative side.

As described above, a part of the power section is replaced with a power IC.
The following effects can be obtained by using the configuration.
Since the output stage is constituted by the independent power IC 111, it is less susceptible to switching noise. Also,
This power IC 111 can be used as a general-purpose output stage,
Compatibility with a conventional output stage can be provided. Since the cascade stage and the switching stage are combined to constitute one power IC 112 (113), the size is reduced. The power ICs 111 to 113 are easy to manufacture because they are almost composed of only resistors and semiconductors. By externally attaching all or a part of the control unit of the circuit, adjustment is facilitated, and versatility of the IC is secured. That is, the cascade stage bias circuit ZD peripheral 4
Since the constants 0 and the hysteresis circuit 50 of the switching stage have different constants depending on the application, all or a part of the hysteresis circuit is externally connected, and a part with little change is taken in the IC.
Versatility is ensured.

The circuit shown in FIG. 13 can also be made into an IC. 13, the current-voltage conversion circuit 3 and the current mirror circuits 6, 7 which require temperature compensation;
If the resistances r1, r2, and r3 that determine the output stage drive gain and the like are formed on the same IC, the variation in characteristics can be improved. FIG. 16 shows an example in which the blocks indicated by the dotted lines in FIG. Note that the IC pin numbers used in FIG. 16 are assigned in FIG. Circuits external to pins other than ICs include semi-fixed resistors (2
0k), a portion Qc for externally adjusting each value of the transistors Qh, Qh ', the resistors r1, r2, r3 and the surrounding voltage dividing resistors.
And a voltage stabilizing color Vc. In the above embodiment, the whole is driven by the common power supply voltages + B and -B. However, as shown in FIG. 6, it is also possible to drive with different power supply voltages necessary and sufficient for each unit.

[0058]

As described above, according to the present invention, the driving voltage of the output stage element is reduced to a voltage corresponding to the input signal level by the main power supply path which is driven by switching. The applied voltage of the output stage element becomes small, and even if the idle current of the output stage element is set to a large value,
Here, the loss (heat generation) is reduced, and the efficiency can be improved. Moreover, since the main power supply path itself is driven by switching, the loss in the main power supply path is small. In addition, when the input power suddenly rises such that the main power supply path cannot follow due to the presence of the smoothing circuit, power is supplied from the auxiliary power supply path having better responsiveness than the main power supply path. It can follow the steep rise of the amplitude input and does not clip the output. Even if the auxiliary power supply path is driven by such a power supply voltage, power is mainly supplied from the main power supply path in normal times, so that the average current value flowing through the auxiliary power supply path can be small, and this auxiliary power supply Road losses are small. As a result, an amplifier circuit having high performance, high efficiency, and good follow-up performance even with a rapid rise of a large amplitude input is realized.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing one embodiment of the present invention.

FIG. 2 is a circuit diagram showing a conventional class A push-pull amplifier.

FIG. 3 is an operation characteristic diagram of the circuit of FIG. 2;

FIG. 4 is a circuit diagram of a current-current conversion circuit used in the present invention.

5 is an operation characteristic diagram of the circuit of FIG.

6 is a circuit diagram showing a conventional amplifier circuit using the circuit of FIG.

[7] of Figure 1 current I _1out, the I _2out, is a characteristic diagram of the I _RL.

8 shows transistors Qb and Qs in the circuit of FIG.
It is the figure which extracted and showed the part.

9 is an operation waveform diagram of the circuit of FIG.

FIG. 10 shows a hysteresis comparator 5 in FIG.
It is a circuit diagram which shows the specific example of 0.

11 is a block diagram conceptually showing a power supply path to transistors Qa and Qa 'in the circuit of FIG.

FIG. 12 is a circuit diagram showing a specific example of the circuit of FIG. 1;
An amplifier circuit may be configured together with the circuits 3 to 15.

FIG. 13 is a circuit diagram showing a specific example of the circuit of FIG. 1;
2. An amplifier circuit is formed together with the circuits shown in FIGS.

FIG. 14 is a circuit diagram showing a specific example of the circuit of FIG.
2. An amplifier circuit is formed together with the circuits shown in FIGS.

FIG. 15 is a circuit diagram showing a specific example of the circuit of FIG. 1;
2 to 14 constitute an amplifier circuit.

FIG. 16 is a circuit diagram in which FIG. 13 is partially integrated into an IC.

[Explanation of symbols]

Reference Signs List 3 current-current conversion circuit 6 first current mirror circuit 7 second current mirror circuit 8 voltage-current conversion circuit 9 voltage synthesis circuit 10 load 21 output circuit 42 main power supply path 44, 44 'power supply 48 auxiliary power supply path 49, 51 addition point, signal path (auxiliary power supply amount control means) Rs, 50 auxiliary current detection resistor, hysteresis comparator (switching control means) L1 smoothing coil (smoothing circuit) Qs switching transistor (switching element) Qb transistor (auxiliary power (Supply amount adjusting element) Tr1 first transistor Tr2 second transistor Tr3 third transistor Tr4 fourth transistor

──────────────────────────────────────────────────続 き Continuation of the front page (56) References JP-A-57-11508 (JP, A) JP-A-60-107905 (JP, A) JP-A-62-214707 (JP, A) JP-A-4- 372212 (JP, A) JP-A-5-63453 (JP, A) JP-A-5-67925 (JP, A) JP-A-5-67926 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H03F 1/02 H03F 1/32 H03F 3/181-3/32 H03G 11/00-11/08

Claims (1)

(57) [Claims] A voltage-current conversion circuit for converting an input signal to be amplified into a voltage-current, an emitter of a first transistor (NPN) and an emitter of a third transistor (PNP) are commonly connected, and a collector of the transistor is connected to its own collector. The emitters of the second transistor (NPN) and the fourth transistor (PNP), whose bases are connected in common, are connected in common, the bases of the first and second transistors are connected in common, and the third and third transistors are connected. The bases of the four transistors are connected in common, the output current of the voltage-current converter is supplied to the common emitter of the first and third transistors, and the constant current is supplied to the second and fourth transistors, respectively. A current-current conversion circuit configured to supply the current; a first current mirror circuit that outputs a current proportional to a collector current of the first transistor; A second current mirror circuit for outputting a current proportional to the collector current of the third transistor; a voltage signal corresponding to the output currents of the first and second current mirror circuits and the input signal to be amplified are synthesized. A voltage synthesizing circuit that amplifies the output voltage of the voltage synthesizing circuit; an output stage element that amplifies the output current of each of the first and second current mirror circuits; An output circuit that combines the output currents of the elements and supplies the load to a load; a main power supply path that supplies a driving power supply to the output stage element; and an on / off switching device that is inserted into the main power supply path to turn on and off the main power supply path A switching element; a smoothing circuit inserted into the main power supply path to smooth an output of the switching element and supply the output to the load; The switching element ratio of the ON period and the OFF period by variably switching,
A switching control unit that mainly supplies the power required by the load from the main power supply path; and a power supply path that supplies a driving power supply to the output stage element separately from the main power supply path. An auxiliary power supply path that can supply power following a change in the level of an input signal with high-speed response, and that is inserted into the auxiliary power supply path in a cascade-connected state with respect to the output-stage element, from the auxiliary power supply path. An amplifier circuit comprising: an auxiliary power supply amount adjusting element for adjusting the amount of power supply to the load.