JP3069342B2 - Fast wave resonant antenna with multilayer ground plane - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a fast wave resonant antenna having a multilayer ground plane, and more particularly to a surface mounting technology (SMT,
The present invention relates to a small-sized high-speed wave resonance antenna having a multi-layer ground plane attached by Surface Mounting Technology).

[0002]

2. Description of the Related Art With the spread of mobile phones, hidden antennas are receiving more and more attention. Hidden antennas are small,
It can be placed on radio frequency circuits with surface mounting technology. Therefore, the reliability is improved and the call quality is improved.

Conventionally, patch microstrip (patch microstrip)
microstrip) is used for hidden antennas. FIG.
As shown in FIG. 2, a ground plane 12, a dielectric substrate 11 formed on the ground plane 12, a patch 13 formed at the center of the upper surface of the dielectric substrate 11, and a And a feed line 14. Such a patch antenna (pat
ch antenna) is often used as a conventional active antenna.

FIG. 15 shows another type of patch antenna. The configuration is similar to the patch antenna of FIG. 14, but instead of the feed line 14, a feed line 15
Extend along the surface of the dielectric substrate 11 and form a via hole (via
hole) and extends down along the edge. With this configuration, a surface-mounted antenna can be created.

[0005]

FIG. 16 shows another type of patch antenna. The configuration is similar to the patch antenna of FIG. 14, but instead of the feed line 14, a probe or coaxial line 16 is used. Coaxial cable
16 needs to be connected to an external coaxial cable by a microwave connector, so it is difficult to connect to another microwave circuit in a surface mount type by the above configuration.

[0006] Previous studies result, the resonance frequency of the microstrip antenna (epsilon _r) inversely proportional to ^1/2 (epsilon _r is the dielectric constant). Due to this restriction condition, FIG.
The microstrip antennas 4 to 16 generally use a dielectric substrate having a relative dielectric constant of 20 or more in order to reduce the size. Further, according to the results of the conventional research, the limited size of the ground plane has a great effect on the microstrip antenna. Therefore, in order for the microstrip to operate properly, it is necessary that the ground area be larger than the area of the metal patch.

Further, by utilizing the resonance phenomenon of a dielectric substance, energy can be coupled to a medium resonator via a microstrip or a slot line, thereby forming a hidden antenna in a general integrated circuit. However, since the size is also inversely proportional to (ε _r ) ^1/2 , the antenna generally uses a dielectric material having a high dielectric constant.

Next, a simplified model of a monopole antenna in a mobile phone will be described. As shown in FIG. 17A, a single-pole antenna 42 is provided in a housing 41 of a mobile phone. The length of the monopole antenna 42 is a free space wavelength λ ₀
About a quarter of FIG. 17 (b) shows a simplified model of another type of antenna in a mobile phone. The total length of the spiral antenna 43 is close to the free-space wavelength λ ₀ , and is not suitable for an antenna of a hidden small mobile phone, like the monopole antenna 42.

Further, a monopole antenna 42 and a helical antenna
In 43, the housing is used as a ground plane, and the ground area is quite large. In a general design, the ground area is about 2λ ₀ ² (λ ₀ : wavelength of free space). However, as the size of the mobile phone is reduced, the ground area of the antenna is reduced, and the performance of the antenna is deteriorated.

[0010]

SUMMARY OF THE INVENTION The present invention has been made in view of the above problems and provides a small antenna having a special design. The antenna includes a suspended microstrip in a bound mode and a fast-wave leakage mode.
leaky mode). Since the mode current and the transverse electric field (magnetic field) in both modes are similar near the microstrip, a fast wave resonant antenna having a multilayer ground plane can be designed based on the resonance phenomenon due to the fast wave leakage mode. .

[0011] The antenna includes a high-speed wave vibrator and a multilayer grounding device. The high-speed wave vibrator includes a cubic dielectric and a microstrip wound on the surface of the cubic dielectric. The microstrip is wound within the narrow surface area of the dielectric and wrapped around the surface of the dielectric to produce the desired radiated field pattern. Furthermore, one end of the microstrip is
The other end is open as a signal supply end.

The multilayer grounding device is provided below the high-speed wave vibrator and mainly includes a plurality of parallel layers and a plurality of via holes. In addition, all the inner surfaces of the grooves formed by the plurality of parallel layers, all the inner surfaces of the via holes, and all the other outer surfaces can all serve as metal ground planes.

The dimensions of the antenna are such that the microstrip in the high-speed vibrator fits within the narrow surface area of the dielectric, and the configuration of the multilayer grounding device can provide a substantial ground area in a limited space. Can be sufficiently reduced. Furthermore, the antenna is mounted directly on the printed circuit board by surface mounting technology. Note that the antenna according to the present invention does not need to use a dielectric material having a high dielectric constant, and is sufficiently applied to a dielectric material having a relative dielectric constant of 2 to 5.

[0014]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1A is a sectional view of the suspended microstrip. Below are the parameters of the microstrip: x ₁ = 300 mm, b = 421.6 m
m, w = 1.6 mm, h = 0.762 mm, ε _r1 = 1.0, ε _r2 = 2.1
, Ε _r3 = 1.0. FIG. 1 (b) shows two modes present in a suspended microstrip under the condition that the metal conductor in FIG. 1 (a) has infinite conductivity.
That is, γ _m = β _m −j · 0 = β _m and γ ₁ = β ₁ −j · α ₁ . γ _m and γ ₁ indicate the propagation constants in the bound mode and the leaky mode, respectively, β _m and β ₁ indicate the phase constants in the bound mode and the leaky mode, respectively,
₁ indicates a damping constant in the leaky mode.

The horizontal and vertical current distributions in both modes are shown in FIGS. 2A and 2B.
As shown in the figure, both modes are very similar in the horizontal and vertical directions of the microstrip. Therefore,
Once one mode is excited, the other mode is also excited.

Further, the lateral electric field in both modes is described.
The (magnetic field) is also very similar near the microstrip. FIG. 3 shows the transverse electric field in leaky mode, with the suspended microstrip at several different altitudes. Below are the parameters for that microstrip: (1) χ _b1
= 299 mm, χ _t1 = 303 mm; (2) χ _b2 = 408 mm, χ _t2 = 41
2 mm; (3) χ _b3 = 677 mm, χ _t3 = 681 mm, y ₁ = 208.3 m
m, y ₂ = 213.3 mm. As shown in FIG. 3, the attenuation constant in the leaky mode is not zero.

When analyzed in detail, the two modes are mutually coupled. That is, the integral of the cross-sectional area of the waveguide, that is, neither Equation 1 nor Equation 2 below is non-zero. Here, Equation 3 shows the electric field of the cross section in the bound mode and the leak mode, respectively, and Equation 4 shows the magnetic field of the cross section in the bound mode and the leak mode, respectively. That is, when the conventional bound mode is excited, when the bound mode propagates, a part of the energy is converted to the leak mode. This leak mode transfers energy into the air. On the other hand, when propagating, part of the energy in the leak mode is also converted to the bound mode. What
In the following formulas (1) to (4), (m) indicates a binding mode.
(1) indicates the leak mode. Energy transforms into bound mode.

(Equation 1)

(Equation 2)

(Equation 3)

(Equation 4)

FIG. 4 shows an ideal configuration of a suspended microstrip consisting of a metal line 81, a dielectric substrate 82, an air region 83 and a ground plane 84. The space above the metal wire 81 is also filled with air.

The antenna according to the present invention is designed based on the above-described principle and the ideal configuration of the suspended microstrip. The present invention provides a fast wave vibrator,
A multi-layer grounding device comprising a multi-layer ground plane and via holes.

FIG. 5 shows a preferred embodiment of the antenna according to the present invention. A indicates a fast wave vibrator, and B indicates
1 shows a multilayer grounding device. In order to clarify the circuit of the fast wave vibrator, the dielectric was extracted from the circuit. Further, in the following description, the length, width, and height of the antenna correspond to the X axis, Y axis, and Z axis of the three-dimensional space, respectively. FIG.
6 is a partially enlarged view of FIG.

The air regions C of Figure 5, it corresponds to the air region 83 located between the dielectric substrate 82 and the ground plane 84 in FIG. 4.

The high-speed wave vibrator A shown in FIG. 6 comprises a cubic dielectric and a helical metal microstrip composed of microstrip members A1, A2, A3 and the like wound on the surface of the cubic dielectric. Is done. Due to resonance, one end of the spiral metal microstrip is open.
Is a Hotan. One end of the microstrip A0 connected to the microstrip member A1 is an input / output end of an antenna signal .

The multilayer grounding device B shown in FIG. 6 is provided below the cubic dielectric (lower left in FIG. 6) of the high-speed wave vibrator A and mainly includes a plurality of parallel layers B1 to B9. In order to further increase the ground area, a plurality of via holes B10 to B17 are provided below the parallel layers B1 to B9 (lower left in FIG. 6) under the condition of considering the structural strength. Further, the inner surfaces of the grooves 1 to 4 formed by the parallel layers B1 to B9, the inner surfaces of the via holes B10 to B17, and all other outer surfaces are all metal ground planes .

Further, the signal input / output terminals of the microstrip A0 in FIG. 6 extend along the surface of the dielectric to the signal input / output terminals 51, and the ground terminals 55, 57 of the multilayer grounding device B.
Along with input of coplanar waveguide /
Form the output end.

FIG. 8 shows an example of an antenna 101 according to the present invention mounted on a circumscribed circuit board 103. In FIG.
(b) As shown in the figure, the input / output terminal 5 of the coplanar waveguide of the antenna 101 according to the present invention is mounted on the circumscribed circuit board 103.
At the positions corresponding to 1, 55, 57, input / output ends 61, 65, 67 of the coplanar waveguide are provided. Among them, 61 is a signal input / output terminal, and 65 and 67 are ground terminals. 51, 55, 57
Are connected to 61, 65, 67 respectively by surface mounting technology. Further, the multilayer grounding apparatus B is connected to the ground plane 105 of the external circuit board 103 via the plurality of via holes 69 and the metal ground plane 70 of the external circuit board 103 by a surface mounting technique.

FIG. 7 is a schematic diagram of FIG. With reference to FIG. 7, design parameters according to the above embodiment are shown below. (1) The width w and interval s of the microstrip are each 0.39
× 10 ⁻³ λ ₀ and 0.17 × 10 ⁻³ λ ₀ . (2) The length of the dielectric 10 is approximately 0.039 λ ₀ , where d = 0.0
32 λ ₀ and g = 7.0 × 10 ⁻³ λ ₀ . (3) The width f of the dielectric 10 is 4.3 × 10 ⁻³ λ ₀ , and the height e is 1.47 × 1
0 ⁻³ λ ₀ . (4) Relative permittivity ε _r = 3.25. (5) Number of spiral microstrip zones N = 57.

According to the above parameters, the volume of the antenna according to the present invention is about 0.25 × 10 ⁻⁶ λ ₀ ³ , and the average length is about 0.
63 × 10 ⁻² λ ₀ . Therefore, the small antenna can be integrated.

Further, the length of the spiral microstrip is about 0.667 λ ₀ . 5.8 10 ^-3 λ ₀ × 57 × 2 +0.17 × 10 ^-3 λ ₀ × 57 = 0.667 λ ₀

Further, the total area of the spiral microstrip is approximately 260 × 10 ^-6 λ ₀ ^2. 0.667 λ ₀ × 3.9 × 10 ^-4 λ ₀ = 260 × 10 ^-6 λ ₀ ²

When a resonance phenomenon occurs in the microstrip, the current distribution in the microstrip is similar to a change in the arc of the cosine function from 0 to π / 2, and will be described in detail later. Since the area occupied by a quarter cycle of the cosine function is 2 / π, the average effective area of the spiral microstrip is 166.
× 10 ^-6 λ ₀ ² . That is, the charge has an effective area of 166 × 10 ^-6 λ
Homogeneously distributed in ₀ ² of the microstrip. 260 × 10 ^-6 λ ₀ ² × (2 / π) = 166 × 10 ^-6 λ ₀ ²

The grounding area of the multilayer grounding device is as follows:
It is estimated to be about 90.6 × 10 ^-6 λ ₀ ^2. When resonating, the positive charge Q
Flows through the signal input / output end 51 of FIG. 6 and enters the spiral microstrip, filling its metal surface. At the same time, a portion of the negative charge -Q flows through the ground ends 55, 57 and fills all the metal surfaces in the multilayer ground device. Other negative charges flow to the grounded ends 65, 67 of the circumscribed circuit board and the ground plane 105. This fills all metal surfaces in the multilayer grounding device. Therefore, at the time of resonance, the spiral microstrip, the multilayer grounding device, and the portion near the grounding end can keep the charge balance together. Therefore,
Although the volume of the antenna according to the present invention is much smaller than the volume of the antenna in the conventional mobile phone,
The contact area is sufficient.

Further, the antenna according to the present invention does not need to use a dielectric material having a high dielectric constant. The relative permittivity ε _r is 2
Only 5 dielectric materials are required.

The important function of the fast wave leakage mode in the antenna according to the present invention will be explained by the following calculations and estimations.

Microwave circuit theory (microwave circuit t)
According to heory), the open end of a transmission line supporting a single mode has a frequency of ４ × λ _g (λ _g : single mode) when there is no fringing field effect (a simple open circuit). If it is an odd multiple of the frequency corresponding to (transmission line wavelength),
A resonance circuit is formed. First resonance frequency (resonant freq
The resonance equation corresponding to (uency) is as follows: l = 1/4 × λ ₀ / β ^* = 1/4 × λ _g (1) where l denotes the length of the microstrip and β ^* is normalized Shows the phase constant. Also, β ^* = β / k ₀ ,
k ₀ = 2π / λ ₀ , β = 2π / λ _g , and k ₀ is a free space wave number.

The equivalent circuit of the antenna according to the present invention is, as shown in FIG.
32, a grounding system 33, and a power supply 34.
Based on the microwave circuit theory, 9 When representing the resonant circuit corresponding to the first resonance frequency, the length of the suspended microstrip 32 becomes 1/4 × λ _g.

Based on the above antenna parameters,
First resonance frequency 260MHz by three-dimensional full-wave electromagnetic field theory
Is obtained. On the other hand, the scattering parameter of the antenna created by the above parameters (scattering parameter)
By measuring the S _11, the Smith chart of FIG. 10,
And the corresponding reflection coefficient diagram at the input end of FIG.

FIG. 10 shows a vector analyzer with a frequency of 240 MHz to 30 MHz.
The scan results up to 0 MHz are shown. Refer to the Smith chart,
Start from a point near the rightmost end corresponding to an open circuit and move clockwise to a point near the leftmost end corresponding to a short circuit. It stops at the point corresponding to 300 MHz, which is in the upper right corner of the Smith chart. According to the above analysis, the frequency corresponding to the point closest to the short circuit (ie, the first resonance frequency) is 259 MHz, corresponding to a phase of 180 °. The difference between this value and the theoretical value is only 1 MHz.

Further, the first resonance frequency can be verified in FIG. As shown in the figure, when the resonance, S ₁₁ is a minimum (about -2.8DB) when 259 MHz, corresponding to a phase 180 °. When the resonance circuit shown in FIG. 9 resonates, the reflection coefficient at the input terminal is always negative. That is, its phase is 180 °. in addition,
Since high-speed wave leaky mode is attenuated, the absolute value of S ₁₁ is less than 1 (<0dB). Therefore, corresponding to the first resonance frequency, the length 1 = 0.66 of the microstrip according to the present invention.
7 By substituting λ ₀ into (1), β ^* becomes 0.375. The β ^*
The phase velocity of the leaky mode corresponding to is: c / β ^* = 2.66c (2) where c represents the speed of light. The leakage mode has a phase velocity c /
Since β ^* is 2.66 times faster than the speed of light, it can be said that it is a fast wave.

Further, according to the three-dimensional full-wave electromagnetic field theory, when the resonance frequency is 260 MHz, the current distribution on one side of the microstrip and on the other side thereof can be calculated. FIGS. 12A and 12B show the results. FIG.
As shown in FIG. 2 (a) and FIG. 12 (b), when the resonance frequency is 260 MHz, the current in the microstrip is
A maximum occurs at the input end and gradually decreases towards the open end, at which point the current goes to zero. That is, the current in the microstrip approximates a cosine function from radian 0 to π / 2. As described above, such resonance can be achieved only in the leakage mode.

In summary, the antenna according to the present invention transmits signals mainly in the fast wave leakage mode.

Further, according to the conventional full-wave integration equation, when the antenna according to the present invention has a resonance frequency of 260 MHz,
FIG. 13 shows the measurement results of the radiation pattern on the −Z plane. Among them, q indicates an angle between a straight line drawn from one point on the YZ plane to the origin and the Z axis. As shown in FIG. 15, this radiation pattern is very similar to that of a monopole antenna on an infinite ground plane.

Although the specific embodiments according to the present invention have been described above, the present invention is not limited to the embodiments. For example, in the antenna of FIG. 5, between the microstrip of the high-speed wave vibrator A and the outer surface of the multilayer grounding device B, the air corresponding to the air region 83 of FIG.
The hollow region C need not be provided. That is, an antenna in which the high-speed wave vibrator A and the multilayer grounding device B are directly connected by a dielectric is also conceivable.

Further, the shape of the microstrip in the high-speed wave vibrator is not limited to the spiral type, but can be changed to another shape according to a desired radiation pattern. For example, a plurality of parallel annular microstrips is also conceivable. The design method is similar to the specific embodiment.

Further, the antenna according to the present invention is not limited to the coplanar waveguide type, but may be a direct input / output type using a feed line. At this time, an input / output terminal is formed at a position corresponding to the input / output terminal of the antenna on the circumscribed circuit board. The input / output terminals of the microstrip in the high-speed wave vibrator are connected to the signal input / output terminals of the circumscribed circuit board by surface mounting technology.

[0045]

It will be explained in advance that various changes can be made without departing from the spirit of the invention and within the scope of the claims of the invention.

[Brief description of the drawings]

1A is a sectional view showing a suspended microstrip, and FIG. 1B is a graph showing propagation constants of a suspended microstrip in a bound mode and a leakage mode.

FIG. 2A is a lateral current distribution diagram in a bound mode and a leak mode, and FIG. 2B is a vertical current distribution diagram in a bound mode and a leak mode.

FIG. 3 is an illustration showing the transverse electric field in a leaky mode at several different altitudes for a suspended microstrip.

FIG. 4 is a perspective sectional view showing an ideal configuration of a suspended microstrip.

FIG. 5 is a perspective sectional view of a preferred embodiment of a small high-speed wave vibration antenna having a multilayer ground plane.

FIG. 6 is a partially enlarged view of FIG. 5;

FIG. 7 is a schematic diagram of FIG. 5;

8A is a perspective view showing a state in which the antenna according to one embodiment of the present invention is mounted on a circumscribed circuit board, and FIG. It is explanatory drawing of some circuits of a circuit board.

FIG. 9 is an equivalent circuit diagram of an antenna according to one embodiment of the present invention.

FIG. 10 is a graph showing measurement results of a one-port Smith chart according to an embodiment of the present invention.

FIG. 11 is a graph showing a measurement result of a single-port scattering parameter according to an embodiment of the present invention.

12A is a current distribution diagram on one side of a microstrip at a resonance frequency of 260 MHz in an antenna according to an embodiment of the present invention, and FIG. 12B is a diagram on the opposite side of FIG. It is a current distribution diagram.

FIG. 13 is a diagram illustrating a radiation pattern on a YZ plane when the antenna according to the embodiment of the present invention has a resonance frequency of 260 MHz.

FIG. 14 is a perspective view showing an example of a conventional patch antenna.

FIG. 15 is a perspective view showing another type of conventional patch antenna for introducing a signal through a via hole.

FIG. 16 is a perspective view showing yet another type of conventional patch antenna for introducing a signal by a probe or a coaxial cable.

17A is a perspective view showing a simplified model of a monopole antenna in a conventional mobile phone, and FIG. 17B is a perspective view showing a simplified model of a spiral antenna in a conventional mobile phone.

[Explanation of symbols]

 Reference Signs List 11 dielectric substrate 12 ground plane 13 metal patch 14 feed line 15 feed line connected to via hole 16 probe (or coaxial line) 41 housing 42 monopole antenna 43 spiral antenna A high-speed wave vibrator A0 to A3 microstrip member B multilayer ground Device B1 to B9 Parallel layer B10 to B17 Via hole 1-4 Groove C Hollow area 51 Input / output end of antenna signal 55, 57 Ground end of antenna 61 Signal input / output end of external circuit board 65, 67 External circuit board Ground end 69 via hole 70 metal ground plane 81 metal wire 82 dielectric substrate 83 air area 84 ground plane 101 antenna according to the present invention 103 external circuit board 105 ground plane of external circuit board

Continuation of the front page (58) Fields surveyed (Int. Cl. ⁷ , DB name) H01Q 11/08 5J045 JICST file (JOIS)

Claims (4)

(57) [Claims] 1. A cubic dielectric and said cubic dielectric
Wrapped around the body surface, the first end is the signal input / output end
A microstrip whose second end is open,
A high-speed wave vibrator provided, provided on a first end side of the high-speed wave vibrator, and parallel to each other.
A plurality of parallel layers and a plurality of via holes,
All the inner surfaces of the grooves by the plurality of parallel layers, the plurality
All internal surfaces of the via hole, and all other
The outer surface is a multi-layer grounding device that can all act as a metal ground plane
Fast wave resonator antenna having a multilayer ground plane, characterized in that it comprises a and. 2. The method according to claim 1, wherein said signal input / output system is a coplanar system.
2. The fast wave resonant antenna having a multilayer ground plane according to claim 1, wherein the ground is a line , and a ground end of a coplanar line is formed in the multilayer grounding device. 3. The high-speed wave resonance antenna having a multilayer ground plane according to claim 1, wherein the microstrip in the high-speed wave vibrator is wound around a surface of a dielectric and forms a spiral shape. 4. A dielectric under a parallel layer in the multilayer grounding device, wherein a plurality of via holes are provided under the parallel layer under a condition considering structural strength. A high-speed wave resonance antenna having the multilayer ground plane according to claim 1.