JP2703711B2 - Control method of PWM inverter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION A voltage source inverter to which the control method of the present invention is applied is not affected by the operation time of a control system.
Since the filter capacitor current can follow the command value, high-performance control characteristics can be obtained. Therefore, the present invention can be used in fields such as industrial machines and home appliances using the voltage-type inverter.

[0002]

2. Description of the Related Art Various control methods have been devised for voltage-type PWM inverters according to their applications. FIG. 2 shows an example of a conventional PWM voltage source inverter control method, and FIG. 3 is a waveform diagram thereof.

As shown in FIG. 3, voltage command values v _u ^* , v
_v ^*, _v is compared with the _w ^* and the triangular wave v _c, make a signal of the base drive circuit through the hysteresis comparator.
The output voltage amplitude value can be changed by adjusting the amplitude value of the voltage command value. At the top of FIG. 3, a triangular wave v _c having a height V _c, the phase voltage command values v _u ^*, v
_v ^*, _v shows a waveform diagram of the status of the comparison with the _w ^*, v _uN, v
_vN, v _wN each represents a voltage waveform between the neutral point of each _{phase, v uv, v vw, v} wu shows the voltage waveforms of respective phases, E _d is the voltage of the DC power source. The inverters are switching elements 2 to 3 in which diodes are connected in anti-parallel, respectively.
7 (for example, transistors) in a three-phase bridge connection, and two series-connected E _d / 2 each having a virtual neutral point.
The DC voltage E _d to power the DC input terminals by the DC power supply 1, 1 'having a voltage, a filter reactor 8, 9 and 10 connected in series with each phase AC output terminal, a filter capacitor 11, which has Y connected, 12 and 13 are connected in parallel to supply AC output to the load.

[0004]

In the three-phase voltage source inverter shown in FIG. 2, the output voltage is determined by the command value and the triangular wave, but a considerable load fluctuation occurs with respect to the fluctuation of the DC input voltage and the load fluctuation. Introducing the feedback control can reduce the error, but may cause new problems such as stability. Also, the ripple of the DC input voltage directly affects the output voltage. Further, as can be seen from FIG. 2, since the three phases are controlled independently, it is difficult to reduce the ripple of the output voltage without increasing the switching frequency.

On the other hand, a method of directly calculating an output voltage pattern of an inverter as a voltage vector using an instantaneous space vector has been devised. By performing so-called instantaneous control, the control characteristics of the inverter are improved. However, when performing instantaneous control, complicated operations are required, so that a microprocessor or digital signal processor (D
SP) is often used. In this case, the detection value is always delayed by the sampling time due to the influence of the calculation time, which causes deterioration in control performance.

[0006]

SUMMARY OF THE INVENTION In a system composed of a PWM inverter and an AC LC filter, a voltage pattern of the inverter is output so that a filter capacitor current follows a command value, and a filter capacitor voltage is controlled. The filter capacitor current is predicted based on information such as the output amount detected at each sampling time and the current voltage pattern. The effect of the operation time of the digital control system is eliminated by performing so-called predictive control. In the case of a periodic non-linear load, the problem can be solved by recording sampling data for one cycle of the load current and performing repetitive control.

The details of the principle of the output voltage control method based on the instantaneous space voltage vector are described in
The method is disclosed in Japanese Patent Application No. 5-26646 filed on May 16, entitled "PWM inverter control method". Here, the concept is used below.

FIG. 1 is a diagram showing the principle of the present invention.
1 shows a configuration of a WM inverter system, and (b) shows a control principle thereof. The same reference numerals as those in FIG. 2 indicate the same parts, and the control circuit controls the base drive circuit to drive the inverter. In FIG. 1B, which is a principle diagram, θ _S is the angle at which the current vector should rotate in one sampling time, and ω is the rotation angle frequency.

Control Principle of Output Voltage The output amount of the PWM inverter is converted into three-phase / two-phase, and the current / voltage can be handled as a base. Also, as is generally known, a three-phase voltage source inverter has eight voltage vectors including two zero voltage vectors. The output voltage / current vector equation of FIG.

(Equation 1) It is obtained as follows. v is an arbitrary output voltage vector, v _c is a capacitor voltage vector, i, _ic and i _L are a filter reactor current vector, a capacitor current vector and a load current vector, respectively.

When the three-phase sinusoidal phase output voltage is considered on two-phase coordinates, a circular locus with a constant speed is drawn. Therefore, the fundamental wave current of the load capacitor must also have a circular locus.

Now, the sampling period is defined as follows.

T _s = T / k (2) where T is the fundamental wave period of the inverter, and k is the number of samplings in one period. That is, the sampling time T
_The current vector may move along the circle of the command value in _S.

Next, the switching area corresponding to the six non-zero voltage vectors is divided into six spatial area sectors every 60 degrees as shown in FIG. The output voltage vector corresponding to each sector is defined in the following table.

[Table 1]

The current can follow the command value within the sampling time by using the two non-zero voltage vectors and the two zero voltage vectors corresponding to each sector described above.

According to equation (1),

(Equation 3) Here, t _o + t _i + t _j + t ₇ = T _s , i, j (i, j
= 1 to 6) are the numbers of the non-zero voltage vectors. Δi _L
Is the change amount of the load current, and may be ignored if there is no sudden change in load.

For example, in the sector I, the voltage vector is expressed as V ₇ → V ₆ → V ₁ → V ₀ or V ₀ → V ₁ → V ₆ → V ₇ as shown in FIG. And the capacitor current can follow the command value within one sampling time.

[0016] Selection of the zero voltage vector, the switching of V _1, V _3, V ₅ from the viewpoint of reducing the number of switching operations is V
Use _0, V _2, switching of V _4, V ₆ use the V _7.

Capacitor Current Prediction Control Since the calculation time is required to calculate the voltage vector to be output and its time width, the actual detection value is always delayed by the sampling time. Particularly, since the filter capacitor current changes at a high frequency, a large error occurs due to the effect of the operation time. Therefore, the current at one sampling destination is predicted from the current detection value and the information on the voltage vector.

According to equation (1), the predicted current value is

(Equation 4) It is obtained as follows. Expression (4) with a suffix of 0 represents the current value.

Next, a method of determining the time width of the output voltage vector of the inverter will be described.

The current command value is developed on the α-β2 phase coordinates,
It is given by the following equation.

(Equation 5) The subscripts a and b in the above equation represent components on the α-β coordinates.
The current amplitude command value I _C ^* is obtained by using the voltage amplitude command value V _C ^* .

(Equation 6)

When equation (4) is expanded on α-β coordinates, the following equation is obtained.

(Equation 7) Here, V is the amplitude value of the output voltage vector, and C _a ,
C _b is a constant. According to the above formula,

(Equation 8) Is obtained. It is necessary to determine t ₀ and t ₇ so as to minimize the current ripple. Here, for simplicity,

(Equation 9) It can be.

In a transient state or the like, t _i or t _j may become negative. In such a case, a voltage vector in the opposite direction may be output. Furthermore, in the case of a sudden load change, it may not be possible to follow within one sampling time.

(Equation 10) It is possible that In such cases it may be selected voltage vector as the current error _{^{(Δi ca 2 + Δi cb 2}} ) 1/2 is minimized.

The current error vector is

[Equation 11] Becomes

(Equation 12) The t _{i and} t _j are obtained by solving the equations (11) and (12) so that the error current vector is minimized.

(Equation 13) Therefore, the output voltage vectors may be output at predetermined times.

According to the equation (4), since the calculation formula for predicting the filter capacitor current includes the change in the load current, even when the load current changes rapidly, the calculation time affects the calculation. The calculation accuracy becomes worse. Thus, for a periodic non-linear load such as a rectifier load, the load current can be recorded for one cycle, and the capacitor current can be accurately predicted using these data in the next cycle.

[0025]

In the digital control system of the PWM inverter, the output characteristics of the PWM inverter can be improved by predicting the capacitor current at one sampling destination from the current detection amount and the output voltage pattern and making the capacitor current follow the command value. .

[0026]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The principle described above is applied to CVCF (constant voltage and constant frequency).
An example applied to an inverter will be specifically described. The configuration of the circuit is the same as that shown in FIG. The load is a Y-connection three-phase resistance load or a three-phase rectifier load.

As shown in FIG. 1, the digital control circuit
It comprises an SP (Digital Signal Processor) and an A / D converter. Here, three voltage inputs (v _C ), two current inputs (i _C ), and three current inputs (i _L ) A
2 shows a / D converter. The A / D converter is controlled by the DSP, and inputs a three-phase load current i _L , a capacitor current i _C , and a capacitor voltage v _C as digital quantities to the DSP every sampling time in addition to the DC input voltage V _dc . The DSP executes a control algorithm based on the aforementioned control principle at high speed.

In the embodiment, the fundamental frequency of the output voltage is 60 Hz.
Then, the sampling frequency is set to 10.8 kHz, and in that case, the switch frequency in the steady state is 5.4 kHz. Good results were obtained for both 100% sudden load change and rectifier load.

[0029]

According to the present invention, the output voltage is calculated by calculating the filter capacitor current one sampling ahead by using the current sampling detection value and the voltage output vector, and making the predicted value follow the command value. Can be controlled without being affected by Therefore, the waveform of the output voltage is greatly improved, and the control accuracy can be significantly improved.

[Brief description of the drawings]

1A and 1B are diagrams illustrating the principle of the present invention, in which FIG. 1A illustrates a configuration of a PWM inverter, and FIG. 1B illustrates a control principle thereof.

FIG. 2 is a principle diagram illustrating an example of a conventional PWM inverter control method.

3 shows a waveform diagram of each part in FIG. 2. FIG.

FIG. 4 is a diagram showing a spatial area sector corresponding to a voltage vector.

FIG. 5 is a diagram showing an output order of voltage vectors in a region I.

[Explanation of symbols]

1 DC power source 2 to 7 switching device 8-10 filter reactor 11 to 13 filter capacitor v _uN, v _vN, v voltage waveform v _uv between the neutral point of _wN phase, v _vw, v _wu voltage between phases waveform ^{_{v u *, v v *,}} v w * voltage command value v crest of the waveform V _c triangular wave _c triangular wave E _d is the voltage V ₀ which DC power supply, V ₇ zero voltage vector V ₁ ~V ₆ nonzero voltage vector v Arbitrary output voltage vector v _c Capacitor voltage vector i Filter reactor current vector i _c Capacitor current vector i _L Load current vector T _S sampling time T Inverter fundamental wave period k Number of samplings per period I _C ^* Current amplitude command value V _C ^* Command value of voltage amplitude C _a , C _b constant V _dc DC input voltage Δi _L Load current change

Claims (1)

(57) [Claims] When digital control is performed in a system composed of a three-phase voltage source inverter and an AC LC filter, an output voltage and a current are detected for each sampling time, and a current detection value and a PWM output pattern are compared. The filter capacitor current one sampling ahead is predicted from the information, and the filter capacitor is determined within the sampling time determined by using the optimum two non-zero voltage vectors and the two zero voltage vectors among the eight output voltage vectors of the inverter. A control method of controlling the output voltage (the voltage of the filter capacitor) without affecting the calculation time by causing the predicted value current to follow the command value current.