JP2024022931A - Driving device of permanent magnet synchronous motor of medical apparatus for dental use and control method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a PM motor that realizes position sensorless control capable of estimating a position at a low-speed range regardless of a saliency structure of a rotor.

SOLUTION: A driving device of a permanent magnet synchronous motor of a medical apparatus for dental use comprises an inverter that drives the permanent magnet synchronous motor, detection means that detects a current flowing through the permanent magnet synchronous motor, and control means that controls the permanent magnet synchronous motor via the inverter using a current value detected by the detection means. The control means performs estimation calculation of a position of a rotor of the permanent magnet synchronous motor based on a current generated by applying a second polyphase alternate current having a higher frequency than a first polyphase alternate current for rotationally driving the permanent magnet synchronous motor to the permanent magnet synchronous motor at a predetermined phase interval.

SELECTED DRAWING: Figure 3

COPYRIGHT: (C)2024,JPO&INPIT

Description

The present invention relates to a drive device and a control method for a permanent magnet synchronous motor of a dental medical device.

Permanent magnet synchronous motors (hereinafter also referred to as PM (Permanent Magnet) motors) are widely used in industries, home appliances, medical equipment, electric vehicles, railways, and the like. In a PM motor, in principle, it is necessary to control the current phase for rotational drive based on the position angle of the rotor, so a configuration in which a position sensor for the rotor is provided is used.

On the other hand, position sensorless control, which does not directly detect the rotor position, has also been put into practical use. Most position sensorless controls utilize the induced voltage generated internally as the PM motor rotates. For example, in applications such as railway vehicles that require high torque from start-up, position sensorless control using a rotor's salient pole structure has been studied and has already been put into practical use. By realizing position sensorless control, there is no need to use a precise position sensor, making it possible to drive the motor in harsh environments. Also, by eliminating the position sensor, the entire device can be made smaller and the risk of sensor failure can be reduced. There are also benefits such as being able to avoid this.

For example, Patent Document 1 describes a position sensorless method that utilizes the magnetic salient polarity of a rotor of a PM motor. In the method of Patent Document 1, a harmonic voltage for estimating the position of the rotor is continuously applied to the estimated phase axis of the rotor, and the estimated position and the actual rotor are determined from the harmonic current generated on the axis orthogonal to the harmonic voltage. Sensorless drive is achieved by calculating the error with the phase and correcting that error.

Patent Document 2 utilizes the magnetic saliency of the rotor of a PM motor, and uses a pulsed voltage generated by an inverter as a harmonic voltage for estimating the position of the rotor. Patent Document 3 utilizes the magnetic saliency of the rotor of a PM motor, and estimates the position of the rotor using the current change rate caused by applied harmonics.

Japanese Patent Application Publication No. 7-245981 Japanese Patent Application Publication No. 8-205578 Japanese Patent Application Publication No. 2002-78391

The methods of Patent Documents 1 to 3 require the rotor to have magnetic saliency, and are generally applicable only to rotors with an embedded magnet structure in which the magnetic saliency is significant. For example, many small and inexpensive PM motors have a surface magnet structure in which inexpensive magnets are attached to the rotor surface. Drive cannot be applied.

For example, in Patent Documents 1 and 3, harmonics are applied on the estimated phase axis of the rotor, and as the estimated axis rotates (i.e., the rotation of the rotor), the harmonic application axis also rotates at the same time. do. In this case, the generated harmonic current is not only affected by the saliency of the rotor, but is also strongly influenced by the magnetic circuit of the stator. As a result, in a motor using a rotor with less saliency, it becomes difficult to estimate the position of the rotor due to changes in the magnetic circuit on the stator side. When attempting to perform position estimation under such circumstances, it becomes necessary to increase the applied harmonic voltage, resulting in an increase in harmonic loss.

Further, in Patent Document 2, harmonics generated by switching operations of an inverter are used as harmonics for position estimation. The harmonics generated by the inverter vary greatly depending on the phase of the voltage output by the inverter and the modulation rate, and are not uniquely determined. Therefore, as in Patent Documents 1 and 3, depending on the conditions, it is affected by the magnetic circuit on the stator side, making it difficult to estimate the rotor position.

For the above-mentioned reasons, the methods of Patent Documents 1 to 3 do not allow the rotation of the rotor in a PM motor that has a structure with little magnetic saliency or a structure with spatial harmonics in the stator core (for example, a concentrated winding structure). Estimating the position was difficult, and position sensorless control was not possible. In particular, in a PM motor that does not have a salient pole structure, the above method cannot be used for position estimation in a low speed range, and in such a case, it is necessary to separately prepare a position sensor.

For example, dental medical equipment is generally intended for delicate work inside the oral cavity. In devices equipped with such PM motors, there is a demand for further size reduction of the entire device. In a dental medical device such as that shown in FIG. 15, an operator performs treatment using a handpiece connected to a PM motor, but from the viewpoint of operability, it is desirable that the PM motor be small. In order to downsize the PM motor, the salient pole structure may not be used. Further, due to the use of dental medical equipment, it is necessary to frequently change the acceleration/deceleration of the rotational speed in a wide range from stop to low speed to high speed. Regarding the motor control of a device having such characteristics, rotation control is performed in a low speed range by detecting the rotor position using a sensor for the above-mentioned reasons. On the other hand, in high-speed ranges, the detection error of the sensor increases, so rotation control is performed by estimating the rotor position by detecting the induced voltage without using a sensor. Such a configuration still requires a position sensor.

In view of the above problems, it is an object of the present invention to realize position sensorless control in a PM motor that allows position estimation in a low speed range, regardless of the structure related to the saliency of the rotor.

In order to solve the above problems, the present invention has the following configuration. That is, a drive device for a permanent magnet synchronous motor of a dental medical device,
an inverter that drives the permanent magnet synchronous motor;
detection means for detecting a current flowing through the permanent magnet synchronous motor;
Control means for controlling the permanent magnet synchronous motor via the inverter using the current value detected by the detection means;
has
The control means applies a second multiphase alternating current having a higher frequency than the first multiphase alternating current to the permanent magnet synchronous motor at a predetermined phase interval for rotationally driving the permanent magnet synchronous motor. Estimating the position of the rotor of the permanent magnet synchronous motor is performed based on the current generated.

Further, another embodiment of the present invention has the following configuration. That is, a method for controlling a permanent magnet synchronous motor of a dental medical device, comprising: an inverter that drives a permanent magnet synchronous motor; and a detection means that detects a current flowing through the permanent magnet synchronous motor.
a control step of controlling the permanent magnet synchronous motor via the inverter using the current value detected by the detection means;
In the control step, applying a second polyphase AC having a higher frequency than the first polyphase AC for rotationally driving the permanent magnet synchronous motor to the permanent magnet synchronous motor at a predetermined phase interval. Estimating the position of the rotor of the permanent magnet synchronous motor is performed based on the current generated.

According to the present invention, in a PM motor, regardless of the structure related to the saliency of the rotor, it is possible to perform position sensorless control that allows position estimation in a low speed range.

FIG. 3 is a diagram for explaining a configuration example of a PM motor and its saliency. FIG. 3 is a diagram for explaining a rotor position detection method using a harmonic superimposition method. FIG. 1 is a diagram showing an example of a circuit configuration according to an embodiment of the present invention. FIG. 2 is a diagram showing an example of a circuit configuration of a position estimating section according to an embodiment of the present invention. FIG. 2 is a diagram showing an example of a circuit configuration according to a conventional method. FIG. 3 is a graph diagram for explaining the phase according to an embodiment of the present invention. FIG. 3 is a diagram showing an example of detection results according to an embodiment of the present invention. FIG. 3 is a diagram showing an example of position estimation according to an embodiment of the present invention. FIG. 3 is a diagram showing an example of position estimation according to an embodiment of the present invention. FIG. 3 is a diagram for explaining a current supply method in a PM motor. FIG. 3 is a diagram for explaining changes in inductance in a PM motor. FIG. 3 is a diagram showing an example of a waveform pattern according to an embodiment of the present invention. FIG. 3 is a diagram showing an example of a waveform pattern according to an embodiment of the present invention. FIG. 3 is a diagram showing an example of a waveform pattern according to an embodiment of the present invention. 1 is a schematic configuration diagram showing a configuration example of a dental medical device to which sensorless control according to an embodiment of the present invention can be applied.

DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, embodiments for carrying out the present invention will be described with reference to the drawings and the like. The embodiment described below is one embodiment for explaining the present invention, and is not intended to be interpreted as limiting the present invention, and the embodiments described in each embodiment Not all configurations are essential for solving the problems of the present invention. Furthermore, in each drawing, correspondence is indicated by assigning the same reference numerals to the same components.

<First embodiment>
[PM motor overview]
First, a schematic configuration of a PM motor 100 to which the present invention can be applied will be described. FIG. 1(a) shows a schematic configuration of a PM motor 100 according to this embodiment. PM motor 100 includes a rotor 101 and a stator 102. The stator 102 has a concentrated winding structure in which a coil of copper wire or the like is wound around a stator iron core having salient poles. Although an example in which the stator 102 includes six stator cores is shown here, the number of stator cores is not limited to this. Further, the coil is not limited to concentrated winding, and may have a distributed winding configuration. The rotor 101 includes permanent magnets. The rotor 101 includes a surface magnet rotor with permanent magnets attached to the surface around the rotor core as shown in FIG. 1(b), and a surface magnet type rotor with permanent magnets attached to the rotor core as shown in FIG. 1(c). An example of this is an embedded magnet type rotor. The example in FIG. 1 shows an example of a four-pole rotor.

The saliency of the surface magnet type rotor shown in FIG. 1(b) and the embedded magnet type rotor shown in FIG. 1(c) will be explained. In a surface magnet type rotor, permanent magnets are located on both the d-axis, which is the direction of magnetic flux created by the magnetic poles (main magnetic flux direction), and the q-axis, which is magnetically orthogonal to the d-axis, so the inductance in the d-axis direction Ld and inductance Lq in the q-axis direction match (Ld=Lq) and have non-saliency. On the other hand, in an embedded magnet type rotor, since a permanent magnet exists only on the d-axis, the inductance in the q-axis direction becomes large (Ld<Lq), and the rotor has saliency. In other words, the structure of the rotor causes a difference in inductance between the d-axis and the q-axis.

Due to the difference in the saliency of the rotor as described above, position sensorless control using the conventional harmonic superimposition method using the saliency shown in Patent Documents 1 to 3 is difficult for PM motors using surface magnet rotors. That is, it cannot be applied to SPM motors. Further, when the stator coil has concentrated winding, the influence of spatial harmonics is large, so that it is difficult to apply sensorless control using the conventional harmonic superimposition method.

Here, the applicant has confirmed that even with the above-described surface magnet type rotor configuration, weak saliency appears depending on the angle of the permanent magnets. However, the change in the magnetic circuit due to the slot structure of the stator as shown in FIG. 1(a) is relatively large, and the weak saliency of the rotor cannot be detected. As a result, with the surface magnet rotor configuration, it is difficult to apply position sensorless control using the conventional harmonic superposition method.

Therefore, in the method according to the present embodiment, in addition to the structure of a built-in magnet type rotor having salient poles as shown in FIG. 1(c), a surface magnet type rotor as shown in FIG. 1(b) is used. This shows a configuration that can be applied even to a configuration having weak saliency (weak salient poles). In the configuration described in this specification, the non-saliency (weakly salient pole) of the surface magnet rotor will be explained as being extremely small compared to the saliency of the embedded magnet rotor. Details will be described later.

[Position estimation method]
FIG. 2 is a diagram for explaining the difference between the conventional rotor position estimation method described in Patent Document 1 and the like and the rotor position estimation method according to the present embodiment.

FIG. 2(a) shows the U-phase, V-phase, and W-phase axes of a three-phase AC motor that is a PM motor. Further, the d-axis indicates the direction of the magnet magnetic flux Φ. θd indicates the difference between the U-phase axis and the d-axis with reference to the position of the U-phase fixed winding. θd corresponds to the position (angle) of the rotor. Note that if the motor is equipped with a sensor, θd can be directly detected, but in the case of sensorless control, θd cannot be directly detected.

FIG. 2(b) shows an outline of rotor position estimation and control in the conventional method. For example, in Patent Document 2, voltage changes of arbitrary harmonics are superimposed on the dc-qc axis, which is the control axis, based on saliency, and the actual phase of the rotor is calculated from the resulting current. An error Δθd between θd, which is the estimated phase, and θdc, which is the estimated phase, is calculated. Sensorless control is realized by controlling the dq axis and the dc-qc axis to match by correcting the angular velocity ω1 so that Δθd becomes 0. At this time, in the conventional method, the superimposed phase of the harmonics is rotated.

FIG. 2C shows an outline of estimation of the rotor position according to this embodiment. In the method according to this embodiment, a harmonic voltage of phase θdh is applied on the dh axis based on a predetermined pattern, regardless of the rotor position, and the rotor position is estimated based on the detection result. . In this embodiment, unlike conventional methods, the position of the rotor can be estimated using a method in which the axis of the harmonic superimposed phase is fixed and not rotated. Details of the pattern etc. will be described later.

[Circuit configuration]
An example of the circuit configuration of this embodiment will be explained using FIGS. 3 and 4. Furthermore, an example of a circuit configuration in a conventional method is shown in FIG. 5 for comparison. FIG. 3 is a diagram showing an example of the configuration of a system including the PM motor control device according to the present embodiment. The system includes a permanent magnet synchronous motor (PM motor 1), an inverter 2, a current detector 3, a control device 4, and a mechanical load 5. The PM motor 1 according to the present embodiment can be used with a rotor having either of the configurations shown in FIGS. 1(b) and 1(c), but here, as shown in FIG. The explanation will be given as a three-phase PM motor equipped with a motor.

The inverter 2 performs a switching operation based on a PWM (Pulse Width Modulation) control signal from the control device 4, converts current from an AC power source (not shown) into three-phase voltage, and applies the voltage to the PM motor 1.

The mechanical load 5 is a load device driven by the PM motor 1. In this embodiment, the operating portion of a dental medical device corresponds. An example of the dental medical device will be described later using FIG. 15. However, the present invention is not limited thereto, and the PM motor control method according to the present invention may be used in industrial equipment, home appliances, railways, electric vehicles, and the like.

The current detector 3 detects the three-phase current flowing due to the voltage applied to the PM motor 1 from the inverter 2 and feeds it back to the control device 4 . Here, an example will be shown in which a current value Iu corresponding to the U phase and a current value Iw corresponding to the W phase among the three phase currents are detected.

The control device 4 performs sensorless vector control to drive the PM motor 1 without using a position sensor (rotation angle sensor). The control device 4 includes a command generator 6, current controllers (ACR) 7a and 7b, an inverse coordinate converter 8a, a coordinate converter 9a, a PWM controller 10, an adder/subtractor 11, a KPLL controller 12, and an integrator 13, 2. It is configured to include a multiplication gain 14 and a position estimator 15.

The command generator 6 generates an excitation current command Idr and a torque current command Iqr corresponding to the motor current. The excitation current command Idr is input to the current controller 7a via the adder/subtractor 11a. At this time, the adder/subtractor 11a subtracts the feedback current value IdFB output from the coordinate converter 9a from the excitation current command Idr, and outputs the result to the current controller 7a. Torque current command Iqr is input to current controller 7b via adder/subtractor 11b. At this time, the adder/subtractor 11b subtracts the feedback current value IqFB output from the coordinate converter 9a from the torque current command Iqr, and outputs the result to the current controller 7b. The current controller 7a controls the voltage command Vd for the d-axis based on the input excitation current command Ibr. The current controller 7b controls the voltage command Vq for the q-axis based on the input torque current command Iqr.

Voltage command Vd from current controller 7a and voltage command Vq from current controller 7b are input to inverse coordinate converter 8a. The inverse coordinate converter 8a converts the input voltage commands Vd and Vq from two phases to three-phase voltage commands Vu0, Vv0, and Vw0 corresponding to the PM motor 1 based on θdc input from the integrator 13. and output. In other words, the inverse coordinate transformer 8a performs coordinate transformation from a two-phase coordinate system to a three-phase coordinate system.

The adder/subtractor 11d adds the voltage command Vu0 corresponding to the U phase and the voltage command Vuh of the U phase output from the position estimator 15, and outputs the result to the PWM controller 10 as a voltage command Vu1. Adder/subtractor 11e adds voltage command Vv0 corresponding to the V phase and voltage command Vvh of V phase output from position estimator 15, and outputs the result to PWM controller 10 as voltage command Vv1. Adder/subtractor 11f adds voltage command Vw0 corresponding to the W phase and voltage command Vwh of W phase output from position estimator 15, and outputs the result to PWM controller 10 as voltage command Vw1. That is, the voltage commands Vuh, Vvh, and Vwh output from the position estimator 15 are superimposed on Vu0, Vv0, and Vw0 for controlling the rotation of the PM motor 1. Details of the voltage commands Vuh, Vvh, and Vwh output from the position estimator 15 will be described later.

Based on θdc input from the integrator 13, the coordinate converter 9a converts the U-phase current value Iu and W-phase current value Iw detected by the current detector 3 from the 3-phase feedback current value to the 2-phase feedback current value. Convert to IdFB, IqFB. That is, the coordinate converter 9a performs coordinate conversion from a three-phase coordinate system to a two-phase coordinate system. The converted feedback current values IdFB and IqFB are output to adders and subtracters 11a and 11b, respectively.

The PWM controller 10 outputs a PWM control signal to the inverter 2 based on the input voltage commands Vu1, Vv1, and Vw1.

The position estimator 15 outputs three-phase voltage commands Vuh, Vvh, and Vwh corresponding to a predefined pattern signal in order to estimate the position of the rotor of the PM motor 1, and correspondingly outputs a current detector. The position of the rotor of the PM motor 1 is estimated from the U-phase current value Iu and the W-phase current value Iw detected in step 3. A specific method of position estimation according to this embodiment will be described later. The position estimator 15 outputs three-phase voltage commands Vuh, Vvh, and Vwh to adders/subtractors 11d, 11e, and 11f, respectively. The position estimator 15 outputs the phase θdc2r to the adder/subtractor 11c as a result of estimating the position of the rotor of the PM motor 1.

The adder/subtractor 11c subtracts the phase from the double gain 14 from the phase θdc2r from the position estimator 15, and outputs the result to the KPLL controller 12. The KPLL controller 12 calculates an angular frequency ω1 for adjusting the position of the rotor based on the rotation angle from the adder/subtractor 11c, and outputs it to the integrator 13. The integrator 13 derives the phase θdc using the angular frequency ω1 from the KPLL controller 12. The phase θdc is output to each of the inverse coordinate converter 8a, the coordinate converter 9a, and the double gain 14. The double gain 14 is an amplifier that doubles the phase θdc, and outputs the amplified value to the adder/subtractor 11c.

(position estimator)
FIG. 4 is a diagram showing an example of the circuit configuration of the position estimator 15 according to this embodiment. As shown using FIG. 3, the position estimator 15 inputs the current values Iu and Iw of the PM motor 1 detected by the current detector 3, and receives a voltage command Vuh corresponding to the harmonic voltage of the three phases. , Vvh, Vwh, and phase θdc2r are output.

In the position estimator 15, a harmonic phase generator 16 generates a harmonic phase θdh and outputs it to an inverse coordinate transformer 8b, a coordinate transformer 9b, a sine signal generator 19, and a cosine signal generator 20, respectively. The harmonic voltage setter 17 sets a voltage value Vh indicating the magnitude of the harmonic voltage to be applied. The voltage value set here is used as Vhd, which is the voltage command for the dh axis. The zero setter 18 sets a voltage value of zero. The voltage value set here is used as the qh-axis voltage command Vhq.

The inverse coordinate converter 8b converts the voltage command Vhd from the harmonic voltage setter 17 and the voltage command Vhq from the zero setter 18 into three-phase converters based on the harmonic phase θdh from the harmonic phase generator 16. is converted into harmonic voltage commands Vuh, Vvh, and Vwh and output. That is, the inverse coordinate transformer 8b performs coordinate transformation from a two-phase coordinate system to a three-phase coordinate system.

The coordinate converter 9b converts the U-phase current value Iu and the W-phase current value Iw detected by the current detector 3 into the dh-qh axis based on the harmonic phase θdh from the harmonic phase generator 16. Convert to two-phase current value. That is, the coordinate converter 9b performs coordinate conversion from a three-phase coordinate system to a two-phase coordinate system. Among these, the harmonic current value Iqh on the qh axis is input to each of integrator 21a and 21b.

The sine signal generator 19 calculates sin (2×θdh) based on the harmonic phase θdh from the harmonic phase generator 16, and outputs it to the integrator 21a. The cosine signal generator 20 calculates cos(2×θdh) based on the harmonic phase θdh from the harmonic phase generator 16, and outputs it to the integrator 21b.

The integrator 21a integrates the harmonic current value Iqh from the sine signal generator 19 and the coordinate converter 9b, and outputs it to the average value calculator 22a. The average value calculator 22a calculates the average value of the output values from the integrator 21a, and outputs it to the arctangent calculator 23 as a current value Iqhsin. The integrator 21b integrates the harmonic current value Iqh from the cosine signal generator 20 and the coordinate converter 9b, and outputs the result to the average value calculator 22b. The average value calculator 22b calculates the average value of the output values from the integrator 21b, and outputs it to the arctangent calculator 23 as a current value Iqhcos.

The arctangent calculator 23 calculates an arctangent (tan ⁻¹ ) using the current value Iqhsin and the current value Iqhcos inputted from the average value calculators 22a and 22b, and outputs it as θdc2r.

(Conventional configuration example)
FIG. 5 is a diagram showing an example of a circuit configuration as a comparative example. Components that overlap with the configuration of the present embodiment shown in FIG. 3 are given the same reference numerals, and description thereof will be omitted. The control device 4Z includes a voltage setting device 95, a change amount extractor 96, an axis error estimator 97, and a voltage setting device 95, a change amount extractor 96, an axis error estimator 97, and A zero setter 98 is provided.

The voltage command from the current controller 7a and the voltage command Vh from the voltage setting device 95 are added by an adder/subtractor 11d, and inputted to the inverse coordinate converter 8a as a voltage command for the d-axis. The voltage command Vh from the voltage setting device 95 corresponds to harmonics that are superimposed using the conventional method.

Feedback current value IqFB from coordinate converter 9a is input to change amount extractor 96. The change amount extractor 96 calculates ΔIqc, which is an error current on the qc axis, based on the feedback current value IqFB, and outputs it to the axis error estimator 97. The axis error estimator 97 uses ΔIqc to calculate Δθdc shown in FIG. 2(b). Then, the adder/subtractor 11c subtracts the value of Δθdc from the value from the zero setter 98 to calculate a negative difference value and outputs it to the KPLL controller 12. In the conventional method, the dq axis and the dc-qc axis are made to coincide by correcting the angular velocity ω1 so that this difference value becomes 0.

[phase]
FIG. 6 shows the relationship among the phases θdc, θdc2r, and θdh in the control device 4 of the PM motor 1 according to the present embodiment. In FIG. 6, the horizontal axis indicates time t, and the vertical axis indicates phase [deg]. As shown in FIGS. 6(a) and 6(b), the phase θdc2r has a frequency twice that of the phase θdc. Further, θdh is an extremely high frequency (harmonic) compared to θdc and θdc2r.

In this embodiment, the multiphase AC (in this example, three-phase AC) output from the inverter 2 as a control signal corresponding to the phase θdc is also referred to as "first polyphase AC." On the other hand, the polyphase AC (three-phase AC in this example) for position estimation corresponding to the phase θdh is also referred to as "second polyphase AC". Note that when position estimation is performed with the rotor of the PM motor stopped, only the second multiphase AC is output from the inverter 2, but when the rotor is rotated (for example, low-speed rotation etc.), when estimating the position of the rotor, the second polyphase alternating current is output in a superimposed state on the first polyphase alternating current.

According to the frequency relationship between FIGS. 6(a) and 6(c), the output frequency of the voltage commands Vu0, Vv0, and Vw0 from the inverse coordinate converter 8 shown in FIG. The output frequency of the commands Vuh, Vvh, and Vwh becomes higher. In other words, the process for position estimation is executed at a shorter cycle than the process related to rotational drive.

FIG. 7 shows an example of a detected waveform in the position estimator 15. In FIG. 7, the horizontal axis indicates time t, which corresponds to FIGS. 7(a) to (d). Further, Ts indicates a control period for position estimation, and is defined in advance. For example, Ts is set to approximately the calculation processing cycle of the control device 4. Therefore, when the processing speed of the control device 4 is high, Ts can be set shorter.

In FIG. 7(a), the vertical axis indicates the phase θdh of the harmonic voltage, and here the step width is set at 60° increments. Therefore, θdh repeats the values of 0°, 60°, 120°, 180°, 240°, and 300°. These harmonic voltages are superimposed on the control voltage applied to the PM motor 1. A voltage command Vh is set only for the dh axis by the harmonic voltage setter 17 and zero setter 18 shown in FIG. Then, the set voltage is converted into harmonic voltage commands Vuh, Vvh, and Vwh based on θdh from the harmonic phase generator 16. As shown in FIG. 3, harmonic voltage commands Vuh, Vvh, and Vwh are added (superimposed) to three-phase voltage commands Vu0, Vv0, and Vw0, respectively, and are input to the PWM controller 10 as Vu1, Vv1, and Vw1. .

FIG. 7(b) shows how the coordinate converter 9b of the position estimator 15 is calculated based on the current values Iu and Iw obtained by the current detector 3 when the position of the rotor of the PM motor 1 is θd=0°. This is a waveform showing the harmonic current value Iqh obtained by In FIG. 7B, the values plotted with black circles (●) correspond to the calculation results of the Fourier integral in each section of Ts.

FIG. 7(c) shows the coordinate converter 9b of the position estimator 15 based on the current values Iu and Iw obtained by the current detector 3 when the position of the rotor of the PM motor 1 is θd=45°. This is a waveform showing the value of the harmonic current value Iqh obtained by As shown in FIGS. 7(b) and 7(c), the phase of the waveform differs depending on the position of the rotor. Furthermore, as shown in FIGS. 7(b) and 7(c), the harmonic current value Iqh is generated at a period twice as long as θdh. In this way, by observing the harmonic current value Iqh and determining its phase, it is possible to estimate the phase θd of the rotor, that is, the position.

FIG. 7(d) shows signal waveforms output by the sine signal generator 19 and the cosine signal generator 20, respectively, based on the harmonic phase θdh shown in FIG. 7(a).

The principle of generation of the waveforms shown in FIGS. 7(b) and 7(c) will be explained in more detail with reference to FIGS. 8 and 9. First, using FIG. 8, the waveform when the rotor position is θd=0° (FIG. 7(b)) will be described. As shown in FIG. 7(a), for each of the six positions of θd=0, 60, 120, 180, 240, and 300 [deg], the superimposed voltage Vh, harmonic current Ih, and magnet magnetic flux Φ in the rotor shows the relationship between The direction of the magnet magnetic flux Φ corresponds to the direction of the d-axis, and the direction perpendicular thereto corresponds to the direction of the q-axis. Further, the superimposed voltage Vh indicates the voltage applied to the PM motor based on the harmonic voltage commands Vuh, Vvh, and Vwh described above.

Here, the explanation will be given assuming that the rotational speed of the rotor of the PM motor 1 is low, and that θd can be considered to be stationary compared to the change in θdh. That is, the description will be made assuming a case where θdh is sequentially switched while the rotor is stopped at a certain rotation angle (position). This is based on the relationship shown in FIGS. 6(a) and 6(c).

FIG. 8(a) shows the state when θdh=0°. In this case, since the harmonic superimposed voltage Vh is applied in a positive direction on the d-axis, the harmonic current Ih is observed only on the dh-axis and is not generated on the qh-axis. That is, the harmonic current value Iqh on the qh axis becomes 0.

FIG. 8(b) shows the state when θdh=60°. In this case, the harmonic current Ih is tilted in the negative direction of the dh axis with respect to the dh axis. Therefore, the harmonic current value Iqh on the qh axis is observed as a negative value. The value here corresponds to the position where the negative value is plotted in FIG. 7(b) (θdh=60°). The slope of the harmonic current Ih at this time is the same for both PM motors with surface magnet rotors and embedded magnet rotors with different salient poles, as shown in FIGS. 1(b) and 1(c). This is a phenomenon that can occur. In the present invention, this phenomenon is applied to estimate the rotor position.

FIG. 8(c) shows the state when θdh=120°. In this case, the harmonic current Ih is tilted in the positive direction of the qh axis with respect to the dh axis. Therefore, the harmonic current value Iqh on the qh axis is observed as a positive value. The value here corresponds to the position where a positive value is plotted in FIG. 7(b) (θdh=120°).

FIG. 8(d) shows the state when θdh=180°. In this case, since the harmonic superimposed voltage Vh is applied in the negative direction on the d-axis, the harmonic current Ih is observed only on the dh-axis and is not generated on the qh-axis. In other words, the harmonic current value Iqh on the qh axis is 0, as in FIG. 8(a). The value here corresponds to the position where 0 is plotted in FIG. 7(b) (θdh=180°).

FIG. 8(e) shows the state when θdh=240°. In this case, the harmonic current Ih tilts in the negative direction of the qh axis with respect to the dh axis. Therefore, Iqh on the qh axis is observed as a negative value. The value here corresponds to the position where the negative value is plotted in FIG. 7(b) (θdh=240°). Moreover, this value is observed to be equivalent to the case where θdh=60°.

FIG. 8(f) shows the state when θdh=300°. In this case, the harmonic current Ih is tilted in the positive direction of the qh axis with respect to the dh axis. Therefore, Iqh on the qh axis is observed as a positive value. The value here corresponds to the position where a positive value is plotted in FIG. 7(b) (θdh=300°). Moreover, this value is observed to be equivalent to the case where θdh=120°. In this way, Iqh can be observed as a periodic value due to the relationship between the value of θdh and the position of the rotor, and the position of the rotor can be estimated based on the observation results.

Similarly, FIG. 9 shows the principle of detecting the waveform when the rotor position is θd=45° (FIG. 7(c)). Since θd=45°, the direction of the magnet magnetic flux Φ (that is, the direction of the d-axis) is different from that in FIG.

FIG. 9(a) shows the state when θdh=0°. In this case, the harmonic current Ih is tilted in the positive direction of the qh axis with respect to the dh axis. Therefore, Iqh on the qh axis is observed as a positive value. The value here corresponds to the position where a positive value is plotted in FIG. 7(c) (θdh=0°).

FIG. 9(b) shows the state when θdh=60°. In this case, the harmonic current Ih tilts in the negative direction of the qh axis with respect to the dh axis. Therefore, the harmonic current value Iqh on the qh axis is observed as a negative value. The value here corresponds to the position where the negative value is plotted in FIG. 7(c) (θdh=60°).

FIG. 9(c) shows the state when θdh=120°. In this case, the harmonic current Ih tilts in the negative direction of the qh axis with respect to the dh axis. Therefore, the harmonic current value Iqh on the qh axis is observed as a negative value. The slope in the negative direction here is larger than θdh=60°, and as a result, the observed value is also smaller than when θdh=60°. The value here corresponds to the position where the negative value is plotted in FIG. 7(c) (θdh=120°).

FIG. 9(d) shows the state when θdh=180°. In this case, the harmonic current Ih is tilted in the positive direction of the qh axis with respect to the dh axis. Therefore, the harmonic current value Iqh on the qh axis is observed as a positive value. The value here corresponds to the position where a positive value is plotted in FIG. 7(c) (θdh=180°). Moreover, this value is observed to be equivalent to the case where θdh=0°.

FIG. 9(e) shows the state when θdh=240°. In this case, the harmonic current Ih tilts in the negative direction of the qh axis with respect to the dh axis. Therefore, the harmonic current value Iqh on the qh axis is observed as a negative value. The value here corresponds to the position where the negative value is plotted in FIG. 7(c) (θdh=240°). Moreover, this value is observed to be equivalent to the case where θdh=60°.

FIG. 9(f) shows the state when θdh=300°. In this case, the harmonic current Ih tilts in the negative direction of the qh axis with respect to the dh axis. Therefore, the harmonic current value Iqh on the qh axis is observed as a negative value. The slope in the negative direction here is larger than θdh=240°, and as a result, the observed value is also smaller than when θdh=240°. The value here corresponds to the position where the negative value is plotted in FIG. 7(c) (θdh=300°). Moreover, this value is observed to be equivalent to the case where θdh=120°.

Note that when θdh is observed for one revolution (360°) in 60° increments, the harmonic current value Iqh can be observed for two periods. At this time, it is not necessary to observe values for two periods, and the position of the rotor may be estimated when one period of harmonic current value Iqh has been observed. Furthermore, in order to improve the estimation accuracy, the position estimation of the rotor may be performed after making the observation period of the harmonic current value Iqh longer (for example, 3 periods or more).

The actual phase θd of the rotor of the PM motor 1 is calculated using the observation result of the harmonic current value Iqh according to the above θdh as phase information. In this embodiment, Fourier integral is used to estimate the phase θd of the rotor based on the harmonic current value Iqh. Fourier integration is realized by the sine signal generator 19, cosine signal generator 20, integrator 21a, 21b, and average value calculators 22a, 22b of the configuration of the position estimator 15 shown in FIG. A series of these processes corresponds to Fourier integration, and as a result, the magnitudes of the sine component and cosine component included in the harmonic current value Iqh are obtained. Then, the phase θdc2r is obtained from these sine and cosine components by the arctangent calculator 23, and this becomes a value corresponding to the actual phase of the rotor.

Regarding the phase θdc2r, which is the estimated position of the rotor estimated by the position estimator 15, since the estimation is performed using harmonic current, it is assumed that noise components are included. Therefore, in this embodiment, in order to suppress the influence of noise, PLL (Phase Locked Loop) control by the KPLL controller 12 is used to provide a constant band to the response, thereby removing the noise component. Here, since the output of the KPLL controller 12 becomes the angular frequency ω1 for driving the PM motor 1, it is possible to estimate the rotational speed of the rotor in addition to estimating the position of the rotor of the PM motor 1. .

Further, the phase θdc2r obtained by applying the harmonic changes at a frequency twice as high as θd, as described above with reference to FIG. Since the saliency of the rotor acts equally on both the north and south poles of the permanent magnets provided in the rotor, the frequency always changes by twice. Therefore, in this embodiment, as shown in FIG. 3, a double gain 14 is provided, the estimated phase θdc is doubled in advance, and the deviation between this and θdc2r is determined by the adder/subtractor 11c. Then, the KPLL controller 12 performs PLL control on the deviation.

[effect]
The effects of the position sensorless drive using the method according to this embodiment will be described below. FIG. 10 shows a configuration example of a PM motor to which the method according to the present embodiment can be applied, and here a two-pole three-phase PM motor is shown. The PM motor has a structure in which three-phase windings of U-phase, V-phase, and W-phase are wound around stator slots. Furthermore, as explained using FIG. 1, the PM motor shown here has a structure that does not have saliency like a surface magnet rotor (that is, the saliency is so weak that changes cannot be detected using conventional methods). It is.

In FIG. 10, white arrows indicate the direction of magnetic flux when each phase is energized, and black arrows indicate the direction of magnetic lines of force when each phase is energized. Further, FIG. 11 is a graph diagram for explaining changes in inductance of the stator and rotor that constitute the PM motor. In FIG. 11, the horizontal axis shows the rotation angle (actual phase) of the rotor that rotates over time, and the vertical axis shows the inductance.

When harmonics are applied to the windings, for example, when current is applied from the V phase and W phase to the U phase (Fig. 10 (a)), and when the current is applied from the W phase to the U phase but not the V phase ( The magnetic circuit of the magnetic flux caused by the current is different from FIG. 10(b)). In the case of the pattern shown in FIG. 10(b), the V phase is an empty coil. The magnetic circuit of the magnetic flux within the PM motor changes depending on the energization state of each phase. As a result, the inductance of the stator appears to fluctuate in the circuit of the PM motor.

When the inductance of the stator changes according to the energization phase, it has periodicity of every 60°, as shown in FIG. 11(a). On the other hand, due to the saliency of the rotor, the inductance of the d-axis and the q-axis changes as shown in FIG. 11(b). Here, it has periodicity of every 180°. Furthermore, as shown in FIGS. 10(a) and 10(b), the range of change in inductance is smaller in the rotor than in the stator. In other words, there is a difference in the change in inductance between the stator and rotor. In estimating the rotor position using saliency as in the conventional method, it is necessary to extract only the change in rotor inductance. However, when the variation width of the rotor inductance is small, the variation is buried in the variation of the stator inductance, making it difficult to detect only the variation of the rotor inductance.

In other words, when harmonics are superimposed on an arbitrary phase as in the conventional method, changes in rotor inductance cannot be detected due to changes in inductance on the stator side, making rotor position estimation impossible. becomes.

On the other hand, in the method according to the present embodiment, the value of the inductance of the stator is fixed by setting the harmonic superposition phase θdh at a predetermined interval, that is, at a period of 60°, as shown in FIG. 7(a). . That is, the phase is fixed at the position plotted by the black circle (●) in FIG. 11(a). In the case of the example shown in FIG. 11(a), it is fixed so that detection is performed at a position where the inductance is highest. As a result, it becomes possible to extract and detect only the change in the inductance of the rotor, and the accuracy of estimating the rotor position can be improved.

Even if the stator is a concentrated winding PM motor with large spatial harmonics, or the rotor is a surface magnet type PM motor with non-salient poles (that is, weak saliency), the rotor position can be estimated. becomes possible. Furthermore, if the stator has a slotless structure, the magnetic flux density distribution may differ depending on the arrangement of the windings, resulting in slight spatial harmonics, but the method according to this embodiment can also be applied to such a structure. be.

Further, in the example shown in FIG. 7(a), θdh is set as 0, 60, 120, etc., but it is not limited to this. As explained using FIG. 11(a), other angles may be used as long as the change in the inductance of the stator can be fixed. Further, in the examples of FIGS. 8 and 9, the position of the rotor is estimated using six patterns, but more patterns may be used to improve the granularity and accuracy. Further, the setting interval of θdh may also vary depending on the structure of the stator and rotor of the PM motor.

Furthermore, when applying a superimposed voltage to the PM motor, a current pattern as shown in FIG. 10(a) (that is, a pattern with no vacant coils) may be used, or a current pattern as shown in FIG. A pattern (that is, a pattern with empty coils) may also be used. In either case, any configuration may be used as long as the change in inductance on the stator side is fixed and only the inductance on the rotor side can be extracted as shown in FIG. 11.

As described above, according to the present embodiment, position sensorless driving becomes possible in a non-salient pole (that is, weak saliency) PM motor, which has been difficult to drive sensorless due to harmonic superposition. Further, even when the PM motor is operating in a low speed range, the position of the rotor can be estimated.

[Waveform]
Examples of harmonic application patterns used in rotor position estimation according to the present invention are shown in FIGS. 12 to 14. Here, an example targeting a two-phase three-pole PM motor will be described. Note that when the number of phases and the number of poles of the PM motor are different, the application pattern may be defined according to the configuration. The application pattern shown below shows an example in which all three stator coils V, U, and W are used and there are no empty coils.

FIG. 12 is a diagram showing an example of a harmonic superimposed waveform according to the present embodiment. The waveform here shows an example of a 180 degree rectangular wave. The horizontal axis in FIG. 12 indicates time, which corresponds to each graph. In FIG. 12, the broken lines correspond to the intervals of Ts, and six sections indicate one period of the waveform, as shown in both directions in the figure. Further, in FIG. 12, the three from the top indicate harmonic voltage commands Vuh, Vvh, and Vwh for the stator coils U-phase, V-phase, and W-phase in order, and are shown here corresponding to the triangular carrier wave. Moreover, Pu, Pv, and Pw indicate pulse signals from the PWM controller 10 corresponding to harmonics for position estimation, respectively corresponding to the U phase, V phase, and W phase. Vuv indicates line voltage in a three-phase circuit. Vun indicates the phase voltage of the U phase.

The harmonic voltage commands Vuh, Vvh, and Vwh are shifted in phase by 120°. In other words, it has a symmetrical three-phase AC voltage configuration in which the voltages of each phase are equal in magnitude and the phase difference is 120°. Further, the voltage commands Vuh, Vvh, and Vwh are configured to rise and fall at the positive and negative peak positions of the triangular carrier wave.

A harmonic pulse signal Pu to the stator coil U, a harmonic pulse signal Pv to the stator coil V, and a harmonic pulse signal Pw to the stator coil W are transmitted to the PWM controller when estimating the rotor position. 10 shows a signal waveform output from the inverter 2, and these pulse signals are defined so that the superimposed voltage during position estimation is constant.

In the example of FIG. 12, the six sections for one period correspond to θdh=300°, 0°, 60°, 120°, 180°, and 240° from the right, which corresponds to ( Corresponds to f), (a), (b), (c), (d), and (e).

13 and 14 are diagrams showing other examples of harmonic superimposed waveforms according to this embodiment. The waveform in FIG. 13 shows an example of a 120 degree rectangular wave. The waveform in FIG. 14 shows an example of a sine wave. 13 and 14 also have a configuration in which the phases are symmetrical. 13 and 14 also show examples in which the carrier wave is a triangular wave, and the voltage commands Vuh, Vvh, and Vwh are configured to rise and fall at the positive and negative peak positions of the triangular carrier wave.

In addition, in the control device 4 according to the present embodiment, regarding rotor position estimation by the position estimator 15, current detection (detection of current values Iu and Iw), voltage update (update of voltage commands Vuh, Vvh, and Vwh), Fourier integral processing is performed. These processes for estimating the position may be performed with priority over control of the rotational operation of the rotor, such as current control and speed control from the command generator 6. In other words, signals related to control for position estimation may be preferentially interrupted and executed.

[Device example]
FIG. 15 is a diagram illustrating a configuration example of a dental medical device, which is an example of a device to which sensorless control of a PM motor according to the present embodiment can be applied. In the dental medical device 1500, a motor section 1502 and a controller 1505 are connected via a connector 1503 and a cable 1504. The PM motor 1 shown in FIG. 3 is provided in the motor section 1502, and, for example, the hand piece 1501 (or a dental bar located at the tip thereof) connected to the motor section 1502 corresponds to the mechanical load 5.

Inverter 2, current detector 3, and control device 4 shown in FIG. 3 are provided in controller 1505. Note that although the example in FIG. 15 shows a configuration in which the motor section 1502 and the controller 1505 are provided separately, they may be integrated. In that case, the circuit configuration shown in FIG. 3 will be provided in one device.

As mentioned above, dental medical equipment is necessarily required to be small in size depending on its purpose of use. For example, in order to reduce the size of the motor unit 1502 in which the PM motor 1 is provided, a sensorless configuration in which the position sensor of the PM motor 1 is omitted is required. Furthermore, when using dental medical equipment, it is necessary to perform rotational operations such as stopping and rotating at low speeds, and to control the rotational speed to frequently accelerate and decelerate.Even under such operating conditions, it is necessary to control the position of the PM motor. There is a need for a method that can appropriately detect this.

The method for estimating the position of the rotor of a PM motor according to the present embodiment is applicable to the above-mentioned device, and its usefulness is obvious.

As described above, according to this embodiment, it is possible to realize position sensorless control in which the position can be estimated in a low speed range in the PM motor, regardless of the structure related to the saliency of the rotor.

<Other embodiments>
In the above embodiment, a PM motor having non-saliency was taken as an example. The application of the present invention is not limited to the motor configuration shown in the above embodiment. For example, it is applicable to slotless motors, stepping motors, etc. In addition, the stator coil configuration can be applied to PM motors such as concentrated winding/distributed winding coil configuration, single-layer winding/multilayer winding coil configuration, and full-pitch winding/short-pitch winding coil configuration. be.

Moreover, the apparatus to which the present invention is applicable is not limited to dental medical equipment. For example, it may be used in PM motor control of equipment that requires operation in a low speed range, equipment that aims to reduce size, and the like. Note that the low speed range here may be, for example, a speed range of 10% or less of the rated rotational speed of the motor.

Furthermore, the motor control method according to this embodiment and other sensorless control methods may be combined in one device. For example, in motor control, the method according to the present invention may be used in a low speed range, and the conventional sensorless control may be used in a high speed range.

In addition, in the present invention, a program or application for realizing the functions of one or more embodiments described above is supplied to a system or device using a network or a storage medium, and one or more computers of the system or device are provided with a program or an application. This can also be realized by a process in which a processor reads and executes a program.

Further, it may be realized by a circuit (for example, an ASIC (Application Specific Integrated Circuit) or an FPGA (Field Programmable Gate Array)) that realizes one or more functions.

As described above, the present invention is not limited to the embodiments described above, and those skilled in the art can combine the configurations of the embodiments with each other, modify and apply them based on the description of the specification and well-known techniques. It is also contemplated by the present invention to do so, and is within the scope for which protection is sought.

As mentioned above, the following matters are disclosed in this specification.
(1) A drive device for a permanent magnet synchronous motor of a dental medical device,
an inverter that drives the permanent magnet synchronous motor;
detection means for detecting a current flowing through the permanent magnet synchronous motor;
Control means for controlling the permanent magnet synchronous motor via the inverter using the current value detected by the detection means;
has
The control means applies a second multiphase alternating current having a higher frequency than the first multiphase alternating current to the permanent magnet synchronous motor at a predetermined phase interval for rotationally driving the permanent magnet synchronous motor. performing an estimation calculation of the position of the rotor of the permanent magnet synchronous motor based on the current generated by the
A drive device characterized by:
According to this configuration, in a dental medical device equipped with a PM motor, regardless of the structure related to the saliency of the rotor, it is possible to realize position sensorless control that allows position estimation in a low speed range.

(2) The drive device according to (1), wherein the second multiphase alternating current is symmetrical between phases.
According to this configuration, it becomes possible to control the application of multiphase alternating current with higher accuracy.

(3) The drive device according to (1), wherein the second multiphase alternating current has a waveform synchronized with a carrier wave for performing a switching operation in the inverter.
According to this configuration, it becomes possible to control the application of multiphase alternating current with higher accuracy.

(4) The multiphase alternating current output from the inverter is three-phase alternating current,
The drive device according to (1), wherein the second polyphase alternating current has an integral multiple of one-third of the frequency of a carrier wave for performing a switching operation by the inverter.
According to this configuration, it is possible to realize position sensorless control that can also perform position estimation in a low speed range for a PM motor with a three-phase configuration.

(5) The drive device according to (1), wherein the second polyphase AC waveform pattern is defined in advance based on a phase configuration of the permanent magnet synchronous motor.
According to this configuration, it is possible to define a pattern for estimating the rotor position according to the phase configuration of the PM motor.

(6) The drive device according to (1), wherein the second polyphase AC waveform pattern is based on any one of a 180° rectangular wave, a 120° rectangular wave, and a sine wave.
According to this configuration, various waveforms can be defined as patterns for estimating the rotor position.

(7) The permanent magnet synchronous motor has a two-pole three-phase configuration,
The drive device according to (1), wherein the second multiphase alternating current includes a pattern of harmonics every 60 degrees.
According to this configuration, it is possible to realize position sensorless control that can also perform position estimation in a low speed range for a PM motor having a two-pole three-phase configuration.

(8) The drive device according to (1), wherein the control means estimates the position of the rotor based on the phase of the current generated by applying it to the permanent magnet synchronous motor at the predetermined phase interval. .
According to this configuration, it is possible to estimate the position of the rotor by suppressing the influence on the stator side and extracting only changes on the rotor side.

(9) The control means:
a first process for controlling the first multiphase alternating current for rotationally driving the permanent magnet synchronous motor;
a second second process for controlling the second polyphase alternating current for estimating the position of the permanent magnet synchronous motor;
Run
The drive device according to (1), wherein the second process is executed at a shorter cycle than the first process, and is executed with priority over the first process.
According to this configuration, in the PM motor, it becomes possible to perform rotor position estimation with priority over rotation control in a shorter cycle.

(10) The second process includes updating a voltage command corresponding to the second multiphase alternating current, and updating a current value applied to the permanent magnet synchronous motor corresponding to the second multiphase alternating current. The drive device according to (9), characterized in that the drive device includes at least one of detection processing and calculation of the position of the rotor based on the current value obtained by the detection processing.
According to this configuration, in the PM motor, as control for estimating the rotor position, it is possible to prioritize updating of the voltage command, detection of the current value, and calculation of the rotor in a short cycle. .

(11) A method for controlling a permanent magnet synchronous motor of a dental medical device, comprising: an inverter that drives a permanent magnet synchronous motor; and a detection means that detects a current flowing through the permanent magnet synchronous motor;
a control step of controlling the permanent magnet synchronous motor via the inverter using the current value detected by the detection means;
In the control step, applying a second polyphase AC having a higher frequency than the first polyphase AC for rotationally driving the permanent magnet synchronous motor to the permanent magnet synchronous motor at a predetermined phase interval. Estimating the position of the rotor of the permanent magnet synchronous motor is performed based on the current generated.
According to this configuration, in a dental medical device equipped with a PM motor, regardless of the structure related to the saliency of the rotor, it is possible to realize position sensorless control that allows position estimation in a low speed range.

Although various embodiments have been described above with reference to the drawings, it goes without saying that the present invention is not limited to such examples. It is clear that those skilled in the art can come up with various changes or modifications within the scope of the claims, and these naturally fall within the technical scope of the present invention. Understood. Further, each of the constituent elements in the above embodiments may be arbitrarily combined without departing from the spirit of the invention.

1 PM motor 2 Inverter 3 Current detector 4 Control device 5 Mechanical load 6 Command generator 7a, 7b Current controller 8a, 8b Inverse coordinate converter 9a, 9b Coordinate converter 10 PWM controller 11a to 11f Adder/subtractor 12 KPLL control 13 Integrator 14 Double gain 15 Position estimator 16 Harmonic phase generator 17 Harmonic voltage setter 18 Zero setter 19 Sine signal generator 20 Cosine signal generator 21a, 21b Integrator 22a, 22b Average value calculator 23 Arctangent calculator 95 Voltage setter 96 Variation amount extractor 97 Axial error estimator 98 Zero setter 100 PM motor 101 Rotor 102 Stator

Claims (11)

A drive device for a permanent magnet synchronous motor for dental medical equipment,
an inverter that drives the permanent magnet synchronous motor;
detection means for detecting a current flowing through the permanent magnet synchronous motor;
Control means for controlling the permanent magnet synchronous motor via the inverter using the current value detected by the detection means;
has
The control means applies a second multiphase alternating current having a higher frequency than the first multiphase alternating current to the permanent magnet synchronous motor at a predetermined phase interval for rotationally driving the permanent magnet synchronous motor. performing an estimation calculation of the position of the rotor of the permanent magnet synchronous motor based on the current generated by the
A drive device characterized by: The drive device according to claim 1, wherein the second multiphase alternating current is symmetrical between phases. The drive device according to claim 1, wherein the second polyphase alternating current has a waveform synchronized with a carrier wave for performing a switching operation in the inverter. The multiphase alternating current output from the inverter is three-phase alternating current,
2. The drive device according to claim 1, wherein the second polyphase alternating current has an integral multiple of one third of the frequency of a carrier wave for performing a switching operation by the inverter. The drive device according to claim 1, wherein the second polyphase AC waveform pattern is predefined based on a phase configuration of the permanent magnet synchronous motor. The drive device according to claim 1, wherein the waveform pattern of the second polyphase alternating current is based on any one of a 180° rectangular wave, a 120° rectangular wave, and a sine wave. The permanent magnet synchronous motor has a two-pole three-phase configuration,
The drive device according to claim 1, wherein the second multiphase alternating current is composed of a pattern of harmonics every 60°. The drive device according to claim 1, wherein the control means estimates the position of the rotor based on the phase of the current generated by applying it to the permanent magnet synchronous motor at the predetermined phase interval. The control means includes:
a first process for controlling the first multiphase alternating current for rotationally driving the permanent magnet synchronous motor;
a second second process for controlling the second polyphase alternating current for estimating the position of the permanent magnet synchronous motor;
Run
The drive device according to claim 1, wherein the second process is executed at a shorter cycle than the first process, and is executed with priority over the first process. The second process includes updating a voltage command corresponding to the second multi-phase alternating current, detecting a current value being applied to the permanent magnet synchronous motor corresponding to the second multi-phase alternating current, The drive device according to claim 9, further comprising at least one of: and calculating the position of the rotor based on the current value obtained by the detection process. A method for controlling a permanent magnet synchronous motor of a dental medical device, comprising: an inverter that drives a permanent magnet synchronous motor; and a detection means that detects a current flowing through the permanent magnet synchronous motor.
a control step of controlling the permanent magnet synchronous motor via the inverter using the current value detected by the detection means;
In the control step, applying a second polyphase AC having a higher frequency than the first polyphase AC for rotationally driving the permanent magnet synchronous motor to the permanent magnet synchronous motor at a predetermined phase interval. performing an estimation calculation of the position of the rotor of the permanent magnet synchronous motor based on the current generated by the
A control method characterized by:

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