JP2021145165A - Semiconductor device - Google Patents

Abstract

To drive a switching element at high speed.SOLUTION: First and second variable voltage sources (110H and 110L) are provided beforehand with respect to first and second transistors (TrH and TrL) constituting a half bridge circuit. The first variable voltage source comprises a voltage generation unit (112H) that outputs a rectangle waveform voltage (VPLS_H) for alternately turning on/off the first transistor. The first variable voltage source generates a driving voltage (VO_H) formed by superimposing a voltage component on the rectangle waveform voltage in accordance with the change of the rectangle waveform voltage when switching the first transistor by supplying the rectangle waveform voltage between a gate and a source of the first transistor, and the first variable voltage source supplies the driving voltage to between the gate and the source of the first transistor. The same is applied to the second variable voltage source and the second transistor.SELECTED DRAWING: Figure 11

Description

The present invention relates to a semiconductor device.

In a half-bridge circuit or the like, it is often required to increase the speed of driving a switching element.

Japanese Patent Application Laid-Open No. 2007-501544

Although some proposals have been made for driving switching elements at high speed, there is room for improvement in the technology for high-speed driving.

An object of the present invention is to provide a semiconductor device that contributes to high-speed driving of a switching element.

The semiconductor device according to the present invention has a first electrode, a second electrode, and a control electrode, and the first electrode and the second electrode are turned on according to the voltage between the control electrode and the second electrode. Alternatively, the switching element controlled to the off state and a variable voltage source for supplying a variable voltage between the control electrode and the second electrode of the switching element are provided, and the variable voltage source alternates the switching elements. It has a voltage generating unit that outputs a rectangular wavy voltage for turning it on or off, and switches the switching element by supplying the rectangular wavy voltage between the control electrode and the second electrode. At that time, a configuration (first) in which a drive voltage in which a voltage component corresponding to a change in the rectangular wave voltage is superimposed on the rectangular wave voltage is generated, and the drive voltage is supplied between the control electrode and the second electrode (first). Configuration).

The semiconductor device according to the first configuration may have a configuration (second configuration) in which the drive voltage is generated by a differentiating circuit using an operational amplifier in the variable voltage source.

In the semiconductor device according to the second configuration, the operational amplifier includes a first input terminal, a second input terminal, and an output terminal that receive the rectangular wavy voltage with reference to the potential of the second electrode of the switching element. The variable voltage source includes the operational amplifier, a feedback resistor provided between the second input terminal and the output terminal of the operational amplifier, and the second input terminal of the operational amplifier and the switching element. A capacitor provided between the second electrode and a capacitor may be provided, and the drive voltage may be output from the output terminal of the operational amplifier when switching the switching element (third configuration).

In the semiconductor device according to the third configuration, the product of the capacitance value of the capacitor and the resistance value of the feedback resistor is the capacitance value of the capacitance between the control electrode and the second electrode of the switching element. It has a value based on the product of the resistance value of the control electrode resistance of the switching element and a value based on the product of the capacitance value of the input capacitance of the switching element and the resistance value of the control electrode resistance of the switching element. However, the control electrode resistance of the switching element may have a configuration (fourth configuration) including the internal resistance of the control electrode of the switching element.

In the semiconductor device according to any one of the first to fourth configurations, the variable voltage source controls the switching element to an off state by maintaining the level of the rectangular wavy voltage at a predetermined level. When a voltage change is applied between the first electrode and the second electrode, a voltage corresponding to the voltage change between the first electrode and the second electrode is supplied between the control electrode and the second electrode. (Fifth configuration) may be used.

In the semiconductor device according to the third configuration, the variable voltage source has the first electrode and the first electrode in an off control section for controlling the switching element to an off state by maintaining the level of the rectangular wavy voltage at a predetermined level. When a voltage change is applied between the second electrodes, a voltage corresponding to the voltage change between the first electrode and the second electrode is supplied between the control electrode and the second electrode, and the variable voltage source is used. In addition to the operational capacitor, the feedback resistor, and the first capacitor as the capacitor, a second capacitor provided between the second input terminal of the operational capacitor and the first electrode of the switching element is further provided. In the off control section, a configuration (sixth configuration) may be used in which a voltage corresponding to a voltage change between the first electrode and the second electrode is output from the output terminal of the operational amplifier.

In the semiconductor device according to the sixth configuration, the product of the capacitance value of the first capacitor and the resistance value of the feedback resistor is the capacitance of the capacitance between the control electrode and the second electrode of the switching element. A value based on the product of the capacitance value and the resistance value of the control electrode resistance of the switching element, or a value based on the product of the capacitance value of the input capacitance of the switching element and the resistance value of the control electrode resistance of the switching element. The product of the capacitance value of the second capacitor and the resistance value of the feedback resistor is the product of the capacitance value of the feedback capacitance of the switching element and the resistance value of the control electrode resistance of the switching element. The control electrode resistance of the switching element may have a value based on the above, and may have a configuration (seventh configuration) including an internal resistance in the control electrode of the switching element.

In the semiconductor device according to the sixth or seventh configuration, the second capacitor may be formed by using the parasitic capacitance of the diode (eighth configuration).

In the semiconductor device according to the eighth configuration, the second capacitor may be formed by a series circuit of the parasitic capacitance of the diode and another capacitance (nineth configuration).

In the semiconductor device according to any one of the third, fourth, and sixth to ninth configurations, a buffer circuit is provided between the output terminal of the operational amplifier and the control electrode of the switching element in the variable voltage source. The output voltage of the operational amplifier may be supplied between the control electrode and the second electrode of the switching element through the buffer circuit (tenth configuration).

In the semiconductor device according to any one of the first to tenth configurations, the switching element may have a configuration made of a wide-gap semiconductor (11th configuration).

In the semiconductor device according to any one of the first to eleventh configurations, a plurality of sets of the switching element and the variable voltage source are provided, and the plurality of sets include the first set and the second set, and the first set is included. The first switching element, which is a set of switching elements, and the second switching element, which is a second set of switching elements, are connected in series with each other, and a predetermined DC voltage is applied to the series circuit of the first switching element and the second switching element. May be applied (12th configuration).

According to the present invention, it is possible to provide a semiconductor device that contributes to high-speed driving of a switching element.

It is a block diagram of the herb bridge circuit which concerns on 1st Embodiment of this invention. It is a figure which shows the parasitic capacitance and parasitic resistance of each transistor which constitutes the herb bridge circuit which concerns on 1st Embodiment of this invention. FIG. 5 is an explanatory diagram of an erroneous arc that may occur in an herb bridge circuit according to the first embodiment of the present invention. It is a model circuit diagram which concerns on the 1st Embodiment of this invention and concerns on an erroneous arc. It is a model circuit diagram which concerns on the 1st Embodiment of this invention and concerns on an erroneous arc. It is a figure for demonstrating the factor which suppresses the high-speed drive of a transistor which concerns on 1st Embodiment of this invention. It is a model circuit diagram which concerns on the 1st Embodiment of this invention and is concerned with high-speed drive. It is an internal block diagram of the variable voltage source of FIG. It is a model circuit diagram which concerns on the 1st Embodiment of this invention and is concerned with high-speed drive. FIG. 6 is a schematic overall configuration diagram of a semiconductor device according to Example EX1_1 belonging to the first embodiment of the present invention. FIG. 5 is a circuit diagram of a semiconductor device according to Example EX1_1 belonging to the first embodiment of the present invention. FIG. 5 is a diagram showing a plurality of voltage waveforms related to a switching operation and a state change of each transistor according to the example EX1_1 belonging to the first embodiment of the present invention. FIG. 5 is a diagram showing a plurality of voltage waveforms related to a switching operation and a state change of each transistor according to the example EX1_1 belonging to the first embodiment of the present invention. FIG. 5 is a diagram showing a state in which a resistor is externally connected to the gate of each transistor according to the embodiment EX1_1 belonging to the first embodiment of the present invention. It is a figure which shows the relationship of each value of a plurality of resistors and a plurality of capacitors, relating to Example EX1_1 which belongs to 1st Embodiment of this invention. It is a figure which shows the formation method of the target capacitor which concerns on Example EX1-2 which belongs to 1st Embodiment of this invention. FIG. 5 is a diagram showing a modified configuration of a variable voltage source according to Example EX1_3 belonging to the first embodiment of the present invention. FIG. 5 is a diagram showing a modified configuration of a variable voltage source according to Example EX1_3 belonging to the first embodiment of the present invention. FIG. 5 is a diagram showing a modified configuration of a variable voltage source according to Example EX1_4 belonging to the first embodiment of the present invention. It is external perspective view of the power module which concerns on 2nd Embodiment of this invention. It is an exploded perspective view of the semiconductor device which concerns on 2nd Embodiment of this invention.

Hereinafter, examples of embodiments of the present invention will be specifically described with reference to the drawings. In each of the referenced figures, the same parts are designated by the same reference numerals, and duplicate explanations regarding the same parts will be omitted in principle. In this specification, for the sake of simplification of description, by describing a symbol or a code that refers to an information, a signal, a physical quantity, an element or a part, etc., the information, a signal, a physical quantity, an element or a part corresponding to the symbol or the code is described. Etc. may be omitted or abbreviated. For example, the high-side transistor referred to by "TrH" described later (see FIG. 1) may be referred to as a high-side transistor TrH or abbreviated as transistor TrH, but they are all the same. Refers to things.

First, some terms used in the description of the embodiments of the present invention will be described. IC is an abbreviation for Integrated Circuit. The ground refers to a conductive portion having a reference potential of 0 V (zero volt) or the potential of 0 V itself. The potential of 0V may be referred to as the ground potential. In the embodiment of the present invention, the voltage shown without any particular reference represents the potential seen from the ground. Level refers to the level of potential, where a high level has a higher potential than a low level for any signal or voltage. For any signal or voltage, a signal or voltage at a high level means that the signal or voltage level is at a high level, and a signal or voltage at a low level means that the signal or voltage level is at a low level. Means that it is in. A level for a signal is sometimes referred to as a signal level, and a level for a voltage is sometimes referred to as a voltage level.

For any transistor configured as a FET (Field Effect Transistor) including a MOSFET, the on state means that the drain and source of the transistor are in a conductive state, and the off state means the drain of the transistor. And it means that there is a non-conduction state (interruption state) between the sources. The same applies to transistors that are not classified as FETs. Unless otherwise specified, the MOSFET may be understood as an enhancement type MOSFET. MOSFET is an abbreviation for "metal-oxide-semiconductor field-effect transistor". The on state or the off state may be understood as a term for a state between a drain and a source. That is, for any transistor configured as a FET, the on state of the transistor is synonymous with the on state between the drain and source of the transistor, and the off state of the transistor is synonymous with the off state between the drain and source of the transistor. be. Hereinafter, the on state and the off state of any transistor may be simply expressed as on and off. For any transistor, switching from the off state to the on state is expressed as turn-on, and switching from the on state to the off state is expressed as turn-off.

<< First Embodiment >>
The first embodiment of the present invention will be described. FIG. 1 shows the configuration of the herb bridge circuit HB according to the first embodiment of the present invention. The herb bridge circuit HB consists of transistors TrH and TrL, which are examples of two switching elements connected in series with each other. Each of the transistors TrH and TrL is configured as an N-channel MOSFET. The source of the transistor TrH and the drain of the transistor TrL are commonly connected to each other. In the semiconductor device including the herb bridge circuit HB, a predetermined DC voltage is applied between the source of the transistor TrL and the drain of the transistor TrH with the source of the transistor TrL on the low potential side. Therefore, the transistor TrH functions as a high-side transistor, and the transistor TrL functions as a low-side transistor. The voltage V _{DS_H} represents the drain-source voltage of the transistor TrH (the drain potential seen from the source potential), and the voltage V _{DS_L} represents the drain-source voltage (drain potential seen from the source potential) of the transistor TrL. show.

In the herb bridge circuit HB, a parallel diode whose forward direction is from the source of the transistor TrH to the drain may be connected in parallel to the transistor TrH, and similarly, the direction from the source to the drain of the transistor TrL is the forward direction. The parallel diode to be used may be connected in parallel to the transistor TrL.

The transistors TrH and TrL are composed of a wide gap semiconductor. Wide-gap semiconductors are, for example, SiC (silicon carbide), GaN (gallium nitride), Ga ₂ O ₃ (gallium oxide), and diamond. However, the transistors TrH and TrL may be made of Si (silicon) or GaAs (gallium arsenide).

As shown in FIG. 2, the transistors TrH and TrL have a parasitic capacitance and an internal gate resistance (internal resistance at the gate). In FIG. 2, the capacitance C _{GD_H} is the gate-drain capacitance of the _{transistor TrH, the capacitance C GS_H} is the gate-source capacitance of the _{transistor TrH, and the capacitance C DS_H} is the drain-source capacitance of the transistor TrH. The capacitances C _{GD_H} , C _{GS_H} and C _{DS_H} are parasitic capacitances existing inside the transistor TrH. In particular, the capacitance C _{GD_H} is referred to as the feedback capacitance of the transistor TrH, and _{the sum of the capacitances C GD_H and CGS_H} is referred to as the input capacitance of the transistor TrH. The resistor R _{GIN_H} is a parasitic resistance existing inside the transistor TrH, and is an internal gate resistance inevitably associated with the gate of the transistor TrH. The current flowing through the internal gate resistor R _{GIN_H} flows through the capacitance C _{GD_H} or C _{GS_H} .

In FIG. 2, the voltage V _{GS_H} represents the gate-source voltage of the transistor TrH. The voltage observed or applied from the outside of the transistor TrH, and the voltage generated at the gate of the transistor TrH with reference to the source potential of the transistor TrH corresponds to the _{voltage VGS_H.} On the other hand, the voltage V _{GSIN_H} represents the internal gate-source voltage of the transistor TrH and is equal to the voltage generated between the two poles of _{the capacitance C GS_H.} Considering including transient state, transistor TrH the internal gate - source voltage _{V GSIN_H} is turned on when it is positive predetermined threshold voltage _{V TH_H} or more, the internal gate - source voltage _{V GSIN_H} a predetermined threshold value It is turned off when the voltage _{is less than V TH_H.} Except for the transient state (that is, _{when the current flowing through the resistor R GIN_H} is zero), the transistor TrH is turned on when the gate-source voltage V _{GS_H is equal to} or higher than the positive predetermined threshold voltage V TH_H. It is turned off when the gate-source voltage V _{GS_H} is less than the predetermined threshold voltage V _{TH_H.}

Similarly, the capacitance C _{GD_L} is the gate-drain capacitance of the _{transistor TrL, the capacitance C GS_L} is the gate-source capacitance of the _{transistor TrL, and the capacitance C DS_L} is the drain-source capacitance of the transistor TrL. The capacitances C _{GD_L} , C _{GS_L} and C _{DS_L} are parasitic capacitances existing inside the transistor TrL. In particular, the capacitance C _{GD_L} is referred to as the feedback capacitance of the transistor TrL, and _{the sum of the capacitances C GD_L and CGS_L} is referred to as the input capacitance of the transistor TrL. The resistor R _{GIN_L} is a parasitic resistance existing inside the transistor TrL, and is an internal gate resistance inevitably associated with the gate of the transistor TrL. The current flowing through the internal gate resistor R _{GIN_L} flows through the capacitance C _{GD_L} or C _{GS_L} .

In FIG. 2, the voltage V _{GS_L} represents the gate-source voltage of the transistor TrL. The voltage observed or applied from the outside of the transistor TrL and generated at the gate of the transistor TrL with reference to the source potential of the transistor TrL corresponds to the _{voltage VGS_L.} On the other hand, the voltage V _{GSIN_L} represents the internal gate-source voltage of the transistor TrL and is equal to the voltage generated between the two poles of _{the capacitance C GS_L.} Considering including transient state, the transistor TrL is, the internal gate - source voltage _{V GSIN_L} is turned on when it is positive predetermined threshold voltage _{V TH_L} above, the internal gate - source voltage _{V GSIN_L} a predetermined threshold value It is turned off when the voltage _{is less than V TH_L.} Except for the transient state (that is, _{when the current flowing through the resistor R GIN_L} is zero), the transistor TrL is _{turned on when the gate-source voltage V GS_L is equal to} or higher than the positive predetermined threshold voltage V TH_L. It is turned off when the gate-source voltage V _{GS_L} is less than the predetermined threshold voltage V _{TH_L.} The threshold voltages V _{TH_H} and V _{TH_L} may or may not match each other.

Although not shown in FIG. 2, the connection node between the source of the transistor TrH and the drain of the transistor TrL is connected to a load such as a coil. Now, in a semiconductor device including a half-bridge circuit HB, consider a situation in which the transistor TrL is turned on while the transistor TrH is controlled to be kept off. At this time, the turn-on of the transistor TrL drain of the transistor TrH - to provide a change in the source voltage _{V DS_H,} current flows to the capacitor _{C GS_H} through the capacitor _{C GD_H} as shown in FIG. 3, thereby the capacitance _{C GS_H} When the voltage between both electrodes _{becomes equal to} or higher than the threshold voltage VTH_H, a phenomenon may occur in which the transistor TrH is erroneously turned on. This phenomenon is called false firing (wrong on).

The false ignition of the transistor TrH _{is caused by a surge-like positive voltage VGSIN_H} (hereinafter referred to as a positive gate surge). Positive gate surge current toward the source from the gate of the transistor TrH when the transistor TrL is turned on is generated by flowing through the capacitance C _{GS_H.}

In order to suppress such an erroneous arc (positive gate surge), a mirror clamp circuit capable of short-circuiting between the gate and the source of the transistor TrH is provided outside the transistor TrH, and the transistor is turned on when the transistor TrL is turned on. A method of short-circuiting between the gate and the source of the transistor is also considered. However, if the internal gate resistance R _{GIN_H} is large, there is a concern that the effect of the mirror clamp circuit will be diminished. In particular, when the transistor TrH is formed by using SiC in which the internal gate resistance R _{GIN_H tends to be large, the concern becomes large.}

[Surge countermeasure function]
Considering this, in the semiconductor device according to the present embodiment, the drain of the transistor TrH with the turn-on of the transistor TrL - detecting a change in the source voltage V _{DS_H,} transistor TrH a voltage corresponding to the change from the outside of the transistor TrH Apply to the gate of. As a result, the internal gate-source voltage _{VGSIN_H} is maintained at zero or near zero, thereby suppressing an erroneous ignition (positive gate surge) of the transistor TrH.

A configuration for suppressing an erroneous ignition (positive gate surge) of the transistor TrH will be examined. FIG. 4 shows a model circuit related to the transistor TrH when the transistor TrL is switched. In the model circuit of FIG. 4, the switching of the transistor TrL in the half-bridge circuit HB is simulated by applying a rectangular wave-shaped voltage as the drain-source voltage V _{DS_H.} In the model circuit of FIG. 4, VS _A represents a variable voltage source. Variable voltage source VS _A supplies a variable voltage _{V OA} relative to the source potential of the transistor TrH external transistor TrH to the gate of the transistor TrH. In the model circuit, the source potential of the transistor TrH is assumed to be zero.

Now, simulating the transistor TrL of the half-bridge circuit HB is turned on, the drain - Consider the situation where the source voltage _{V DS_H} increases momentarily, represents the current flowing through the capacitor _{C GD_H} at _{I A} in this situation. Then, the following equation (A1) holds from the circuit equation.
I _A x R _{GIN_H} + V _OA- V _{GSIN_H} = 0 ... (A1)

Assuming that no current flows through the _{capacitance C GS_H} in this situation, the _{internal gate-source voltage V GSIN_H,} which represents the voltage between the two poles of _{the capacitance C GS_H} , becomes zero, so the equation (A1) is equivalent to the following equation (A2). Is.
_{V OA = -I A × R GIN_H} ··· (A2)

_Then, if a _{"V GSIN_H} = _0", the current _{I A,} because represented by the product of the time derivative of the capacitance _{C GD_H} and the voltage _{V DS_H (dV DS_H / dt)} , holds the following formula (A3) .. This is because the variable voltage _{V OA} is taken the value of the right side of formula (A3), indicates that the current does not flow through the capacitor _{C GS_H.}
V _OA = -R _{GIN_H} x C _{GD_H} x (dV _{DS_H} / dt) ... (A3)

_{A variable voltage VOA} corresponding to the equation (A3) can be generated by using a differentiating circuit. FIG. 5 shows a model circuit including a differentiating circuit DIF _A which is an example of the variable voltage source VS _A. The differentiating circuit DIF _A is composed of a resistor R _DIFA , a capacitor C _DIFA and an operational amplifier A _DIFA . The inverting input terminal of the operational amplifier A _DIFA is connected to the drain of the transistor TrH via _{the capacitor C DIFA.} The non-inverting input terminal of the operational amplifier A _DIFA is connected to the source of the transistor TrH. Inverting input terminal and output terminal of the operational amplifier _{A Difa} operational amplifier _{A Difa} is connected via a resistor _{R Difa.} Then, the output terminal of the operational amplifier A _DIFA is connected to the gate of the transistor TrH. Voltage at the output terminal of the operational amplifier _{A Difa} corresponds to the variable voltage _{V OA.} As described above, since the resistor R _{GIN_H} is an internal gate resistor existing inside the transistor _TrH , the output terminal of the operational amplifier A _{DIFA is connected to the portion of the transistor TrH where the voltage V GSIN_H} should be applied via the internal gate resistor R _{GIN_H.} Will be done.

Drain - a current flowing through the capacitor _{C Difa} and resistor _{R Difa} in situations where the source voltage _{V DS_H} increases instantaneously, denoted by _{I A} '. The operational amplifier A _DIFA operates so as to make the potential difference between the inverting input terminal and the non-inverting input terminal zero. The function of the virtual short circuit due to the operational amplifier _{A Difa,} 'is, while the current _{I A' V OA = -R DIFA} × I A is the time derivative of the capacitance value and the voltage _{V DS_H} capacitor _{C DIFA (dV DS_H} / Since it is represented by the product of dt), the following equation (A4) holds. Note that in the model circuit of FIG. 5, and the polarity of the operational amplifier A inverting input terminal from going to the output terminal direction of the current I _A of _{Difa 'positive.
V OA = -R DIFA × I A} '
= -R _DIFA x C _DIFA x (dV _{DS_H} / dt) ... (A4)

_{Therefore, if} the resistance value of the resistor R DIFA and the capacitance value of the capacitor C _DIFA are determined so that the following equation (A5) is satisfied from the comparison of the above equations (A3) and (A4), the model of FIG. _No current flows through the capacitance C GS_H in the circuit.
R _DIFA x C _DIFA = R _{GIN_H} x C _{GD_H} ... (A5)

In the above equations (A1) to (A5), for convenience of explanation, a symbol representing a voltage is used as a symbol representing the voltage value of the voltage. The same applies to current, resistance, and the like. That is, in the above formulas (A1) to (A5),
The symbol V _OA represents the voltage value of the output voltage V _OA of the variable voltage source VS _A and the differentiating circuit DIF _A.
The symbol V _{GSIN_H} represents the voltage value of the internal gate-source voltage V _{GSIN_H.}
Symbol _I A, _{I A} ', respectively, the current _I A, _{I A'} represents the current value of,
The symbol R _{GIN_H} represents the resistance value of the internal gate resistor R _{GIN_H.}
The symbol C _{GD_H} represents the capacitance value of the gate-drain capacitance C _{GD_H.}
The symbol R _DIFA represents the resistance value of the resistor R _DIFA.
The symbol C _DIFA represents the capacitance value of the capacitor C _DIFA.
The symbol (dV _{DS_H} / dt) represents the value of the time derivative of the _{voltage V DS_H.}

_{In the half-bridge circuit HB, if a differentiating circuit DIF A} designed to satisfy the equation (A5) is used, false firing (positive gate surge) of the transistor TrH when the transistor TrL turns on is effectively suppressed. can.

Further, in the half-bridge circuit HB, when the differentiating circuit DIF _A is not used, a current from the source of the transistor TrH toward the gate _{flows through the capacitance C GS_H when the transistor TrL turns off, thereby causing a} surge-like negative voltage V. _{GSIN_H} (hereinafter referred to as negative gate surge) occurs. Negative gate surges in the transistor TrH can lead to destruction of the transistor TrH. _{However, if the differentiating circuit DIF A} whose constant is designed to satisfy the equation (A5) is used in the half-bridge circuit HB, a negative gate surge does not occur in the transistor TrH when the transistor TrL turns off. With the differentiation circuit DIF _A which is constant designed to satisfy equation (A5), at the time the change occurs in the voltage _{V DS_H,} the current flowing through the no capacitance _{C GS_H} relation to the polarity of the variation of the voltage _{V DS_H} remains zero Because it is done.

Of the transistors TrH and TrL, the positive and negative gate surges that can occur in the transistor TrH have been considered, but the same measures can be taken for the positive and negative gate surges that can occur in the transistor TrL.

In summary, the semiconductor device according to this embodiment has the following surge countermeasure functions. _{In the surge countermeasure function, by applying the differentiating circuit DIF A} to each of the transistors TrH and TrL, positive and negative gate surges are generated in the transistor TrH, and positive and negative gate surges are generated in the transistor TrL. Suppress.

[High-speed drive function]
On the other hand, when paying attention to the switching of the transistor TrH or TrL, it is also important to perform the switching at high speed. For the purpose of embodying the description, the switching of the transistor TrH will be considered by focusing on the transistor TrH among the transistors TrH and TrL. By supplying a rectangular wavy voltage between the gate and the source of the transistor TrH, the transistor TrH can be turned on and off alternately. If a rectangular wavy voltage can be applied to the internal gate-source voltage _{VGSIN_H} without delay, switching will occur at high speed. However (see FIG. 6), if no measures are taken, the voltage drop will occur in _{the resistance component (including the internal gate resistance R GIN_H} ) located between the rectangular wave-shaped voltage supply source and the gate of the transistor TrH. , Switching (switching between on / off) takes time.

Consider a configuration for high-speed switching of Transis TrH. FIG. 7 shows a model circuit related to the transistor TrH when switching the transistor TrH. In the model circuit of FIG. 7, VS _B represents a variable voltage source. Variable voltage source VS _B supplies a variable voltage _{V OB} relative to the source potential of the transistor TrH external transistor TrH to the gate of the transistor TrH. In the model circuit of FIG. 7, a DC voltage is applied between the gate and the source of the transistor TrH, and it is assumed that the source potential of the transistor TrH is zero. Further, in the model circuit of FIG. 7, the capacitances C _{GD_H} and C _{DS_H} are ignored for the sake of simplification of the description. As shown in FIG. 8, the variable voltage V _OB corresponds to the sum of the rectangular wavy voltage V _{OB 1} and the variable voltage V _{OB 2.} A variable voltage source VS _B can be formed by a series circuit of a voltage generating unit VS _B1 that outputs a rectangular wavy voltage V _OB1 and a voltage source VS _B2 that outputs a variable voltage V _OB2.

Now, the model circuit of FIG. 7, considering that turning on the transistor TrH by supplying the current I _B from the variable voltage source VS _B toward the gate of the transistor TrH. In the model circuit of FIG. 7, first, the circuit equation according to the equation (B1) holds.
_{V GSIN_H = V OB -I B ×} R GIN_H ··· (B1)

At this time, in _{order to obtain} "V GSIN_H = V _OB1 ", since "V _OB = V _OB1 + V _OB2 ", the following equation (B2) may be satisfied.
_{V OB2 = I B × R GIN_H} ··· (B2)

In the model circuit of FIG. 7, the current _{I B} is represented by the product of the time derivative of the capacitance _{C GS_H} and the voltage _{V GSIN_H (dV GSIN_H / dt)} . Therefore, if the variable voltage V _OB satisfies the following equation (B3), the optimum "V _{GSIN_H} = V _OB1 " for high-speed switching is achieved.
V _OB = V _OB1 + V _{OB2
= V OB1 + I B × R} GIN_H
= V _OB1 + R _{GIN_H} × C _{GS_H} × (dV _{GSIN_H} / dt) ・ ・ ・ (B3)

_{A variable voltage VOB} corresponding to Eq. (B3) can be generated by using a differentiating circuit. FIG. 9 shows a model circuit including a differentiating circuit DIF _B which is an example of the variable voltage source VS _B. The differentiating circuit DIF _B includes a resistor R _DIFB , a capacitor C _DIFB, and an operational amplifier A _DIFB . The inverting input terminal of the operational amplifier A _DIFB is connected to the source of the transistor TrH via _{the capacitor C DIFB.} The non-inverting input terminal of the operational amplifier _{A DIFB} is connected to the voltage generator _{VS B1,} rectangular waveform voltage _{V OB1} relative to the source potential of the transistor TrH is input to the non-inverting input terminal of the operational amplifier _{A DIFB.} Inverting input terminal and output terminal of the operational amplifier _{A DIFB} of the operational amplifier _{A DIFB} is connected via a resistor _{R DIFB.} Then, the output terminal of the operational amplifier A _DIFB is connected to the gate of the transistor TrH. Voltage at the output terminal of the operational amplifier _{A DIFB} corresponds to the variable voltage _{V OB.} As described above, since the resistor R _{GIN_H} is an internal gate resistor existing inside the transistor _TrH , the output terminal of the operational amplifier A _{DIFB is connected to the portion of the transistor TrH where the voltage V GSIN_H} should be applied via the internal gate resistor R _{GIN_H.} Will be done.

In the model circuit of FIG. 9 represents a current flowing through the resistor _{R DIFB} at _{I B} '. The operational amplifier A _DIFB operates so as to make the potential difference between the inverting input terminal and the non-inverting input terminal zero. The function of the virtual short circuit due to the operational amplifier _{A DIFB, V OB = V OB1} + I B '× R DIFB, are, on the one hand the current _{I B'} is the time derivative of the capacitance value and the voltage _{V OB1} of the capacitor _{C DIFB (dV OB1} Since it is expressed as the product of / dt), the following equation (B4) holds. Note that in the model circuit of FIG. 9, and the polarity of the operational amplifier A direction of the current I _B toward the inverting input terminal from the output terminal of _{DIFB 'positive.
V OB = V OB1 + I B} '× R DIFB
= V _OB1 + R _DIFB × C _DIFB × (dV _OB1 / dt) ・ ・ ・ (B4)

_{Therefore, if} the resistance value of the resistor R DIFB and the capacitance value of the capacitor C _DIFB are determined so that the following equation (B5) is satisfied from the comparison of the above equations (B3) and (B4), the model of FIG. 9 In the circuit, the optimum "V _{GSIN_H} = V _OB1 " for high-speed switching will be achieved.
R _DIFB x C _DIFB = R _{GIN_H} x C _{GS_H} ... (B5)

In the model circuits of FIGS. 7 and 9, the existence of the gate-drain capacitance C _{GD_H} of the transistor TrH is ignored, but in order to actually change the internal gate-source voltage V _{GSIN_H} , only the _{capacitance C GS_H is used.} It is also necessary to charge and discharge the capacitance C _{GD_H.} Therefore, instead of the above equation (B5), the _{values of the resistor R DIFB} and the capacitor C _DIFB may be determined so that the following equation (B5') is satisfied. As described above, "(C _{GS_H} + C _{GD_H} )" corresponds to the input capacitance of the transistor TrH.
R _DIFB x C _DIFB = R _{GIN_H} x (C _{GS_H} + C _{GD_H} ) ... (B5')

In the above equations (B1) to (B5) and (B5'), for convenience of explanation, a symbol representing a voltage is used as a symbol representing the voltage value of the voltage. The same applies to current, resistance, and the like. That is, in the above formulas (B1) to (B5) and (B5'),
The symbol V _OB represents the voltage value of the output voltage V _OB of the variable voltage source VS _B and the differentiating circuit DIF _B.
The symbols V _OB1 and V _OB2 represent the voltage values of the voltages V _OB1 and V _{OB 2, respectively.}
The symbol V _{GSIN_H} represents the voltage value of the internal gate-source voltage V _{GSIN_H.}
Symbol _I B, _{I B} ', respectively, the current _I B, _{I B'} represents the current value of,
The symbol R _{GIN_H} represents the resistance value of the internal gate resistor R _{GIN_H.}
The symbol C _{GS_H} represents the capacitance value of the gate-source capacitance C _{GS_H.}
The symbol C _{GD_H} represents the capacitance value of the gate-drain capacitance C _{GD_H.}
The symbol R _DIFB represents the resistance value of the resistor R _DIFB.
The symbol C _DIFB represents the capacitance value of the capacitor C _DIFB.
The symbol (dV _{GSIN_H} / dt) represents the value of the time derivative of the _{voltage V GSIN_H.}
The symbol (dV _OB1 / dt) represents the value of the time derivative of the _{voltage V OB1.}

_{In the half-bridge circuit HB, by using the differentiating circuit DIF B} whose constant is designed to satisfy the equation (B5) or the equation (B5'), the transistor TrH can be driven at high speed (that is, switched at high speed). ..

Of the transistors TrH and TrL, the circuit configuration for driving the transistor at high speed has been described by paying attention to the transistor TrH, but the same circuit configuration can be adopted for the transistor TrL.

In summary, the semiconductor device according to this embodiment has the following high-speed drive functions. In the high-speed drive function, the differentiating circuit DIF _B is applied to each of the transistors TrH and TrL to drive the transistors TrH and TrL at high speed, respectively.

The first embodiment includes the following Examples EX1-1-1 to EX1_4. The detailed circuit and the like of the semiconductor device according to the first embodiment will be described in Examples EX1_1 to EX1_4. Unless otherwise specified and without contradiction, the above-mentioned matters in the first embodiment are applied to the following Examples EX1_1 to EX1_4, and in each embodiment, the matters inconsistent with the above-mentioned matters in the first embodiment. May give priority to the description in each embodiment. As long as there is no contradiction, the matters described in any of the examples EX1-1 to EX1_4 can be applied to any other embodiment (that is, any two or more of the plurality of examples). It is also possible to combine examples).

[Example EX1_1]
Example EX1_1 will be described. FIG. 10 shows the overall configuration of the semiconductor device 1 according to the example EX1_1. The DC voltage source 2 and the external power supply 3 are connected to the semiconductor device 1. The semiconductor device 1 includes a power module PM and a control module CM. The power module PM includes a half-bridge circuit HB including a series circuit of the above-mentioned transistors TrH and TrL.

The semiconductor device 1 is provided with terminals P _TM , O1 _TM , O2 _TM , N _TM , DH _TM , GH _TM , SH _TM , GL _TM, and SL _TM . The terminals O1 _TM and O2 _TM correspond to the output terminals of the power module PM. The drain of the transistor TrH is connected to the terminal _{P TM} and _{DH TM.} The source of the transistor TrL is connected to the terminal _{N TM} and _{SL TM.} The source of the transistor TrH and the drain of the transistor TrL are connected to each other and are also commonly connected to the _{terminals O1 TM} , O2 _TM and SH _TM. The gate of the transistor TrH is _{connected to the terminal GH TM,} and the gate of the transistor TrL is connected to the _{terminal GL TM.}

A predetermined DC voltage from the DC voltage source 2, and a terminal N _TM to the low potential side is applied between the terminals N _TM and P _TM. Terminal _NTM is grounded. The power module PM is controlled by the control module CM, for example, the DC power supplied from the DC voltage source 2 is converted into AC power, and the obtained AC power is converted from the output terminals O1 _TM and O2 _TM to the output terminals O1 _TM and O2 _TM. It supplies a load such as a coil connected to. Here, the number of output terminals of the power module PM is two, but the number of output terminals of the power module PM may be one or three or more.

The control module CM operates based on the electric power supplied from the external power source 3. Here, it is assumed that the DC voltage _VIN is supplied to the control module CM from the external power supply 3. The external power supply 3 and the DC voltage source 2 can be a common voltage source.

The control module CM includes a drive control unit 10H which is a high-side drive control unit, a drive control unit 10L which is a low-side drive control unit, a control signal generation unit 20, and a power supply circuit 30.

The drive control unit 10H is _{connected to the terminals DH TM} , GH _TM and SH _TM , and switches and drives the transistor TrH by controlling the on / off of the transistor TrH according to the control signal supplied from the control signal generation unit 20. The drive control unit 10L is _{connected to the terminals SH TM} , GL _TM, and SL _TM , and switches and drives the transistor TrL by controlling the on / off of the transistor TrL according to the control signal supplied from the control signal generation unit 20. The control signal generation unit 20 generates a control signal for switching and driving the transistors TrH and TrL based on the signal supplied from the external device of the semiconductor device 1. The power supply circuit 30 generates a power supply voltage (drive voltage) necessary for driving the drive control unit 10H, the drive control unit 10L, and the control signal generation unit 20 based on _{the DC voltage VIN} from the external power supply 3, and the drive control unit. It is supplied to 10H, the drive control unit 10L, and the control signal generation unit 20. The power supply circuit 30 is preferably an isolated power supply circuit.

FIG. 11 shows a specific circuit configuration example of the semiconductor device 1. In the semiconductor device 1 of FIG. 11, the high-side drive control unit 10H includes a high-side variable voltage source 110H, and the low-side drive control unit 10L includes a low-side variable voltage source 110L.

The variable voltage source 110H for the high side is a voltage source that supplies a variable voltage between the gate and the source of the transistor TrH, and is an operational amplifier 111H, a voltage generating unit 112H, a resistor 113H, a capacitor 114H, a capacitor 115H, and an output unit 116H. To be equipped. The resistor 113H functions as a feedback resistor. The operational amplifier 111H has two input terminals, an inverting input terminal, a non-inverting input terminal, and an output terminal. The operational amplifier 111H outputs a voltage signal obtained by amplifying the voltage (potential difference) between the inverting input terminal and the non-inverting input terminal from its own output terminal, so that the operational amplifier 111H cooperates with the feedback resistor 113H to inverting input terminal and non-inverting input terminal. Bring the voltage (potential difference) between them close to zero. The operational amplifier 111H, the resistor 113H, and the capacitor 114H _{form a first differentiating circuit for the high side corresponding to the differentiating circuit DIF A} in FIG. 5, and the operational amplifier 111H, the resistor 113H, and the capacitor 115H _{correspond to the differentiating circuit DIF B} in FIG. A second differentiating circuit for the high side is formed.

The inverting input terminal of the operational amplifier 111H is commonly connected to one end of the resistor 113H, one end of the capacitor 114H, and one end of the capacitor 115H. The other end of the resistor 113H is connected to the output terminal of the operational amplifier 111H. The other end of the capacitor 114H is connected to the terminal _{DH TM} (i.e. is connected to the drain of the transistor TrH through the terminal _{DH TM).} The other end of the capacitor 115H is connected to the terminal _{SH TM} (i.e. is connected to the source of the transistor TrH through a terminal _{SH TM).}

The voltage generating unit 112H is inserted between the non-inverting input terminal and the terminal _{SH TM} of the operational amplifier 111H. Voltage generator 112H generates a voltage _{V PLS_H} based on the control signal CNT_H, it supplies a voltage _{V PLS_H} to the non-inverting input terminal of the operational amplifier 111H based on the potential (i.e. the source potential of the transistor TrH) terminal _{SH TM.}

The voltage V _{PLS_H} is a rectangular wavy voltage for alternately turning on or off the transistors TrH, and _{the levels of the voltage V PLS_H} are alternately low level, which is the first predetermined level, and high, which is the second predetermined level. Become a level. The value of the low-level voltage V _{PLS_H} is zero when viewed from the source potential of the transistor TrH. Therefore, when the voltage V _{PLS_H} is at the low level, the potential at the non-inverting input terminal of the operational amplifier 111H coincides with the source potential of the transistor TrH. When the voltage V _{PLS_H} is at a high level, the potential at the non-inverting input terminal of the operational amplifier 111H becomes higher than the source potential of the transistor TrH by the amplitude of _{the voltage V PLS_H.} The amplitude of the voltage _{V PLS_H} is greater than the threshold voltage _{V TH_H} transistor TrH. Therefore, in the transistor TrH, when the gate potential is _{higher by the amplitude of the voltage V PLS_H} when viewed from the source potential, the transistor TrH is turned on. For example, the threshold voltage V _{TH_H} is 2.7 V, and the amplitude of the _{voltage V PLS_H is 18 V.} It is possible that the low-level voltage _{VPLS_H} has a positive or negative minute voltage value when viewed from the source potential of the transistor TrH.

The voltage at the output terminal of the operational amplifier 111H is _{represented by the symbol “VO_H} ”. Further, the output unit of the variable voltage source 110H is represented by the reference numeral “116H”. In the configuration of FIG. 11, the output unit 116H is also the output unit of the differentiating circuit constituting the variable voltage source 110H, and is equal to the output terminal of the operational amplifier 111H. The output unit 116H is connected to the _{terminal GH TM.}

The variable voltage source 110L for the low side is a voltage source that supplies a variable voltage between the gate and the source of the transistor TrL, and includes an operational amplifier 111L, a voltage generating unit 112L, a resistor 113L, a capacitor 114L, a capacitor 115L, and an output unit 116L. Be prepared. The resistor 113L functions as a feedback resistor. The operational amplifier 111L has two input terminals, an inverting input terminal, a non-inverting input terminal, and an output terminal. The operational amplifier 111L outputs a voltage signal obtained by amplifying the voltage (potential difference) between the inverting input terminal and the non-inverting input terminal from its own output terminal, so that the operational amplifier 111L cooperates with the feedback resistor 113L to display the inverting input terminal and the non-inverting input terminal. Bring the voltage (potential difference) between them close to zero. The operational amplifier 111L, the resistor 113L and the capacitor 114L _{form a first differentiating circuit for the low side corresponding to the differentiating circuit DIF A} of FIG. 5, and the operational amplifier 111L, the resistor 113L and the capacitor 115L form the low side corresponding to _{the differentiating circuit DIF B of FIG.} Second differentiating circuit is formed.

The inverting input terminal of the operational amplifier 111L is commonly connected to one end of the resistor 113L, one end of the capacitor 114L, and one end of the capacitor 115L. The other end of the resistor 113L is connected to the output terminal of the operational amplifier 111L. The other end of the capacitor 114L is connected to the terminal _{SH TM} (i.e. is connected to the drain of the transistor TrL through a terminal _{SH TM).} The other end of the capacitor 115L is connected to the terminal _{SL TM} (i.e. connected to the source of transistor TrL through the terminal _{SL TM).}

The voltage generating unit 112L is inserted between the non-inverting input terminal and the terminal _{SL TM} of the operational amplifier 111L. Voltage generating unit 112L generates the voltage _{V PLS_L} based on the control signal CNT_L, supplies a voltage _{V PLS_L} to the non-inverting input terminal of the operational amplifier 111L in reference to the potential (i.e. the source potential of the transistor TrL) terminal _{SL TM.}

The voltage V _{PLS_L} is a rectangular wavy voltage for alternately turning on or off the transistors TrL, and _{the levels of the voltage V PLS_L} are alternately low level, which is the third predetermined level, and high, which is the second predetermined level. Become a level. The value of the low level voltage V _{PLS_L} is zero when viewed from the source potential of the transistor TrL. Therefore, when the voltage V _{PLS_L} is at the low level, the potential at the non-inverting input terminal of the operational amplifier 111L coincides with the source potential of the transistor TrL. When the voltage V _{PLS_L} is at a high level, the potential at the non-inverting input terminal of the operational amplifier 111L becomes higher than the source potential of the transistor TrL by the amplitude of _{the voltage V PLS_L.} The amplitude of the voltage _{V PLS_L} is greater than the threshold voltage _{V TH_L} of the transistor TrL. Therefore, in the transistor TrL, when the gate potential is _{higher by the amplitude of the voltage V PLS_L} when viewed from the source potential, the transistor TrL is turned on. For example, the threshold voltage V _{TH_L} is 2.7 V, and the amplitude of the _{voltage V PLS_L is 18 V.} It is possible that the low-level voltage _{VPLS_L} has a positive or negative minute voltage value when viewed from the source potential of the transistor TrL.

The voltage at the output terminal of the operational amplifier 111L is _{represented by the symbol “VO_L} ”. Further, the output unit of the variable voltage source 110L is represented by the reference numeral “116L”. In the configuration of FIG. 11, the output unit 116L is also the output unit of the differentiating circuit constituting the variable voltage source 110L, and is equal to the output terminal of the operational amplifier 111L. The output unit 116L is connected to the _{terminal GL TM.}

_{The control signal generation unit 20 generates control signals CNT_H and CNT_L based on the signal S IN} supplied from the external device of the semiconductor device 1, and outputs the control signals CNT_H and CNT_L to the voltage generation units 112H and 112L, respectively. Each of the control signals CNT_H and CNT_L is a binarized signal that takes a high level or a low level signal level. When the control signal CNT_H is high level and low level, the voltage V _{PLS_H} is also high level and low level, respectively. When the control signal CNT_L is high level and low level, the voltage V _{PLS_L} is also high level and low level, respectively.

The power supply circuit 30 generates power supply voltages VCS1_H, VCS2_H, VCS1_L and VCS2_L based _{on the DC voltage VIN} supplied from the external power supply 3.

The power supply voltages VCS1_H and VCS2_H are the power supply voltages on the positive and negative sides of the operational amplifier 111H, and the operational amplifier 111H is driven based on the power supply voltages VCS1_H and VCS2_H (VCC1_H> VCS2_H). Therefore, the output voltage _{VO_H} of the operational amplifier 111H has a potential of not less than or equal to the negative side power supply voltage VCS2_H and not more than or equal to the positive side power supply voltage VCS1_H. In order to achieve surge capability and high-speed driving function for the transistor TrH, so that the voltage _{V O_H} viewed from the source potential of the transistor TrH can also have a positive even potential negative potential, the voltage value of the power supply voltage VCC1_H and VCC2_H set Will be done.

The power supply voltages VCS1_L and VCS2_L are the power supply voltages on the positive and negative sides of the operational amplifier 111L, and the operational amplifier 111L is driven based on the power supply voltages VCS1_L and VCS2_L (VCC1_L> VCS2_L). Thus, the output voltage _{V O_L} operational amplifier 111L have a power supply voltage and the positive supply voltage VCC1_L following potential than VCC2_L negative side. In order to achieve surge capability and high-speed driving function for the transistor TrL, so that the voltage _{V O_L} viewed from the source potential of the transistor TrL can be had positive even potential negative potential, the voltage value of the power supply voltage VCC1_L and VCC2_L set Will be done.

Although not particularly shown, the power supply voltage for the voltage generating units 112H and 112L is also generated in the power supply circuit 30 based on the _{DC voltage VIN.}

--Low side off control section (Fig. 12) -
With reference to FIG. 12, consider a low-side-off control section in which the transistor TrL is maintained in the off state by maintaining the _{voltage V PLS_L at a low level.} In the low side-off control section, the transistor TrH is switched by varying _{the voltage V PLS_H between low and high levels.}

Operational amplifier 111H, by the function of the resistor 113H and the differentiating circuit comprising a capacitor 115H, the output voltage _{V O_H} becomes one to rectangular waveform voltage _{V PLS_H} superimposed the voltage component corresponding to the change of the voltage _{V PLS_H.} Voltage component corresponding to the change of the voltage _{V PLS_H} has a positive or negative voltage value in a transient state immediately after the variation of the voltage _{V PLS_H,} after change of the voltage _{V PLS_H,} in a stable state in which sufficient time has elapsed, It becomes zero. Thus, the output voltage _{V O_H} in a stable state corresponds to the voltage _{V PLS_H.} In steady state, the transistor TrH if the voltage _{V PLS_H} at a low level is off, the transistor TrH if the voltage _{V PLS_H} high level is on.

That is, when the variable voltage source _{110H switches the transistor TrH by supplying the rectangular wave-shaped voltage V PLS_H} between the gate and the source of the transistor TrH _{, the variable voltage source 110H responds to} the change of the voltage V _{PLS_H with respect to the rectangular wave-shaped voltage V PLS_H.} A drive voltage is generated by superimposing the voltage components, and the drive voltage is applied as an output voltage _{VO_H} between the gate and the source of the transistor TrH (in other words, the drive voltage is applied to the gate of the transistor TrH with reference to the source potential of the transistor TrH. ). Due to the superimposed voltage component, _{the waveform of the internal gate-source voltage V GSIN_H} matches or approximates the waveform of the voltage V _{PLS_H} , including the transient state, and high-speed driving of the transistor TrH is realized.

On the other hand, in the low-side-off control section, switching of the transistor TrH causes a voltage change between the drain and the source of the transistor TrL. However, this time, the variable voltage source 110L includes an operational amplifier 111L, by the function of the resistor 113L and the differentiating circuit comprising a capacitor 114L, the drain of the transistor TrL - generates a voltage corresponding to the voltage change between the source as the output voltage _{V O_L,} The output voltage _{VO_L} is applied between the gate and the source of the transistor TrL (in other words, it is applied to the gate of the transistor TrL with reference to the source potential of the transistor TrL). As a result, positive and negative gate surges with respect to the transistor TrL due to switching of the transistor TrH are suppressed.

--High side off control section (Fig. 13)--
With reference to FIG. 13, consider a high-side-off control section in which the transistor TrH is maintained in the off state by maintaining the _{voltage V PLS_H at a low level.} In the high side-off control section, the transistor TrL is switched by varying _{the voltage V PLS_L between low and high levels.}

Operational amplifier 111L, by the function of the resistor 113L and the differentiating circuit comprising a capacitor 115L, the output voltage _{V O_L} becomes one to rectangular waveform voltage _{V PLS_L} superimposed the voltage component corresponding to the change of the voltage _{V PLS_L.} Voltage component corresponding to the change of the voltage _{V PLS_L} has a positive or negative voltage value in a transient state immediately after the variation of the voltage _{V PLS_L,} after change of the voltage _{V PLS_L,} in a stable state in which sufficient time has elapsed, It becomes zero. Thus, the output voltage _{V O_L} in a stable state corresponds to the voltage _{V PLS_L.} In steady state, the transistor TrL if the voltage _{V PLS_L} at a low level is off, the transistor TrL if the voltage _{V PLS_L} high level is on.

That is, when the variable voltage source 110L _{switches the transistor TrL by supplying the rectangular wave-shaped voltage V PLS_L} between the gate and the source of the transistor TrL _{, the variable voltage source 110L responds to} the change of the voltage V _{PLS_L with respect to the rectangular wave-shaped voltage V PLS_L.} A drive voltage is generated by superimposing the voltage components, and the drive voltage is applied as an output voltage _{VO_L} between the gate and the source of the transistor TrL (in other words, the drive voltage is applied to the gate of the transistor TrL with reference to the source potential of the transistor TrL. ). Due to the superimposed voltage component, _{the waveform of the internal gate-source voltage V GSIN_L} , including the transient state, matches or approximates the waveform of the voltage V _{PLS_L} , and high-speed driving of the transistor TrL is realized.

On the other hand, in the high side-off control section, the switching of the transistor TrL causes a voltage change between the drain and the source of the transistor TrH. _{However, at this time, the variable voltage source 110H} generates a voltage corresponding to the voltage change between the drain and the source of the transistor TrH as the output voltage VO_H by the function of the differential circuit including the operational capacitor 111H, the resistor 113H and the capacitor 114H. The output voltage _{VO_H} is applied between the gate and the source of the transistor TrH (in other words, it is applied to the gate of the transistor TrH with reference to the source potential of the transistor TrH). As a result, positive and negative gate surges with respect to the transistor TrH due to switching of the transistor TrL are suppressed.

The operational amplifiers 111H and 111L, respectively, need to respond to transient changes associated with switching at high speed. Therefore, it is desirable that each of the operational amplifiers 111H and 111L is composed of a current feedback type operational amplifier capable of realizing high-speed operation. However, it is also possible to configure each of the operational amplifiers 111H and 111L with operational amplifiers that are not classified as current feedback type operational amplifiers.

--Constant design--
The constant design of the variable voltage source 110H will be described. The capacitance values of the capacitors 114H and 115H are _{represented by "C 114H} " and "C _115H ", respectively, and the resistance value of the resistor _113H is represented by "R 113H". In the transistor TrH, gate - drain capacitance _{C GD_H,} gate - represented by the capacitance value of the source capacitance _{C GS_H, respectively, "C GD_H""C GS_H} " ( see FIG. 2). Further, the resistance value of the gate resistance of the transistor TrH is represented _{by "RG_H".}

The gate resistance (control electrode resistance) of the transistor TrH having a resistance value R _{G_H typically refers to the internal gate resistance R GIN_H} itself of the transistor TrH (see FIG. 15). The internal gate resistance R _{GIN_H} is, for example, about 1Ω. However, as shown in FIG. 14, a resistor R _{EX_H} may be externally connected to the gate of the transistor TrH. In this case, will be the resistance R _{EX_H} to be externally connected to the gate of the transistor TrH, the series combined resistance of the inner gate resistor R _{GIN_H} functions as a gate resistance of a transistor TrH, of the series combined resistance The resistance value is _{understood as the resistance value RG_H} (see FIG. 15). In FIG 14, although the resistance _{R EX_H} between the gate terminal _{GH TM} and transistor TrH provided, sometimes the resistance _{R EX_H} is provided between the terminal _{GH TM} and an output section 116H.

_{The values C 114H} and R _113H may be designed based on the characteristics of the transistor TrH so that the product (C _114H × R _113H ) and the product (C _{GD_H} × _{RG_H) match.} As a result, the positive and negative gate surges related to the transistor TrH are properly suppressed. The coincidence of their products is understood as a concept having a certain width including an error. Related to this, the product (C _114H x R _113H ) and the product (C _{GD_H} x _{RG_H} ) do not necessarily have to be _{exactly the same} , and the product (C 114H x R _113H ) is the product (C _{GD_H).} It may have a value close to the product (C _{GD_H} × _{RG_H ) based on × RG_H).} Even in this case, the effect of suppressing positive and negative gate surges related to the transistor TrH can be obtained. In summary, for example, as shown in FIG. 15, it is _{preferable that} “(C 114H × R _113H ) = k _H1 (C _{GD_H} × _{RG_H} )”. The coefficient k _H1 has a predetermined value in the range of 0.5 or more and 1.5 or less, for example.

The capacitance value of the gate-source capacitance C _{GS_H} of the transistor TrH or the capacitance value of the input capacitance of the transistor TrH (that is, the sum of the _{capacitances C GS_H} and C _{GD_H} ) is represented by _{"CG_H" for convenience.} In this case, the _{values C 115H} and R _113H may be designed based on the characteristics of the transistor TrH so that _{the product (C 115H} × R _113H ) and the product ( _{CG_H} × _{RG_H) match.} As a result, high-speed driving of switching of the transistor TrH is realized. The coincidence of their products is understood as a concept having a certain width including an error. Although in this connection the product _{(C 115H} × _{R 113H)} and product _{(C G_H} × _{R G_H)} and completely is not always necessary to match the product _{(C 115H} × _{R 113H)} is the product _{(C G_H} It may have a value close to the product ( _{CG_H} × _{RG_H ) based on × RG_H).} Even in this case, high-speed driving of the transistor TrH is realized. In summary, for example, as shown in FIG. 15, it is _{preferable that} “(C 115H × R _113H ) = k _H2 ( _{CG_H} × R _{G_H} )”. The coefficient k _H2 has a predetermined value in the range of 0.5 or more and 1.5 or less, for example.

The same constant design is applied to the low-side variable voltage source 110L. The capacitance values of the capacitors 114L and 115L are _{represented by "C 114L} " and "C _115L ", respectively, and the resistance value of the resistor _113L is represented by "R 113L". In the transistor TrL, the gate - drain capacitance _{C GD_L,} gate - represented by the capacitance value of the source capacitance _{C GS_L, respectively, "C GD_L""C GS_L} " ( see FIG. 2). Further, the resistance value of the gate resistance of the transistor TrL is represented _{by "RG_L".}

The gate resistance (control electrode resistance) of the transistor TrL having a resistance value R _{G_L typically refers to the internal gate resistance R GIN_L} itself of the transistor TrL (see FIG. 15). The internal gate resistance R _{GIN_L} is, for example, about 1Ω. However, as shown in FIG. 14, a resistor R _{EX_L} may be externally connected to the gate of the transistor TrL. In this case, will be the resistance R _{EX_L} to be externally connected to the gate of the transistor TrL, the series combined resistance of the inner gate resistor R _{GIN_L} functions as a gate resistance of the transistor TrL, the series combined resistance The resistance value is _{understood as the resistance value RG_L} (see FIG. 15). In FIG 14, although the resistance _{R EX_L} between the gate terminal _{GL TM} and transistor TrL is provided, there is also the resistance _{R EX_L} is provided between the terminal _{GL TM} and an output unit 116L.

_{The values C 114L} and R _113L may be designed based on the characteristics of the transistor TrL so that the product (C _114L × R _113L ) and the product (C _{GD_L} × _{RG_L) match.} As a result, the positive and negative gate surges related to the transistor TrL are properly suppressed. The coincidence of their products is understood as a concept having a certain width including an error. Related to this, the product (C _114L x R _113L ) and the product (C _{GD_L} x RG_L) do not necessarily have to be _{exactly the same , and the product (C 114L} x R _113L ) is the product (C _{GD_L).} It may have a value close to the product (C _{GD_L} × _{RG_L ) based on × RG_L).} Even in this case, the effect of suppressing positive and negative gate surges related to the transistor TrL can be obtained. In summary, for example, as shown in FIG. 15, it is preferable that _{“(C 114L × R 113L} ) = k _L1 (C _{GD_L} × _{RG_L} )”. The coefficient k _L1 has a predetermined value in the range of 0.5 or more and 1.5 or less, for example.

The capacitance value of the gate-source capacitance C _{GS_L} of the transistor TrL or the capacitance value of the input capacitance of the transistor TrL (that is, the sum of the _{capacitances C GS_L} and C _{GD_L} ) is represented by _{"CG_L" for convenience.} In this case, the _{values C 115L} and R _113L may be designed based on the characteristics of the transistor TrL so that _{the product (C 115L} × R _113L ) and the product ( _{CG_L} × _{RG_L) match.} As a result, high-speed driving of switching of the transistor TrL is realized. The coincidence of their products is understood as a concept having a certain width including an error. Related to this, the product (C _115L x R _113L ) and the product ( _{CG_L} x RG_L) do not necessarily have to be _{exactly the same , and the product (C 115L} x R _113L ) is the product ( _{CG_L).} It may have a value close to the product ( _{CG_L} × _{RG_L ) based on × RG_L).} Even in this case, high-speed driving of the transistor TrL is realized. In summary, for example, as shown in FIG. 15, it is _{preferable that} “(C 115L × R _113L ) = k _L2 ( _{CG_L} × _{RG_L} )”. The coefficient k _L2 has a predetermined value in the range of 0.5 or more and 1.5 or less, for example.

[Example EX1_2]
Example EX1_2 will be described. A specific capacitor provided in the semiconductor device 1 is referred to as a target capacitor for convenience.

The target capacitor may be a capacitive element composed of a ceramic capacitor or the like, but the target capacitor may be formed by using the parasitic capacitance of the diode. In the configuration of FIG. 11, each of the capacitors 114H and 114L can be the target capacitors. _{The capacitor 114H is provided to simulate the behavior of the capacitance CGD_H} , which is the parasitic capacitance of the transistor TrH, and forms the capacitor 114H with the parasitic capacitance of the diode rather than forming the capacitor 114H with a ceramic capacitor or the like. There is a possibility that the gate surge of the transistor TrH can be suppressed more effectively. The same applies to the capacitor 114L.

An example of a configuration in which a target capacitor is formed using the parasitic capacitance of a diode is given. Now, as shown in FIG. 16 (a), consider a case where one end of the target capacitor C _X is the other end of the target capacitor C _X is connected to the node ND2 together connected to node ND1. When capacitor 114H is the subject capacitor _{C X,} the node ND1 corresponds to the inverting input terminal of the operational amplifier 111H, node ND2 is equivalent to the terminal _{DH TM.} When capacitor 114L is the subject capacitor _{C X,} the node ND1 corresponds to the inverting input terminal of the operational amplifier 111L, node ND2 is equivalent to the terminal _{SH TM.}

For example, as shown in FIG. 16 (b), provided a diode _{D X} in the semiconductor device 1, may be used parasitic capacitance PC _X of diode _{D X} as a target capacitor _{C X.} In FIG. 16 (b), the anode of the diode _{D X,} the cathode, respectively, are connected to the node ND1, ND2. As shown in FIGS. 16 (c) or 16 (d), the parasitic capacitance PC _X may be used as the capacitor 114H or 114L, which is an example of the target capacitor C _X.

If required the interruption of direct current component from the node ND1 to the node ND2, as shown in FIG. 16 (e), provided a series circuit of a diode _{D X} and a capacitor _{C Y} between nodes ND1 and ND2, the series The target capacitor _CX may be formed in the circuit. In the arrangement of FIG. 16 (e), the cathode of the diode _{D X} is connected to node ND2, the anode of the diode _{D X} is connected to the node ND1 through the capacitor _{C Y.} Alternatively, connect the anode of the diode D _X to the node ND1, the cathode of the diode D _X may be connected to the node ND2 through the capacitor C _Y. In any case, it is possible to form the subject capacitor _{C X} at serial combined capacitance of the parasitic capacitance of the diode _{D X} PC _X and a capacitor _{C Y} (other volume), shown in FIG. 16 (f) or (g) as a series combined capacitance of the parasitic capacitance PC _X and a capacitor _{C Y} (other volume) of the diode _{D X,} it may be used as a capacitor 114H or 114L is an example of a target capacitor _{C X.}

Capacitor C _Y may be a capacitor formed in the ceramic capacitor or the like, in this case, may capacitance value of the capacitor C _Y is sufficiently larger than the capacitance value of the parasitic capacitance PC _X. Thus, the series combined capacitance Most would be produced depending only on the parasitic capacitance PC _X, it can be brought closer to a state where the target capacitor C _X is formed only by the parasitic capacitance PC _X of diode D _X. However, may be a parasitic capacitance of the further diode and the capacitor C _Y itself diode D _X (in this case, the forward of the forward and another diode of the diode D _X and reverse).

Any capacitor (for example, capacitors 115H and 115L) different from the capacitors 114H and 114L provided in the semiconductor device 1 may be the target capacitor _CX .

[Example EX1_3]
Example EX1_3 will be described.

The operational amplifier 111H may not have a high current capacity from the viewpoint that high-speed operation is required. Therefore, based on the configuration of the variable voltage source 110H shown in FIG. 11, a buffer circuit 117H for increasing the current capacity of the variable voltage source 110H is added to the variable voltage source 110H as shown in FIG. 17A. You may.

The buffer circuit 117H of FIG. 17A includes a transistor 117Ha configured as an NPN bipolar transistor and a transistor 117Hb configured as a PNP bipolar transistor, and further has a resistor 117Hc. It is also possible to omit the resistor 117Hc. The collector of the transistor 117Ha is connected to the terminal to which the power supply voltage VCS1_H is applied, and the collector of the transistor 117Hb is connected to the terminal to which the power supply voltage VCS2_H is applied. Each base of the transistors 117Ha and 117Hb is connected to the output terminal of the operational amplifier 111H via a resistor 117Hc, and each emitter of the transistors 117Ha and 117Hb is connected to the output unit 116H of the variable voltage source 110H. _{That is, when the variable voltage source 110H of} FIG. 17A is used, the output voltage VO_H from the output terminal of the operational amplifier 111H is applied between the gate and the source of the transistor TrH through the buffer circuit 117H.

Transistors 117Ha and 117Hb can also be configured by MOSFET or the like. If the negative gate surge is not so problematic, the buffer circuit 117H may be configured with only the transistor 117Hb. In this case, the transistor 117Ha may be simply deleted based on the configuration of FIG. 17A.

The same may apply to the variable voltage source 110L. That is, based on the configuration of the variable voltage source 110L shown in FIG. 11, a buffer circuit 117L for increasing the current capacity of the variable voltage source 110L is added to the variable voltage source 110L as shown in FIG. 17B. You may.

The buffer circuit 117L of FIG. 17B includes a transistor 117La configured as an NPN bipolar transistor and a transistor 117Lb configured as a PNP bipolar transistor, and further has a resistor 117Lc. It is also possible to omit the resistor 117Lc. The collector of the transistor 117La is connected to the terminal to which the power supply voltage VCS1_L is applied, and the collector of the transistor 117Lb is connected to the terminal to which the power supply voltage VCS2_L is applied. Each base of the transistors 117La and 117Lb is connected to the output terminal of the operational amplifier 111L via a resistor 117Lc, and each emitter of the transistors 117La and 117Lb is connected to the output unit 116L of the variable voltage source 110L. That is, when the variable voltage source 110L of FIG. 17B is used, the output voltage _{VO_L} from the output terminal of the operational amplifier 111L is applied between the gate and the source of the transistor TrL through the buffer circuit 117L.

Transistors 117La and 117Lb can also be configured by MOSFET or the like. If the negative gate surge is not so problematic, the buffer circuit 117L may be configured with only the transistor 117Lb. In this case, the transistor 117La may be simply deleted based on the configuration of FIG. 17B.

Further, the configuration of FIG. 17A may be modified to the configuration of FIG. 18A. That is, based on the configuration of FIG. 17A, one end of the resistor 113H is connected to the inverting input terminal of the operational amplifier 111H, and the other end of the resistor 113H is not the output terminal of the operational amplifier 111H but the output unit 116H of the variable voltage source 110H. You may try to connect to.
Similarly, the configuration of FIG. 17 (b) may be modified to the configuration of FIG. 18 (b). That is, based on the configuration of FIG. 17B, one end of the resistor 113L is connected to the inverting input terminal of the operational amplifier 111L, and the other end of the resistor 113L is not the output terminal of the operational amplifier 111L but the output unit 116L of the variable voltage source 110L. You may try to connect to.

[Example EX1_4]
Example EX1_4 will be described.

In the variable voltage source 110H, a phase compensation element and a protection circuit for the operational amplifier 111H may be added. Specifically, for example, with respect to the variable voltage source 110H shown in any one of the examples EX1-1 to EX1_3, as shown in FIG. 19A, a capacitor is used. 113H_C, resistors 114H_R and 115H_R may be added, and the protection circuit 111H_D may be added. In the variable voltage source 110H of FIG. 19A, the capacitor 113H_C and the resistor 114H_R function as a first phase compensating element for compensating the signal phase in the first differentiating circuit (111H, 113H, 114H), and the capacitor 113H_C. And the resistor 115H_R functions as a second phase compensating element for compensating the signal phase in the second differentiating circuit (111H, 113H, 115H).

The capacitor 113H_C is connected in parallel to the resistor 113H. The resistor 114H_R is connected in series with the capacitor 114H, and a series circuit of the resistor 114H_R and the capacitor 114H is provided between _{the inverting input terminal of the operational amplifier 111H and the terminal DH TM.} Which of the resistor 114H_R and the capacitor 114H may be arranged on the _{terminal DH TM side.} Resistance 115H_R are serially connected to the capacitor 115H, a series circuit of a resistor 115H_R and capacitor 115H is provided between the inverting input terminal and the terminal _{SH TM} of the operational amplifier 111H. Resistance 115H_R and of the capacitor 115H, one can may be disposed in the terminal _{SH TM} side. The protection circuit 111H_D is composed of two diodes connected between the inverting input terminal and the non-inverting input terminal of the operational amplifier 111H, and the forward directions of the two diodes are opposite to each other.

Similarly, in the variable voltage source 110L, a phase compensation element and a protection circuit for the operational amplifier 111L may be added. Specifically, for example, with respect to the variable voltage source 110L shown in any one of the examples EX1-1 to EX1_3, as shown in FIG. 19B, a capacitor is used. 113L_C, resistors 114L_R and 115L_R may be added, and the protection circuit 111L_D may be added. In the variable voltage source 110L of FIG. 19B, the capacitor 113L_C and the resistor 114L_R function as a first phase compensating element for compensating the signal phase in the first differentiating circuit (111L, 113L, 114L), and the capacitor 113L_C. And the resistor 115L_R functions as a second phase compensating element for compensating the signal phase in the second differentiating circuit (111L, 113L, 115L).

The capacitor 113L_C is connected in parallel to the resistor 113L. Resistance 114L_R are serially connected to the capacitor 114L, a series circuit of a resistor 114L_R and capacitor 114L is provided between the inverting input terminal and the terminal _{SH TM} of the operational amplifier 111L. Resistance 114L_R and of the capacitor 114L, either is or may be arranged in the terminal _{SH TM} side. Resistance 115L_R are serially connected to the capacitor 115L, a series circuit of a resistor 115L_R and capacitor 115L is provided between the inverting input terminal and the terminal _{SL TM} of the operational amplifier 111L. Resistance 115L_R and of the capacitor 115L, either is or may be arranged in the terminal _{SL TM} side. The protection circuit 111L_D is composed of two diodes connected between the inverting input terminal and the non-inverting input terminal of the operational amplifier 111L, and the forward directions of the two diodes are opposite to each other.

<< Second Embodiment >>
A second embodiment of the present invention will be described. In the second embodiment, the structure of the semiconductor device 1 whose circuit configuration is shown in the first embodiment will be described.

FIG. 20 is a perspective view of the power module PM. For convenience of explanation, the X-axis, Y-axis, and Z-axis that are orthogonal to each other are defined. The X-axis, Y-axis and Z-axis intersect at the origin, and the polarity of the position in the X-axis direction, the polarity of the position in the Y-axis direction, and the polarity of the position in the Z-axis direction change between positive and negative with the origin as the boundary. .. Here, it is considered that the origin is at the center or the center of gravity of the power module PM having a substantially rectangular parallelepiped shape. The power module PM includes power terminals 511 to 514, signal terminals 521 to 525, a case 530, and a top plate 540. The transistors TrH and TrL are built in a housing formed by the case 530 and the top plate 540.

Power terminals 511 and 512, respectively, terminals _P TM in FIG. _10, corresponds to the _{N TM,} positive output terminal of the DC voltage source 2 is connected to the output terminal of the negative side. The power terminals 511 and 512 are supported by the case 530. The power supply terminals 511 and 512 are formed of, for example, a thin metal plate made of copper. The surface of each thin metal plate may be nickel-plated. The power supply terminals 511 and 512 are arranged apart from each other along the Y-axis direction. The power supply terminals 511 and 512 have the same shape as each other. A part of each of the power supply terminals 511 and 512 is exposed to the outside of the power module PM. Each exposed portion of the power supply terminals 511 and 512 is provided with a connection hole penetrating along the Z axis. A fastening member such as a bolt is inserted into the connection hole. The power supply terminal 511 is connected to the drain of the transistor TrH inside the power module PM, and the power supply terminal 512 is connected to the source of the transistor TrL inside the power module PM.

The power supply terminals 513 and 514 _{correspond to the terminals O1 TM} and O2 _TM in FIG. 10, respectively, and are connected to a load such as a motor arranged outside the semiconductor device 1. The power terminals 513 and 514 are supported by the case 530. The power supply terminals 513 and 514 are formed of, for example, a thin metal plate made of copper. The surface of each thin metal plate may be nickel-plated. The power supply terminals 513 and 514 are arranged apart from each other along the Y-axis direction. The power supply terminals 513 and 514 have the same shape as each other. A part of each of the power supply terminals 513 and 514 is exposed to the outside of the power module PM. Each exposed portion of the power supply terminals 513 and 514 is provided with a connection hole penetrating along the Z axis. A fastening member such as a bolt is inserted into the connection hole. The power supply terminal 513 is connected to the source of the transistor TrH inside the power module PM, and the power supply terminal 514 is connected to the drain of the transistor TrL inside the power module PM. Instead of the power supply terminals 513 and 514, a single power supply terminal in which the power supply terminals 513 and 514 are combined may be provided.

The first power supply terminal row including the power supply terminals 511 and 512 and the second power supply terminal row including the power supply terminals 513 and 514 are arranged apart from each other in the X-axis direction. Here, it is assumed that the first power supply terminal row is located on the positive side of the X-axis and the second power supply terminal row is located on the negative side of the X-axis. The first power supply terminal row is arranged at the positive end of the X-axis in the case 530, and the second power supply terminal row is arranged at the negative end of the X-axis in the case 530.

The signal terminals 521, 522, 523, 524, and 525 _{correspond to the terminals GH TM} , SH _TM , GL _TM , SL _TM , and DH _TM in FIG. 10, respectively, and inside the power module PM, the gate and transistor of the transistor TrH. It is connected to the source of TrH, the gate of transistor TrL, the source of transistor TrL, and the drain of transistor TrH. The signal terminals 521 to 525 are supported by the case 530. A part of each of the signal terminals 521 to 525 is exposed to the outside of the power module PM. Each exposed portion of the signal terminals 521 to 525 projects from the top plate 540 along the Z-axis direction. The direction of protrusion from the top plate 540 at each signal terminal is defined as the direction from the negative side to the positive side of the Z axis, and the direction is defined as upward. Each of the signal terminals 521 to 525 is, for example, a metal rod made of copper as a constituent material. The surface of each metal rod is tin-plated. The signal terminals 521 to 525 have the same shape as each other.

The signal terminals 521, 522, and 525 are arranged side by side while being separated from each other along the X-axis direction to form a first signal terminal sequence. The signal terminals 523 and 524 are arranged side by side while being separated from each other along the X-axis direction to form a second signal terminal row. The first signal terminal row and the second signal terminal row are arranged so as to be separated from each other in the Y-axis direction. Here, it is assumed that the first signal terminal row is located on the positive side of the Y-axis and the second signal terminal row is located on the negative side of the Y-axis. The first signal terminal row is arranged at the positive end of the Y-axis in the case 530, and the second signal terminal row is arranged at the negative end of the Y-axis in the case 530. In the first signal terminal row, the signal terminal 522 is located between the signal terminals 521 and 525, and the distance between the signal terminals 522 and 525 is larger than the distance between the signal terminals 521 and 522.

In a plan view when the power module PM is observed from above, the signal terminals 521, 522 and 525 are located on the negative side of the X axis when viewed from the center of the power module PM, and the signal terminals 523 and 524 are located on the negative side of the X axis. It is located on the positive side of the X-axis when viewed from the center of the power module PM.

In a plan view when the power module PM is observed from above, the signal terminals 521 and 523 are arranged at positions approximately symmetrical with respect to the center point CN of the power module PM, and the signal terminals 522 and 524 are the centers of the power module PM. It is arranged at a position approximately symmetrical with respect to the point CN. When the smallest rectangular parallelepiped that can accommodate the power module PM is defined as a virtual rectangular parallelepiped, the center of the virtual rectangular parallelepiped corresponds to the center point CN. The center point CN may be understood as the center or the center of gravity of the power module PM.

The transistors TrH and TrL are arranged side by side along the X-axis direction. The transistor TrH is located on the negative side of the X-axis when viewed from the center point CN, and the transistor TrL is located on the positive side of the X-axis when viewed from the center point CN. The transistors TrH and TrL are arranged at positions approximately symmetrical with respect to the center point CN, passing through the center point CN and approximately line-symmetrical with respect to a straight line parallel to the Y axis, passing through the center point CN, and passing through the center point CN and the Y axis and. It is arranged at a position approximately plane-symmetrical with respect to a plane parallel to the Z-axis.

Case 530 is a container for accommodating transistors TrH and TrL, and has a box-like shape without a lid. The case 530 is made of an electrically insulating material. For example, the case 530 is formed of a synthetic resin having electrical insulation and excellent heat resistance, such as PPS (polyphenylene sulfide).

The top plate 540 is a lid that closes the internal region of the power module PM formed by the case 530. The top plate 540 is made of a synthetic resin having electrical insulation. The control module CM is arranged on the top plate 540.

FIG. 21 shows an exploded perspective view of the semiconductor device 1. The semiconductor device 1 is formed by arranging the control module CM on the top plate 540 and combining the power module PM and the control module CM. In FIG. 21, the power module PM in the state before they are combined is formed. And the control module CM are shown.

The control module CM includes a circuit board 600. Although not shown in FIG. 21, each circuit component shown in the first embodiment is mounted on the circuit board 600, and between each circuit component and each terminal shown in the first embodiment. A circuit pattern that realizes the connection is formed. The circuit board 600 is provided with connection holes 611 to 615 arranged at five positions corresponding to the positions of the signal terminals 521 to 525. In the circuit board 600, lands are formed around each of the connection holes 611 to 615. After arranging the circuit board 600 on the top plate 540 so that the signal terminals 521 to 525 are inserted into the connection holes 611 to 615, the circuit board 600 is fixed to the case 530 and formed around the connection holes 611 to 615. The lands are made conductive to the signal terminals 521 to 525, respectively, through a soldering step. As a result, the signal terminals 521 to 525 are electrically connected to necessary locations on the circuit board 600. Through holes are formed at each of the four corners of the substantially rectangular circuit board 600, and bolt holes having internal threads are formed in the case 530 at positions corresponding to the four through holes. ing. The circuit board 600 is fixed to the case 530 by using a through hole of the circuit board 600, a bolt hole of the case 530, and a bolt (not shown).

Of the two surfaces of the circuit board 600, the surface relatively far from the top plate 540 is the component mounting surface. Each circuit component forming the control module CM is mounted on the component mounting surface. The center point of the component mounting surface is referred to as the center point CNa. The center point CNa and the above-mentioned center point CN are located on one straight line parallel to the Z-axis direction.

On the component mounting surface of the circuit board 600, the variable voltage source 110H is arranged in the area 631 and the variable voltage source 110L is arranged in the area 632. Region 631 is located between the center point CNa and the connection holes 611 and 612 into which the signal terminals 521 and 522 are inserted, and region 632 is the connection hole 613 and in which the center point CNa and the signal terminals 523 and 524 are inserted. It is located between 614 and 614. The region 631 is located on the negative side of the X-axis and the positive side of the Y-axis when viewed from the center point CNa, and the region 632 is located on the positive side of the X-axis and the negative side of the Y-axis when viewed from the center point CNa. Regions 631 and 632 may be considered to be located approximately symmetrically with respect to the center point CNa.

Although not particularly shown in FIG. 21, the control signal generation unit 20 is arranged in a region including the center point CNa on the component mounting surface of the circuit board 600. On the component mounting surface of the circuit board 600, a region that does not overlap with each of the above-mentioned regions (for example, a region on the negative side of the X-axis and the negative side of the Y-axis when viewed from the center point CNA, or a region on the X-axis when viewed from the center point CNa. The power supply circuit 30 is arranged on the positive side and the positive side of the Y-axis). _{Further, a connector for receiving a signal (including the above-mentioned signal S IN} ) supplied from an external device of the semiconductor device 1 may be mounted on a component mounting surface of the circuit board 600.

Based on the above contents, other suitable arrangement examples and various modified arrangement examples are shown below.

The variable voltage source 110H may be arranged as close as possible to the signal terminal 521 (hence the connection hole 611), and similarly, the variable voltage source 110L should be placed as close as possible to the signal terminal 523 (hence the connection hole 613). It is good to arrange it.

A single semiconductor IC (one-chip semiconductor IC) may be arranged in a region including the center point CNa on the component mounting surface of the circuit board 600.

The control signal generation unit 20 may be included in the single semiconductor IC, and the variable voltage sources 110H and 110L may be configured by discrete components outside the single semiconductor IC.

Alternatively, the variable voltage sources 110H and 110L may be included in the single semiconductor IC in addition to the control signal generation unit 20.

Further, the variable voltage sources 110H and 110L may be mounted not on the circuit board 600 but on the circuit board on which the transistors TrH and TrL are mounted, which are arranged below the top plate 540. In this case, it can be understood that the variable voltage sources 110H and 110L are built in the power module PM instead of the control module CM.

<< Third Embodiment >>
A third embodiment of the present invention will be described. In the third embodiment, the deformation techniques and the like applicable to the first and second embodiments will be described.

In the first and second embodiments, it is assumed that the semiconductor device 1 includes only one half-bridge circuit HB, but by providing the semiconductor device 1 with a plurality of half-bridge circuits HB, a full-bridge circuit or three A phase bridge circuit may be formed. In this case, it is preferable that the drive control units 10H and 10L and the control signal generation unit 20 are provided for each half-bridge circuit HB.

According to the configuration shown in the first embodiment, the semiconductor device 1 having both the surge countermeasure function and the high-speed drive function can be formed, but in the case where the influence of the gate surge is small, the surge countermeasure function is deleted from the semiconductor device 1. It is also possible to do. That is, the capacitors 114H and 114L may be deleted from the semiconductor device 1 of FIG.

An example of configuring the variable voltage source 110H using a differentiating circuit has been described above, but the configuration of the variable voltage source 110H is arbitrary as long as the above-mentioned functions to be realized by the variable voltage source 110H can be realized. The same applies to the variable voltage source 110L.

The semiconductor device 1 can be applied to any device that requires one or more half-bridge circuits. For example, the semiconductor device 1 can be applied to an inverter circuit for driving a motor or an isolated DC / DC converter.

The channel types of FETs (Field Effect Transistors) shown in each embodiment are examples, so that P-channel type FETs are changed to N-channel type FETs, or N-channel type FETs are P-channels. The configuration of the circuit containing the FET can be modified so that it is changed to a type FET. Further, any FET may be configured by HEMT (High Electron Mobility Transistor).

Further, each transistor illustrated in the above-described embodiment may be any kind of transistor. For example, the transistor shown as a MOSFET can be replaced with a junction FET, an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor. Any transistor has a first electrode, a second electrode and a control electrode. In the FET, one of the first and second electrodes is a drain, the other is a source, and the control electrode is a gate. In the IGBT, one of the first and second electrodes is a collector, the other is an emitter, and the control electrode is a gate. In a bipolar transistor that does not belong to an IGBT, one of the first and second electrodes is a collector, the other is an emitter, and the control electrode is the base.

However, the above-mentioned transistors TrH and TrL are preferably voltage-controlled transistors such as FETs or IGBTs including MOSFETs. A voltage-controlled transistor is a transistor in which the drain-source is controlled to be conductive or non-conductive (in other words, the current flowing between the drain and source is controlled) according to the gate-source voltage. , It is a transistor in which the collector-emitter is controlled in a conductive state or a non-conductive state (in other words, the current flowing between the collector and the emitter is controlled) according to the gate-emitter voltage.

<< Consideration of the present invention >>
The present invention embodied in the above-described embodiment will be considered.

The semiconductor device according to one aspect of the present invention (hereinafter, referred to as semiconductor device W) has a first electrode, a second electrode, and a control electrode, and has a voltage between the control electrode and the second electrode (for example, _{VGS_H} ). A variable voltage is applied between the switching element (for example, TrH) in which the first electrode and the second electrode are controlled to be on or off, and the control electrode and the second electrode of the switching element. A voltage generating unit including a variable voltage source (for example, 110H) to be supplied, and the variable voltage source outputs a _{rectangular wavy voltage (for example, VPLS_H) for alternately turning on or off the switching elements.} (For example, 112H), when the switching element is switched by supplying the rectangular wavy voltage between the control electrode and the second electrode, a voltage component (for example) corresponding to a change in the rectangular wavy voltage is used. It is characterized in _{that a drive voltage (for example, VO_H ) is generated by superimposing a voltage V PLS_H (corresponding to time differentiation of voltage V PLS_H} ) on the rectangular wavy voltage, and the drive voltage is applied between the control electrode and the second electrode. ..

Regarding the semiconductor device W, in the variable voltage source, the drive voltage may be generated by a differentiating circuit (for example, 111H, 113H, 115H) using an operational amplifier.

More specifically, for example, in the semiconductor device W, the operational amplifier (for example, 111H) receives the _{rectangular wavy voltage (for example, VPLS_H) with reference to the potential at the second electrode of the switching element.} It has (+), a second input terminal (-), and an output terminal, and the variable voltage source includes the operational amplifier (for example, 111H), the second input terminal (-) of the operational amplifier, and the output. It has a feedback resistor (for example, 113H) provided between the terminals and a capacitor (for example, 115H) provided between the second input terminal of the operational amplifier and the second electrode of the switching element, and the switching. When switching the elements, it is preferable to output the drive voltage (for example, _{VO_H} ) from the output terminal of the operational amplifier.

At this time, in the semiconductor device W, for example (see FIG. 15), the product of the capacitance value of the capacitor and the resistance value of the feedback resistance (for example, C _115H × R _113H ) is the control electrode of the switching element. and a value based on the product of the resistance value of the control electrode resistance of the electrostatic capacitance value and the switching element of the capacitance between the second electrode (e.g., _{k H2 (C GS_H × R G_H} )), or the input of the switching element has a value based on the product of the resistance value of the control electrode resistance of the switching element and the capacitance value of the capacitance (e.g., _{k H2 (C GS_H + C GD_H} ) × R G_H), the control electrode resistance of the switching element (e.g. _{RG_H} ) may include an internal resistance (for example, _{RGIN_H} ) in the control electrode of the switching element.

The embodiments of the present invention can be appropriately modified in various ways within the scope of the technical idea shown in the claims. The above embodiments are merely examples of the embodiments of the present invention, and the meanings of the terms of the present invention and the constituent requirements are not limited to those described in the above embodiments. The specific numerical values shown in the above description are merely examples, and as a matter of course, they can be changed to various numerical values.

1 Semiconductor device PM power module CM control module HB half bridge circuit TrH, TrL transistor (switching element)
10H High-side drive control unit 10L Low-side drive control unit 20 Control signal generator 30 Power supply circuit 110H, 110L Variable voltage source 111H, 111L Operational amplifier 112H, 112L Voltage generator 113H, 113L Resistance (feedback resistance)
114H, 114L Capacitor 115H, 115L Capacitor 116H, 116L Output

Claims (12)

Switching having a first electrode, a second electrode, and a control electrode, in which the first electrode and the second electrode are controlled to be on or off according to the voltage between the control electrode and the second electrode. With the element
A variable voltage source for supplying a variable voltage between the control electrode and the second electrode of the switching element is provided.
The variable voltage source has a voltage generating unit that outputs a rectangular wavy voltage for alternately turning the switching element on or off, and transfers the rectangular wavy voltage between the control electrode and the second electrode. When switching the switching element, a drive voltage is generated in which a voltage component corresponding to a change in the rectangular wave-shaped voltage is superimposed on the rectangular wave-shaped voltage, and the drive voltage is applied to the control electrode and the first. A semiconductor device characterized by supplying between two electrodes. The semiconductor device according to claim 1, wherein the drive voltage is generated by a differentiating circuit using an operational amplifier in the variable voltage source. The operational amplifier has a first input terminal, a second input terminal, and an output terminal that receive the rectangular wave-shaped voltage with reference to the potential of the second electrode of the switching element.
The variable voltage source includes the operational amplifier and
A feedback resistor provided between the second input terminal and the output terminal of the operational amplifier,
It has a capacitor provided between the second input terminal of the operational amplifier and the second electrode of the switching element, and outputs the drive voltage from the output terminal of the operational amplifier when switching the switching element. The semiconductor device according to claim 2, wherein the semiconductor device is characterized by the above. The product of the capacitance value of the capacitor and the resistance value of the feedback resistor is
A value based on the product of the capacitance value of the capacitance between the control electrode and the second electrode of the switching element and the resistance value of the control electrode resistance of the switching element, or
It has a value based on the product of the capacitance value of the input capacitance of the switching element and the resistance value of the control electrode resistance of the switching element.
The semiconductor device according to claim 3, wherein the control electrode resistance of the switching element includes an internal resistance in the control electrode of the switching element. The variable voltage source responds to a voltage change between the first electrode and the second electrode in an off control section in which the switching element is controlled to an off state by maintaining the rectangular wavy voltage level at a predetermined level. The semiconductor device according to any one of claims 1 to 4, wherein a voltage corresponding to a voltage change between the first electrode and the second electrode is supplied between the control electrode and the second electrode. .. The variable voltage source responds to a voltage change between the first electrode and the second electrode in an off control section in which the switching element is controlled to an off state by maintaining the rectangular wavy voltage level at a predetermined level. At that time, a voltage corresponding to the voltage change between the first electrode and the second electrode is supplied between the control electrode and the second electrode.
In addition to the operational amplifier, the feedback resistor, and the first capacitor as the capacitor, the variable voltage source is provided between the second input terminal of the operational amplifier and the first electrode of the switching element. The semiconductor device according to claim 3, further comprising a capacitor, and outputting a voltage corresponding to a voltage change between the first electrode and the second electrode from the output terminal of the operational amplifier in the off control section. .. The product of the capacitance value of the first capacitor and the resistance value of the feedback resistor is
A value based on the product of the capacitance value of the capacitance between the control electrode and the second electrode of the switching element and the resistance value of the control electrode resistance of the switching element, or
It has a value based on the product of the capacitance value of the input capacitance of the switching element and the resistance value of the control electrode resistance of the switching element.
The product of the capacitance value of the second capacitor and the resistance value of the feedback resistor is a value based on the product of the capacitance value of the feedback capacitance of the switching element and the resistance value of the control electrode resistance of the switching element. Have and
The semiconductor device according to claim 6, wherein the control electrode resistance of the switching element includes an internal resistance in the control electrode of the switching element. The semiconductor device according to claim 6 or 7, wherein the second capacitor is formed by using the parasitic capacitance of the diode. The semiconductor device according to claim 8, wherein the second capacitor is formed by a series circuit of the parasitic capacitance of the diode and another capacitance. In the variable voltage source, a buffer circuit is provided between the output terminal of the operational amplifier and the control electrode of the switching element.
The semiconductor device according to any one of claims 3, 4, and 6 to 9, wherein the output voltage of the operational amplifier is supplied between the control electrode and the second electrode of the switching element through the buffer circuit. The semiconductor device according to claim 10, wherein the switching element is made of a wide-gap semiconductor. A plurality of sets of the switching element and the variable voltage source are provided, and the plurality of sets include the first set and the second set.
The first switching element, which is the first set of switching elements, and the second switching element, which is the second set of switching elements, are connected in series with each other, and are predetermined with respect to the series circuit of the first switching element and the second switching element. The semiconductor device according to any one of claims 1 to 11, wherein a DC voltage is applied.

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