JP2021141454A - Reception device, transmission device, reception method, and transmission method - Google Patents

Abstract

To demodulate an OFDM signal multiplexed by an asynchronous LDM method.SOLUTION: According to the embodiment, a reception device receives a signal transmitted from a transmission device. The reception device includes a reception unit and a demodulation unit. The reception unit receives a reception signal including a first OFDM signal having a first symbol length and a first power level and a second OFDM signal having a second symbol length longer than the first symbol length and a second power level lower than the first power level. The demodulation unit demodulates the second OFDM signal by subtracting a signal generated by convolving the first OFDM signal and a delay profile from the reception signal.SELECTED DRAWING: Figure 7

Description

Embodiments of the present invention relate to a receiving device, a transmitting device, a receiving method and a transmitting method.

As a wireless signal transmission method, a layered division multiplexing (LDM) method has been proposed in which two different signals are multiplexed with different powers and transmitted in the same band at the same time in order to improve frequency utilization efficiency. There is. In this LDM method, when the signal to be multiplexed is an OFDM (Orthogonal Frequency Division Multiplexing) signal, the two OFDM signals have the same symbol length and symbol switching timing, and the two OFDM signals have the same symbol length and symbol switching timing. Is also classified as an asynchronous method that is not equal. In the synchronous method, since the symbol lengths of both signals are equal, the transmission efficiency of both signals is also equal, and the transmission efficiency of one signal cannot be increased.

In the asynchronous method, the symbol length of each OFDM signal is a symbol length suitable for each OFDM signal, so that the transmission efficiency of one signal is improved as compared with the synchronous method. However, the control of the asynchronous method is more complicated than the control of the synchronous method, and the demodulation algorithm similar to the synchronous method cannot demodulate the multiplexed OFDM signal.

Hiromasa Okada, 6 outsiders, "A device for improving various problems when applying LDM to terrestrial digital broadcasting-Examination of new broadcasting system reception area expansion method and synchronization method-", Technical Report of the Institute of Imaging Media, September 7, 2018 , Vol. 42, No. 28

An object to be solved by the present invention is to provide a receiving device, a transmitting device, a receiving method, and a transmitting method capable of demodulating an OFDM signal multiplexed by an asynchronous LDM method.

According to one embodiment, the receiving device receives the signal transmitted from the transmitting device. The receiving device includes a receiving unit and a demodulating unit. The receiver has a first OFDM signal having a first symbol length and a first power level, and a second symbol length longer than the first symbol length and the first power level. Receives a received signal that includes a second OFDM signal with a lower second power level. The demodulation unit demodulates the second OFDM signal by subtracting the signal generated by convolving the first OFDM signal and the delay profile from the received signal.

FIG. 1 is a diagram showing an outline of a signal transmitted by the LDM method according to the embodiment. FIG. 2 is a diagram showing an outline of an LDM type transmission device according to an embodiment. FIG. 3 is a diagram showing an example of power adjustment of the LDM method according to the embodiment. FIG. 4 is a diagram showing a case where a reception device receives a reception signal including a composite transmission signal multiplexed by a synchronous LDM method in an environment in which a delay wave is received after a direct wave. FIG. 5 is a diagram showing a case where a reception device receives a reception signal including a composite transmission signal multiplexed by an asynchronous LDM method in an environment in which a delay wave is received after a direct wave. FIG. 6 is a diagram showing an example of the configuration of the transmission device according to the embodiment. FIG. 7 is a diagram showing an example of the configuration of the receiving device according to the embodiment. FIG. 8 is a diagram for explaining a delay profile.

Some embodiments will be described with reference to the drawings.
It should be noted that the disclosure is merely an example, and those skilled in the art can easily conceive of appropriate changes while maintaining the gist of the invention are naturally included in the scope of the present invention. Further, the drawings may be represented schematically as compared with the embodiments in order to clarify the description, but the drawings are merely examples and do not limit the interpretation of the present invention. Further, in the present specification and each figure, components exhibiting the same or similar functions as those described above with respect to the above-mentioned figures may be designated by the same reference numerals, and duplicate detailed description may be omitted.

First, the LDM method will be described as a premise of the embodiment. In studying the next-generation standard for current terrestrial digital television broadcasting called the Integrated Services Digital Broadcasting-Terrestrial (ISDB-T) method, it is possible to improve frequency utilization efficiency or coexist with the current method. Therefore, the use of the LDM method has been proposed. In the LDM system, as shown in FIG. 1, there are two layers, an upper layer (UL) (also referred to as a high power layer) and a lower layer (LL) (also referred to as a low power layer) modulated by the same or different methods. Two different signals are multiplexed and transmitted at the same time and in the same band. For example, an example of an upper layer signal may be a signal conforming to the ISDB-T system, and an example of a lower layer signal is a next-generation terrestrial digital television broadcasting system (for example, called a 4K8K system) in which ISDB-T is advanced. It may be a signal conforming to). The latter is sometimes referred to as a Super Hi-Vision (SHV) signal. The power level of the upper layer signal is higher than the power level of the lower layer signal, and the power ratio between the two layers is called an injection level, and may be, for example, 23 dB. The ISDB-T signal may not be transmitted, but a signal conforming to the 4K8K system may be transmitted in the upper layer and the lower layer. It should be noted that the signal of not only two layers but also three or more layers may be multiplexed and transmitted.

FIG. 2 is a diagram showing the concept of LDM. The ISDB-T signal 2 in the time domain is input to the power adjustment unit 3, and the power level is adjusted based on the normalization coefficient β. The SHV signal 5 in the time domain is input to the power adjustment unit 6, and the power level is adjusted based on the product of the normalization coefficient β and the scaling coefficient α. The outputs of the power adjusting unit 3 and the power adjusting unit 6 are added by the adding unit 4, and the combined transmission signal of the ISDB-T signal and the SHV signal is transmitted.

The injection level is a value representing the ratio of the average power of the ISDB-T signal to the average power of the SHV signal. The injection level is determined when setting the receiver. The higher the injection level, the larger the power ratio of the ISDB-T signal in the synthetic transmission signal, and the smaller the injection level, the smaller the power ratio of the ISDB-T signal in the synthetic transmission signal. The normalization coefficient β and the scaling coefficient α, which are the power adjustment coefficients, are determined as shown in FIG. 3 according to the injection level.

In embodiments, the signal is Orthogonal Frequency Division Multiplexing (OFDM) modulated.

The above-mentioned LDM method is classified into two methods, a synchronous method and an asynchronous method.
In the synchronous LDM method, the symbol lengths of both signals are the same, the symbol switching timings are the same, and the guard interval (GI) positions added to both signals are also the same, so control is not complicated, but SHV. The signal transmission efficiency is the same as the ISDB-T signal transmission efficiency.

In the asynchronous LDM method, the symbol lengths of both signals are different, the switching timing of the symbols of both signals is different, and the position of the GI added to both signals is also different, so that the control is more complicated than the synchronous method. , The transmission efficiency of the SHV signal can be made higher than the transmission efficiency of the ISDB-T signal.

Although the symbol length of the SHV signal is different from the symbol length of the ISDB-T signal, the case where the symbol length is an integral multiple of the symbol length of the ISDB-T signal may be referred to as a quasi-synchronous method. The case where the symbol lengths of both signals are different will be collectively referred to as an asynchronous method.

Here, as shown in FIGS. 4 and 5, it is assumed that the receiving signal is received by the receiving device in an environment in which the delayed wave is received after the direct wave (leading wave). FIG. 4 shows a case where a composite transmission signal multiplexed by the synchronous LDM method is transmitted from the transmission device. FIG. 5 shows a case where a composite transmission signal multiplexed by the asynchronous LDM method is transmitted from the transmission device.

As shown in FIG. 4, when the composite transmission signal multiplexed by the synchronous LDM method is transmitted in an environment in which the delayed wave is received after the direct wave, the synchronous LDM method described above. As described above, the symbol lengths of the ISDB-T signal and the SHV signal are the same, the symbol switching timings are the same, and the positions of the GIs added to both signals are the same, so that the receiving device is equal to the symbol lengths of both signals. By performing a Fourier transform (FFT) with an FFT size, only a lower layer signal, that is, an SHV signal can be obtained.

On the other hand, as shown in FIG. 5, when the composite transmission signal multiplexed by the asynchronous LDM method is transmitted in an environment where the delayed wave is received after the direct wave, the asynchronous LDM method is used. As described above, the symbol lengths of the ISDB-T signal and the SHV signal are different, the switching timing of the symbols is also different, and the position of the GI added to both signals is also different. Therefore, for example, in the receiving device, the symbol of the SHV signal Even if FFT is performed with an FFT size equal to the length, intersymbol interference occurs, and it is not possible to obtain only an SHV signal as in the synchronous method.

In the following, in an environment where a delay wave is received after a direct wave, when a composite transmission signal multiplexed by the asynchronous LDM method is transmitted from the transmission device, the lower layer signal is demodulated from the reception signal. The receiving device that can be used will be mainly described.

In this embodiment, it is assumed that the symbol length of the ISDB-T signal is, for example, 8K, and the symbol length of the SHV signal is, for example, 32K. Further, in the present embodiment, it is assumed that the symbol length of the GI added to each of the ISDB-T signal and the SHV signal is 1K.

In the present specification, 8K pieces (8K pieces) generated by performing an inverse Fourier transform (IFFT) with an IFFT size of 8K on an OFDM symbol consisting of 8K (8 × 1024 = 8192) subcarriers. The symbol length (for example, 1008 μsec) of the time symbol composed of the components of (sample) is referred to as the symbol length of 8K, and has an IFFT size of 32K for an OFDM symbol consisting of 32K (32 × 1024 = 32768) subcarriers. The symbol length (for example, 4032 μsec) of a time symbol composed of 32K (sample) components generated by performing an IFFT is referred to as a 32K symbol length. In the transmitting device, IFFT is performed on the signal in the frequency domain, and the signal in the frequency domain is converted into the signal in the time domain. When IFFT is performed on the ISDB-T signal in the frequency domain with an IFFT size of 8K, the ISDB-T signal in the time domain is obtained. When the IFFT is performed on the SHV signal in the frequency domain with an IFFT size of 32K, the SHV signal in the time domain is obtained. The signal in which the ISDB-T signal and the SHV signal in the time domain are multiplexed is the composite transmission signal.

FIG. 6 is a block diagram showing an example of the configuration of the transmission device according to the embodiment. Information bit generation units 12 and 42 are provided. The information bit generation units 12 and 42 may be realized by a device of a higher layer. The information bit generation unit 12 generates the information bit string of the upper layer (UL) signal, here the ISDB-T signal, and the information bit generation unit 42 generates the information bit string of the lower layer (LL) signal, here the SHV signal.

For the upper layer, the information bit string of the ISDB-T signal is input to the coding unit 14. The coding unit 14 is also referred to as an external coding unit. The coding unit 14 can adopt, for example, a Reed-Solomon (RS) coding method. Hereinafter, the coding unit 14 is referred to as an RS coding unit 14. The RS coding unit 14 can perform shortened RS coding, for example, and correct random errors up to 8 bytes out of 204 bytes.

The coded bit string output from the RS coding unit 14 is input to the energy spreading unit 16. Energy diffusion is performed using a pseudo-random code sequence (Pseudo-Random Binary Sequence: PRBS). The PRBS generation circuit in this case consists of, for example, 15 D-type flip-flops connected in series, and the output of the 14th flip-flop and the output of the 15th flip-flop are added to the first flip-flop. Entered.

A bitwise exclusive OR is taken between the signal excluding the synchronization bytes and the PRBS. The initial value of the PRBS generation circuit is set to "100101010000000" from the lowest order, and is initialized for each OFDM frame.

The output bit string of the energy diffusion unit 16 is input to the byte interleaving (IL) unit 18. The byte IL unit 18 performs convolution byte interleaving on a signal of 204 bytes. The interleave depth is, for example, 12 bytes. The bite IL portion 18 has 12 paths. Path 0 has a delay amount of 0. The capacity of the FIFO shift register of path 1 is 17 bytes, the capacity of the FIFO shift register of path 2 is 17 × 2 = 34 bytes, and similarly, the capacity of the FIFO shift register of path 11 is 17 × 11 = 187. It is a byte. The input and output are cyclically switched to pass 0, path 1, path 2, ... path 11, path 0, path 1, ... For each byte.

The output bit string of the byte IL unit 18 is input to the convolutional coding unit 22. The convolutional coding unit 22 is also referred to as an internal coding unit. The convolutional coding unit 22 performs puncture-chard convolutional coding using, for example, a high-speed length k = 7 and a coding rate of 1/2 as a mother code. The generation polynomial of the mother code is G1 = 171 (decimal number) for the X output and G2 = 133 (decimal number) for the Y output.

The coded bit string output from the convolutional coding unit 22 is input to the bit interleaving (IL) unit 24. The output of the bit IL unit 24 is input to the mapping unit 26. The mapping unit 26 maps a designated multi-valued number of coded bit strings in the frequency domain. The bit IL unit 24 and the mapping unit 26 form a subcarrier modulation unit. For subcarrier modulation, any of the following methods may be adopted.

QPSK (Quadrature Phase Shift Keying) modulation: The input signal is set to 2 bits / symbol, QPSK mapping is performed, and multi-bit I-axis data and Q-axis data are output. Bit 0 and bit 1 after the series-parallel conversion are input to the QPSK mapping circuit, and a delay element of, for example, 120 bits is inserted into bit 1, and bit interleaving is performed.

16QAM (16 Quadrature Amplitude Modulation) Modulation: The input signal is 4 bits / symbol, 16QAM mapping is performed, and multi-bit I-axis data and Q-axis data are output. Bits 0 to 3 after the series-parallel conversion are input to the 16QAM mapping circuit, but a delay element of, for example, 40 bits is inserted in bit 1, a delay element of, for example, 80 bits is inserted in bit 2, and a delay element of, for example, 80 bits is inserted in bit 3. A 120-bit delay element is inserted and bit interleaving is performed.

64QAM modulation: The input signal is 6 bits / symbol, 64QAM mapping is performed, and a plurality of bits of I-axis data and Q-axis data are output. Bits 0 to 5 after serial-parallel conversion are input to the 64QAM mapping circuit, but a delay element of, for example, 24 bits is inserted into bit 1, a delay element of, for example, 48 bits is inserted into bit 2, and a delay element of, for example, 48 bits is inserted into bit 3. A 72-bit delay element is inserted, a delay element of, for example, 96 bits is inserted into bit 4, a delay element of, for example, 120 bits is inserted into bit 5, and bit interleaving is performed.

The mapping unit 26 leaves blanks for the positions of the subcarriers corresponding to the diffusion pilot (SP) signal, the TMCC (Transmission and Multiplexing Control) signal, and the AC (Auxiliary Channel) signal.

The SP signal is a known signal that is inserted once into 12 subcarriers in the subcarrier direction and once into 4 symbols in the symbol direction. The SP signal is inserted only into the ISDB-T signal. The TMCC signal is a signal for transmitting control information. The AC signal is an extension signal for transmitting additional information related to broadcasting. The subcarriers of the TMCC signal and the AC signal are randomly arranged in the frequency direction in order to reduce the influence of the periodic dip of the propagation path characteristics due to multipath.

The output of the mapping unit 26 is input to the time / frequency interleaving (IL) unit 28. First, the time / frequency IL unit 28 performs time interleaving processing in the data segment in modulation symbol units (I-axis and Q-axis units). Next, the time / frequency IL unit 28 performs intra-segment interleaving processing by performing subcarrier rotation processing and subcarrier randomization processing.

The output of the time / frequency IL unit 28 is input to the pilot addition unit 32. The pilot addition unit 32 adds an SP signal, a TMCC signal, and an AC signal to the subcarrier symbol to form a data segment.

The output of the pilot addition unit 32 is input to the IFFT unit 34. The IFFT unit 34 performs an IFFT on the data segment of the ISDB-T signal with an IFFT size equal to the symbol length (8K), and obtains a bit string in the time domain.

The output of the Fourier unit 34 is input to the guard interval (GI) addition unit 36. The GI addition unit 36 adds data having a specified time length to the front of the symbol as it is from the rear side in terms of time among the output data after the IFFT.

As a result, a transmission signal in the upper layer of the time domain can be obtained. The output of the GI addition unit 36 is input to the LDM multiplexing unit 38 as an ISDB-T signal.

For the lower layers, the information bit string of the SHV signal is input to the energy diffusion unit 44. Similar to the energy diffusion unit 16, the energy diffusion unit 44 performs energy diffusion using the pseudo-random coding sequence PRBS.

The output bit string of the energy diffusion unit 44 is input to the coding unit. As the coding unit, a convolutional coding method, a BCH (Bose Chain Hocquengem) coding method, an LDPC (Low Density Parity Check) coding method, or the like can be adopted. Here, a BCH coding method and an LDPC coding method are adopted. Therefore, the output bit string of the energy diffusion unit 44 is input to the BCH coding unit 46, and the output bit string of the BCH coding unit 46 is input to the LDPC coding unit 48.

In FIG. 6, the coding methods of the upper layer and the lower layer are different methods, but the coding methods of the upper layer and the lower layer may be the same method. When the coding methods of the upper layer and the lower layer are the same, the error correction capability and the code length of the coding methods of the upper layer and the lower layer may be the same or different.

The coded bit string output from the LDPC coding unit 48 is input to the bit interleaving (IL) unit 52. The output of the bit IL unit 52 is input to the mapping unit 54.

The mapping unit 54 maps a designated multi-valued number of coded bit strings in the frequency domain. The bit IL unit 52 and the mapping unit 54 form a subcarrier modulation unit. The subcarrier modulation of the upper layer and the lower layer may be the same method or may be different methods. When the subcarrier modulation of the upper layer and the lower layer is the same method, the bit interleaving length may be the same or different.

The output of the mapping unit 54 is input to the time / frequency interleaving (IL) unit 56. The time / frequency IL unit 56 performs the same processing as the time / frequency IL unit 28. The interleave lengths of the upper hierarchy and the lower hierarchy may be the same or different.

The output of the time / frequency IL unit 56 is input to the IFFT unit 58. The IFFT unit 58 performs an IFFT on the data segment of the SHV signal with an IFFT size equal to the symbol length (32K) to obtain a bit string in the time domain.

The output of the IFFT unit 58 is input to the GI addition unit 62. The GI addition unit 62 adds data having a specified time length to the front of a symbol having a symbol length of 32K as it is from the rear side in terms of time among the output data after the IFFT.

As a result, a transmission signal in the lower layer of the time domain can be obtained. The output of the GI addition unit 62 is input to the LDM multiplexing unit 38 as an SHV signal.

The LDM multiplexing unit 38 is configured as shown in FIG. 2, in which the ISDB-T signal is the upper layer, the SHV signal is the lower layer, and the power level of the lower layer signal is lower than the power level of the upper layer signal. A combined transmission signal is generated by adding a power difference to both signals and adding them. Here, assuming that the transmission ISDB-T signal is Txi and the transmission SHV signal is Txs, the combined transmission signal Tx is expressed by the equation (1) described later.

The combined transmission signal is supplied to the transmission unit 40. The transmission unit 40 transmits a composite transmission signal via a transmission antenna or the like (not shown).

Next, the receiving device will be described with reference to FIG. 7. In the receiving device according to the embodiment, the receiving signal Rx represented by the equation (2) described later is received.

As shown in the above equation (2), the received signal Rx received by the receiving device includes the combined transmission signal Tx transmitted from the transmitting unit 40 of the transmitting device and the delay profile H which is a propagation path characteristic in the time domain. Noise N is added to the convoluted version of. Since the delay profile H will be described later, detailed description thereof will be omitted here.

The received signal Rx received by a receiving antenna or the like (not shown) is input to the GI removing unit 102 and also input to the receiving replica canceling unit 138. The GI removing unit 102 regards the received signal Rx as an ISDB-T signal having a symbol length of 8K, and removes the GI between the symbols having a symbol length of 8K from the received signal Rx.

The output of the GI removing unit 102 is input to the FFT unit 104. The FFT unit 104 FFTs the input signal with an FFT size equal to the symbol length of 8K, and obtains a signal in the frequency domain having a symbol length of 8K. Since the power of the SHV signal is low, the SHV signal can be regarded as noise, and this signal can be regarded as an ISDB-T signal. The output of the FFT unit 104 is input to the propagation path estimation unit 106, the frequency equalization unit 112, and the delay profile estimation unit 132.

The propagation path estimation unit 106 extracts the SP signal from the input frequency domain signal. From the extracted SP signal, the frequency characteristics of the propagation path corresponding to the 5617 subcarriers (in other words, the propagation path characteristics of the ISDB-T signal) are estimated. The output of the propagation path estimation unit 106 is input to the frequency equalization unit 108.

The frequency equalization unit 108 of the upper layer performs frequency equalization processing on the output of the FFT unit 104 according to the estimation result of the propagation path estimation unit 106, and obtains an output compensated for the propagation path characteristics. The output of the frequency equalization unit 108 is input to the time / frequency deinterleave (De-IL) unit 110.

The time / frequency De-IL unit 112 performs deinterleave processing corresponding to the time / frequency IL unit 28 of the transmitter, and outputs a signal corresponding to the input to the time / frequency IL unit 28, that is, the output of the mapping unit 26. obtain. The output of the time / frequency De-IL unit 112 is input to the log-likelihood ratio (LLR) calculation unit 114.

The LLR calculation unit 114 performs demodulation processing corresponding to the mapping unit 26 of the transmission device, calculates the LLR of the demodulation result, and obtains an input to the mapping unit 26, that is, a signal corresponding to the output of the bit IL unit 24. .. The output of the LLR calculation unit 114 is input to the bit De-IL unit 116.

The bit De-IL unit 116 performs deinterleaving processing corresponding to the bit IL unit 24 of the transmission device, and obtains a signal corresponding to an input to the bit IL unit 24, that is, an output of the convolutional coding unit 22. The output of the bit De-IL unit 116 is input to the Viterbi decoding unit 118.

The Viterbi decoding unit 118 performs decoding processing corresponding to the convolutional coding unit 22 of the transmission device, and obtains a signal corresponding to the input to the convolutional coding unit 22, that is, the output of the byte IL unit 18. The output of the Viterbi decoding unit 118 is input to the byte De-IL unit 122.

The byte De-IL unit 122 performs deinterleaving processing corresponding to the byte IL unit 18 of the transmission device, and obtains a signal corresponding to an input to the byte IL unit 18, that is, an output of the energy diffusion unit 16. The output of the byte De-IL unit 122 is input to the energy reverse diffusion unit 124.

The energy reverse diffusion unit 124 performs energy reverse diffusion processing corresponding to the energy diffusion unit 16 of the transmission device, and obtains a signal corresponding to an input to the energy diffusion unit 16, that is, an output of the RS coding unit 14. The output of the energy reverse diffusion unit 124 is input to the RS decoding unit 126.

The RS decoding unit 126 performs RS decoding processing corresponding to the RS coding unit 14 of the transmission device, and obtains an input to the RS coding unit 14, that is, an information bit string of the ISDB-T signal in the upper layer. The output of the RS decoding unit 126 is input to the transmission replica generation unit 128.

The transmission replica generation unit 128 uses the information bit string of the input ISDB-T signal to use the RS coding unit 14, the energy diffusion unit 16, the byte IL unit 18, the convolutional coding unit 22, and the bit IL unit 24 of the transmission device. , Mapping unit 26, time / frequency IL unit 28, pilot addition unit 32, IFFT unit 34, and GI addition unit 36 are subjected to various processes to generate a transmission ISDB-T signal Txi (transmission replica) in the upper layer. The output of the transmission replica generation unit 128, that is, the generated transmission ISDB-T signal Txi is input to the delay profile estimation unit 132 and the reception replica generation unit 136.

The delay profile estimation unit 132 estimates the delay profile H, which is a propagation path characteristic in the time domain. Since the process of estimating the delay profile H by the delay profile estimation unit 132 will be described later, detailed description thereof will be omitted here. The output of the delay profile estimation unit 132, that is, the estimated delay profile H is input to the FFT unit 134 and the reception replica generation unit 136.

The FFT unit 134 FFTs the delay profile H from the delay profile estimation unit 132 with an FFT size of 32K, and estimates the frequency characteristics of the propagation path used for frequency equalization. The output of the FFT unit 134 is input to the frequency equalization unit 146.

The reception replica generation unit 136 adjusts the power level of the transmission ISDB-T signal Txi from the transmission replica generation unit 128 based on the normalization coefficient β. As shown in the equation (3) described later, the reception replica generation unit 136 convolves the power-adjusted transmission ISDB-T signal βTxi and the delay profile H from the delay profile estimation unit 132, and receives ISDB-T signal Rxi ( (Receive replica) is generated.

The output of the reception replica generation unit 136, that is, the generated reception ISDB-T signal Rxi is input to the reception replica cancellation unit 138.

As shown in the equation (4) described later, the reception replica canceling unit 138 subtracts the reception ISDB-T signal Rxi from the reception replica generation unit 136 from the reception signal Rx received by a reception antenna or the like (not shown).

Thereby, the influence of the ISDB-T signal can be removed from the received signal Rx, and the received signal in the lower layer of the time domain, that is, the received SHV signal Rxs can be obtained. The output of the reception replica canceling unit 138, that is, the receiving SHV signal Rxs is input to the GI removing unit 142.

The GI removing unit 142 removes the GI between symbols having a symbol length of 32K from the received SHV signal Rxs. The output of the GI removing unit 142 is input to the FFT unit 144.

The FFT unit 144 FFTs the input signal with an FFT size of 32K, and obtains a signal in a frequency region having a symbol length of 32K. The output of the FFT unit 144 is input to the frequency equalization unit 146.

The frequency equalization unit 146 performs frequency equalization processing on the output of the FFT unit 144 according to the estimation result of the FFT unit 134, and obtains an output compensated for the propagation path characteristics. The frequency equalization unit divides the output whose propagation path characteristics are compensated by the product of the normalization coefficient β and the scaling coefficient α to adjust the power. The output of the frequency equalization unit 146 is input to the time / frequency De-IL unit 148.

The time / frequency De-IL unit 148 performs deinterleave processing corresponding to the time / frequency IL unit 56 of the transmitter, and outputs a signal corresponding to the input to the time / frequency IL unit 56, that is, the output of the mapping unit 54. obtain. The output of the time / frequency De-IL unit 148 is input to the LLR calculation unit 152.

The LLR calculation unit 152 performs demodulation processing corresponding to the mapping unit 54 of the transmission device, calculates the LLR of the demodulation result, and obtains an input to the mapping unit 54, that is, a signal corresponding to the output of the bit IL unit 52. .. The output of the LLR calculation unit 152 is input to the bit De-IL unit 154.

The bit De-IL unit 154 performs deinterleaving processing corresponding to the bit IL unit 52 of the transmission device, and obtains a signal corresponding to an input to the bit IL unit 52, that is, an output of the LDPC coding unit 48. The output of the bit De-IL unit 154 is input to the LDPC decoding unit 156.

The LDPC decoding unit 156 performs a decoding process corresponding to the LDPC coding unit 48 of the transmission device, and obtains a signal corresponding to an input to the LDPC coding unit 48, that is, an output of the BCH coding unit 46. The output of the LDPC decoding unit 156 is input to the BCH decoding unit 158.

The BCH decoding unit 158 performs decoding processing corresponding to the BCH coding unit 46 of the transmission device, and obtains a signal corresponding to an input to the BCH coding unit 46, that is, an output of the energy diffusion unit 44. The output of the BCH decoding unit 158 is input to the energy reverse diffusion unit 162.

The energy reverse diffusion unit 162 performs the energy reverse diffusion processing corresponding to the energy diffusion unit 44 of the transmission device, and obtains an input to the energy diffusion unit 44, that is, an information bit string of the SHV signal in the lower layer.

Next, the delay profile H and its estimation method will be described.
The delay profile H is a propagation path characteristic in the time domain. For example, as shown in FIG. 8, the delay time from the reception of the direct wave (leading wave) to the reception of the delayed wave in the receiving device and the delay thereof. It shows the complex amplitude of the wave. The origin in FIG. 8 corresponds to the time when the direct wave was received. When a plurality of delayed waves are received in the receiving device, the delay profile H indicates the delay time from the reception of the direct wave to the reception of the plurality of delayed waves, and the complex amplitude of each delayed wave. In this embodiment, it is assumed that the delay profile H is an 8K size row vector. Further, in the present embodiment, it is assumed that there is no delay exceeding GI. In the present embodiment, since it is assumed that the symbol length of the GI is 1K, zero is assigned to the value after the 1024th row vector of the 8K size, and the delay profile H is [h0, It is defined as h1, ..., h1023, 0, 0, ..., 0]. In the present embodiment, it is assumed that the delay profile H is a row vector of 8K size, but the present embodiment is not limited to this, and for example, the delay profile H may be a row vector of 1K size equal to the symbol length of GI. good. Further, in the present embodiment, it is assumed that there is no delay exceeding GI, but the present invention is not limited to this, and a delay exceeding GI may be provided. In this case, the number of unknowns (variables) included in the row vector of the delay profile H may be increased according to the maximum value of the delay. Further, in the present embodiment, it is assumed that the delay profile H is a row vector, but the delay profile H is not limited to this, and the delay profile H is the delay time from the reception of the direct wave to the reception of the delay wave. And any complex amplitude of the delayed wave may be used.

The delay profile H is estimated by the delay profile estimation unit 132. Specifically, first, the delay profile estimation unit 132 FFTs the transmission ISDB-T signal Txi from the transmission replica generation unit 128 with an FFT size equal to the symbol length of 8K, and the transmission signal in the frequency domain with the symbol length of 8K. Obtain a Txif. Here, assuming that the output of the FFT unit 104, that is, the received signal in the frequency domain having a symbol length of 8K is Rxf, the delay profile estimation unit 132 receives the received signal Rxf in the frequency domain as shown in the equation (5) described later. Is divided by the transmission signal Txif in the frequency domain to obtain the propagation path estimated value Hf in the frequency domain.

In the present embodiment, the delay profile estimation unit 132 obtains the propagation path estimated value Hf in the frequency domain based on the above equation (5), but the present invention is not limited to this, and for example, the delay profile estimation unit 132. The 132 may obtain the propagation path estimation value Hf in the frequency domain by acquiring the output of the propagation path estimation unit 106.

Here, the ISDB-T signal has an FFT size of 8K, but the symbols are molded with both ends set to zero. Therefore, in the propagation path estimated value Hf in the frequency domain, the central portion of the 8K samples Only 5617 samples have values. As described above, when the FFT size and the actual number of samples are different, the delay profile H cannot be accurately estimated even if the propagation path estimated value Hf in the frequency domain is IFFTed in 8K size.

Therefore, the delay profile estimation unit 132 estimates the delay time by using the MUSIC (MUSIC Signal Classification) algorithm, and estimates the complex amplitude of the delay wave by using the LS (Least Squares) algorithm to estimate the delay profile. Estimate H with high accuracy.

In the following, first, a method of estimating the delay time using the MUSIC algorithm will be described.
Here, as an example, it is assumed that K delay waves arrive after the direct wave. In this case, the correlation matrix Rxx (hereinafter, simply referred to as the correlation matrix Rxx of the propagation path estimation value) corresponding to the 5617 samples having values in the propagation path estimated value Hf in the frequency domain has the eigenvalues λ and Using the eigenvector e, it is expressed as in the equation (6) described later. Here, it is assumed that the correlation matrix Rxx of the propagation path estimated value is incoherent.

In equation (6) above, I is the identity matrix, S is the correlation matrix of the delay profile, σ ² is the noise variance, and H is the Hermitian transpose. Further, i shown in the above equation (6) is the number of samples having a value in the propagation path estimated value Hf. That is, in this embodiment, L represented by the above formula (6) is 5617. Here, A shown in the above equation (6) is a vector called a mode vector, and is expressed as in the equation (7) described later.

The a (τK) shown in the above equation (7) can be expressed as the equation (8) described later.

ΤK shown in the above equations (7) and (8) is the delay time of the delayed wave arriving at the Kth position. Further, Δf shown in the above equation (8) is a subcarrier interval. Here, since it is assumed that K delay waves arrive, the eigenvalue λ shown in the above equation (6) becomes a noise dispersion value after the K + 1th order. Therefore, the number K of the delayed wave can be determined by calculating the eigenvalue λ.

When the number of delayed waves is K, the mode vector A has a property of being orthogonal to the K + 1th and subsequent noise vectors of the eigenvectors e. Therefore, the equation (9) described later is defined as a MUSIC spectrum, and this MUSIC spectrum is calculated while changing τK sequentially to obtain τK whose denominator becomes zero. The delay time can be estimated.

The method of estimating the delay time using the MUSIC algorithm is, for example, "Marian Ozewicz," On Application of MUSIC Algorithm to Time Delivery Estimation in OFDM Channel, "in IEET TRANSACTION. It is explained in "JUNE 2005".

Further, when the correlation matrix Rxx of the propagation path estimation value is coherent, the delay time of the K-th arrival delay wave, that is, τK can be estimated by recovering the rank by spatial averaging or the like. The method of estimating the delay time when the correlation matrix Rxx of the propagation path estimation value is coherent is described in, for example, "Nobuyoshi Kikuma," Adaptive Signal Processing in Array Antenna ", Science and Technology Publishing, 1998".

The delay profile estimation unit 132 executes a process based on the delay time estimation method using the MUSIC algorithm described above, and estimates the delay time.

Next, a method of estimating the complex amplitude of the delayed wave using the LS algorithm will be described.
First, among the unknowns (that is, h0 to h1023) included in the row vector of the delay profile H, zero is substituted for the unknowns estimated to have no delay wave as a result of the estimation using the MUSIC algorithm described above. Then, the value FH ^T obtained by multiplying the DFTmatrix of 8K size transpose ^{H T} of the delay profile H, the equation (5) the smallest is the square error of the propagation path characteristics Hf in the frequency domain obtained by using a Find H (row vector) such that Note that F shown in the formula (10) described later is an 8K size DFT (discrete Fourier transform) matrix.

The T shown in the above equation (10) represents transposition.

Minimum value determined by the equation (10) described above, since the zero is differentiated the equation (10), calculates the equation (11) described later, by obtaining the transposition H ^T of the delay profile, the delay wave Complex amplitude can be estimated. Note that F_inv shown in equation (11) is an inverse matrix of F.

The above equation (11) may be calculated by extracting only the columns estimated to have delayed waves. In this case, F_inv is a pseudo-inverse matrix. Further, the estimation of the complex amplitude of the delayed wave may be performed using only the 5617 lines having a value in the propagation path estimated value Hf in the frequency domain.

The delay profile estimation unit 132 executes processing based on the method for estimating the complex amplitude of the delayed wave using the LS algorithm described above, and estimates the complex amplitude of the delayed wave.

As described above, the delay profile estimation unit 132 uses the MUSIC algorithm to estimate the delay time of the incoming delayed wave, and uses the LS algorithm to estimate the complex amplitude of the delayed wave. Can be done. In other words, the delay profile estimation unit 132 can estimate the delay profile H by using the MUSIC algorithm and the LS algorithm. According to this, since the reception replica canceling unit 138 can execute the process based on the above equation (4), the influence of the ISDB-T signal can be removed from the received signal Rx, and the time domain can be removed. The received signal in the lower layer of the above, that is, the received SHV signal Rxs having a symbol length of 32K can be obtained. According to this, the receiving device can demodulate the SHV signal from the received signal Rx multiplexed by the asynchronous LDM method.

In the present embodiment, the delay profile estimation unit 132 uses the MUSIC algorithm to estimate the delay time of the incoming delayed wave, and then uses the LS algorithm to estimate the complex amplitude of the delayed wave. However, the delay profile H is estimated, but the delay profile estimation unit 132 estimates the complex amplitude of the delay wave by using the LS algorithm without estimating the delay time by using the MUSIC algorithm. Then, the delay profile H may be estimated. In this case, the delay profile estimation unit 132 may perform estimation using the LS algorithm after leaving all the unknowns included in the row vector of the delay profile H as unknowns.

Further, in the present embodiment, it is assumed that the propagation path estimated value Hf in the frequency domain is 8K size, but the present invention is not limited to this, and for example, by reducing the number of subcarriers used in the MUSIC algorithm, The size of the propagation path estimated value Hf in the frequency domain may be reduced.

Further, in the present embodiment, the case where the delay profile H is the propagation path characteristic in the time domain has been described, but the present invention is not limited to this, and the delay profile H may be the propagation path response in the frequency domain. In this case, the received ISDB-T signal Rxi (reception replica) to be subtracted from the received signal Rx may be obtained by multiplying the transmitted ISDB-T signal Txi by the propagation path response in the frequency domain.

According to one embodiment described above, the receiving device receives the synthetic transmission signal transmitted from the transmitting device. The receiving device receives a received signal including an ISDB-T signal having a symbol length of 8K and an SHV signal having a symbol length of 32K and a power level lower than that of the ISDB-T signal. The receiving device demodulates the SHV signal by subtracting the signal generated by convolving the ISDB-T signal and the delay profile from the received signal. According to this, it becomes possible to provide a receiving device, a transmitting device, a receiving method, and a transmitting method capable of demodulating an OFDM signal multiplexed by an asynchronous LDM method.

Although some embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other embodiments, and various omissions, replacements, and changes can be made without departing from the gist of the invention. These embodiments and modifications thereof are included in the scope and gist of the invention, and are also included in the scope of the invention described in the claims and the equivalent scope thereof.

102 ... GI removal unit (8K), 104 ... FFT unit (8K), 106 ... propagation path estimation unit (8K), 108 ... frequency equalization unit (8K), 112 ... time / frequency De-IL unit, 114 ... LLR Calculation unit, 116 ... Bit De-IL unit, 118 ... Bitabi decoding unit, 122 ... Byte De-IL unit, 124 ... Energy reverse diffusion unit, 126 ... RS decoding unit, 128 ... Transmission replica generation unit, 132 ... Delay profile estimation Unit, 134 ... FFT unit (32K), 136 ... Received replica generation unit, 138 ... Received replica canceling unit, 142 ... GI removal unit (32K), 144 ... FFT unit (32K), 146 ... Frequency equalization unit (32K) , 148 ... Time / frequency De-IL unit, 152 ... LLR calculation unit, 154 ... Bit De-IL unit, 156 ... LDPC decoding unit, 158 ... BCH decoding unit, 162 ... Energy reverse diffusion unit.

Claims (12)

A receiving device that receives a signal transmitted from a transmitting device.
A first OFDM signal having a first symbol length and a first power level and a second having a second symbol length longer than the first symbol length and lower than the first power level. A receiver that receives a receive signal, including a second OFDM signal of power level,
A receiving device including a demodulator that demodulates the second OFDM signal by subtracting the signal generated by convolving the first OFDM signal and the delay profile from the received signal. The demodulation unit
The receiver according to claim 1, wherein the delay profile is estimated using the Least Squares algorithm. The demodulation unit
The reception according to claim 1, wherein the delay profile is estimated by estimating the delay time of the delayed wave using the MUSIC Signal Classification algorithm and estimating the complex amplitude of the delayed wave using the Last Square algorithm. Device. In at least one period of receiving the received signal, the interval of the guard interval added to the first OFDM signal and the interval of the guard interval added to the second OFDM signal are different. The receiving device according to any one of claims 1 to 3. The first OFDM signal includes a signal conforming to the current terrestrial digital television broadcasting system.
The receiving device according to any one of claims 1 to 4, wherein the second OFDM signal includes a signal conforming to the next-generation terrestrial digital television broadcasting system. A first OFDM signal having a first symbol length and a first power level and a second having a second symbol length longer than the first symbol length and lower than the first power level. It is provided with a transmitter for transmitting a transmission signal generated by addition to a second OFDM signal of a power level.
The transmitter unit subtracts a signal generated by convolving the first OFDM signal and the delay profile from the received signal corresponding to the transmitted signal, and demodulates the second OFDM signal. A transmission device that transmits the transmission signal to the user. A receiving method applied to a receiving device that receives a signal transmitted from a transmitting device.
A first OFDM signal having a first symbol length and a first power level and a second having a second symbol length longer than the first symbol length and lower than the first power level. The step of receiving a received signal, including a second OFDM signal of the power level,
A receiving method comprising: subtracting a signal generated by convolving the first OFDM signal and the delay profile from the received signal to demodulate the second OFDM signal. The receiving method according to claim 7, further comprising a step of estimating the delay profile using the Least Squares algorithm. A claim further comprising a step of estimating the delay profile by estimating the delay time of the delayed wave using the Multiple SIgnal Classification algorithm and estimating the complex amplitude of the delayed wave using the Last Square algorithm. 7. The receiving method according to 7. In at least one period of receiving the received signal, the interval of the guard interval added to the first OFDM signal and the interval of the guard interval added to the second OFDM signal are different. The receiving method according to any one of claims 7 to 9. The first OFDM signal includes a signal conforming to the current terrestrial digital television broadcasting system.
The receiving method according to any one of claims 7 to 10, wherein the second OFDM signal includes a signal conforming to the next-generation terrestrial digital television broadcasting system. It is a transmission method applied to the transmission device.
A first OFDM signal having a first symbol length and a first power level and a second having a second symbol length longer than the first symbol length and lower than the first power level. It comprises a step of transmitting a transmit signal generated by addition of the power level to a second OFDM signal.
The step of transmitting the transmission signal is to demodulate the second OFDM signal by subtracting the signal generated by convolving the first OFDM signal and the delay profile from the reception signal corresponding to the transmission signal. A transmission method comprising transmitting the transmission signal to one receiving device.

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