JP2019017176A - Linear motor system - Google Patents

Abstract

To provide a linear motor system that can drive a linear motor efficiently at a mechanical resonance frequency considering load without a need for a sensor for detecting an induction voltage that changes depending on variation in load.SOLUTION: A linear motor system 100 comprises a linear motor 104 that has a movable element 6 at least connected with a coil 8 applied with an AC voltage and an elastic body 23. A resonance frequency of the movable element 6 is varied by the elastic body 23 and load of the linear motor 104. The linear motor system 100 has a linear motor drive device 102 that detects a fundamental wave amplitude of a current flowing in the coil 8, and increases a phase difference between the AC voltage Vm and the current Im flowing in the coil 8, in accordance with increase in the fundamental wave amplitude.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a linear motor system, and more particularly to a linear motor system including a linear motor having a mover to which an elastic body is connected.

There is known a linear motor that drives a mover connected to an elastic body at a mechanical resonance frequency in a system of the elastic body and the mover. Since the mechanical resonance frequency varies depending on the friction of the mover and the load connected to the mover, it is desirable to estimate the resonance frequency effectively.
For example, Patent Document 1 discloses a configuration in which a phase of an induced voltage from a search coil is detected, a phase difference with a current phase flowing through a linear motor is detected, and the resonance frequency of the piston is matched according to the phase difference. Yes. Patent Document 1 describes a configuration in which the piston stroke is kept constant by correcting the voltage value of the output voltage by a value corresponding to the frequency of the output voltage.

Japanese Patent Laid-Open No. 11-351143

In Patent Document 1, a search coil is provided in the winding of the linear compressor, and the phase of the induced voltage is detected. However, when the search coil is used, not only the wiring becomes complicated, but also it is easily affected by noise, so it is not easy to control the resonance frequency with high accuracy. In addition, there is no specific configuration for correcting the voltage value of the output voltage by a value corresponding to the frequency of the output voltage and keeping the stroke of the piston of the linear compressor constant. Is not easy.
Accordingly, the present invention provides a linear motor system that can drive a linear motor with high efficiency at a mechanical resonance frequency including a load without requiring a sensor that detects an induced voltage that changes in accordance with a change in the load. .

In order to solve the above problems, a linear motor system according to the present invention includes a linear motor having at least a winding to which an AC voltage is applied and a mover to which an elastic body is connected, and depending on a load of the elastic body and the linear motor. A linear motor system in which a resonance frequency of a mover fluctuates, and includes a linear motor driving device that adjusts a phase difference between the AC voltage and a current flowing through the winding so that the driving frequency of the AC voltage becomes a resonance frequency. It is characterized by that.
The linear motor system according to the present invention includes a linear motor having at least a winding to which an AC voltage is applied and a mover to which an elastic body is connected, and a resonance frequency of the mover by a load of the elastic body and the linear motor. Is a linear motor system that has a linear motor driving device that increases a phase difference between the AC voltage and the current flowing through the winding in accordance with an increase in the load.
Furthermore, a linear motor system according to the present invention includes a linear motor having at least a winding to which an AC voltage is applied and a mover to which an elastic body is connected, and resonance of the mover by a load of the elastic body and the linear motor. A linear motor system in which the frequency varies, detecting a fundamental wave amplitude of a current flowing through the winding, and increasing a phase difference between the AC voltage and a current flowing through the winding in accordance with an increase in the fundamental wave amplitude. And a linear motor driving device.

According to the present invention, there is provided a linear motor system capable of driving a linear motor with high efficiency at a mechanical resonance frequency including a load without requiring a sensor for detecting an induced voltage that changes in accordance with a change in load. It becomes possible.
Problems, configurations, and effects other than those described above will be clarified by the following description of embodiments.

1 is an overall schematic configuration diagram of a linear motor system according to a first embodiment of the present invention. It is a perspective view of the structural example of an armature. It is a schematic diagram which shows the longitudinal cross-section of a magnetic pole, and the flow of magnetic flux. It is explanatory drawing of the polarity which generate | occur | produces in a magnetic pole tooth. It is explanatory drawing of the external mechanism connected to a needle | mover. It is explanatory drawing of the relationship between a drive frequency and a stroke. It is explanatory drawing of the phase relationship of the position of a needle | mover, and the speed | rate of a needle | mover, and the phase relationship of an applied voltage and a motor current. It is a vector diagram of applied voltage and current. It is explanatory drawing which shows the structural example of the phase difference detector which comprises the control part shown in FIG. It is explanatory drawing of the relationship between a drive frequency and a phase difference detector output. It is explanatory drawing which shows the structural example of the drive frequency regulator which comprises the control part shown in FIG. It is explanatory drawing which shows the structural example of the induced voltage component creator which comprises the control part shown in FIG. It is explanatory drawing which shows the structural example of the stroke controller which comprises the induced voltage component preparation device shown in FIG. It is explanatory drawing which shows the structural example of a position estimation part. It is explanatory drawing which shows the structural example of the load current detector which comprises the control part shown in FIG. It is explanatory drawing which shows the other structural example of the load current detector which comprises the control part shown in FIG. It is explanatory drawing which shows the structural example of the voltage drop component preparation device which comprises the control part shown in FIG. It is explanatory drawing which shows the structural example of the voltage command value preparation device which comprises the control part shown in FIG. It is a vector diagram which shows the vector sum in the voltage command value preparation machine at the time of light load and heavy load. It is explanatory drawing which shows the time change of an alternating voltage command phase and a drive frequency command value. It is a figure which shows the structural example of the power converter circuit which comprises the linear motor drive device shown in FIG. It is a longitudinal cross-sectional view of the hermetic compressor of Example 2 which concerns on the other Example of this invention. It is a whole schematic block diagram of the linear motor system of Example 2. It is explanatory drawing which shows the structural example of the load current detector which comprises the control part shown in FIG. It is explanatory drawing which shows the structural example of the voltage drop component preparation device which comprises the control part shown in FIG. It is explanatory drawing which shows the structural example of the voltage command value preparation device which comprises the control part shown in FIG. It is a circuit diagram of the air suspension system of Example 3 which concerns on the other Example of this invention. It is the schematic of the vehicle carrying the air suspension system shown in FIG. It is explanatory drawing which shows the structural example of a load current detector.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings. Similar components are denoted by the same reference numerals, and redundant description is omitted.
The various components of the present invention do not necessarily have to be independent of each other. A plurality of components are formed as a single member, and a single component is formed of a plurality of members. It is allowed that a certain component is a part of another component, a part of a certain component overlaps a part of another component, and the like.

  In this embodiment, for convenience of explanation, the terms front-rear direction, left-right direction, and up-down direction that are orthogonal to each other are used, but the gravitational direction is not necessarily parallel to the lower direction. It can be parallel to other directions.

<Linear motor system 100>
FIG. 1 is an overall schematic configuration diagram of a linear motor system according to a first embodiment of the present invention. The linear motor system 100 includes a linear motor driving device 101 and a linear motor 104. As will be described later, the linear motor 104 has an armature 9 and a mover 6 that move relative to each other.
The linear motor driving apparatus 101 includes a current detector 107, a control unit 102, and a power conversion circuit 105.
In this embodiment, the mover 6 moves in the vertical direction, but the armature 9 and the mover 6 (field element) may move relative to each other, and the armature 9 may move in the vertical direction. In the following, the case where the movable element 6 reciprocates in the vertical direction will be described as an example, but the direction of reciprocation is not limited to the vertical direction. For example, the movable element 6 may be configured to reciprocate in the horizontal direction, or the movable element 6 may be configured to reciprocate in a direction having an arbitrary angle with respect to the vertical direction. The same applies to the armature 9.
The control unit 102 outputs an output voltage command value to the power conversion circuit 105 or a drive signal (pulse signal) for driving the power conversion circuit 105 according to the detection result of the current detector 107. Details of the control unit 102 will be described later.
Although details will be described later, the power conversion circuit 105 is an example of a power conversion unit that converts the voltage of the DC voltage source 120 (FIG. 21) and outputs an AC voltage. Note that a DC current source may be used instead of the DC voltage source 120.

<Linear motor 104>
FIG. 2 is a perspective view of the linear motor 104 (a perspective view of a configuration example of the armature). The linear motor 104 of the present embodiment has a mover 6 that can move relative to the armature 9 in the direction (front-rear direction) in which the permanent magnets 2 (2a, 2b) are arranged. The armature 9 has two magnetic poles 7 that face each other in the vertical direction through a gap, and a winding 8 wound around the magnetic pole 7. The mover 6 is disposed in this gap. The magnetic pole 7 has magnetic pole teeth 70 (also referred to as teeth) as end faces facing the mover 6.
The armature 9 can apply a force in the front-rear direction (hereinafter referred to as thrust) to the mover 6. For example, as will be described later, the thrust can be controlled so that the mover 6 reciprocates in the front-rear direction.
The mover 6 includes two flat permanent magnets 2 (2a, 2b) magnetized in the vertical direction. The rear permanent magnet 2a and the front permanent magnet 2b are magnetized in opposite directions. In this embodiment, the rear permanent magnet 2a has an N pole on the upper side, and the front permanent magnet 2b has an S pole on the upper side. In FIG. 2, the permanent magnets 2a and 2b are shown, but the mover 6 is not shown. As the mover 6, for example, a flat plate with a flat permanent magnet 2 fixed can be adopted.
The control unit 102 outputs a drive signal so that the mover 6 reciprocates in a range where the permanent magnets 2 a and 2 b face the armature 9.

  FIG. 3 is a cross-sectional view taken along the line A-A ′ of FIG. 2 (A-A ′ cross-sectional view). As shown in FIG. 3, the magnetic pole 7 and the yoke 7e are integrally formed of a magnetic material such as iron to constitute a magnetic circuit. The arrow lines in FIG. 3 show an example of magnetic flux lines when current is passed through the two windings 8. The direction of the flow of magnetic flux is not limited to that shown in the figure because it can be reversed depending on the direction of the current flowing through the winding 8. The magnetic pole teeth 70 are magnetized by the magnetic flux lines.

[Thrust applied to the mover 6]
FIG. 4 is a diagram for explaining the thrust received by the mover 6 due to the magnetization of the magnetic pole teeth 70. The polarities of the magnetic pole teeth 70 generated by the current flowing in the winding 8 are represented by “N” and “S” attached in the vicinity of the magnetic pole teeth 70 in the drawing. Further, in FIG. 4, the white arrow indicates the direction of the current flowing through the winding 8. The left diagram of FIG. 4 shows that the upper magnetic pole tooth 70a is magnetized to “S” and the lower magnetic pole tooth 70b is magnetized to “N” by the current flowing through the winding 8, so that the movable element 6 exerts a force in the forward direction. An example is shown in which the mover 6 has moved forward. The right diagram in FIG. 4 shows that the upper magnetic pole tooth 70a is magnetized to “N” and the lower magnetic pole tooth 70b is magnetized to “S” by the current flowing through the winding 8, so that the movable element 6 exerts a force in the backward direction. In this example, the mover 6 is moved backward.

  In this way, by applying a voltage or current to the winding 8, magnetic flux is supplied to the magnetic circuit including the two magnetic poles 7, and the two opposing magnetic pole teeth 70 (magnetic pole teeth set) can be magnetized. By applying an alternating voltage or current such as a sine wave or rectangular wave (square wave) as the voltage or current, a thrust for reciprocating the mover 6 can be applied. Thereby, the movement of the mover 6 can be controlled.

Note that the thrust applied to the mover 6 can be changed by changing the amplitude of the applied alternating current or alternating voltage. Moreover, the displacement of the mover 6 can be changed as desired by appropriately changing the thrust applied to the mover 6 using a known method. Here, when the mover 6 reciprocates (for example, a movement that occurs in the mover 6 by sequentially repeating the magnetization of the magnetic pole teeth 70 as shown in the left and right diagrams of FIG. 4), it changes in an alternating waveform. The amount of change in the displacement of the mover 6 is called a stroke.
Since the magnetic pole teeth 70 are magnetic bodies, a magnetic attractive force that attracts the permanent magnet 2 acts. In the present embodiment, since the two magnetic pole teeth 70 are opposed to each other with a gap so as to sandwich the mover 6, the resultant magnetic attraction force acting on the mover 6 can be reduced.

[Mechanism outside the mover 6]
FIG. 5 is an explanatory diagram of an external mechanism connected to the mover 6. For example, an external mechanism constituted by a resonance spring 23 (assist spring) that is a coil spring is connected to one end of the mover 6, and its spring force It is a figure explaining the mechanism in which the needle | mover 6 is returned by FIG. One end of the resonance spring 23 is connected to the mover 6 via the intermediate portion 24, and the other end is fixed to the base portion 25. Further, a side portion 26 that extends substantially parallel to the extending direction of the resonance spring 23 and guides or supports the resonance spring 23 is provided. When the linear motor 104 is reciprocated, acceleration and deceleration are repeated each time the moving direction of the mover 6 changes. During deceleration, the velocity energy of the mover 6 is converted into electrical energy (regenerative operation), but loss occurs due to the resistance of the wiring to the linear motor 104. On the other hand, as shown in FIG. 5, when a resonance spring 23 (assist spring) is added to the mover 6 and the mover 6 is reciprocated at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant, the mover 6 is movable. The speed energy of the child 6 can be effectively used, and a highly efficient linear motor drive system can be configured. Instead of the resonance spring 23, for example, a leaf spring or an elastic body such as rubber having an appropriate Young's modulus and expanding and contracting similarly to the case of using a coil spring may be used. With this configuration, the movable element 6 (field element 6) is configured as a movable element (field element) moving type that moves in the vertical direction, but an elastic body is connected to the armature 9 instead of the movable element 6. The armature 9 may be configured as an armature moving type that moves the armature 9 in the vertical direction.

FIG. 6 is an explanatory diagram of the relationship between the driving frequency and the stroke, and shows the relationship between the driving frequency of the AC voltage on the horizontal axis and the stroke of the mover 6 on the vertical axis. The amplitude of the AC voltage at each drive frequency is the same. As can be seen from FIG. 6, the stroke of the mover 6 increases sharply near the resonance frequency, and the stroke decreases as the distance from the resonance frequency decreases. The resonance frequency is given by the square root of the value obtained by dividing the spring constant k of the resonance spring 23 by the mass m of the mover 6, but this value is an approximate value depending on the system of the linear motor 104.
Thus, by applying an AC voltage having a resonance frequency or a driving frequency in the vicinity of the resonance frequency, it is possible to vibrate with a large stroke (large energy). That is, when controlling the linear motor 104 in which an elastic body such as the resonance spring 23 is added to the mover 6, it is important to detect or estimate the resonance frequency of the mover 6. Even when the stroke of the mover 6 is controlled as desired, it is important to detect or estimate the resonance frequency of the mover 6.

[Phase relationship during driving]
FIG. 7 is an explanatory diagram of the phase relationship between the position of the mover and the speed of the mover, and the phase relationship between the applied voltage and the motor current. When the linear motor 104 is driven, the upper diagram of FIG. 7 shows the time variation of the position and speed of the mover 6, and the lower diagram of FIG. 7 shows the relationship between the applied voltage waveform and the time variation of the current flowing through the linear motor 104. . Note that the upper diagram and the lower diagram in FIG. 7 are waveforms at the same timing. FIG. 8 is a diagram showing the AC waveform of FIG. 7 as a vector. When the mover 6 connected to an external mechanism having the resonance spring 23 is reciprocated, the displacement of the mover 6 changes in a sine wave shape. Since the speed of the mover 6 is a time derivative of the displacement, it changes in a cosine wave shape. Therefore, they can be shown as vectors on two orthogonal axes. 7 and 8, it can be seen that the speed of the mover 6, the applied voltage, and the motor current are substantially in phase.
When the resonance spring 23 is added to the mover 6 and the mover 6 is reciprocated at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant, the phase of the position of the mover 6 is the winding 8. It is known that there is a phase difference of 90 degrees with respect to each of the applied voltage Vm, the motor current Im, and the phase of the speed of the mover 6. That is, when any of these relationships is established, it can be estimated that the drive is performed at the resonance frequency.

Depending on the value of the winding resistance and winding inductance of the linear motor 104 or the load element added to the mover 6, the phase relationship among the voltage Vm applied to the winding 8, the motor current Im, and the speed of the mover 6 is Since the phase difference is not necessarily 90 degrees, it is desirable to include a control unit that changes according to conditions. At this time, in particular, it is desirable to take into account a change in the motor current Im caused by a load variation.
When the mass of the mover 6 deviates from the assumption due to manufacturing variations, or when the mass connected to the resonance spring 23 changes due to the load element added to the mover 6, the resonance frequency changes. In addition, when the load element added to the mover 6 has position dependency, the resonance frequency changes during driving. Even in such a case, in order to obtain a desired stroke, it is preferable to detect or estimate a resonance frequency that varies depending on conditions with high accuracy. Hereinafter, a method for detecting or estimating the resonance frequency, control of the drive frequency, and control of the voltage and current applied to the winding 8 will be described.

<Control unit 102>
As shown in FIG. 1, the control unit 102 uses a phase difference detector 130 and a phase difference estimated value dltθ ^ (hereinafter simply referred to as a phase difference) that is an output of the phase difference detector 130 as a phase difference command value dltθ ^* . A drive frequency adjuster 131 that adjusts the drive frequency command value ω ^* to follow, an integrator 140, a load current detector 136, an induced voltage component creator 135 that creates an induced voltage component, and a voltage drop that creates a voltage drop component component generator 137, the voltage command value generator 103 outputs a voltage command value Vm ^*, and, by comparing the voltage command value Vm ^* and the triangular wave carrier signal, a drive signal for driving the power conversion circuit 105 which outputs a voltage The output PWM signal generator 134 is configured.

The control unit 102 inputs the motor current Im from the current detector 107. The motor current Im is an AC waveform. The motor current Im input to the control unit 102 is input to the phase difference detector 130 and the load current detector 136 that constitute the control unit 102. The phase difference detector 130 outputs a reference phase θ ^* that is a phase command value, which will be described in detail later, and a phase difference dltθ ^ of the motor current Im. The phase difference dltθ ^ output from the phase difference detector 130 is input to the drive frequency adjuster 131. The drive frequency adjuster 131 outputs a frequency command value ω ^* based on the input phase difference dltθ ^. An applied voltage Vm based on the frequency command value ω ^* is output to the linear motor 104.

When the mass of the mover 6 and the spring constant of the resonance spring 23 deviate from the assumption, the resonance frequency changes from the assumed value, but the influences are the applied voltage Vm to the winding 8, the motor current Im, and the movement. Since it can be seen from the phase relationship of the speed of the child 6, by controlling based on these phase relationships, a highly efficient linear motor system can be configured. Even when the load element added to the mover 6 has a position dependency, the control is performed based on the phase relationship among the voltage Vm applied to the winding 8, the motor current Im, and the speed of the mover 6. An efficient linear motor system can be configured.
Hereinafter, the configuration or operation of each of the above-described units constituting the control unit 102 will be described.

<Reference phase generator>
The reference phase θ ^*, which is the phase command value of the present embodiment, is obtained by integrating the drive frequency command value ω ^* , which is the output of the drive frequency adjuster 131 in FIG. 1, with the integrator 140 as a reference phase generator. That is, the reference phase theta ^* applied voltage Vm (θ ^*) is the phase theta ^* at each time of the wave having a driving frequency command value omega ^* corresponding to the target frequency. Thus, reference the phase theta ^* as is used driving frequency command value of the drive frequency regulator 131 omega ^*, for example, may be fixed to the mechanical resonance frequency of the vibration body including the movable member 6 in the present embodiment.
While the drive frequency command value ω ^* is constant, the reference phase θ ^* is, for example, a sawtooth wave having a range of [−π, π], [0, 2π] or a wider range for each time. Or may increase linearly with respect to time. As will be described later, when the drive frequency command value ω ^* changes, the sawtooth wave and the linear increase shape change accordingly (the inclination changes).

<Phase difference detector 130>
When the mover 6 is reciprocating, the position xm, speed, and motor current Im of the mover 6 are periodic functions. Since the periodic function can be expressed by a Fourier series, if the position xm of the movable element 6 is expressed using a Fourier transform equation, it can be defined as the following equation (1).

Here, _{x 0} is the DC offset value, _{a n} and _{b n} are the Fourier coefficients of the order n is obtained by the following formula (2) and (3).

Here, T ₀ is the period of the fundamental wave (period in which the movable element 6 reciprocates), that is, the reciprocal of the primary frequency (drive frequency). In a so-called spring mass system in which a resonance spring 23 is added to the mover 6, the mechanical resonance frequency determined from the mass of the mover 6 and the spring constant is the dominant component. Therefore, attention should be paid to the fundamental wave.

  When controlling to drive the mover 6 at the resonance frequency, the high-order component is not important, and attention should be paid to the primary component, that is, the drive frequency component. In particular, the phase of the primary frequency component (drive frequency component) at the position xm of the mover 6 is important. The position xm of the movable element 6 with respect to the sinusoidal applied voltage Vm can be obtained by the following equation (4) by the arc tangent of the first-order Fourier coefficient.

  In Expression (4), the integration range is −2π to 0. This is because when the phase difference detector 130 is realized by a semiconductor integrated circuit such as a microcomputer or a DSP (Digital Signal Processor), only past information can be acquired.

In the above formulas (1) to (4), the phase of the position xm of the mover 6 with respect to the reference phase θ ^* is calculated. Similarly, by using the motor current Im, the following formula (5) is obtained. Thus, the phase of the motor current Im with respect to the reference phase θ ^* can be calculated.

FIG. 9 is an explanatory diagram illustrating a configuration example of the phase difference detector 130 included in the control unit 102 illustrated in FIG. 1, and is an explanatory diagram when Expression (5) is illustrated in a block diagram. A reference phase θ ^* that is a phase command value is input to each of the sine calculator 81 that outputs the sine of the input value and the cosine calculator 82 that outputs the cosine of the input value, and the reference phase θ ^* (phase command value) is output. Get the sine and cosine. A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92. When the outputs are integrated by integrators 94a and 94b, first-order Fourier coefficients of sine and cosine are obtained. That is, since higher-order frequency components can be eliminated from the drive frequency ω of the Fourier expansion, it can be configured robustly against higher-order noise.
The outputs of the integrators 94 a and 94 b are input to the arc tangent device 86. The arc tangent device 86 outputs an arc tangent value based on the input sine and cosine components. The arc tangent unit 86 of this embodiment outputs an arc tangent value of the phase with the numerator as the output of the integrator 94a and the denominator as the output of the integrator 94b, but even if a value obtained by reversing the numerator and denominator is output. good.

FIG. 10 is an explanatory diagram of the relationship between the drive frequency and the output from the phase difference detector 130. That is, it is a diagram showing the relationship between the drive frequency (horizontal axis) of the AC voltage and the output value (phase difference dltθ ^ (θpos ^)) (vertical axis) of the arctangent 86. As can be seen from FIG. 10, in this embodiment, when the drive frequency is the resonance frequency, 0 ° is output from the arc tangent device 86. The value output from the arc tangent device 86 is greater than 0 ° when the drive frequency is higher than the resonance frequency, and less than 0 ° when the drive frequency is lower than the resonance frequency. Thus, (in this embodiment, the motor current Im) input AC signal to the phase difference detector 130 with respect to the reference phase theta ^* can obtain the phase difference Dltshita ^ of primary frequency components of the reference phase theta ^*, resonance The frequency can be estimated. For example, when the phase difference dltθ ^ is greater than 0 °, the phase difference dltθ ^ is controlled to decrease, and when the phase difference dltθ ^ is less than 0 °, it may be controlled to increase. It is preferable to control the phase difference dltθ ^ when the reference phase θ ^* and the fundamental frequency ω have the same value as the target value dltθ ^* .

  An incomplete integrator can be used in place of the integrators 94a and 94b. The incomplete integrator is a kind of low-pass filter (low-pass filter) and can be configured similarly to the first-order lag filter. Of course, the low-pass filter may be configured with other known configurations such as a secondary delay filter, not limited to the primary delay filter. In addition, instead of or in addition to the incomplete integrator, a high-pass filter (not shown) can be provided before the integrators 94a and 94b (or the incomplete integrator). The cutoff frequency of the high pass filter can be set to 10 Hz or 5 Hz or less, for example.

Thus, when the phase difference detector 130 obtains the phase θ of the motor current Im with respect to the AC voltage command value V ^* using the arctangent of the ratio of the first-order Fourier coefficient of the drive frequency component, the phase difference detector 130 is obtained. Only the primary frequency component of the motor current Im, which is an AC signal input to the motor, has a large sensitivity. That is, for example, even when a DC offset or higher-order noise is superimposed on the motor current Im, the primary frequency component of the motor current Im that is an AC signal input to the phase difference detector 130 with respect to the reference phase θ ^* . The phase dltθ can be obtained more accurately. In addition, when the high-pass filter is provided as described above, it can be configured to be robust even at a frequency smaller than the drive frequency ω.
Therefore, as a method for detecting the motor current Im, particularly effective control is adopted when a system in which noise is easily superimposed, for example, a system in which the inductance position of the inductance is large or a system in which another device exists in the vicinity is adopted. realizable. In this way, it is possible to detect or estimate the resonance frequency with high accuracy and to realize highly efficient linear motor driving.

<Drive frequency adjuster 131>
FIG. 11 is an explanatory diagram illustrating a configuration example of the drive frequency adjuster 131 included in the control unit 102 illustrated in FIG. The drive frequency adjuster 131 obtains the difference between the phase difference command value dltθ ^* (for example, 0 °) and the phase difference dltθ ^ obtained by the phase difference detector 130 by the subtractor 91, and the multiplier 92b supplies the difference to the proportional gain Kp_adtr. Is multiplied by the integration gain Ki_adtr by the multiplier 92c, and the result of integration control by which the result is integrated by the integrator 94c is added by the adder 90, and a frequency command is further added to the addition result. The drive frequency command value ω ^* is output by adding the initial value (ω ₀ ).
The frequency command initial value (ω ₀ ) may be obtained from a host controller (not shown) or may be set in advance, for example, 0 °. Further, the drive frequency adjuster 131 of the present embodiment has a configuration of proportional integral control, but other control configurations such as proportional control and integral control can also be applied.

[Realization of high-efficiency drive]
The operation of the phase difference detector 130 and the drive frequency adjuster 131 when the linear motor 104 is driven at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant will be described.
For example, when the mass of the mover 6 is heavier than the design value, the actual resonance frequency is lower than the design value. That is, when the initial value of the drive frequency is determined using the mass design value of the mover 6 (when the initial value of the drive frequency command value ω ^* is determined using the design value), the actual resonance frequency is used. Drive at a high frequency. At this time, the phase difference dltθ ^ obtained by the phase difference detector 130 is larger than the phase difference command value dltθ ^* . Therefore, the drive frequency adjuster 131 executes control to decrease the drive frequency command value ω ^*, and as a result, the drive frequency command value ω ^* matches the actual resonance frequency. Therefore, the velocity energy of the mover 6 can be effectively used, and the linear motor 104 can be driven with high efficiency.

<Induced voltage component generator 135>
FIG. 12 is an explanatory diagram illustrating a configuration example of the induced voltage component generator 135 that configures the control unit 102 illustrated in FIG. 1. The induced voltage component generator 135 generates a voltage command value V _m1 ^* corresponding to the induced voltage generated according to the speed of the mover 6.
The upper diagram in FIG. 12 is a so-called open command type configuration example that does not constitute a stroke controller. The induced voltage component generator 135 integrates a stroke command value l ^* obtained from a higher-level controller (not shown) and the like and a drive frequency command value ω ^* output from the drive frequency adjuster 131 as a reference phase generator. The reference phase θ ^* obtained by integration by the device 140 is input.
The reference phase θ ^* is input to the cosine calculator 82b that outputs the cosine of the input value, and the cosine output, the stroke command value l ^*, and the reference phase θ ^* are multiplied by the multiplier 92d to obtain the speed command value vm. ^{Get *} . The speed command value vm ^* is multiplied by an induced voltage constant Ke ^* by a multiplier 92e to generate a voltage command value (V _m1 ^* ) corresponding to the induced voltage.

The speed command value vm ^* can also be obtained from the differentiation of the command value xm ^* of the position of the mover 6, and the arithmetic expression that is the basis of the configuration of FIG.

The speed command value vm ^* can be expressed as a sine wave having the stroke command value l ^* as an amplitude. When the sine wave is differentiated with respect to time, it becomes a cosine wave. Therefore, if the configuration shown in the upper diagram of FIG. 12 is used, the differentiation operation can be omitted, and mounting on the control unit 102 becomes easy.

  The lower diagram of FIG. 12 is a configuration example of a so-called close command type constituting the stroke controller. The induced voltage component generator 135a feeds back a stroke detection value or a stroke estimation value by a known position sensor, and the stroke controller 153 controls the stroke command value.

<Stroke controller 153>
FIG. 13 is an explanatory diagram showing a configuration example of the stroke controller 153 constituting the induced voltage component generator 135a shown in the lower diagram of FIG. As shown in FIG. 13, the stroke controller 153 obtains the difference between the stroke command value l ^* and the detected stroke value or the estimated stroke value by a subtractor 91a, and multiplies it by a proportional gain Kp_astr by a multiplier 92d. The controlled operation result is multiplied by the integral gain Ki_astr by the multiplier 92e, and the result of integration control by integrating the result by the integrator 94d is added by the adder 90a, and the adjusted stroke command value l ^** is output. To do. Although the stroke controller 153 of this embodiment has a configuration of proportional-integral control, other control configurations such as proportional control and integral control can be applied.

  In the case of estimating the position, for example, the position estimated value xm ^ is obtained by the following equation (7) using the voltage Vm applied to the linear motor 104 and the current Im flowing through the linear motor 104.

In Expression (7), Vm ^* is a voltage command value Vm ^* applied to the linear motor 104.

FIG. 14 is an explanatory diagram illustrating a configuration example of the position estimation unit, and is an explanatory diagram when Expression (7) is illustrated in a block diagram. As shown in FIG. 14, the position estimation unit 308 inputs the voltage command value Vm ^* and the motor current Im, and multiplies the motor current Im by the winding resistance value Rm ^* of the linear motor 104 to obtain the voltage command value Vm ^*. Further, the result obtained by calculating the reduced result by the integrator is obtained. Further, a result obtained by multiplying the motor current Im by the winding inductance value Lm ^* of the linear motor 104 is obtained and subtracted from the result calculated by the integrator, and this reduced result is divided by the induced voltage constant Ke ^* . As a result, the estimated position value xm ^ is output. In addition to the above, a known synchronous motor position estimation method can be applied to the position estimation unit 308. If a stroke is calculated based on the position estimated value xm ^ and input to the trolley detector 153 shown in FIG. 13, it can be controlled to a desired stroke.

<Load current detector 136>
FIG. 15 is an explanatory diagram showing a configuration example of the load current detector 136 constituting the control unit 102 shown in FIG. The load current detector 136 of the present embodiment extracts the amplitude of the load current component using a Fourier transform equation.
Similarly to the configuration of the phase difference detector 130 shown in FIG. 9, the load current detector 136 includes a sine calculator 81 that outputs the sine of the input value and a cosine calculator 82 that outputs the cosine of the input value. The reference phase θ ^* , which is a phase command value, is input to obtain the sine and cosine for the reference phase θ ^* . A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92. When the outputs are integrated by integrators 94e and 94f, respectively, first-order Fourier coefficients of sine and cosine are obtained. That is, since higher-order frequency components can be eliminated from the drive frequency ω of the Fourier expansion, it can be configured robustly against higher-order noise.

  The outputs of the integrators 94e and 94f are squared and input to the square root calculator 96. That is, the square root sum of squares of the sine component and the cosine component, which are the first order Fourier coefficients of the sine and cosine, is obtained, and the amplitude of the fundamental current is obtained. As the load increases, the amplitude of the fundamental wave current also increases, so that the load current Im_ld can be detected with the configuration of FIG.

FIG. 16 is an explanatory diagram illustrating another configuration example of the load current detector that configures the control unit 102 illustrated in FIG. 1. The load current detector 136b inputs the reference phase θ ^* that is the phase command value to the cosine calculator 82 that outputs the cosine of the input value, and obtains the cosine with respect to the reference phase θ ^* . By multiplying the obtained cosine and the motor current Im by the multiplier 92, a cos component (Im_cos) of the fundamental frequency of the motor current Im is extracted. Next, the cos component (Im_cos) of the fundamental frequency of the motor current Im is low-pass filtered (low-pass filter) processed by the first-order lag filter 141, and is output as the load current Im_ld.
In addition, although the structural example of the primary delay filter was shown in FIG. 16, you may comprise a low-pass filter not only with a primary delay filter but with other known structures, such as a secondary delay filter.

  As shown in FIGS. 7 and 8, when the position of the mover is a sine wave, the cosine component is dominant in the load current. Therefore, the configuration of FIG. 16 can detect the load current by filtering the extracted cosine component. In the case of the configuration of FIG. 16, it is effective for reducing the calculation load.

<Voltage drop component generator 137>
FIG. 17 is an explanatory diagram illustrating a configuration example of the voltage drop component generator 137 configuring the control unit 102 illustrated in FIG. 1. The voltage drop component generator 137 receives the load current Im_ld detected by the load current detector 136 and the reference phase θ ^*, and a voltage command value (V _m2) corresponding to the voltage drop due to the resistance and inductance of the linear motor 104. ^* ) Is output.

The load current Im_ld is multiplied by a preset value or estimated resistance value Rm ^* and inductance value Lm ^* by a multiplier 92. Next, the value obtained by multiplying the load current Im_ld and the resistance value Rm ^* is multiplied by the output of the cosine calculator 82 that outputs the cosine of the reference phase θ ^{* as} the input value, and the load current Im_ld and the inductance value Lm ^* are multiplied. The value multiplied at 92 is multiplied by the output of the sine calculator 81a that outputs the negative sine of the reference phase θ ^{* as} the input value. Further, these are added by an adder 90 and output as a voltage command value (V _m2 ^* ) corresponding to the voltage drop.

<Voltage command value generator 103>
FIG. 18 is an explanatory diagram illustrating a configuration example of the voltage command value generator 103 that configures the control unit 102 illustrated in FIG. 1. As shown in FIG. 18, the voltage command value generator 103 receives a voltage command value (V _m1 ^* ) corresponding to the induced voltage input from the induced voltage component generator 135 and the voltage drop component generator 137. The voltage command value (V _m2 ^* ) corresponding to the voltage drop is added by the adder 90 and output as an AC voltage command value (V _m ^* ). Since the two voltage command values (V _m1 ^* and V _m2 ^* ) to be input are both AC waveforms, the voltage command value generator 103 is equivalent to outputting a vector sum (vector addition).
The voltage command value generator 103 has a voltage phase calculator 138 and outputs an AC voltage command phase θ _Vm ^* which is a phase of the output AC voltage command value V _m ^* .

FIG. 19 is a vector diagram showing a vector sum in the voltage command value generator at the time of light load and heavy load, and is an explanatory diagram showing vector addition in the voltage command value generator 103 as a vector diagram. The left diagram of FIG. 19 is a vector diagram when the load is light, that is, when the load current is small, and the right diagram of FIG. 19 is a vector diagram when the load is heavy, that is, when the load current is large. Since the stroke command value is the same value in both the left diagram and the right diagram in FIG. 19, the voltage command value (V _m1 ^* ) corresponding to the induced voltage is the same vector. In the left and right diagrams of FIG. 19, the counterclockwise direction is positive.
In this way, the voltage command value (V _m2 ^* ) corresponding to the voltage drop due to the resistance and inductance is added to the voltage command value (V _m1 ^* ) corresponding to the induced voltage in accordance with the load current. the amplitude of the voltage command value Vm ^* is increased, minute AC voltage command phase (theta _{Vm *),} the phase advances, the AC voltage command value of the single-phase V _m ^* is output. That is, the voltage command value V _m ^* is appropriately controlled according to the load condition even when the stroke of the mover 6 is the same, in other words, even when the speed command value is the same.

The voltage Vm ^* applied to the linear motor 104 can be adjusted by changing any one of the stroke command value l ^* , the phase command value θ ^* , and the speed command value vm ^* . Therefore, by adjusting the amplitude and frequency of the applied voltage, it is possible to control the drive frequency to the resonance frequency and to control the stroke.

<Voltage phase calculator 138>
The voltage phase calculator 138 outputs the AC voltage command phase (θ _Vm ^* ) using, for example, an arctangent calculator. Although there are various definitions of the voltage phase, in this embodiment, as shown in FIG. 19, the vertical axis (FIG. 19) rotated 90 degrees counterclockwise from the mover position (horizontal axis in FIG. 19) changing in a sine wave shape. The vertical axis is the phase relative to the vertical axis.
That is, the AC voltage command phase (θ _Vm ^* ) is near zero in the no-load state, and is a positive value in the heavy load state.

[How to give the phase difference command value]
In describing the configuration of the drive frequency adjuster 131, the phase difference command value dltθ ^{* is set} to, for example, 0 ° for the sake of simplicity. In this embodiment, as shown in FIG. It is the output AC voltage command phase (.theta.V _m ^*) as a phase difference command value dltθ ^*, and inputs the driving frequency adjuster 131.
Even if the voltage command value corresponding to the voltage drop (V _m2 ^* ) is vector-added by the voltage command value generator 103, the phase difference command value dltθ ^* is fixed to 0 °. In this case, the difference between the phase difference command value dltθ ^* obtained by the subtractor 91 (FIG. 11) constituting the reduction drive frequency adjuster 131 and the phase difference dltθ ^ obtained by the phase difference detector 130 becomes a negative value, and the drive frequency adjustment The device 131 executes control for increasing the drive frequency command value ω *. As a result, convergence occurs at a frequency higher than the mechanical resonance frequency determined from the mass of the mover 6 and the spring constant.

On the other hand, as in this embodiment, when the negative value of the AC voltage command phase (θV _m ^* ) that is the output of the voltage command value generator 103 is input to the drive frequency adjuster 131 as the phase difference command value dltθ ^* , the vector As a result of the addition, the drive frequency adjuster 131 is not controlled to increase the drive frequency command value ω ^* in order to consider the advanced voltage phase, and is a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant. To converge.

As a result, the amplitude of the voltage command value V _m ^* increases in accordance with the load change (load current), and even if the phase advances by the AC voltage command phase (θV _m ^* ), the drive frequency is higher than the resonance frequency. Without being highly controlled, it is possible to drive with high efficiency even under a wide range of load conditions.
In FIG. 18, the output of the voltage phase calculator 138 is multiplied by −1 by the multiplier 92, but this is not necessary depending on the definition of the voltage phase.

[Followability to load]
FIG. 20 is an explanatory diagram showing temporal changes in the AC voltage command phase and the drive frequency command value. Changes in the AC voltage command phase (θV _m ^* ) and the drive frequency command value when the load changes with time. FIG. The AC voltage command phase (θV _m ^* ) in FIG. 20 is based on a vertical axis (vertical axis in FIG. 19) rotated 90 degrees counterclockwise from a mover position (horizontal axis in FIG. 19) that changes in a sine wave shape. Is shown as a phase.
As described above, the AC voltage command phase and the drive frequency command value are respectively θ1 → θ2 → θ3 as the load changes from L1 → L2 → L3 (L2>L1> L3) while the linear motor 104 is driven. (Θ2>θ1> θ3) and f1 → f2 → f3 (f2>f1> f3).

<PWM signal generator 134>
The PWM signal generator 134 constituting the control unit 102 shown in FIG. 1 uses a known pulse width modulation by comparing the triangular wave carrier signal and the voltage command value Vm ^*, and drives according to the voltage command value Vm ^*. A signal is generated, and the generated drive signal is output to the power conversion circuit 105.

<Power conversion circuit 105>
FIG. 21 is a diagram illustrating a configuration example of the power conversion circuit 105 included in the linear motor driving apparatus 101 illustrated in FIG. The full bridge circuit 126 switches the DC voltage source 120 according to the drive signal input by the control unit 102 and outputs a voltage to the linear motor 104. The full bridge circuit 126 includes four switching elements 122, a first upper and lower arm (hereinafter referred to as U phase) having switching elements 122a and 122b connected in series, and a second upper and lower arm having switching elements 122c and 122d. And an arm (hereinafter referred to as a V phase). The switching element 122 performs a switching operation according to the pulsed gate signals (124a to 124d) output from the gate driver circuit 123 based on the voltage command value Vm ^* generated by the control unit 102 and a drive signal by pulse width modulation. it can.
By controlling the conduction state (ON / OFF) of the switching element 122, a voltage corresponding to the AC voltage of the DC voltage source 120 can be output to the winding 8. Note that a DC current source may be used instead of the DC voltage source 120. As the switching element 122, for example, a semiconductor switching element such as IGBT or MOS-FET can be employed.

[Connection with linear motor 104]
The linear motor 104 is connected between the switching elements 122 a and 122 b of the first upper and lower arms and between the switching elements 122 c and 122 d of the second upper and lower arms of the power conversion circuit 105. Although FIG. 21 shows an example in which the windings 8 of the upper and lower armatures 9 are connected in parallel, the windings 8 can also be connected in series.

[Current detector 107]
For example, a current detector 107 such as a CT (current transformer) can be provided on the U-phase lower arm and the V-phase lower arm. Thereby, the current Im flowing through the winding 8 of the linear motor 104 can be detected.
As the current detector 107, for example, instead of CT, a phase shunt current method is adopted in which a shunt resistor 125 is added to the lower arm of the power conversion circuit 105 and a current flowing through the linear motor 104 is detected from a current flowing through the shunt resistor 125. it can. In place of or in addition to the current detector 107, a single shunt current detection method for detecting the current on the AC side of the power conversion circuit 105 from the DC current flowing in the shunt resistor 125 added to the DC side of the power conversion circuit 105. It may be adopted. The single shunt current detection method utilizes the fact that the current flowing through the shunt resistor 125 changes with time depending on the energization state of the switching element 122 constituting the power conversion circuit 105.

  As described above, according to this embodiment, a linear motor can be driven with high efficiency at a mechanical resonance frequency including a load without requiring a sensor for detecting an induced voltage that changes in accordance with a change in load. A linear motor system can be provided. Specifically, by controlling the voltage amplitude and voltage phase according to the load and adjusting the drive frequency in consideration of the voltage phase, the linear motor can be driven at a mechanical resonance frequency including the load, resulting in high efficiency. A simple linear motor system can be configured.

  The configuration of the present embodiment can be the same as that of the first embodiment except for the following points. The present embodiment relates to a hermetic compressor 50 as an example of a device on which a linear motor system 200 described later is mounted.

<Sealed compressor 50>
FIG. 22 is a longitudinal sectional view of a hermetic compressor according to a second embodiment of the present invention, and is an example of a longitudinal sectional view of a hermetic compressor 50 having a linear motor 104. The hermetic compressor 50 is a reciprocating compressor in which the compression element 20 and the electric element 30 are arranged in the hermetic container 3. The compression element 20 and the electric element 30 are elastically supported in the sealed container 3 by a support spring 49. The electric element 30 includes a mover 6 and an armature 9.
The compression element 20 includes a cylinder block 1 that forms a cylinder 1a, a cylinder head 16 that is assembled to an end face of the cylinder block 1, and a head cover 17 that forms a discharge chamber space. The working fluid supplied into the cylinder 1a is compressed by the reciprocating motion of the piston 4, and the compressed working fluid is sent to a discharge pipe (not shown) communicating with the outside of the compressor. For example, air or a refrigerant of a refrigeration cycle can be adopted as the working fluid.

A piston 4 is attached to one end of the mover 6. In this embodiment, the movable body 6 and the piston 4 reciprocate to compress and expand the working fluid. This corresponds to a load in which work required for compression and expansion fluctuates. A compression element 20 is disposed at one end of the electric element 30. The cylinder block 1 has a guide rod that guides the reciprocating motion of the mover 6 along the front-rear direction.
When a resonance spring 23 (not shown in FIG. 22) is added to the mover 6 and the mover 6 is reciprocated at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant, resonance by the compression element 20 is performed. It is necessary to consider the influence on the frequency. That is, since the spring action of the working fluid is applied by the suction pressure of the compression element 20 and the pressure of the discharge space, the frequency at which the resonance state is reached changes. That is, when the pressure of the cylinder 1a is high, it is equivalent to a high spring constant of the resonance spring 23 added to the mover 6, and the resonance frequency becomes high. On the contrary, when the pressure of the cylinder 1a is low, the spring constant of the resonance spring 23 added to the mover 6 becomes dominant, and the resonance frequency is a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant. Close to.

  Thus, when the linear motor 104 is used as the power of the compression element 20, the resonance frequency changes depending on the conditions of the compression element 20 (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency in accordance with changes in the load and the resonance frequency.

  The influence of the change in the resonance frequency can be seen from the phase relationship between the voltage Vm applied to the winding 8, the motor current Im, and the speed of the mover 6. Therefore, a highly efficient linear motor system can be configured by controlling based on these phase relationships.

FIG. 23 is an overall schematic configuration diagram of the linear motor system 200 of the present embodiment. As shown in FIG. 23, the linear motor system 200 has the same configuration as that of the linear motor system 100 in Example 1 described above, but a load current detector 136a and a voltage drop component creator 137a that creates a voltage drop component. And the voltage command value generator 103a that outputs the voltage command value Vm ^* is different. Hereinafter, the load current detector 136a, the voltage drop component generator 137a, and the voltage command value generator 103a will be described. Since other configurations are the same as those of the linear motor system 100 in the first embodiment, the description thereof is omitted.

<Load current detector 136a>
FIG. 24 is an explanatory diagram showing a configuration example of the load current detector 136a that constitutes the control unit 202 of the linear motor system 200 shown in FIG. As shown in FIG. 24, the load current detector 150a receives the motor current Im from the current detector 107 and the reference phase θ ^* that is a phase command value, and the cosine component (Im_cos) of the fundamental frequency of the motor current Im. ) And sin component (Im_sin) are extracted and output.
A sine calculator 81 which outputs the sine of the input value, each of the cosine calculator 82 which outputs the cosine of the input value, receives a reference phase theta ^* a phase command value, the sine and cosine with respect to the reference phase theta ^* obtain. A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92. The output is subjected to low-pass filter (low-pass filter) processing by a first-order lag filter 141 to obtain first-order Fourier coefficients of sine and cosine, and load current Im_sin as a sin component (Im_sin) of the fundamental frequency of motor current Im, The load current Im_cos is output to the voltage drop component generator 137a as the cos component (Im_cos) of the fundamental frequency of the motor current Im. That is, since higher-order frequency components can be eliminated from the drive frequency ω of the Fourier expansion, it can be configured robustly against higher-order noise. In the case of the configuration shown in FIG. 24, calculation of a vector sum, which will be described later, can be facilitated, which is desirable for mounting on the control unit 202.

<Voltage drop component generator 137a>
FIG. 25 is an explanatory diagram showing a configuration example of a voltage drop component generator 137a that constitutes the control unit 202 of the linear motor system 200 shown in FIG. As shown in FIG. 25, the voltage drop component generator 137a receives the load current Im_sin and the load current Im_cos detected by the load current detector 136a, and a voltage command corresponding to the voltage drop due to the resistance and inductance of the linear motor 104. A value (V _m2 ^* Im_sin) and a voltage command value (V _m2 ^* Im_cos) corresponding to the voltage drop are output.

The voltage drop component generator 137a multiplies the load current Im_cos by the preset or estimated resistance value Rm ^* of the linear motor 104 by the multiplier 92, and uses the voltage command value generator 103a as a voltage command value (V _m2 ^* _cos). Output to. Further, the voltage drop component generator 137a multiplies the load current Im_sin by the inductance value Lm ^* of the linear motor 104 set or estimated in advance by the multiplier 92, and generates a voltage command value as a voltage command value (V _m2 ^* _sin). To the device 103a.

<Voltage command value generator 103a>
FIG. 26 is an explanatory diagram showing a configuration example of the voltage command value generator 103a that constitutes the control unit 202 of the linear motor system 200 shown in FIG. As shown in FIG. 26, the voltage command value generator 103a includes a voltage command value V _m1 ^* corresponding to the induced voltage generated according to the speed of the mover 6 output from the induced voltage component generator 135, and a voltage drop component. The voltage command value (V _m2 ^* _cos) output from the generator 137a is added by the adder 90 and output. Thus, by adding the cos component and the sin component separately, it is equivalent to outputting a vector sum (vector addition) in the voltage command value generator 103a.

Although there are various definitions of the voltage phase, in this embodiment, as shown in FIG. 19 in the above-described first embodiment, 90 degrees counterclockwise from the mover position (horizontal axis in FIG. 19) changing in a sine wave shape. The phase is based on the rotated vertical axis (vertical axis in FIG. 19). Therefore, the output value of the induced voltage component generator 135 (voltage command value V _m1 ^* corresponding to the induced voltage generated according to the speed of the mover 6) and the voltage command value (V) that is the output of the voltage drop component generator 137a. _m2 ^* _cos) is added by the adder 90.

The voltage command value generator 103 a includes a voltage phase calculator 138 and a square root calculator 96. The square root calculator 96 obtains the addition result of the adder 90 and the square root of the voltage command value (V _m2 ^* _sin) output from the voltage drop component generator 137a, and generates the AC voltage command value (Vm ^* ) as a PWM signal. Output to the device 134. Further, the voltage phase calculator 138 receives the addition result from the adder 90 and the voltage command value (V _m2 ^* _sin) output from the voltage drop component generator 137a, and uses, for example, an alternating current using an arctangent calculator. The voltage command phase (θV _m ^* ) is output to the drive frequency adjuster 131.

  As described above, according to the present embodiment, in the linear motor 104 of the hermetic compressor 50, the voltage amplitude and the voltage phase are controlled according to the load, and the drive frequency is adjusted in consideration of the voltage phase. The linear motor can be driven at a mechanical resonance frequency including the load, and a highly efficient linear motor system can be configured.

The configuration of this embodiment can be the same as that of Embodiment 1 or 2 except for the following points. The present embodiment relates to an air suspension system 300 as an example of a device equipped with a linear motor system.
27 is a circuit diagram of an air suspension system 300 according to a third embodiment of the present invention, and FIG. 28 is a schematic diagram of a vehicle equipped with the air suspension system 300 shown in FIG. However, in FIG. 28, only a distribution point 309N to be described later and components on the air suspensions 301 and 302 side are illustrated.

  As shown in FIG. 27, the air suspension system 300 includes two air suspensions 301 and 302, a compressor 303 using a linear motor 104 as a drive source, an intake filter 304, a first tank 305, an air dryer 307, and valves. There are three check valves 308, 315, 317, a supply / discharge switching valve 310, two suspension control valves 311, 312, a return passage opening / closing valve 314, and an exhaust passage opening / closing valve 319. The air suspension system 300 connects these through a passage through which air can flow.

  As shown in FIG. 28, the air suspension system 300 is mounted on a vehicle 400, for example, and controls air pressure in the air chambers 301C and 302C (FIG. 27) of the air suspensions 301 and 302. For example, the left wheel 410L and the right wheel 410R of the vehicle 400 are provided with an axle 420 that connects these hubs and the like. For example, air suspensions 301 and 302 are provided between the wheel 410 side and the vehicle body 430 side, such as between the left wheel 410L and the right wheel 410R and the vehicle body 430, or between the hub and the vehicle body 430, and the air chambers 301C, The vehicle height can be adjusted by controlling the air pressure in 302C.

  As shown in FIG. 28, the air suspensions 301 and 302 may be attached between the axle 420 on the wheel 410 side and the vehicle body 430 of the vehicle 400, and the suspension arm that connects the wheel 410 and the vehicle body 430. It may be mounted between the vehicle (the wheel 410 side) and the vehicle body 430 or between the hub of the wheel 410 (the wheel 410 side) and the vicinity of the vehicle body 430 mounting portion (the vehicle body 430 side) of the upper arm of the suspension. As described above, the air suspensions 301 and 302 may be provided so as to support the wheel 410 and the vehicle body 430. For example, the air suspensions 301 and 302 can be provided between the wheel 410 and the vehicle body 430 in the vertical direction. It is not restricted to the aspect attached to 430.

  In this embodiment, an air suspension system 300 having two air suspensions will be described. However, the number of air suspensions included in the air suspension system 300 is not particularly limited as long as it is one or more. The number of air suspensions can be made equal to the number of wheels, for example. For example, in the case of a four-wheeled vehicle, a total of four air suspensions can be provided, two on two front wheels and two on two rear wheels. In the present embodiment, an example in which the cushioning cylinders 301A and 302A and the air chambers 301C and 302C serving as air springs are integrated is shown. However, as is known on the large vehicle or rear suspension side, the cushioning cylinders are known. (Hydraulic shock absorbers) 301A, 302A and air springs may be provided independently.

  As shown in FIG. 27, in the air suspensions 301 and 302, air chambers 301C and 302C are formed between the buffer cylinders 301A and 302A and the piston rods 301B and 302B, respectively. ing. Each of the air chambers 301 </ b> C and 302 </ b> C is connected to a passage described later, and the pressure and the vehicle height are controlled by the operation of the air suspension system 300.

  The compressor 303 can compress the air sucked from the suction port 303C and discharge the compressed air from the discharge port 303D. The compressor 303 includes a compressor main body 303A and a linear motor 104. A pressure sensor for measuring the pressure at the suction port 303C or the discharge port 303D or both ports is provided.

  When the resonance spring 23 is added to the mover 6 of the linear motor 104 and the mover 6 is reciprocated at a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant, it is shown in FIG. As described above, it is necessary to consider the influence of the compression element 20 on the resonance frequency. That is, since the spring action of the working fluid is applied by the suction pressure of the compression element 20 and the pressure of the discharge space, the frequency at which the resonance state is reached changes. That is, when the pressure of the cylinder 1a is high, this is equivalent to a high spring constant of the resonance spring 23 added to the mover 6, and the resonance frequency becomes high. On the contrary, when the pressure of the cylinder 1a is low, the spring constant of the resonance spring 23 added to the mover 6 becomes dominant, and the resonance frequency is a mechanical resonance frequency determined from the mass of the mover 6 and the spring constant. Close to.

  Thus, when the linear motor 104 is used as the power of the compression element, the resonance frequency changes depending on the conditions of the compression element (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency in accordance with changes in the load and the resonance frequency.

<Load current detector 136b>
FIG. 29 is a diagram illustrating a configuration example of the load current detector 136b. The load current detector 150b receives the motor current Im, and extracts and outputs a cos component (Im_cos) and a sin component (Im_sin) of the fundamental frequency of the motor current Im.

A sine calculator 81 which outputs the sine of the input value, each of the cosine calculator 82 which outputs the cosine of the input value, receives a reference phase theta ^* a phase command value, the sine and cosine with respect to the reference phase theta ^* Get. A value obtained by multiplying each of the sine and cosine by the motor current Im is output from the multiplier 92. The output is subjected to a low-pass filter (low-pass filter) process by a first-order lag filter 141a to obtain first-order Fourier coefficients for the sine and cosine. That is, since higher-order frequency components can be eliminated from the drive frequency ω of the Fourier expansion, it can be configured robustly against higher-order noise. The filter time constant (T_ld) or the cut-off frequency of the first-order lag filter 141a can be changed from the outside.

  In the configuration of the load current detector 136b shown in FIG. 29, the filter time constant (T_ld) or the cutoff frequency of the first-order lag filter 141a can be changed according to the pressure of the suction port 303C or the discharge port 303D of the compressor 303. ing. Thereby, the influence on the load current detection can be reduced even by a change in pressure.

  As described above, according to the present embodiment, in the air suspension system 300, the voltage amplitude and the voltage phase are controlled according to the load, and the drive frequency is adjusted in consideration of the voltage phase. The linear motor can be driven at a specific resonance frequency, and a highly efficient linear motor system can be configured.

The present invention is not limited to the above-described embodiments, and includes various modifications. For example, the above-described embodiments have been described in detail for easy understanding of the present invention, and are not necessarily limited to those having all the configurations described.
Each of the above-described configurations, functions, processing units, processing procedures, and the like may be realized by hardware by designing a part or all of them with, for example, an integrated circuit. Each of the above-described configurations, functions, and the like may be realized by software by interpreting and executing a program that realizes each function by the processor.

In the first to third embodiments, the linear motor 104 has been described as a single-phase machine, but the configuration of the present invention can be applied to a three-phase machine, and similar effects can be obtained.
The power conversion circuit 105 may be configured to output current. In this case, a current command value generator may be provided instead of the voltage command value generator 103.

DESCRIPTION OF SYMBOLS 1 ... Cylinder block 1a ... Cylinder 2 ... Permanent magnet 3 ... Sealed container 4 ... Piston 6 ... Movable element 7 ... Magnetic pole 8 ... Winding 9 ... Armature 16 ... Cylinder head 17 ... Head cover 20 ... Compression element 23 ... Resonant spring (assist Spring)
DESCRIPTION OF SYMBOLS 30 ... Electric element 50 ... Hermetic compressor 100, 200 ... Linear motor system 101, 201 ... Linear motor drive device 102, 202 ... Control part 103 ... Voltage command value generator 104 ... Linear motor 105 ... Power conversion circuit 107 ... Current Detector 126 ... Full bridge circuit 130 ... Phase difference detector 131 ... Drive frequency adjuster 134 ... PWM signal generator 135 ... Induced voltage component generator 136 ... Load current detector 137 ... Voltage drop component generator 140 ... Integrator

Claims (11)

A linear motor having at least a winding to which an AC voltage is applied and a mover to which an elastic body is connected;
A linear motor system in which the resonance frequency of the mover varies depending on the load of the elastic body and the linear motor,
A linear motor system comprising: a linear motor driving device that adjusts a phase difference between the AC voltage and a current flowing through the winding so that a driving frequency of the AC voltage becomes a resonance frequency. A linear motor having at least a winding to which an AC voltage is applied and a mover to which an elastic body is connected;
A linear motor system in which the resonance frequency of the mover varies depending on the load of the elastic body and the linear motor,
A linear motor system comprising a linear motor driving device that increases a phase difference between the AC voltage and a current flowing through the winding in accordance with an increase in the load. A linear motor having at least a winding to which an AC voltage is applied and a mover to which an elastic body is connected;
A linear motor system in which the resonance frequency of the mover varies depending on the load of the elastic body and the linear motor,
A linear motor driving device that detects a fundamental wave amplitude of a current flowing through the winding and increases a phase difference between the AC voltage and a current flowing through the winding according to an increase in the fundamental wave amplitude. Linear motor system. The linear motor system according to claim 3,
The linear motor driving device is:
A voltage drop component generator that outputs a first voltage command value corresponding to a voltage drop of the resistance and inductance of the winding using the fundamental wave amplitude;
An induced voltage component generator that outputs a second voltage command value corresponding to the induced voltage generated according to the speed of the mover;
A voltage command value generator for adding a vector of the first voltage command value and the second voltage command value and outputting a third voltage command value;
A linear motor system, wherein a voltage corresponding to the third voltage command value is applied to the winding. The linear motor system according to claim 3,
The linear motor driving device is:
A fundamental wave cosine component that is a component orthogonal to the mover position of the fundamental wave current flowing through the winding is detected, and the magnitude of the fundamental wave cosine component is used to correspond to the voltage drop of the resistance and inductance of the winding. A voltage drop component generator that outputs a first voltage command value
An induced voltage component generator that outputs a second voltage command value corresponding to the induced voltage generated according to the speed of the mover;
A voltage command value generator for adding a vector of the first voltage command value and the second voltage command value and outputting a third voltage command value;
A linear motor system, wherein a voltage corresponding to the third voltage command value is applied to the winding. In the linear motor system according to claim 4 or 5,
The linear motor driving device is:
A load current detector that outputs a load current that is a value after low-pass filter processing of the fundamental wave amplitude or the magnitude of the fundamental cosine component;
The linear motor system, wherein the voltage drop component generator outputs the first voltage command value based on the load current. The linear motor system according to claim 3,
The linear motor driving device is:
A voltage drop component generator for outputting a sine component of the first voltage command value and a cosine component of the first voltage command value corresponding to the voltage drop of the resistance and inductance of the winding using the fundamental wave amplitude; ,
An induced voltage component generator that outputs a second voltage command value corresponding to the induced voltage generated according to the speed of the mover;
The cosine component of the first voltage command value and the second voltage command value are vector-added to output a third voltage command value, and the voltage command phase is output based on the sine component of the first voltage command value. A voltage command value generator,
A linear motor system, wherein a voltage corresponding to the third voltage command value and the voltage command phase is applied to the winding. The linear motor system according to claim 7,
The linear motor driving device is:
A load current detector that outputs a sine component of a load current that is a value after low-pass filter processing of the magnitude of the fundamental wave amplitude or the fundamental wave cosine component and a cosine component of the load current;
The voltage drop component generator outputs a sine component of the first voltage command value and a cosine component of the first voltage command value based on a sine component of the load current and a cosine component of the load current. Linear motor system. The linear motor system according to claim 8,
A linear motor system, wherein a piston is connected to the mover, and the piston is a compressor that compresses fluid or air as a load. The linear motor system according to claim 9, wherein
A linear motor system, wherein a cutoff frequency of the low-pass filter process is changed according to a suction or discharge pressure of the compressor. In the linear motor system according to any one of claims 4, 5, 7, and 8,
In a plurality of air chambers that are interposed between the vehicle body side and the wheel side and adjust the vehicle height according to the supply and discharge of air,
A linear motor system, wherein a piston is connected to the movable element, and the piston is an air suspension system that supplies air compressed by a compressor that compresses fluid or air as a load.

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