JP2017216814A - Non-contact power supply device, program, control method for non-contact power supply device, and non-contact power transmission system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To supply stable power regardless of a relative positional relation between a primary side coil and a secondary side coil.SOLUTION: A frequency control unit 233 adjusts the magnitude of output power by performing control so that frequencies of first drive signals G1 to G4 are varied discretely. A phase difference control unit 234 adjusts the magnitude of output power by performing control so that phase differences, phase delays of second drive signals G5 to G8 with respect to the first drive signals G1 to G4, are varied discretely. A determination unit 235 determines a pitch width of a control target value composed of at least one of a frequency controlled by the frequency control unit 233 and a phase difference controlled by the phase difference control unit 234, depending on a resonant characteristic estimated by an estimation unit 232.SELECTED DRAWING: Figure 1

Description

  The present invention generally relates to a non-contact power supply apparatus, a program, a control method for the non-contact power supply apparatus, and a non-contact power transmission system, and more particularly, to a non-contact power supply apparatus, a program, and a non-contact power supply that perform non-contact power supply to a load. The present invention relates to a device control method and a non-contact power transmission system.

  2. Description of the Related Art Conventionally, a non-contact power feeding apparatus that supplies power to a load in a non-contact manner using electromagnetic induction has been proposed (see, for example, Patent Document 1).

  The non-contact power supply device described in Patent Literature 1 includes a primary coil (power supply coil) that supplies electric power by generating a magnetic field, and is used for power supply to a moving body such as an electric vehicle. The electric vehicle includes a non-contact power receiving device. The non-contact power receiving device includes a secondary side coil (power receiving coil) and a storage battery, and accumulates the power supplied from the primary side coil to the secondary side coil in the non-contact power feeding device in the storage battery.

JP2013-243929A

  By the way, in the non-contact power feeding device as described above, the primary side coil and the secondary side coil are determined by the relative positional relationship between the primary side coil of the non-contact power feeding device and the secondary side coil of the load (moving body). The coupling coefficient between changes. For this reason, there is a possibility that the output power output from the non-contact power supply apparatus varies depending on the relative positional relationship between the primary side coil and the secondary side coil, and stable power supply cannot be performed.

  The present invention has been made in view of the above reasons, and a non-contact power supply device, a program, and a non-contact power supply device capable of supplying stable power regardless of the relative positional relationship between the primary side coil and the secondary side coil It is an object of the present invention to provide a control method and a non-contact power transmission system.

  A contactless power supply device according to one aspect of the present invention includes an inverter circuit, a primary coil, a power correction circuit, an element control unit, an estimation unit, a frequency control unit, a phase difference control unit, and a determination unit. . The inverter circuit includes a plurality of conversion switch elements that are electrically connected between a pair of input points and a pair of output points. The inverter circuit converts a DC voltage applied to the pair of input points into an AC voltage and outputs the AC voltage from the pair of output points by switching of the plurality of conversion switch elements. The primary side coil is electrically connected between the pair of output points, and supplies the output power in a non-contact manner to the secondary side coil when the AC voltage is applied. The power correction circuit is electrically connected between at least one of the pair of output points and the primary coil, and includes a correction capacitor and a plurality of correction switch elements. The power correction circuit charges and discharges the correction capacitor by switching the plurality of correction switch elements. The element control unit controls the plurality of conversion switch elements with a first drive signal, and controls the plurality of correction switch elements with a second drive signal. The estimation unit estimates a resonance characteristic representing a relationship between the frequency of the first drive signal and the magnitude of the output power. The frequency control unit adjusts the magnitude of the output power by controlling the frequency of the first drive signal so as to change discretely. The phase difference control unit adjusts the magnitude of the output power by controlling so as to discretely change a phase difference that is a phase delay of the second drive signal with respect to the first drive signal. The determining unit is configured to estimate the step size of the control target value including at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit. Determined according to the resonance characteristics.

  A program according to one embodiment of the present invention causes a computer used for a non-contact power supply device to function as an element control unit, an estimation unit, a frequency control unit, a phase difference control unit, and a determination unit. The non-contact power feeding device includes an inverter circuit, a primary side coil, and a power correction circuit. The inverter circuit includes a plurality of conversion switch elements that are electrically connected between a pair of input points and a pair of output points. The inverter circuit converts a DC voltage applied to the pair of input points into an AC voltage and outputs the AC voltage from the pair of output points by switching of the plurality of conversion switch elements. The primary side coil is electrically connected between the pair of output points, and supplies the output power in a non-contact manner to the secondary side coil when the AC voltage is applied. The power correction circuit is electrically connected between at least one of the pair of output points and the primary coil. The power correction circuit includes a correction capacitor and a plurality of correction switch elements. The power correction circuit charges and discharges the correction capacitor by switching the plurality of correction switch elements. The element control unit controls the plurality of conversion switch elements with a first drive signal, and controls the plurality of correction switch elements with a second drive signal. The estimation unit estimates a resonance characteristic representing a relationship between the frequency of the first drive signal and the magnitude of the output power. The frequency control unit adjusts the magnitude of the output power by controlling the frequency of the first drive signal so as to change discretely. The phase difference control unit adjusts the magnitude of the output power by controlling so as to discretely change a phase difference that is a phase delay of the second drive signal with respect to the first drive signal. The determining unit is configured to estimate the step size of the control target value including at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit. Determined according to the resonance characteristics.

  A method for controlling a non-contact power feeding device according to one embodiment of the present invention includes an element control step, an estimation step, a frequency control step, a phase difference control step, and a determination step. The non-contact power feeding device includes an inverter circuit, a primary side coil, and a power correction circuit. The inverter circuit includes a plurality of conversion switch elements that are electrically connected between a pair of input points and a pair of output points. The inverter circuit converts a DC voltage applied to the pair of input points into an AC voltage and outputs the AC voltage from the pair of output points by switching of the plurality of conversion switch elements. The primary side coil is electrically connected between the pair of output points, and supplies the output power in a non-contact manner to the secondary side coil when the AC voltage is applied. The power correction circuit is electrically connected between at least one of the pair of output points and the primary coil. The power correction circuit includes a correction capacitor and a plurality of correction switch elements. The power correction circuit charges and discharges the correction capacitor by switching the plurality of correction switch elements. In the element control step, the plurality of conversion switch elements are controlled by a first drive signal, and the plurality of correction switch elements are controlled by a second drive signal. In the estimation step, a resonance characteristic representing a relationship between the frequency of the first drive signal and the magnitude of the output power is estimated. In the frequency control step, the magnitude of the output power is adjusted by controlling the frequency of the first drive signal so as to change discretely. In the phase difference control step, the magnitude of the output power is adjusted by controlling the phase difference, which is the phase delay of the second drive signal with respect to the first drive signal, to be discretely changed. In the determining step, the step size of the control target value including at least one of the frequency controlled in the frequency control step and the phase difference controlled in the phase difference control step is estimated in the estimation step. Determined according to the resonance characteristics.

  A non-contact power transmission system according to an aspect of the present invention includes the above-described non-contact power feeding device and a non-contact power receiving device having the secondary side coil. The contactless power receiving device is configured such that the output power is supplied in a contactless manner from the contactless power feeding device.

  The present invention has an advantage that stable power supply is possible regardless of the relative positional relationship between the primary side coil and the secondary side coil.

FIG. 1 is a circuit diagram showing a non-contact power transmission system according to an embodiment of the present invention. FIG. 2 is a waveform diagram showing a first drive signal and a second drive signal of the non-contact power feeding device according to the embodiment of the present invention. FIG. 3 is a graph showing an example of resonance characteristics in the above-described non-contact power feeding apparatus. 4A and 4B are graphs showing examples of resonance characteristics in the above-described contactless power supply device. FIG. 5A is a graph showing an example of the phase difference characteristic in the case of initial phase advancement in the non-contact power feeding apparatus same as above, and FIG. 5B is a graph showing an example of the phase difference characteristic in the case of initial delay phase in the non-contact power feeding apparatus same as above. is there. FIG. 6A is an explanatory diagram showing a first charging mode of the power correction circuit in the above-described non-contact power feeding apparatus. FIG. 6B is an explanatory diagram showing a first discharge mode of the power correction circuit in the above-described contactless power supply device. FIG. 6C is an explanatory diagram illustrating a second charging mode of the power correction circuit in the above-described contactless power supply device. FIG. 6D is an explanatory diagram illustrating a second discharge mode of the power correction circuit in the non-contact power supply apparatus according to the embodiment. FIG. 7 is a waveform diagram of the first drive signal, the primary side current, and the second drive signal when the voltage-current phase difference in the non-contact power feeding apparatus is 90 degrees. FIG. 8 is a waveform diagram of the first drive signal, the primary side current, and the second drive signal when the voltage-current phase difference is 45 degrees in the non-contact power feeding apparatus same as above. FIG. 9 is a flowchart showing output power control in the above-described contactless power supply apparatus. FIG. 10 is a graph showing the frequency characteristics of output power at the time of frequency control in the above non-contact power feeding apparatus. FIG. 11A is a graph showing an example of phase difference-coil current characteristics in the above non-contact power feeding apparatus. FIG. 11B is an enlarged graph of the area A1 in FIG. 11A. FIG. 12 is a flowchart showing the operation of the search mode in the above non-contact power feeding apparatus. FIG. 13 is a circuit diagram showing a configuration of a power correction circuit according to a modification of the embodiment of the present invention.

  The contactless power supply device according to the present embodiment supplies power to a load in a contactless manner. The non-contact power feeding device is configured such that the primary side coil provided in the non-contact power feeding device and the secondary side coil provided in the load are electromagnetically coupled (at least one of electric field coupling and magnetic field coupling). Power is supplied to the load by transmitting power from the coil to the secondary coil. This type of non-contact power feeding apparatus constitutes a non-contact power transmission system together with a non-contact power receiving apparatus provided in a load.

(1) Overview of contactless power transmission system First, an overview of a contactless power transmission system will be described with reference to FIG.

  The non-contact power transmission system 1 includes a non-contact power feeding device 2 having a primary side coil L1 and a non-contact power receiving device 3 having a secondary side coil L2. The non-contact power receiving device 3 is configured so that output power is supplied from the non-contact power feeding device 2 in a non-contact manner. The “output power” here is power output from the non-contact power feeding device 2 and is contactless from the primary coil L1 to the secondary coil L2 by applying an AC voltage to the primary coil L1. This is the power supplied.

  In the present embodiment, a case where the non-contact power receiving device 3 is mounted on an electric vehicle as a load will be described as an example. The electric vehicle includes a storage battery 4 and travels using electric energy stored in the storage battery 4. The non-contact power receiving device 3 mounted on the electric vehicle is used as a charging device for the storage battery 4. Here, an electric vehicle that travels using a driving force generated by an electric motor will be described as an example of an electric vehicle. However, the electric vehicle is not limited to an electric vehicle, and may be a two-wheeled vehicle (electric motorcycle), an electric bicycle, or the like.

  The non-contact power feeding device 2 charges the storage battery 4 of the electric vehicle by supplying the non-contact power receiving device 3 with power supplied from a commercial power source (system power source) or a power generation facility such as a solar power generation facility. The power supplied to the non-contact power supply device 2 may be either AC power or DC power, but in the present embodiment, the non-contact power supply device 2 is electrically connected to the DC power source 5 and contactless. A case where DC power is supplied to the power supply device 2 will be described as an example. When AC power is supplied to the non-contact power supply device 2, the non-contact power supply device 2 is provided with an AC / DC converter that converts alternating current into direct current.

  The non-contact power supply device 2 is installed in a parking lot such as a commercial facility, a public facility, or a housing complex. The non-contact power supply device 2 has at least a primary side coil L1 installed on the floor or the ground, and supplies power in a non-contact manner to the non-contact power reception device 3 of the electric vehicle parked on the primary side coil L1. At this time, the secondary coil L2 of the non-contact power receiving device 3 is electromagnetically coupled to the primary coil L1 by being positioned above the primary coil L1. Therefore, the output power from the primary side coil L1 is transmitted (power transmission) to the secondary side coil L2. The primary coil L1 is not limited to a configuration that is installed so as to be exposed from the floor or the ground, but may be installed so as to be embedded in the floor or the ground.

  The non-contact power receiving device 3 includes a secondary coil L2, a pair of secondary capacitors C21 and C22, a rectifier circuit 31, and a smoothing capacitor C2. The rectifier circuit 31 includes a diode bridge having a pair of AC input points and a pair of DC output points. One end of the secondary side coil L2 is electrically connected to one AC input point of the rectifier circuit 31 via the first secondary side capacitor C21, and the other end of the secondary side coil L2 is connected to the second secondary coil L2. It is electrically connected to the other AC input point of the rectifier circuit 31 via the secondary capacitor C22. The smoothing capacitor C2 is electrically connected between the pair of DC output points of the rectifier circuit 31. Furthermore, both ends of the smoothing capacitor C2 are electrically connected to a pair of output terminals T21 and T22. The storage battery 4 is electrically connected to the pair of output terminals T21 and T22.

  Thereby, the non-contact power receiving device 3 rectifies the AC voltage generated between both ends of the secondary coil L2 by receiving the output power from the primary coil L1 of the non-contact power feeding device 2 at the secondary coil L2. The DC voltage is obtained by rectifying by the circuit 31 and further smoothing by the smoothing capacitor C2. The non-contact power receiving device 3 outputs (applies) the DC voltage thus obtained to the storage battery 4 from the pair of output terminals T21 and T22.

  In the present embodiment, the non-contact power feeding device 2 includes a power correction circuit 22 that forms a resonance circuit (hereinafter referred to as “primary resonance circuit”) together with the primary coil L1, and a pair of primary capacitors C11 and C12. ing. That is, the primary side resonance circuit includes a primary side coil L1, a power correction circuit 22, and a pair of primary side capacitors C11 and C12. Further, in the non-contact power receiving device 3, the secondary coil L2 forms a resonance circuit (hereinafter referred to as “secondary resonance circuit”) together with the pair of secondary capacitors C21 and C22. That is, the secondary side resonance circuit includes a secondary side coil L2 and a pair of secondary side capacitors C21 and C22. The non-contact power transmission system 1 according to the present embodiment employs a magnetic field resonance method (magnetic resonance method) in which power is transmitted by resonating a primary side resonance circuit and a secondary side resonance circuit. That is, in the non-contact power transmission system 1, even when the primary side coil L1 and the secondary side coil L2 are relatively separated by matching the resonance frequencies of the primary side resonance circuit and the secondary side resonance circuit, The output power of the contact power supply device 2 can be transmitted with high efficiency.

  The primary side coil L1 and the secondary side coil L2 in the present embodiment may be solenoid type coils in which a conducting wire is spirally wound around the core, but the conducting wire is wound spirally on a plane. A spiral coil is preferable. Spiral type coils (circular coils) have the advantage that unwanted radiation noise is less likely to occur than solenoid type coils. Further, the use of the spiral type coil has an advantage that the range of operating frequencies usable in the inverter circuit is expanded as a result of reducing unnecessary radiation noise.

(2) Overview of Non-Contact Power Supply Device Next, an overview of the non-contact power supply device will be described with reference to FIG.

  The contactless power supply device 2 according to the present embodiment includes an inverter circuit 21, a power correction circuit 22, a control circuit 23, a measurement unit 24, a primary coil L1, and a pair of primary capacitors C11 and C12. ing.

  The inverter circuit 21 includes a plurality (four in this case) of conversion switch elements Q1 to Q4 that are electrically connected between the pair of input points 211 and 212 and the pair of output points 213 and 214. . The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 to an AC voltage by switching of the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214.

  The primary coil L1 is electrically connected between the pair of output points 213 and 214. The primary side coil L1 supplies output electric power to the secondary side coil L2 in a non-contact manner by applying an AC voltage.

  The power correction circuit 22 is electrically connected between the output point 213 and the primary coil L1, and includes a correction capacitor C1 and a plurality (four in this case) of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. In other words, the power correction circuit 22 adjusts the magnitude of the capacitance component of the primary side resonance circuit between the output point 213 and the primary side coil L1 by switching the plurality of correction switch elements Q5 to Q8. Thereby, the magnitude of the output power transmitted from the primary side resonance circuit of the non-contact power feeding device 2 including the primary side coil L1 to the secondary side resonance circuit of the non-contact power receiving device 3 including the secondary side coil L2 is corrected. Is done.

  The control circuit 23 has functions as an element control unit 231, an estimation unit 232, a frequency control unit 233, a phase difference control unit 234, and a determination unit 235.

  The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8. The estimation unit 232 estimates a resonance characteristic that represents the relationship between the frequency of the first drive signals G1 to G4 and the magnitude of the output power.

  Here, “resonance characteristics” means frequency characteristics of output power of the non-contact power feeding device 2. The “resonance characteristics” will be described in detail later in the section “(4.1) No power correction circuit” in “(4) Basic operation”.

  The frequency control unit 233 adjusts the magnitude of the output power by controlling the frequencies of the first drive signals G1 to G4 to change discretely. The phase difference control unit 234 adjusts the magnitude of the output power by controlling the phase difference, which is the phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to be discretely changed. To do.

  Here, the “phase difference” is a phase delay of the second drive signals G6 and G7 with respect to the first drive signals G1 and G4 or a phase delay of the second drive signals G5 and G8 with respect to the first drive signals G2 and G3. is there. This point will be described in detail later in the section “(4.2) With power correction circuit” in “(4) Basic operation”.

  The determination unit 235 sets the step size of the control target value including at least one of the frequency controlled by the frequency control unit 233 and the phase difference controlled by the phase difference control unit 234 to the resonance characteristic estimated by the estimation unit 232. Decide accordingly.

  Here, the “step size” means the minimum change amount of the control target value (at least one of the frequency and the phase difference) that changes discretely. For example, the frequencies of the first drive signals G1 to G4 are controlled by the frequency control unit 233 so as to change discretely. Therefore, when the magnitude of the output power is adjusted by the frequency control unit 233, the control target value (frequency) that is the control target of the frequency control unit 233 changes with the “step size” as the minimum unit. Will do. Similarly, the phase difference that is the phase delay of the second drive signals G6, G7 (G5, G8) with respect to the first drive signals G1, G4 (G2, G3) is discretely changed by the phase difference control unit 234. Is controlled to do. Therefore, when the magnitude of the output power is adjusted by the phase difference control unit 234, the control target value (phase difference) that is the target of control by the phase difference control unit 234 has the “step size” as a minimum unit. Will change.

  According to the above configuration, the contactless power feeding device 2 according to the present embodiment has a magnitude of output power by at least one of frequency control by the frequency control unit 233 and phase difference control by the phase difference control unit 234. Can be adjusted. Therefore, even if the relative positional relationship between the primary side coil L1 and the secondary side coil L2 changes and the coupling coefficient between the primary side coil L1 and the secondary side coil L2 changes, the non-contact power feeding device Second, by adjusting the magnitude of the output power, it becomes possible to supply stable power.

  Moreover, in the non-contact power feeding device 2, the step size of the control target value (at least one of the frequency and the phase difference) that is the control target when adjusting the output power is the resonance characteristic estimated by the estimation unit 232. Will be decided accordingly. Therefore, for example, when the output power of the non-contact power feeding device 2 is far from the target value determined by the non-contact power receiving device 3, the output power is set to the target value by relatively increasing the step size of the control target value. It is possible to reduce the time required for the processing for approaching. When the output power of the non-contact power feeding device 2 approaches the target value, the step size of the control target value is made relatively small, thereby generating ripples due to fluctuation of the output power near the target value, Can be significantly prevented from exceeding the target value. In other words, in the non-contact power feeding device 2, the resolution of the control target value that is the control target when adjusting the output power is not fixed, but is determined according to the resonance characteristics, thereby adjusting the output power. The output power can be stabilized while shortening the time required for. Therefore, the non-contact power feeding device 2 has an advantage that stable power can be supplied regardless of the relative positional relationship between the primary side coil L1 and the secondary side coil L2.

(3) Circuit Configuration Next, a specific circuit configuration of the contactless power supply device 2 according to the present embodiment will be described with reference to FIG.

  The contactless power supply device 2 according to the present embodiment includes a pair of input terminals T11 and T12. A DC power source 5 is electrically connected to the pair of input terminals T11 and T12.

  The inverter circuit 21 is a full bridge inverter circuit in which four conversion switch elements Q1 to Q4 are connected in a full bridge. That is, the inverter circuit 21 has a first arm and a second arm that are electrically connected in parallel between the pair of input points 211 and 212, and the first arm and the second arm are four conversion switches. It consists of elements Q1 to Q4. The first arm is composed of a series circuit of a (first) conversion switch element Q1 and a (second) conversion switch element Q2, and the second arm is a (third) conversion switch element Q3 (fourth). A) a series circuit with a conversion switch element Q4. The midpoint of the first arm (connection point of the conversion switch elements Q1 and Q2) and the midpoint of the second arm (connection point of the conversion switch elements Q3 and Q4) are a pair of output points 213 and 214. In the present embodiment, each of the four conversion switch elements Q1 to Q4 is an n-channel depletion type MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor).

  More specifically, the pair of input points 211 and 212 has a pair of inputs such that the first input point 211 is on the positive side of the DC power source 5 and the second input point 212 is on the negative side of the DC power source 5. The terminals T11 and T12 are electrically connected. The drains of the conversion switch elements Q1 and Q3 are electrically connected to the first input point 211. The sources of the conversion switch elements Q2 and Q4 are electrically connected to the second input point 212. A connection point between the source of the conversion switch element Q1 and the drain of the conversion switch element Q2 is the first output point 213 of the inverter circuit 21. The connection point between the source of the conversion switch element Q3 and the drain of the conversion switch element Q4 is the second output point 214 of the inverter circuit 21.

  The “input point” and “output point” in the present embodiment may not have an entity as a component (terminal) for connecting an electric wire or the like. For example, a lead of an electronic component or a conductor included in a circuit board It may be a part of

  Between the drains and sources of the four conversion switch elements Q1 to Q4, four diodes D1 to D4 are electrically connected to correspond to the four conversion switch elements Q1 to Q4 in a one-to-one relationship. . Each of the diodes D1 to D4 is connected in a direction in which the drain side of each of the conversion switch elements Q1 to Q4 is a cathode. Here, the diodes D1 to D4 are parasitic diodes of the conversion switch elements Q1 to Q4.

  The power correction circuit 22 includes a correction capacitor C1 and four correction switch elements Q5 to Q8. The power correction circuit 22 includes a third arm and a fourth arm that are electrically connected in parallel between the pair of output points 213 and 214 of the inverter circuit 21, and the third arm and the fourth arm are 4 in number. It is composed of two correction switch elements Q5 to Q8. The third arm is composed of a series circuit of a (first) correction switch element Q5 and a (third) correction switch element Q7, and the fourth arm is a (second) correction switch element Q6 and (fourth). A) a series circuit with a correction switch element Q8. A correction capacitor C1 is electrically connected between the midpoint of the third arm (connection point of the correction switch elements Q5 and Q7) and the midpoint of the fourth arm (connection point of the correction switch elements Q6 and Q8). Connected. In this embodiment, each of the four correction switch elements Q5 to Q8 is an n-channel depletion type MOSFET.

  More specifically, the source of the correction switch element Q5 and the drain of the correction switch element Q6 are electrically connected to the first output point 213 of the inverter circuit 21 via the first primary capacitor C11. ing. Further, the source of the correction switch element Q7 and the drain of the correction switch element Q8 are electrically connected to the second output point 214 via the second primary side capacitor C12 and the primary side coil L1. . One end of the correction capacitor C1 is electrically connected to a connection point between the drain of the correction switch element Q5 and the drain of the correction switch element Q7. The other end of the correction capacitor C1 is electrically connected to a connection point between the source of the correction switch element Q6 and the source of the correction switch element Q8. The capacity of the correction capacitor C1 is sufficiently larger than the capacity of the primary side capacitors C11 and C12 in the primary side resonance circuit. For example, if the capacity of the primary side capacitors C11 and C12 is on the order of [nF], the correction capacitor C1 Is in the order of [μF].

  Between the respective drains and sources of the four correction switch elements Q5 to Q8, four diodes D5 to D8 are electrically connected so as to correspond to the four correction switch elements Q5 to Q8 on a one-to-one basis. . The diodes D5 to D8 are connected in such a direction that the drain side of each of the correction switch elements Q5 to Q8 is a cathode. Here, the diodes D5 to D8 are parasitic diodes of the correction switch elements Q5 to Q8.

  The control circuit 23 has functions as an element control unit 231, an estimation unit 232, a frequency control unit 233, a phase difference control unit 234, and a determination unit 235. The control circuit 23 includes, for example, a microcomputer as a main configuration. The microcomputer realizes the function as the control circuit 23 by executing a program recorded in the memory of the microcomputer by a CPU (Central Processing Unit). That is, the functions of the element control unit 231, the estimation unit 232, the frequency control unit 233, the phase difference control unit 234, and the determination unit 235 are realized when the microcomputer processor executes the program. The program may be recorded in advance in a memory of a microcomputer, may be provided by being recorded on a recording medium such as a memory card, or may be provided through an electric communication line.

  The element control unit 231 outputs first drive signals G1 to G4 for switching on and off the conversion switch elements Q1 to Q4 of the inverter circuit 21. The four first drive signals G1 to G4 correspond one to one with the four conversion switch elements Q1 to Q4. Here, the element control unit 231 controls the corresponding conversion switch elements Q1 to Q4 by outputting the first drive signals G1 to G4 to the gates of the corresponding conversion switch elements Q1 to Q4, respectively. Yes.

  In addition, the element control unit 231 outputs second drive signals G5 to G8 for switching each of the four correction switch elements Q5 to Q8 of the power correction circuit 22 on and off. The four second drive signals G5 to G8 correspond one to one with the four correction switch elements Q5 to Q8. Here, the element control unit 231 controls the corresponding correction switch elements Q5 to Q8 by outputting the second drive signals G5 to G8 to the gates of the corresponding correction switch elements Q5 to Q8, respectively. Yes.

  In the present embodiment, the control circuit 23 (element control unit 231) applies the first drive signal G1 to G4 and the second drive signal to the gates of the conversion switch elements Q1 to Q4 and the correction switch elements Q5 to Q8. Although G5 to G8 are directly output, this is not restrictive. For example, the non-contact power feeding device 2 further includes a drive circuit, and the drive circuit receives the first drive signals G1 to G4 and the second drive signals G5 to G8 from the control circuit 23 (element control unit 231) and performs conversion. The switch elements Q1 to Q4 and the correction switch elements Q5 to Q8 may be driven.

  Functions of the control circuit 23 other than the element control unit 231 (estimation unit 232, frequency control unit 233, phase difference control unit 234, and determination unit 235) will be described in the section “(6) Output power control”.

  The primary coil L <b> 1 is electrically connected in series with the pair of primary capacitors C <b> 11 and C <b> 12 and the power correction circuit 22 between the pair of output points 213 and 214 of the inverter circuit 21. One end of the primary side coil L1 is electrically connected to the first output point 213 of the inverter circuit 21 via the power correction circuit 22 and the first primary side capacitor C11. The other end of the primary coil L1 is electrically connected to the second output point 214 of the inverter circuit 21 via the second primary capacitor C12.

  The measurement part 24 measures the magnitude | size of the electric current which flows into the primary side coil L1 as a measured value. Between the primary side coil L1 and the 2nd primary side capacitor | condenser C12, the current sensor 25 which consists of a current transformer, for example is provided. The measuring unit 24 receives the output of the current sensor 25, and measures the magnitude of the current flowing through the primary coil L1 as a measured value. The measurement unit 24 is configured to output the measurement value to the control circuit 23. The control circuit 23 monitors the magnitude of the output power output from the primary coil L1 using the measurement value measured by the measurement unit 24.

(4) Basic Operation Next, the basic operation of the contactless power supply device 2 of the present embodiment will be described with reference to FIGS. 1 and 2. In FIG. 2, with the horizontal axis as a time axis, the signal waveforms of the first drive signals “G1, G4”, “G2, G3”, the second drive signals “G5, G8”, “G6, G7” are shown in order from the top. ing. Note that “ON” and “OFF” in FIG. 2 indicate ON / OFF of the corresponding switch element (conversion switch element, correction switch element).

(4.1) No power correction circuit Here, first, when there is no power correction circuit 22, only the primary coil L1 and the pair of primary capacitors C11 and C12 are electrically connected between the pair of output points 213 and 214. The operation of the non-contact power feeding device 2 will be described assuming a connection. The operation of the non-contact power feeding device 2 in this case is performed when the power correction circuit 22 stops operating in the circuit configuration of FIG. This is equivalent to the operation of the non-contact power feeding device 2 when it is fixed.

  As shown in FIG. 2, the element controller 231 includes first drive signals G1 and G4 corresponding to the conversion switch elements Q1 and Q4 and first drive signals G2 and G3 corresponding to the conversion switch elements Q2 and Q3. , Signals having opposite phases (phase difference 180 degrees) are generated. Thus, in the inverter circuit 21, the pair of the first conversion switch element Q1 and the fourth conversion switch element Q4, the pair of the second conversion switch element Q2 and the third conversion switch element Q3, Are controlled to turn on alternately.

  As a result, a voltage (AC voltage) whose polarity (positive / negative) is periodically inverted is generated between the pair of output points 213 and 214 of the inverter circuit 21. In short, the inverter circuit 21 converts the DC voltage applied to the pair of input points 211 and 212 to an AC voltage and outputs the AC voltage from the pair of output points 213 and 214 by switching of the plurality of conversion switch elements Q1 to Q4. . Hereinafter, with respect to the output voltage of the inverter circuit 21, the voltage at which the first output point 213 of the pair of output points 213 and 214 has a high potential is referred to as “positive polarity”, and the second output point 214 has a high potential. This voltage is called “negative polarity”. That is, the output voltage of the inverter circuit 21 is positive when the conversion switch elements Q1 and Q4 are on, and is negative when the conversion switch elements Q2 and Q3 are on.

  Thus, when the inverter circuit 21 outputs an alternating voltage from the pair of output points 213 and 214, an alternating current flows through the primary coil L1 electrically connected between the pair of output points 213 and 214, and the primary The side coil L1 generates a magnetic field. Thereby, the non-contact power feeding device 2 can supply output power from the primary side coil L1 to the secondary side coil L2 of the non-contact power receiving device 3 in a non-contact manner.

  By the way, when there is no electric power correction circuit 22, in the non-contact electric power feeder 2, the primary side coil L1 comprises a primary side resonance circuit with a pair of primary side capacitors C11 and C12. Therefore, the magnitude of the output power output from the primary side coil L1 changes according to the operating frequency of the inverter circuit 21 (that is, the frequency of the first drive signals G1 to G4), and the operating frequency of the inverter circuit 21 is the primary side. A peak is reached when it matches the resonant frequency of the resonant circuit.

  Here, when the relative positional relationship between the primary side coil L1 and the secondary side coil L2 changes and the coupling coefficient between the primary side coil L1 and the secondary side coil L2 changes, the non-contact power feeding device 2 The frequency characteristic of the output power (hereinafter referred to as “resonance characteristic”) changes. The “resonance characteristic” here, that is, the frequency characteristic of the output power, represents the relationship between the operating frequency of the inverter circuit 21 and the magnitude of the output power. That is, the relationship between the frequency of the AC voltage applied to the primary side resonance circuit from the pair of output points 213 and 214 of the inverter circuit 21 and the magnitude of the output power output from the primary side coil L1 of the primary side resonance circuit is It is expressed as a resonance characteristic. FIG. 3 shows a change in the resonance characteristics of the non-contact power feeding device 2 when the relative positional relationship between the primary side coil L1 and the secondary side coil L2 changes. In FIG. 3, the horizontal axis is frequency (operating frequency of the inverter circuit 21), and the vertical axis is output power of the non-contact power feeding device 2. The resonance characteristics of the non-contact power feeding device 2 are indicated by “X1” and “X2”.

  Here, as shown in FIG. 3, it is assumed that the frequency band that can be used as the operating frequency of the inverter circuit 21 (hereinafter referred to as “permitted frequency band F1”) is limited. The permitted frequency band F1 is defined by laws such as the Radio Law. In this case, a frequency that is less than the lower limit value fmin and exceeds the upper limit value fmax of the permitted frequency band F1 cannot be used as the operating frequency of the inverter circuit 21. In such a case, if the resonance characteristic of the non-contact power feeding device 2 is in a state indicated by “X1” in FIG. 3, for example, the output of the non-contact power feeding device 2 can be adjusted no matter how the operating frequency of the inverter circuit 21 is adjusted. There is a possibility that electric power does not have the required size (hereinafter referred to as “target value”).

  For example, as shown in FIG. 4A, when the resonance frequency fr0 of the primary side resonance circuit is out of the permitted frequency band F1, the magnitude of the output power of the non-contact power feeding device 2 does not reach the peak, and as a result, the target value The output power may be insufficient with respect to P1. For example, as shown in FIG. 4B, even when the resonance frequency fr0 of the primary side resonance circuit is within the permitted frequency band F1, the peak of the output power of the non-contact power feeding device 2 does not reach the target value P1, and as a result The output power may be insufficient with respect to the target value P1. That is, in the examples of FIGS. 4A and 4B, the power in the hatched portion (shaded area) is insufficient with respect to the target value P1.

  Therefore, the non-contact power feeding device 2 according to the present embodiment has a function of correcting the magnitude of the output power so as to satisfy the target value P1 by including the power correction circuit 22.

(4.2) With Power Correction Circuit Next, when the power correction circuit 22 is provided as shown in FIG. 1, that is, between the pair of output points 213 and 214, the primary coil L1 and the pair of primary capacitors C11 and C12. The operation of the non-contact power feeding device 2 when the power correction circuit 22 is electrically connected will be described.

  As shown in FIG. 2, the element controller 231 includes second drive signals G6 and G7 corresponding to the correction switch elements Q6 and Q7 and second drive signals G5 and G8 corresponding to the correction switch elements Q5 and Q8. , Signals having opposite phases (phase difference 180 degrees) are generated. Thereby, in the power correction circuit 22, a pair of the second correction switch element Q6 and the third correction switch element Q7, and a pair of the first correction switch element Q5 and the fourth correction switch element Q8. And are controlled to turn on alternately. In the present embodiment, the element control unit 231 uses the same frequency for the first drive signals G1 to G4 and the second drive signals G5 to G8.

  When the correction switch elements Q5 and Q8 are on and the correction switch elements Q6 and Q7 are off, if the output voltage of the inverter circuit 21 is positive, the correction is performed via the correction switch elements Q5 and Q8. A voltage is applied to the capacitor C1. That is, the primary coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 via the correction capacitor C1 (hereinafter also referred to as “first state”).

  On the other hand, when the correction switch elements Q5 and Q8 are on and the correction switch elements Q6 and Q7 are off, if the output voltage of the inverter circuit 21 is negative, a current flows through the diode D7 and the correction switch element Q5. . That is, the primary coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 without passing through the correction capacitor C1 (hereinafter also referred to as “second state”). . In other words, both ends of the correction capacitor C1 are bypassed by the diode D7 and the correction switch element Q5.

  In addition, when the correction switch elements Q6 and Q7 are on and the correction switch elements Q5 and Q8 are off, if the output voltage of the inverter circuit 21 is positive, a current flows through the correction switch element Q6 and the diode D8. . That is, the primary coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 without passing through the correction capacitor C1 (hereinafter also referred to as “third state”). . In other words, both ends of the correction capacitor C1 are bypassed by the correction switch element Q6 and the diode D8.

  On the other hand, if the output voltage of the inverter circuit 21 is negative while the correction switch elements Q6 and Q7 are on and the correction switch elements Q5 and Q8 are off, correction is performed via the correction switch elements Q6 and Q7. A voltage is applied to the capacitor C1. That is, the primary coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 via the correction capacitor C1 (hereinafter also referred to as “fourth state”). The polarity of the voltage applied to the correction capacitor C1 is the same in the first state and the fourth state. That is, in any of the first state and the fourth state, the voltage at which the connection point between the drain of the correction switch element Q5 and the drain of the correction switch element Q7 becomes a high potential is used for the correction. Applied to capacitor C1.

  As described above, the power correction circuit 22 is electrically connected between the pair of output points 213 and 214 with the primary coil L1 being electrically connected via the correction capacitor C1 and not via the correction capacitor C1. The state connected to is switched. Thereby, the magnitude of the capacitance component of the primary side resonance circuit between the pair of output points 213 and 214 and the primary side coil L1 apparently changes.

  Therefore, the contactless power supply device 2 of the present embodiment can change the magnitude of the output power by adjusting the magnitude of the capacitance component of the primary side resonance circuit by the power correction circuit 22. As a result, as described above, when the output power of the non-contact power feeding device 2 is insufficient with respect to the target value P1, the power correction circuit 22 corrects the magnitude of the output power so as to satisfy the target value P1. It is possible. In other words, the power correction circuit 22 can correct the magnitude of the output power by charging / discharging the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8.

  In the present embodiment, as described above, the first drive signals G1, G4 corresponding to the conversion switch elements Q1, Q4 and the first drive signals G2, G3 corresponding to the conversion switch elements Q2, Q3 are as follows. The signals are opposite in phase (phase difference 180 degrees). Similarly, the second drive signals G6 and G7 corresponding to the correction switch elements Q6 and Q7 and the second drive signals G5 and G8 corresponding to the correction switch elements Q5 and Q8 are opposite in phase (phase difference 180 degrees). Signal.

  Here, the phase difference θ between the first drive signal and the second drive signal in the present embodiment is the phase delay of the second drive signals G6, G7 with respect to the first drive signals G1, G4, or the first drive signal G2. , G3 is the phase delay of the second drive signals G5, G8 (see FIG. 2). That is, there is a 180 degree difference between the phase delay of the second drive signals G6 and G7 with respect to the first drive signals G1 and G4 and the phase delay of the second drive signals G5 and G8 with respect to the first drive signals G1 and G4. Therefore, the value of the phase difference θ differs depending on which phase delay is the phase difference θ. Therefore, in the present embodiment, the phase delay of the second drive signals G6 and G7 with respect to the first drive signals G1 and G4 or the phase delay of the second drive signals G5 and G8 with respect to the first drive signals G2 and G3 is determined as a phase difference. It is defined as θ.

  Here, if the first drive signals G1, G4 and the second drive signals G6, G7 are all “on”, the primary coil L1 is connected between the pair of output points 213, 214 in the power correction circuit 22. A third state is established in which the correction capacitor C1 is not electrically connected. If the first drive signals G2 and G3 and the second drive signals G5 and G8 are both “on”, the primary coil L1 is corrected between the pair of output points 213 and 214 in the power correction circuit 22. It becomes the 2nd state electrically connected without passing through capacitor C1. That is, in the present embodiment, the first drive signal and the second drive are such that the primary coil L1 is electrically connected between the pair of output points 213 and 214 without passing through the correction capacitor C1. A phase difference for a combination with a signal is defined as a phase difference θ.

(5) Advanced Phase Mode and Late Phase Mode Next, the advanced phase mode and the delayed phase mode will be described.

(5.1) Without power correction circuit Here, first, as in the column of “(4) Basic operation”, when there is no power correction circuit 22, that is, between the pair of output points 213 and 214, the primary coil L1 and A case where only a pair of primary capacitors C11 and C12 are electrically connected will be described.

  In this case, the inverter circuit 21 operates in one of the operation modes of the slow phase mode and the fast phase mode, for example, according to the relationship between the operation frequency of the inverter circuit 21 and the resonance frequency of the primary side resonance circuit.

  The phase advance mode is a mode in which the inverter circuit 21 operates in a state in which the phase of the output current of the inverter circuit 21 (current flowing through the primary coil L1) is ahead of the phase of the output voltage of the inverter circuit 21. In the phase advance mode, the switching operation of the inverter circuit 21 is hard switching. Therefore, in the phase advance mode, the switching loss of the conversion switch elements Q1 to Q4 is likely to increase, and stress is easily applied to the conversion switch elements Q1 to Q4.

  On the other hand, the slow phase mode is a mode in which the inverter circuit 21 operates in a state where the phase of the output current of the inverter circuit 21 (current flowing through the primary coil L1) is delayed from the phase of the output voltage of the inverter circuit 21. . In the slow phase mode, the switching operation of the inverter circuit 21 is soft switching. Therefore, in the slow phase mode, the switching loss of the conversion switch elements Q1 to Q4 can be reduced, and stress is not easily applied to the conversion switch elements Q1 to Q4. Therefore, it is preferable that the inverter circuit 21 operates in the slow phase mode rather than the fast phase mode.

(5.2) With power correction circuit Next, when there is the power correction circuit 22, that is, between the pair of output points 213 and 214, the primary coil L1, the pair of primary capacitors C11 and C12, and the power correction circuit 22. The case where is electrically connected will be described.

  In this case, similarly to the inverter circuit 21, the power correction circuit 22 operates in one of the operation modes of the fast phase mode and the slow phase mode. The power correction circuit 22 also preferably operates in the slow phase mode instead of the fast phase mode.

  Further, when the power correction circuit 22 is provided, the operation modes (the slow phase mode and the fast phase mode) of the inverter circuit 21 and the power correction circuit 22 are the first drive signals G1 to G4 and the second drive signals G5 to G8. It has been confirmed that it changes in accordance with the phase difference θ. Further, the relationship between the operation mode of the inverter circuit 21 and the phase difference θ is that the operation of the inverter circuit 21 is performed without the power correction circuit 22, that is, under the conditions described in the above “(5.1) No power correction circuit”. Varies depending on the mode. In other words, depending on whether the operation mode of the inverter circuit 21 determined by the relationship between the operation frequency of the inverter circuit 21 and the resonance frequency of the primary side resonance circuit is the slow phase mode or the fast phase mode, the operation mode of the inverter circuit 21 and the phase difference θ The relationship changes.

  5A and 5B show the characteristics of the output power of the non-contact power feeding device 2 with respect to the phase difference θ when the inverter circuit 21 is in the fast phase mode and in the slow phase mode without the power correction circuit 22, respectively. (Phase difference characteristic) is shown. 5A and 5B, the horizontal axis represents the phase difference θ between the first drive signals G1 to G4 and the second drive signals G5 to G8, and the vertical axis represents the output power of the non-contact power feeding device 2.

  That is, when the inverter circuit 21 is in the phase advance mode without the power correction circuit 22 (hereinafter referred to as “initial phase advance”), the output power of the non-contact power feeding device 2 is, for example, as shown in FIG. It changes according to the phase difference θ. In the example of FIG. 5A, the output power of the non-contact power feeding device 2 is maximized and maximized when the phase difference θ is 90 degrees, and minimized and minimized when the phase difference θ is 180 degrees. Change. The principle that the output power of the non-contact power supply device 2 changes depending on the phase difference θ will be described in the section “(6.2) Principle of output power control by phase difference control” below. Here, the phase difference θ (0 degrees to 360 degrees) is divided into four sections, 0 degrees to 90 degrees is the first section Z1, 90 degrees to 180 degrees is the second section Z2, and 180 degrees to 270 degrees is the third section. The section Z3, 270 to 360 degrees is set as the fourth section Z4. Then, the relationship between the operation mode (slow phase mode, phase advance mode) of the inverter circuit 21 and each section of the phase difference θ is as shown in Table 1.

  In short, in the case of “initial phase advance”, the inverter circuit 21 operates in the slow phase mode in which the phase difference θ is 90 degrees to 180 degrees in the first section Z1 to the fourth section Z4. Only the second section Z2.

  On the other hand, when the inverter circuit 21 is in the slow phase mode without the power correction circuit 22 (hereinafter referred to as “initial delay phase”), the output power of the non-contact power feeding device 2 is, for example, as shown in FIG. It changes according to the phase difference θ. In the example of FIG. 5B, the output power of the non-contact power feeding device 2 is maximized and maximized when the phase difference θ is 270 degrees, and minimized and minimized when the phase difference θ is 180 degrees. Change. In this case, the relationship between the respective operation modes (the slow phase mode and the fast phase mode) of the inverter circuit 21 and the power correction circuit 22 and the respective sections of the phase difference θ is as shown in Table 2.

  In short, in the case of “initial delay phase”, the inverter circuit 21 operates in the delay phase mode because the phase difference θ of the first section Z1 to the fourth section Z4 is 0 degree to 180 degrees, 270 The first section Z1, the second section Z2, and the fourth section Z4, which are degrees to 360 degrees. In addition, in the case of “initial delay phase”, the power correction circuit 22 operates in the delay phase mode because the phase difference θ of the first section Z1 to the fourth section Z4 is 0 degrees to 90 degrees, 180 degrees to The three sections are the first section Z1, the third section Z3, and the fourth section Z4, which are 360 degrees. That is, in the case of “initial delay phase”, both the inverter circuit 21 and the power correction circuit 22 operate in the delay phase mode because the phase difference θ is 0 in the first segment Z1 to the fourth segment Z4. There are two sections of the first section Z1 and the fourth section Z4, which are degrees to 90 degrees and 270 degrees to 360 degrees.

(6) Output power control Next, the operation of “output power control” for adjusting the magnitude of output power in the non-contact power feeding device 2 of the present embodiment will be described.

(6.1) Frequency Control and Phase Difference Control The control circuit 23 outputs two methods, “frequency control” performed by the frequency control unit 233 and “phase difference control” performed by the phase difference control unit 234. It is comprised so that the magnitude | size of electric power may be adjusted. The frequency control is control for discretely changing the frequency of the first drive signals G1 to G4, that is, the operating frequency f1 of the inverter circuit 21. On the other hand, the phase difference control discretely changes the phase difference θ, which is the phase delay of the second drive signals G6, G7 (G5, G8) with respect to the first drive signals G1, G4 (G2, G3). Control.

  That is, the control circuit 23 controls the output frequency by controlling the operating frequency f1 in the frequency control and the phase difference θ in the phase difference control to be the control target values and discretely changing the control target values. Adjust the size. In the frequency control, the frequency control unit 233 discretely changes the operating frequency f1, which is a control target value, with the increment for the operating frequency f1 as a minimum unit. In the phase difference control, the phase difference control unit 234 discretely changes the phase difference θ, which is a control target value, with the increment for the phase difference θ as a minimum unit. Hereinafter, the step size of the control target value (operating frequency f1) in frequency control, that is, the minimum change amount of the operating frequency f1 is also referred to as “frequency step size Δf”. Further, the step width of the control target value (phase difference θ) in the phase difference control, that is, the minimum change amount of the phase difference θ is also referred to as “phase difference step width Δθ”.

  In the present embodiment, the control circuit 23 first adjusts the magnitude of the output power by controlling the frequency controller 233 to change the frequencies (operating frequency f1) of the first drive signals G1 to G4. Take control. That is, as described in the column “(4.1) No power correction circuit” in “(4) Basic operation”, the magnitude of the output power output from the primary side coil L1 depends on the operation of the inverter circuit 21. It changes according to the frequency f1 (see FIG. 3). Therefore, the frequency control unit 233 adjusts the frequency of the first drive signals G1 to G4, thereby adjusting the operating frequency f1 of the inverter circuit 21 and adjusting the magnitude of the output voltage.

  Here, when the frequency band (permitted frequency band F1) that can be used as the operating frequency f1 of the inverter circuit 21 is limited, the operating frequency f1 that can be adjusted by frequency control is limited to the permitted frequency band F1. Is done. Then, when the magnitude of the output power adjusted by the frequency control is less than the predetermined target value P1, the control circuit 23 performs the phase difference control by the phase difference control unit 234. That is, the phase difference control unit 234 performs the phase difference control for controlling the phase difference θ when the magnitude of the output power adjusted by the frequency control unit 233 is less than the predetermined target value P1, thereby outputting the output. The power is configured to approach the target value P1. As described above, when the output power is insufficient with respect to the target value P1 only by the frequency control (see FIGS. 4A and 4B), the control circuit 23 compensates for the shortage by the phase difference control.

  The phase difference control unit 234 changes the phase difference θ, which is the phase delay of the second drive signals G6, G7 (G5, G8) with respect to the first drive signals G1, G4 (G2, G3), within a specified range. By controlling, the magnitude of the output power of the non-contact power feeding device 2 is adjusted. The “specified range” here is a control range of the phase difference θ in the phase difference control, that is, a range in which the phase difference θ is changed in the phase difference control. In short, the phase difference control unit 234 limits the movable range of the phase difference θ to a specified range and performs control so that the phase difference θ is discretely changed within the specified range. Here, the specified range includes at least one range of 270 to 360 degrees and 90 to 180 degrees.

  That is, as apparent from FIGS. 5A and 5B, the output power of the non-contact power feeding device 2 changes according to the phase difference θ, and therefore the phase difference control unit 234 changes the phase difference θ to increase the output power. The height can be adjusted. However, the phase difference θ between the first drive signals G1 to G4 and the second drive signals G5 to G8 is also applied to the operation modes (late phase mode and phase advance mode) of the inverter circuit 21 and the power correction circuit 22 as described above. Affect. Therefore, in order to operate the inverter circuit 21 and the power correction circuit 22 in the slow phase mode, it is necessary to limit the range of the phase difference θ.

  First, the case of “initial phase advance” will be described with reference to FIG. 5A. In this case, as described above, the inverter circuit 21 operates in the slow phase mode only in the second section Z2 in which the phase difference θ is 90 degrees to 180 degrees among the first section Z1 to the fourth section Z4. . Therefore, in the case of “initial phase advancement”, it is preferable that the specified range that is the control range of the phase difference θ in the phase difference control is a range of 90 degrees to 180 degrees. As a result, when the phase difference control unit 234 controls the phase difference θ to change within the specified range and adjusts the magnitude of the output power of the non-contact power feeding device 2, the inverter circuit 21 is in the slow phase mode. Can work.

  Next, the case of “initial delay phase” will be described with reference to FIG. 5B. In this case, as described above, the inverter circuit 21 operates in the slow phase mode in the three sections of the first section Z1, the second section Z2, and the fourth section Z4. However, in the first section Z1 in which the phase difference θ is 0 degree to 90 degrees, the magnitude of the output power hardly changes even if the phase difference θ changes. Therefore, in order to adjust the magnitude of the output power by adjusting the phase difference θ, the second section Z2 in which the phase difference θ is 90 degrees to 180 degrees and the fourth section in which the phase difference θ is 270 degrees to 360 degrees. It is necessary to adjust the phase difference θ within the two sections of the section Z4. Therefore, if the specified ranges are in the range of 270 ° to 360 ° and 90 ° to 180 °, the phase difference control unit 234 controls the phase difference θ to change within the specified range, and the magnitude of the output power is increased. When adjusted, the inverter circuit 21 can operate in the slow phase mode.

  Further, in the case of “initial delay phase”, both the inverter circuit 21 and the power correction circuit 22 operate in the delay mode because the phase difference θ is 0 degree to 90 degrees and 270 degrees to 360 degrees as described above. There are two sections, a first section Z1 and a fourth section Z4. As described above, since the magnitude of the output power hardly changes even if the phase difference θ changes in the first section Z1, the phase difference θ of the two sections of the first section Z1 and the fourth section Z4. Only the fourth section Z4 can adjust the magnitude of the output power by adjustment. Therefore, in the case of “initial delay phase”, the specified range that is the control range of the phase difference θ in the phase difference control is more preferably a range of 270 degrees to 360 degrees. Thus, when the phase difference control unit 234 controls the phase difference θ to change within a specified range and adjusts the magnitude of the output power of the non-contact power feeding device 2, the inverter circuit 21 and the power correction circuit 22 are controlled. Both can operate in slow phase mode.

  In short, in the case of “initial phase advance”, the specified range is preferably a range of 90 degrees to 180 degrees (second section Z2). On the other hand, in the case of “initial delay phase”, the specified range is preferably a range of 90 to 180 degrees (second section Z2) or a range of 270 to 360 degrees (fourth section Z4). In the case of “initial delay phase”, the specified range is more preferably a range of 270 degrees to 360 degrees (fourth section Z4).

  In the section where the inverter circuit 21 operates in the slow phase mode (at least one of the second section Z2 and the fourth section Z4), as shown in FIGS. 5A and 5B, the output power is reduced as the phase difference θ decreases. Becomes bigger. Therefore, for example, if the specified range is the second section (90 degrees to 180 degrees) Z2, the phase difference θ gradually decreases from the upper limit value (180 degrees) to the lower limit value (90 degrees) of the specified range. It is preferable that the phase difference control unit 234 gradually reduce the phase difference θ within a specified range. Similarly, if the specified range is the fourth section (270 degrees to 360 degrees) Z4, the phase difference θ gradually decreases from the upper limit value (360 degrees) to the lower limit value (270 degrees) of the specified range. It is preferable that the phase difference control unit 234 gradually reduce the phase difference θ within a specified range. Thereby, as the phase difference control unit 234 gradually changes (decreases) the phase difference θ within the specified range, the output power of the non-contact power feeding device 2 gradually increases.

(6.2) Principle of output power control by phase difference control Hereinafter, the magnitude of the output power of the non-contact power feeding device 2 is controlled by the phase difference control unit 234 controlling the phase difference θ to change within a specified range. The principle by which is adjusted will be described with reference to FIGS.

  The output power of the non-contact power feeding device 2 varies depending on the voltage applied to the primary side coil L1 of the primary side resonance circuit. The voltage applied to the primary coil L1 is a combination of the output voltage of the inverter circuit 21 and the voltage across the power correction circuit 22 (the voltage between the source of the correction switch element Q5 and the source of the correction switch element Q7). Voltage. Therefore, when the output voltage of the inverter circuit 21 and the voltage at both ends of the power correction circuit 22 have the same polarity and strengthen each other, the voltage applied to the primary coil L1 increases, and the output of the non-contact power feeding device 2 Electric power is increased. In this case, as the voltage across the correction capacitor C1 increases, the voltage across the power correction circuit 22 increases and the voltage applied to the primary coil L1 increases. Becomes bigger. Therefore, the phase difference control unit 234 changes the balance between charging and discharging of the correction capacitor C1 by adjusting the phase difference θ, and changes the voltage across the correction capacitor C1, thereby changing the contactless power feeding device 2. Vary the output power.

  Here, whether the correction capacitor C1 is charged or discharged depends on the on / off of the plurality of correction switch elements Q5 to Q8 and the direction of the output current of the inverter circuit 21. Since the output current of the inverter circuit 21 is a current flowing through the primary side coil L1, it is also referred to as “primary side current I1” hereinafter. The direction of the primary current I1 flowing from the first output point 213 to the second output point 214 through the primary side capacitor C11, the power correction circuit 22, the primary side coil L1, and the primary side capacitor C12, that is, in FIG. The direction of the primary current I1 indicated by the arrow is referred to as “positive direction”. The direction of the primary side current I1 flowing from the second output point 214 to the first output point 213 through the primary side capacitor C12, the primary side coil L1, the power correction circuit 22, and the primary side capacitor C11, that is, in FIG. The direction opposite to the primary current I1 indicated by the arrow is referred to as “negative direction”.

  6A to 6D show a combination pattern of on / off of the plurality of correction switch elements Q5 to Q8 and the direction of the primary current I1. In FIGS. 6A to 6D, thick arrows represent current paths, and correction switch elements with dotted circles represent elements in an on state.

  FIG. 6A shows a state of the power correction circuit 22 in which the correction switch elements Q6 and Q7 are on, the correction switch elements Q5 and Q8 are off, and the primary current I1 in the negative direction is flowing (hereinafter referred to as the power correction circuit 22). "First charging mode"). FIG. 6B shows the state of the power correction circuit 22 in which the correction switch elements Q6 and Q7 are on, the correction switch elements Q5 and Q8 are off, and the primary current I1 in the positive direction is flowing (hereinafter, referred to as the power correction circuit 22). "First discharge mode"). FIG. 6C shows a state of the power correction circuit 22 in which the correction switch elements Q5 and Q8 are on, the correction switch elements Q6 and Q7 are off, and the primary current I1 in the positive direction flows (hereinafter, referred to as “the power correction circuit 22”). "Second charging mode"). FIG. 6D shows the state of the power correction circuit 22 in which the correction switch elements Q5 and Q8 are on, the correction switch elements Q6 and Q7 are off, and the primary current I1 in the negative direction flows (hereinafter referred to as the power correction circuit 22). "Second discharge mode"). In the first charging mode shown in FIG. 6A and the second charging mode shown in FIG. 6C, the correction capacitor C1 is charged. On the other hand, the correction capacitor C1 is discharged in the first discharge mode shown in FIG. 6B and the second discharge mode shown in FIG. 6D.

  Next, with reference to FIGS. 7 and 8, the relationship between the phase difference θ and the balance between charging and discharging of the correction capacitor C1 will be described. 7 and 8, the waveforms of the first drive signal “G1, G4”, the primary side current “I1”, and the two types of second drive signals “G6, G7” are shown in order from the top with the horizontal axis as the time axis. Represents. The two types of second drive signals here have different phase differences θ. Note that “ON” and “OFF” in FIGS. 7 and 8 indicate ON / OFF of the corresponding switch element (conversion switch element, correction switch element).

  FIG. 7 illustrates a case where the phase delay of the primary current I1 with respect to the output voltage of the inverter circuit 21 (hereinafter referred to as “voltage-current phase difference”) φ is 90 degrees in the case of “initial delay phase”. doing. In FIG. 7, as waveforms of the two types of second drive signals “G6 and G7”, a waveform when the phase difference θ is 360 degrees and a waveform when the phase difference θ is 320 degrees are shown in order from the top. Further, in FIG. 7, when the phase difference θ is 360 degrees, the first charging mode period is “Tca1”, the first discharging mode period is “Tda1”, the second charging mode period is “Tca2”, The period of the two discharge mode is represented by “Tda2”. Similarly, in the case where the phase difference θ is 320 degrees, the first charge mode period is “Tcb1”, the first discharge mode period is “Tdb1”, the second charge mode period is “Tcb2”, and the second discharge mode. Is represented by “Tdb2”.

  As is apparent from FIG. 7, when the phase difference θ is 360 degrees, the correction capacitor C1 is charged in one cycle of the second drive signal (hereinafter referred to as “charging time”), and the correction capacitor. The time during which C1 is discharged (hereinafter referred to as “discharge time”) is substantially balanced. That is, if the phase difference θ is 360 degrees, the sum of “Tca1” and “Tca2” and the sum of “Tda1” and “Tda2” have substantially the same length. On the other hand, if the phase difference θ is 320 degrees, the charging time exceeds the discharging time in one cycle of the second drive signal. That is, if the phase difference θ is 320 degrees, the sum of “Tcb1” and “Tcb2” is longer than the sum of “Tdb1” and “Tdb2”.

  From the above, when the phase difference θ changes from 360 degrees to approach 270 degrees, the balance between the charging time and the discharging time is broken in one cycle of the second drive signal, and the proportion of the charging time gradually increases. . In FIG. 7, for convenience of explanation, the description of the change in the voltage / current phase difference φ is omitted, but actually, when the phase difference θ changes, the voltage / current phase difference φ also changes to the initial value along with the change in the phase difference θ. It changes from (90 degrees here). That is, when the charging time exceeds the discharging time, a current flows into the correction capacitor C1, and this current phase advances by 90 degrees with respect to the phase of the voltage across the correction capacitor C1. That is, since the correcting capacitor C1 functions as a phase advance capacitor, the voltage / current phase difference φ, which is the phase delay of the primary current I1 with respect to the output voltage of the inverter circuit 21, is reduced, and the output power is increased. As a result, as the phase difference θ approaches from 270 degrees to 270 degrees, the voltage / current phase difference φ decreases and the output power of the non-contact power feeding device 2 gradually increases.

  In other words, as the phase difference θ decreases, the voltage / current phase difference φ decreases to maintain a balance between the charging time and the discharging time. That is, when the state where the charging time exceeds the discharging time continues, the voltage across the correction capacitor C1 diverges. In order to prevent such divergence from occurring, in practice, when the phase difference θ decreases, the voltage / current phase difference φ decreases, and the balance between the charging time and the discharging time is maintained.

  FIG. 8 illustrates the case of “initial delay phase” in which the voltage-current phase difference φ is 45 degrees. In FIG. 8, as the waveforms of the two types of second drive signals “G6, G7”, the waveforms when the phase difference θ is 315 degrees and the waveforms when the phase difference θ is 290 degrees are shown in order from the top. Further, in FIG. 8, in the case where the phase difference θ is 315 degrees, the first charging mode period is “Tca1”, the first discharging mode period is “Tda1”, the second charging mode period is “Tca2”, The period of the two discharge mode is represented by “Tda2”. Similarly, in the case where the phase difference θ is 290 degrees, the first charge mode period is “Tcb1”, the first discharge mode period is “Tdb1”, the second charge mode period is “Tcb2”, and the second discharge mode. Is represented by “Tdb2”.

  If the voltage-current phase difference φ is 45 degrees, as is apparent from FIG. 8, even if the phase difference θ is 315 degrees, the charging time and the discharging time are substantially balanced in one cycle of the second drive signal. . That is, even if the phase difference θ is 315 degrees, the sum of “Tca1” and “Tca2” and the sum of “Tda1” and “Tda2” have substantially the same length. On the other hand, if the phase difference θ is 290 degrees, the charging time exceeds the discharging time in one cycle of the second drive signal. That is, if the phase difference θ is 290 degrees, the sum of “Tcb1” and “Tcb2” is longer than the sum of “Tdb1” and “Tdb2”.

  From the above, not only when the voltage-current phase difference φ is 90 degrees, but in the case of “initial delay phase”, if the phase difference θ changes from 360 degrees to 270 degrees, one cycle of the second drive signal , The balance between the charging time and the discharging time is broken, and the proportion of the charging time gradually increases. However, when the voltage-current phase difference φ is 90 degrees, the charging time exceeds the discharge time when the phase difference θ is 320 degrees, whereas when the voltage-current phase difference φ is 45 degrees, the phase difference θ is 315. Even when the temperature is high, the charging time and the discharging time are balanced. As described above, when the phase difference θ is gradually reduced from 360 degrees, the phase difference θ corresponding to the inflection point where the balance between the charging time and the discharging time is broken and the output power starts to increase (hereinafter referred to as “change start”). The “point” is different depending on the voltage-current phase difference φ. The change start point shifts closer to 270 degrees when the voltage-current phase difference φ is 45 degrees, that is, when the voltage-current phase difference φ is smaller.

  That is, in the case of “initial delay phase”, even if there is a difference due to the voltage / current phase difference φ, there is a change start point within a specified range (for example, 270 ° to 360 °). Therefore, if the phase difference control unit 234 gradually decreases the phase difference θ from the upper limit value (360 degrees) to the lower limit value (270 degrees) of the specified range, after the phase difference θ reaches the change start point, non- The output power of the contact power supply device 2 gradually increases. Actually, as described above, the voltage / current phase difference φ also changes with the change of the phase difference θ, and after the phase difference θ reaches the change start point, the voltage / current phase difference φ becomes smaller. Thus, the balance between the charging time and the discharging time is maintained.

  7 and 8 exemplify the case of “initial delay phase”, but in the case of “initial advance phase”, the phase of the primary side current I1 is shifted by 180 degrees with reference to the example of “initial delay phase”. Is equivalent to That is, in the example of FIGS. 7 and 8, if the phase of the primary current I1 is shifted by 180 degrees, an example of “initial phase advance” is obtained. Therefore, even in the case of “initial phase advancement”, even if there is a difference due to the voltage / current phase difference φ, there is a change start point within a specified range (180 degrees to 90 degrees). Therefore, if the phase difference control unit 234 gradually decreases the phase difference θ from the upper limit value (180 degrees) to the lower limit value (90 degrees) of the specified range, after the phase difference θ reaches the change start point, non- The output power of the contact power supply device 2 gradually increases.

  Based on the principle as described above, the phase difference control unit 234 controls the phase difference θ to change within a specified range in both the “initial phase delay” and the “initial phase advance”, thereby making contactless. The magnitude of the output power of the power feeding device 2 is adjusted.

(6.3) Overall Flow of Output Power Control Hereinafter, the overall flow of “output power control” of the present embodiment will be described with reference to the flowchart of FIG.

  When the output power control is started, the control circuit 23 first compares the predetermined target value P1 with the magnitude of the output power (S1). If the output power is within the allowable error range (± several percent) of the target value P1 (S1: rating), the output power control is terminated.

  On the other hand, if the output power is below the lower limit of the allowable error range of the target value P1 (S1: insufficient), the control circuit 23 first adjusts the output power by the frequency control unit 233. Specifically, the frequency control unit 233 compares the operating frequency (that is, the frequency of the first drive signals G1 to G4) f1 of the inverter circuit 21 with the lower limit value fmin of the permitted frequency band F1 (S11). If the operating frequency f1 is higher than the lower limit value fmin (S11: Yes), the frequency control unit 233 controls the operating frequency f1 of the inverter circuit 21 to be lowered. At this time, the frequency step Δf, which is the step size of the control target value (operating frequency f1) controlled by the frequency control unit 233, is determined by the determination unit 235 (S12). The process S12 for determining the frequency step size Δf will be described in detail in the column “(6.4) Resolution”. The frequency control unit 233 lowers the operating frequency f1 of the inverter circuit 21 by the frequency increment Δf determined in the process S12 (S13), and returns to the operation of the process S1. By repeating these processes (S11 to S13), the control circuit 23 can decrease the operating frequency f1 by the frequency increment Δf and gradually increase the output power, that is, approach the target value P1. The initial value of the operating frequency f1 is, for example, the upper limit value fmax of the permitted frequency band F1.

  When the operating frequency f1 becomes equal to or lower than the lower limit value fmin (S11: No), the control circuit 23 next adjusts the output power by the phase difference control unit 234. Specifically, the phase difference control unit 234 compares the phase difference θ with the lower limit value of the specified range (S14). Here, if the specified range is a range of 270 to 360 degrees, the initial value of the phase difference θ is 360 (degrees), and the lower limit value of the specified range is 270 (degrees). If the specified range is in the range of 90 degrees to 180 degrees, the initial value of the phase difference θ is 180 (degrees), and the lower limit value of the specified range is 90 (degrees). If the phase difference θ is equal to or greater than the lower limit value (S14: No), the phase difference control unit 234 controls the phase difference θ to be small. At this time, the phase difference step size Δθ, which is the step size of the control target value (phase difference θ) controlled by the phase difference control unit 234, is determined by the determination unit 235 (S15). The process S15 for determining the phase difference increment Δθ will be described in detail in the column “(6.4) Resolution”. The phase difference control unit 234 reduces the phase difference θ by the phase difference increment Δθ determined in step S15 (S16), and returns to the operation of step S1. By repeating these processes (S14 to S16), the control circuit 23 can decrease the phase difference θ by the phase difference increment Δθ and gradually increase the output power, that is, approach the target value P1.

  When the phase difference θ becomes less than the lower limit value in the phase difference control (S14: Yes), the control circuit 23 determines that an error has occurred (S17) and ends the output power control.

  If the output power exceeds the upper limit of the allowable error range of the target value P1 (S1: excess), the control circuit 23 first adjusts the output power by the phase difference control unit 234. Specifically, the phase difference control unit 234 compares the phase difference θ with the upper limit value of the specified range (S21). Here, if the specified range is in the range of 270 to 360 degrees, the upper limit value of the specified range is 360 (degrees), and if the specified range is in the range of 90 to 180 degrees, the upper limit value of the specified range is 180 (degrees). It becomes. If the phase difference θ is less than or equal to the upper limit value (S21: No), the phase difference control unit 234 controls the phase difference θ so as to increase. At this time, the phase difference step size Δθ, which is the step size of the control target value (phase difference θ) controlled by the phase difference control unit 234, is determined by the determination unit 235 (S22). The process S22 for determining the phase difference step width Δθ will be described in detail in the column “(6.4) Resolution”. The phase difference control unit 234 increases the phase difference θ by the phase difference increment Δθ determined in step S22 (S23), and returns to the operation of step S1. By repeating these processes (S21 to S23), the control circuit 23 can increase the phase difference θ by the phase difference increment Δθ and gradually decrease the output power, that is, approach the target value P1.

  When the phase difference θ exceeds the upper limit value (S21: Yes), the control circuit 23 next adjusts the output power by the frequency control unit 233. Specifically, the frequency control unit 233 compares the operating frequency (that is, the frequency of the first drive signals G1 to G4) f1 of the inverter circuit 21 with the upper limit value fmax of the permitted frequency band F1 (S24). If the operating frequency f1 is lower than the upper limit value fmax (S24: Yes), the frequency control unit 233 controls to increase the operating frequency f1 of the inverter circuit 21. At this time, a frequency step Δf that is a step of the control target value (operation frequency f1) controlled by the frequency control unit 233 is determined by the determination unit 235 (S25). The process S25 for determining the frequency step size Δf will be described in detail in the column “(6.4) Resolution”. The frequency control unit 233 increases the operating frequency f1 of the inverter circuit 21 by the frequency increment Δf determined in the process S25 (S26), and returns to the operation of the process S1. By repeating these processes (S24 to S26), the control circuit 23 can increase the operating frequency f1 by the frequency increment Δf and gradually decrease the output power, that is, approach the target value P1.

  When the operating frequency f1 becomes equal to or higher than the upper limit value fmax in frequency control (S24: No), the control circuit 23 determines that an error has occurred (S27) and ends the output power control.

  By the way, during the period in which the control circuit 23 adjusts the output power by the frequency control unit 233, that is, the period in which the frequency control is performed, all the correction switch elements Q5 to Q8 are fixed to be on for the power correction circuit 22. It is preferable to disable the function of the power correction circuit 22. As a result, the non-contact power feeding device 2 is in a state equivalent to “(4.1) No power correction circuit” (see the column “(4) Basic operation”). In this case, the control circuit 23 sets the phase difference θ to the initial value 360 (degrees) and starts the operation of the power correction circuit 22 before starting the adjustment of the output power by the phase difference control. According to this configuration, when the magnitude of the output power reaches the target value P1 only by the frequency control, the control circuit 23 does not operate the power correction circuit 22, and thus the efficiency (power conversion efficiency) by the power correction circuit 22 Can be avoided.

  Further, when the control circuit 23 adjusts the output power by phase difference control, the upper limit value of the specified range is set as an initial value, and the phase difference θ is gradually reduced from the initial value (the upper limit value of the specified range) This is not an essential configuration for the non-contact power feeding device 2. For example, when the specified range is a range of 270 degrees to 360 degrees, the control circuit 23 sets a value exceeding the upper limit value of the specified range (for example, 370 degrees) as an initial value, and gradually decreases the phase difference θ from the initial value. May be. Alternatively, when the specified range is a range of 270 degrees to 360 degrees, the control circuit 23 sets a value smaller than the upper limit value of the specified range (for example, 315 degrees) as an initial value and gradually increases the phase difference θ from the initial value. It may be small. In any case, in the region between the change start point existing within the specified range and the lower limit value of the specified range, the output power of the non-contact power feeding device 2 changes according to the change of the phase difference θ.

(6.4) Resolution If the step size (frequency step size Δf, phase difference step size Δθ) of the control target value that is the control target when adjusting the output power is constant, the following problems There is.

  As a first problem, when the output power of the non-contact power feeding device 2 is far from the target value P1, there is a possibility that the time required for the process for bringing the output power close to the target value P1 may be increased. That is, if the step size of the control target value is constant, even if the output power of the non-contact power feeding device 2 is far from the target value P1, the control target value is changed step by step, and the output power is gradually changed to the target value P1. Since it needs to be close to, the process may take time. In particular, when the non-contact power receiving device 3 charges the storage battery 4 after the output power reaches the target value P1, the time required to start charging the storage battery 4 becomes longer.

  As a second problem, when the step size of the control target value is set to a relatively large value in order to solve the first problem, ripples may occur due to fluctuations in the output power near the target value P1. The output power may significantly exceed the target value P1. In other words, if the step size of the control target value is a relatively large value, the time required for processing to bring the output power closer to the target value P1 can be reduced, but the occurrence of ripples and the target value P1 in the output power can be reduced. May lead to a significant excess.

  In short, when the step size of the control target value is relatively large, the amount of change in output power due to one frequency control or phase difference control also becomes relatively large. When the change amount of the output power at this time is larger than the width of the allowable error range (± several%) of the target value P1, the output power exceeds the upper limit of the allowable error range and falls below the lower limit of the allowable error range. The state may be repeated. As a result, the output power continues to fluctuate in the vicinity of the target value P1, and “ripple” may occur in the power received by the non-contact power receiving apparatus 3. In particular, when the non-contact power receiving device 3 applies a DC voltage to the storage battery 4, ripples are generated in the non-contact power receiving device 3 from the viewpoint of the influence on the charging control of the storage battery 4 and the life of the storage battery 4. It is desirable to suppress as much as possible.

  Similarly, if the step size of the control target value is relatively large, the amount of change in output power due to a single frequency control or phase difference control also becomes relatively large. Therefore, when the output power is increased, the output power fluctuates rapidly. Thus, the output power may greatly exceed the target value P1. In particular, when the non-contact power receiving device 3 applies a DC voltage to the storage battery 4, from the viewpoint of stress on the charging circuit of the storage battery 4, a large excess of the target value P <b> 1 in the output power is suppressed as much as possible. It is desirable.

  Therefore, in the non-contact power feeding device 2 according to the present embodiment, in order to solve these problems, the step size (frequency step size Δf, phase difference step size) of the control target value that is the control target when adjusting the output power. Δθ) is determined according to the resonance characteristics estimated by the estimation unit 232. That is, as described in the section “(6.3) Overall flow of output power control”, the frequency step width Δf and the phase difference step width Δθ are determined by the determination unit 235. For this reason, the frequency step width Δf and the phase difference step width Δθ are not constant and change appropriately. In other words, in the non-contact power feeding device 2, the resolution (resolution) of the control target value can be appropriately changed instead of being constant. The change in resolution can be realized by, for example, high resolution processing. If the resolution of the control target value increases, the step size (frequency step size Δf, phase difference step size Δθ) of the control target value decreases.

  Hereinafter, the process (S12, S15, S22, S25 in FIG. 9) for determining the step size (frequency step size Δf, phase difference step size Δθ) of the control target value will be described in detail.

  First, the estimation unit 232 estimates “resonance characteristics” representing the relationship between the operating frequency f1 (the frequency of the first drive signals G1 to G4) and the magnitude of the output power. The estimation unit 232 estimates the resonance characteristics in advance before the output power control starts. In the present embodiment, the estimation unit 232 gradually changes the phase difference θ, and between the primary side coil L1 and the secondary side coil L2 obtained from the change in the measurement value of the measurement unit 24 accompanying the change in the phase difference θ. The resonance characteristic is estimated based on the coupling coefficient. Thereby, in the estimation part 232, the resonance characteristic as shown, for example in FIG. 4A is estimated. The resonance characteristics estimated by the estimation unit 232 are held in the memory of the control circuit 23 (microcomputer). The procedure for estimating the resonance characteristics in the estimation unit 232 will be described in detail in the section “(8) Search mode”.

  Then, the determination unit 235 determines the step size of the control target value according to the resonance characteristics estimated by the estimation unit 232. As an example, when the control target value is the operating frequency f1, that is, when determining the frequency step size Δf, the determination unit 235 determines the step size (frequency step size Δf) of the control target value as follows. To do.

  That is, the determination unit 235 determines the frequency step width Δf according to the resonance characteristics so that the frequency step width Δf decreases as the change rate of the magnitude of the output power with respect to the change in frequency increases. For example, in the case of the resonance characteristics as shown in FIG. 4A, the rate of change in the magnitude of the output power with respect to the change in frequency is such that the closer to the resonance frequency fr0 (that is, the lower the frequency) in the permitted frequency band F1, It is getting bigger. Therefore, for example, when the frequency is gradually decreased from the upper limit value fmax of the permitted frequency band F1 in the resonance characteristics, the determination unit 235 changes the output power magnitude with respect to the change in frequency (that is, the slope of the graph). Is extracted as the switching frequency fc1. Then, the determination unit 235 determines the frequency step Δf according to the frequency so that the frequency step Δf is smaller in the frequency band lower than the switching frequency fc1 than in the frequency band higher than the switching frequency fc1. As a result, at the time of frequency control, the frequency step Δf is switched between at least the binary frequency step Δf with the switching frequency fc1 as a boundary.

  FIG. 10 shows an example in which the frequency step Δf is switched at the switching frequency fc1 during frequency control. In FIG. 10, the horizontal axis represents the frequency (operating frequency f1 of the inverter circuit 21), and the vertical axis represents the output power of the non-contact power feeding device 2, and represents the relationship between the operating frequency f1 and the output power during frequency control. That is, when gradually reducing the operating frequency f1, the frequency control unit 233 applies the first value α1 as the frequency step Δf up to the switching frequency fc1, and is smaller than the first value α1 below the switching frequency fc1. The second value α2 is applied as the frequency step Δf. In short, the frequency control unit 233 decreases the operating frequency f1 by the first value α1 for the points p1 to p8 in FIG. 10 and decreases the operating frequency f1 to the second value α2 (< Reduce by α1). As a result, in the frequency band Fb in which the change rate (ΔP / Δf) of the magnitude of the output power with respect to the change in the operating frequency f1 is greater than or equal to the threshold, the change rate (ΔP / Δf) is less than the threshold. The frequency step size Δf becomes small.

  For example, the control circuit 23 (microcomputer) has a PWM (Pulse Width Modulation) timer function, and a PWM signal having an operating frequency f1 according to a value (period register value) set in a period register of the PWM timer. Is output. At this time, if the step size of the period register value is “n / 255” (n is an integer), by adjusting “n” in the range of “1” to “255”, the step size of the operating frequency f1 (frequency The step size Δf) varies. As an example, it is assumed that the CPU clock frequency of the control circuit 23 (microcomputer) is 80 [MHz] and the initial value of the operating frequency f1 is 90 [kHz]. In this case, if the period register value changes from “444” in increments of “1” (= 255/255), the operation frequency f1 becomes 89.7976 [kHz] when the period register value becomes “445”. At this time, the frequency step width Δf is 202.4 [Hz], and the resolution becomes the lowest. On the other hand, if the period register value changes in increments of “1/255” from “444”, the operating frequency f1 becomes 89.9992 [kHz] when the period register value becomes “444 + 1/255”. At this time, the frequency step width Δf is 0.8 [Hz], and the resolution becomes the highest. That is, the frequency step Δf (0.8 [Hz]) when the resolution is the highest is 1/255 of the frequency step Δf (202.4 [Hz]) when the resolution is the lowest.

  On the other hand, when the control target value is the phase difference θ, that is, when the phase difference step size Δθ is determined, the determination unit 235, as an example, determines the control target value step size (phase difference step size) as follows. Δθ) is determined.

  That is, the determination unit 235 resonates the phase difference step width Δθ so that the phase difference step width Δθ decreases as the difference value between the maximum value of the output power that can be adjusted only by frequency control and the target value P1 decreases. Determine according to characteristics. For example, if the resonance characteristic is as shown in FIG. 4A, the maximum value of the output power that can be adjusted by frequency control within the permitted frequency band F1 is insufficient with respect to the target value P1. Therefore, for example, the determination unit 235 obtains the maximum value of the output power when the frequency is reduced to the lower limit value fmin of the permitted frequency band F1 in the resonance characteristics, and the difference value between the maximum value and the target value P1, that is, The shortage of the maximum value with respect to the target value P1 is extracted. Then, the determination unit 235 switches the binary phase difference increment Δθ in accordance with the comparison result between the difference value and the threshold value. For example, if the difference value is less than the threshold value, the phase difference step size Δθ is switched depending on the comparison result between the difference value and the threshold value so that the phase difference step size Δθ is reduced. Thereby, the phase difference increment Δθ at the time of phase difference control is determined according to the resonance characteristics.

  As an example, when the difference value is equal to or greater than the threshold value, the determination unit 235 sets the phase difference increment Δθ to 0.4054 [degrees] and sets the resolution to the minimum. In this case, the phase difference θ changes in increments of 0.4054 [degrees]. On the other hand, when the difference value is less than the threshold value, the determination unit 235 sets the phase difference increment Δθ to 0.0016 [degrees] and maximizes the resolution. In this case, the phase difference θ changes in increments of 0.0016 [degrees]. That is, the phase difference increment Δθ (0.0016 [degrees]) when the resolution is the highest is 1/255 of the phase difference increment Δθ (0.4054 [degrees]) when the resolution is the lowest.

(7) Start-up process The contactless power supply device 2 of the present embodiment soft-starts the inverter circuit 21 as described below at the time of start-up when the inverter circuit 21 and the power correction circuit 22 start operation.

  When the inverter circuit 21 is activated, the control circuit 23 sets the duty ratio of the first drive signals G1 to G4 for controlling the conversion switch elements Q1 to Q4 from 0 (zero) to a predetermined value (for example, 0.5). By gradually increasing, the soft start of the inverter circuit 21 is realized. Thereby, the sudden change of the voltage and electric current input into the non-contact electric power feeder 2 is suppressed, and the stress added to a circuit element can be reduced. Hereinafter, the process in which the control circuit 23 soft-starts the inverter circuit 21 by changing the duty ratios of the first drive signals G1 to G4 as described above is referred to as “start-up process”.

  In the contactless power supply device 2 of the present embodiment, all the correction switch elements Q5 to Q8 are fixed to be on with respect to the power correction circuit 22 while the control circuit 23 performs the starting process, and the function of the power correction circuit 22 is maintained. Disable. As a result, the non-contact power feeding device 2 is in a state equivalent to “(4.1) No power correction circuit” (see the column “(4) Basic operation”).

  When the start-up process is completed, that is, when the duty ratio of the first drive signals G1 to G4 reaches a predetermined value (for example, 0.5), the control circuit 23 also starts the operation of the power correction circuit 22. In short, the control circuit 23 starts control of the correction switch elements Q5 to Q8 with the second drive signals G5 to G8 after the start-up process is completed. As a result, the non-contact power feeding device 2 is in a state equivalent to “(4.2) With power correction circuit” (see the column “(4) Basic operation”). At this time, if the specified range is a range of 270 degrees to 360 degrees, the control circuit 23 sets the phase difference θ to the initial value 360 (degrees) and starts the operation of the power correction circuit 22.

(8) Search Mode In the present embodiment, the control circuit 23 is arranged between the primary side coil L1 and the secondary side coil L2 in addition to the normal mode (including the startup process) in which the output power control as described above is performed. Has a search mode for estimating the coupling coefficient. In the search mode, the control circuit 23 gradually changes the phase difference θ within a specified range, and estimates the coupling coefficient based on the change in the measurement value (measurement value of the measurement unit 24) accompanying the change in the phase difference θ. Is configured to do. The measured value here is the magnitude of a current (hereinafter also referred to as “coil current”) flowing through the primary coil L1, and is measured by the measuring unit 24. Hereinafter, the search mode will be described in detail.

  As shown in FIGS. 11A and 11B, the relationship between the phase difference θ of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4 and the coil current is the relationship between the primary side coil L1 and the secondary side coil L2. It has been confirmed that it varies depending on the coupling coefficient. 11A and 11B, the horizontal axis represents the phase difference θ between the first drive signals G1 to G4 and the second drive signals G5 to G8, and the vertical axis represents the coil current (current flowing through the primary coil L1). 11A and 11B, “Y1” to “Y5” represent the relationship between the phase difference θ and the coil current when the coupling coefficients are different. FIG. 11B is an enlarged view of region A1 in FIG. 11A.

  Here, “Y1” to “Y5” in FIGS. 11A and 11B become Y5, Y4, Y3, Y2, and Y1 when arranged in order from the side with the largest coupling coefficient. As is apparent from FIG. 11B, when the phase difference θ is gradually reduced from the upper limit value (360 degrees in this case) of the specified range, the phase difference θ corresponding to the inflection point at which the coil current starts to increase is the coupling coefficient. It depends on. As the coupling coefficient between the primary coil L1 and the secondary coil L2 increases, the phase difference θ at which the coil current starts to increase decreases. Therefore, if the relationship between the phase difference θ and the coil current as shown in FIGS. 11A and 11B is used, the control circuit 23 can detect the coupling from the change in the measured value (the magnitude of the coil current) accompanying the change in the phase difference θ. It is possible to estimate the coefficients.

  Specifically, the control circuit 23 records the relationship between the phase difference θ and the coil current as shown in FIGS. 11A and 11B in advance as characteristic data, for example, in a table format in the memory of the microcomputer. The coupling coefficient can be estimated using the characteristic data. Since such characteristic data varies depending on, for example, the vehicle type when the load is an electric vehicle, it is preferable that characteristic data of a plurality of vehicle types is recorded. In addition, the non-contact power feeding device 2 may acquire characteristic data from the non-contact power receiving device 3 by performing communication with the non-contact power receiving device 3.

  From the coupling coefficient, the control circuit 23 determines the resonance characteristics (that is, the operating frequency of the inverter circuit 21 and the output of the non-contact power feeding device 2) as described in the column “(1) With power correction circuit” of “Basic operation”. The relationship with power) can be further estimated. As a result, the control circuit 23 can estimate the frequency range in which the inverter circuit 21 operates in the slow phase mode (that is, does not enter the fast phase mode), for example, for the operating frequency f1 of the inverter circuit 21. Therefore, the control circuit 23 preferably performs the above-described operation in the search mode before starting the operation in the normal mode. Thereby, the control circuit 23 can set the initial value of the operating frequency f1 when starting the operation in the normal mode within the frequency range in which the inverter circuit 21 operates in the slow phase mode. In this case, the lower limit value of the operating frequency f1 in the frequency control described above is the larger of the lower limit value of the frequency range in which the inverter circuit 21 operates in the slow phase mode and the lower limit value fmin of the permitted frequency band F1. .

  Hereinafter, the overall flow of the “search mode” of the present embodiment will be described with reference to the flowchart of FIG. The process shown in FIG. 12 is performed before process S1 in the flowchart of FIG. 9 and before the startup process.

  When the search mode starts, the control circuit 23 first sets the phase difference θ to an initial value (here, 360 degrees) (S31). Next, the control circuit 23 decrements the phase difference θ (θ-1) (S32), and acquires the measured value of the coil current from the measuring unit 24 (S33). Next, the control circuit 23 obtains the difference between the latest measured value of the coil current and the previous measured value of the coil current as an increase amount of the coil current, and compares the increase amount of the coil current with a threshold value (S34). ). At this time, if the increase amount of the coil current is less than the threshold (S34: No), the operation of the control circuit 23 returns to the process S32.

  On the other hand, if the increase amount of the coil current is equal to or greater than the threshold value (S34: Yes), the control circuit 23 compares the current phase difference θ with a characteristic table (table-type characteristic data) (S35). As a result, the control circuit 23 can estimate the coupling coefficient, and can also estimate the coupling coefficient to the resonance characteristic. Next, the control circuit 23 returns the phase difference θ to the initial value (360 degrees here) (S36), and ends the search mode.

(9) Modifications The above embodiment is merely an example of the present invention, and the present invention is not limited to the above embodiment, and deviates from the technical idea according to the present invention even if it is other than the above embodiment. Various changes can be made in accordance with the design or the like as long as they are not. Below, the modification of the said embodiment is enumerated.

  The power correction circuit 22 is not limited to the configuration using the four correction switch elements Q5 to Q8 as in the above-described embodiment, but includes, for example, two correction switch elements Q9 and Q10 as shown in FIG. May be. In the power correction circuit 22 shown in FIG. 13, each of the correction switch elements Q9 and Q10 is a double-gate semiconductor switch element having two gates. The first correction switch element Q9 is electrically connected in series with the correction capacitor C1. The second correction switch element Q10 is electrically connected in parallel to the series circuit of the correction switch element Q9 and the correction capacitor C1. The second drive signal G7 and the second drive signal G8 are input to the two gates of the correction switch element Q9, respectively. The second drive signal G5 and the second drive signal G6 are input to the two gates of the correction switch element Q10, respectively. The power correction circuit 22 shown in FIG. 13 has the two correction switch elements Q9 and Q10 controlled by the second drive signals G5 to G8, and functions equivalently to the power correction circuit 22 shown in FIG.

  Moreover, the load to which output power is supplied in a non-contact manner from the non-contact power supply device 2 (that is, the power is supplied) is not limited to an electric vehicle, and includes, for example, an electric device including a storage battery such as a mobile phone or a smartphone, or a storage battery. There may be no electrical equipment such as lighting fixtures.

  Further, the transmission method of the output power from the non-contact power supply device 2 to the non-contact power reception device 3 is not limited to the magnetic field resonance method described above, and may be, for example, an electromagnetic induction method or a microwave transmission method.

  Each of the conversion switch elements Q1 to Q4 and each of the correction switch elements Q5 to Q8 may be composed of other semiconductor switching elements such as a bipolar transistor or an IGBT (Insulated Gate Bipolar Transistor).

  The diodes D1 to D4 are not limited to the parasitic diodes of the conversion switch elements Q1 to Q4, and may be externally attached to the conversion switch elements Q1 to Q4. Similarly, the diodes D5 to D8 are not limited to the parasitic diodes of the correction switch elements Q5 to Q8, and may be externally attached to the correction switch elements Q5 to Q8.

  The measurement unit 24 is not limited to the configuration provided separately from the control circuit 23, and may be provided integrally with the control circuit 23. Furthermore, since the measurement unit 24 only needs to measure the magnitude of the current flowing through the primary coil L1, the current sensor 25 is not limited to between the primary coil L1 and the second primary capacitor C12, but the primary coil L1. On the path of the current flowing through

  The inverter circuit 21 may be a voltage-type inverter that can convert a DC voltage into an AC voltage and output it, and is not limited to a full-bridge inverter circuit in which four conversion switch elements Q1 to Q4 are connected in a full bridge. The inverter circuit 21 may be a half-bridge inverter circuit, for example.

  Further, the power correction circuit 22 may be operating during a period in which the control circuit 23 adjusts the output power by the frequency control unit 233, that is, a period in which frequency control is performed. In this case, the phase difference θ is set to an initial value of 360 (degrees). In this case, the frequency control unit 233 changes the frequencies of the second drive signals G5 to G8 together with the frequencies of the first drive signals G1 to G4.

  Further, the power correction circuit 22 only needs to be electrically connected between at least one of the pair of output points 213 and 214 and the primary coil L1, and is replaced between the output point 213 and the primary coil L1. In addition, the output point 214 and the primary coil L1 may be electrically connected.

  In addition, the functions of the element control unit 231, the estimation unit 232, the frequency control unit 233, the phase difference control unit 234, and the determination unit 235 are not limited to be implemented in the control circuit 23, and at least some of these functions are It may be provided separately from the control circuit 23.

  Further, the determination unit 235 is configured to determine the step size for only one of the operating frequency f1 and the phase difference θ, which is a control target value that is a control target when adjusting the output power. Also good. That is, the determination unit 235 estimates the step size of the control target value including at least one of the frequency controlled by the frequency control unit 233 and the phase difference θ controlled by the phase difference control unit 234 by the estimation unit 232. What is necessary is just to determine according to a resonance characteristic. For example, when the determining unit 235 determines only the step size (frequency step size Δf) for the operating frequency f1, the step size (phase difference step size Δθ) for the phase difference θ is a fixed value.

  Further, the determination unit 235 is not limited to the configuration in which the step size is switched between the binary step sizes, and may be switched between the step sizes of three or more values. Furthermore, for example, the determination unit 235 may use a value obtained by multiplying a feature value extracted from the resonance characteristics by a predetermined coefficient as the step size. In this case, the step size is adjusted steplessly, that is, continuously.

  Furthermore, when determining the step width (phase difference step width Δθ) for the phase difference θ, the determination unit 235 may directly determine (designate) the phase difference step width Δθ, or indirectly. The phase difference step width Δθ may be determined. That is, the phase difference θ is a phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4. Therefore, for example, if any one of the frequency (operation frequency f1) of the first drive signals G1 to G4 and the delay time of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4 changes, the phase difference The step size Δθ changes. Therefore, the determination unit 235 may directly change the phase difference increment Δθ, but is not limited thereto, and for example, changes the delay time of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4. Thus, the phase difference increment Δθ may be indirectly changed.

(10) Summary As described above, the contactless power supply device 2 includes the inverter circuit 21, the primary coil L1, the power correction circuit 22, the element control unit 231, the estimation unit 232, and the frequency control unit 233. The phase difference control unit 234 and the determination unit 235 are provided. The inverter circuit 21 includes a plurality of conversion switch elements Q1 to Q4 that are electrically connected between the pair of input points 211 and 212 and the pair of output points 213 and 214. The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 to an AC voltage by switching of the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214. The primary side coil L1 is electrically connected between a pair of output points 213 and 214, and supplies output power to the secondary side coil L2 in a non-contact manner by applying an AC voltage. The power correction circuit 22 is electrically connected between at least one of the pair of output points 213 and 214 and the primary coil L1, and includes a correction capacitor C1 and a plurality of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8. The estimation unit 232 estimates a resonance characteristic that represents the relationship between the frequency (the operating frequency f1) of the first drive signals G1 to G4 and the magnitude of the output power. The frequency control unit 233 adjusts the magnitude of the output power by controlling the frequencies of the first drive signals G1 to G4 to change discretely. The phase difference control unit 234 controls the magnitude of the output power by controlling the phase difference θ, which is the phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to discretely change. Adjust. The determination unit 235 uses the estimation unit 232 to estimate the step size of the control target value including at least one of the frequency controlled by the frequency control unit 233 and the phase difference θ controlled by the phase difference control unit 234. To be decided.

  According to this configuration, the contactless power supply device 2 adjusts the magnitude of the output power by at least one of the frequency control by the frequency control unit 233 and the phase difference θ control by the phase difference control unit 234. Is possible. Therefore, even if the relative positional relationship between the primary side coil L1 and the secondary side coil L2 changes and the coupling coefficient between the primary side coil L1 and the secondary side coil L2 changes, the non-contact power feeding device Second, by adjusting the magnitude of the output power, it becomes possible to supply stable power. Moreover, in the non-contact power feeding device 2, the resonance characteristic in which the step size of the control target value (at least one of the frequency and the phase difference θ) that is the control target when adjusting the output power is estimated by the estimation unit 232. It is decided according to. Therefore, for example, when the output power of the non-contact power feeding device 2 is far from the target value P1 determined by the non-contact power receiving device 3, the output power is set to the target value by relatively increasing the step size of the control target value. It is possible to reduce the time required for the processing for approaching P1. When the output power of the non-contact power feeding device 2 approaches the target value P1, the generation of ripples due to fluctuations in the output power near the target value P1, by making the step size of the control target value relatively small, It is possible to suppress the output power from significantly exceeding the target value P1. Therefore, the non-contact power feeding device 2 has an advantage that stable power can be supplied regardless of the relative positional relationship between the primary side coil L1 and the secondary side coil L2.

  Further, the determination unit 235 sets the frequency step size controlled by the frequency control unit 233 to the resonance characteristic so that the step size of the frequency becomes smaller as the rate of change of the output power with respect to the frequency change becomes larger. It is preferable to be configured to determine accordingly. According to this configuration, when the rate of change in the magnitude of the output power with respect to the change in frequency is relatively small, the frequency step Δf is relatively large, and the time required for processing to bring the output power close to the target value P1 Can be shortened. On the other hand, when the rate of change in the magnitude of the output power with respect to the change in frequency is relatively large, the frequency step size Δf becomes relatively small, and ripples are generated and the output power greatly exceeds the target value P1. It can be suppressed. However, this configuration is not an essential configuration for the non-contact power feeding device 2. For example, the determination unit 235 may reduce the frequency step size as the change rate of the output power with respect to the frequency change decreases. Good.

  The phase difference control unit 234 controls the phase difference θ when the magnitude of the output power adjusted by the frequency control unit 233 is less than a predetermined target value P1, thereby reducing the magnitude of the output power. It is preferably configured to approach the target value P1. According to this configuration, the non-contact power feeding device 2 can adjust the output power in two stages of frequency control and phase difference control, so that the adjustment range of the output power can be widened. However, this configuration is not an essential configuration for the non-contact power feeding device 2. For example, the phase difference control unit 234 controls the phase difference θ before the frequency control unit 233 adjusts the magnitude of the output power. The magnitude of the output power may be adjusted.

  Moreover, it is preferable that the non-contact electric power feeder 2 is further provided with the measurement part 24 which measures the magnitude | size of the electric current which flows into the primary side coil L1 as a measured value. In this case, the estimation unit 232 gradually changes the phase difference θ, and based on the coupling coefficient between the primary side coil L1 and the secondary side coil L2 obtained from the change in the measured value accompanying the change in the phase difference θ. It is preferably configured to estimate resonance characteristics. According to this configuration, even if the coupling coefficient between the primary side coil L1 and the secondary side coil L2 is unknown, the estimation unit 232 can estimate the resonance characteristics from the measurement value of the measurement unit 24. . However, this configuration is not an essential configuration for the contactless power supply device 2, and the measurement unit 24 may be omitted.

  The non-contact power transmission system 1 includes a non-contact power feeding device 2 and a non-contact power receiving device 3 having a secondary coil L2. The non-contact power receiving device 3 is configured so that output power is supplied from the non-contact power feeding device 2 in a non-contact manner. According to this configuration, there is an advantage that stable power supply is possible regardless of the relative positional relationship between the primary side coil L1 and the secondary side coil L2.

  Further, when the control circuit 23 has a microcomputer as a main configuration as in the present embodiment, the program recorded in the memory of the microcomputer causes the control circuit 23 to function as the control circuit 23. That is, the program is a program for causing a computer used for the non-contact power supply apparatus 2 to function as the element control unit 231, the estimation unit 232, the frequency control unit 233, the phase difference control unit 234, and the determination unit 235. The non-contact power feeding device 2 here includes an inverter circuit 21, a primary side coil L <b> 1, and a power correction circuit 22.

  The inverter circuit 21 includes a plurality of conversion switch elements Q1 to Q4 that are electrically connected between the pair of input points 211 and 212 and the pair of output points 213 and 214. The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 to an AC voltage by switching of the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214. The primary side coil L1 is electrically connected between a pair of output points 213 and 214, and supplies output power to the secondary side coil L2 in a non-contact manner by applying an AC voltage. The power correction circuit 22 is electrically connected between at least one of the pair of output points 213 and 214 and the primary coil L1, and includes a correction capacitor C1 and a plurality of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8. The estimation unit 232 estimates a resonance characteristic that represents the relationship between the frequency (the operating frequency f1) of the first drive signals G1 to G4 and the magnitude of the output power. The frequency control unit 233 adjusts the magnitude of the output power by controlling the frequencies of the first drive signals G1 to G4 to change discretely. The phase difference control unit 234 controls the magnitude of the output power by controlling the phase difference θ, which is the phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to discretely change. Adjust. The determination unit 235 uses the estimation unit 232 to estimate the step size of the control target value including at least one of the frequency controlled by the frequency control unit 233 and the phase difference θ controlled by the phase difference control unit 234. To be decided. According to this program, even if the dedicated control circuit 23 is not used, a function equivalent to that of the non-contact power feeding device 2 can be realized, and depending on the relative positional relationship between the primary side coil L1 and the secondary side coil L2. Therefore, there is an advantage that stable power supply is possible.

  Further, the non-contact power feeding device 2 including the inverter circuit 21, the primary coil L1, and the power correction circuit 22 is controlled by the following control method, so that the above non-contact power supply device 2 can be used without using the dedicated control circuit 23. A function equivalent to that of the contact power supply device 2 can be realized. That is, the control method of the non-contact power feeding device 2 includes an element control step, an estimation step, a frequency control step, a phase difference control step, and a determination step. The non-contact power feeding device 2 here includes an inverter circuit 21, a primary side coil L <b> 1, and a power correction circuit 22.

  The inverter circuit 21 includes a plurality of conversion switch elements Q1 to Q4 that are electrically connected between the pair of input points 211 and 212 and the pair of output points 213 and 214. The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 to an AC voltage by switching of the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214. The primary side coil L1 is electrically connected between a pair of output points 213 and 214, and supplies output power to the secondary side coil L2 in a non-contact manner by applying an AC voltage. The power correction circuit 22 is electrically connected between at least one of the pair of output points 213 and 214 and the primary coil L1, and includes a correction capacitor C1 and a plurality of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8. In the estimation step, a resonance characteristic representing a relationship between the frequency of the first drive signals G1 to G4 (operation frequency f1) and the magnitude of the output power is estimated. In the frequency control step, the magnitude of the output power is adjusted by controlling the frequencies of the first drive signals G1 to G4 to be discretely changed. In the phase difference control step, the magnitude of the output power is adjusted by controlling the phase difference θ, which is the phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to be discretely changed. To do. In the determination step, the step size of the control target value consisting of at least one of the frequency controlled in the frequency control step and the phase difference θ controlled in the phase difference control step is determined according to the resonance characteristics estimated in the estimation step. To do. According to this control method, there is an advantage that stable power supply is possible regardless of the relative positional relationship between the primary side coil L1 and the secondary side coil L2.

DESCRIPTION OF SYMBOLS 1 Non-contact electric power transmission system 2 Non-contact electric power feeder 3 Non-contact electric power receiving apparatus 21 Inverter circuit 211,212 A pair of input point 213,214 A pair of output point 22 Power correction circuit 231 Element control part 232 Estimation part 233 Frequency control part 234 rank Phase difference control unit 235 Determination unit 24 Measurement unit C1 Correction capacitor G1 to G4 First drive signal G5 to G8 Second drive signal L1 Primary side coil L2 Secondary side coil Q1 to Q4 Conversion switch element Q5 to Q8 Correction switch element Q9, Q10 Correction switch element f1 (operation) frequency θ phase difference Δf (frequency) step width Δθ (phase difference) step width

Claims (7)

A plurality of conversion switch elements electrically connected between a pair of input points and a pair of output points, and a direct current applied to the pair of input points by switching of the plurality of conversion switch elements An inverter circuit that converts the voltage into an alternating voltage and outputs the voltage from the pair of output points;
A primary coil that is electrically connected between the pair of output points and supplies output power to the secondary coil in a non-contact manner by applying the AC voltage;
A correction capacitor and a plurality of correction switch elements are electrically connected between at least one of the pair of output points and the primary coil, and the correction is performed by switching the plurality of correction switch elements. A power correction circuit that charges and discharges a capacitor for use,
An element control unit that controls the plurality of conversion switch elements with a first drive signal and controls the plurality of correction switch elements with a second drive signal;
An estimation unit that estimates a resonance characteristic representing a relationship between the frequency of the first drive signal and the magnitude of the output power;
A frequency controller that adjusts the magnitude of the output power by controlling the frequency of the first drive signal so as to change discretely;
A phase difference control unit that adjusts the magnitude of the output power by controlling to discretely change a phase difference that is a phase delay of the second drive signal with respect to the first drive signal;
According to the resonance characteristic estimated by the estimation unit, a step size of the control target value including at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit. A non-contact power feeding device comprising: a determination unit that determines the determination. The determination unit
The step size of the frequency controlled by the frequency control unit is set as the resonance characteristic so that the step size of the frequency becomes smaller as the change rate of the magnitude of the output power with respect to the change of the frequency becomes larger. It is comprised so that it may determine according to these. The non-contact electric power feeder of Claim 1 characterized by the above-mentioned. The phase difference control unit
When the magnitude of the output power adjusted by the frequency control unit is less than a predetermined target value, the magnitude of the output power is made closer to the target value by controlling the phase difference The contactless power feeding device according to claim 1, wherein the contactless power feeding device is provided. A measurement unit that measures the magnitude of the current flowing through the primary coil as a measurement value;
The estimation unit includes
The resonance characteristic is estimated based on a coupling coefficient between the primary side coil and the secondary side coil obtained from a change in the measurement value accompanying a change in the phase difference by gradually changing the phase difference. It is comprised by these. The non-contact electric power feeder of any one of Claims 1-3 characterized by the above-mentioned. A plurality of conversion switch elements electrically connected between a pair of input points and a pair of output points, and a direct current applied to the pair of input points by switching of the plurality of conversion switch elements An inverter circuit that converts the voltage into an alternating voltage and outputs the voltage from the pair of output points;
A primary coil that is electrically connected between the pair of output points and supplies output power to the secondary coil in a non-contact manner by applying the AC voltage;
A correction capacitor and a plurality of correction switch elements are electrically connected between at least one of the pair of output points and the primary coil, and the correction is performed by switching the plurality of correction switch elements. A computer used in a non-contact power feeding device including a power correction circuit that charges and discharges a capacitor
An element control unit that controls the plurality of conversion switch elements with a first drive signal and controls the plurality of correction switch elements with a second drive signal;
An estimation unit that estimates a resonance characteristic representing a relationship between the frequency of the first drive signal and the magnitude of the output power;
A frequency control unit for adjusting the magnitude of the output power by controlling the frequency of the first drive signal so as to change discretely;
A phase difference control unit that adjusts the magnitude of the output power by controlling the phase difference, which is a phase delay of the second drive signal with respect to the first drive signal, to discretely change;
And the step size of the control target value consisting of at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit is set as the resonance characteristic estimated by the estimation unit. A decision unit to decide accordingly,
Program to function as. A plurality of conversion switch elements electrically connected between a pair of input points and a pair of output points, and a direct current applied to the pair of input points by switching of the plurality of conversion switch elements An inverter circuit that converts the voltage into an alternating voltage and outputs the voltage from the pair of output points;
A primary coil that is electrically connected between the pair of output points and supplies output power to the secondary coil in a non-contact manner by applying the AC voltage;
A correction capacitor and a plurality of correction switch elements are electrically connected between at least one of the pair of output points and the primary coil, and the correction is performed by switching the plurality of correction switch elements. A method for controlling a non-contact power feeding device including a power correction circuit that charges and discharges a capacitor for a power supply,
An element control step of controlling the plurality of conversion switch elements with a first drive signal and controlling the plurality of correction switch elements with a second drive signal;
An estimation step for estimating a resonance characteristic representing a relationship between the frequency of the first drive signal and the magnitude of the output power;
A frequency control step of adjusting the magnitude of the output power by controlling the frequency of the first drive signal so as to change discretely;
A phase difference control step of adjusting the magnitude of the output power by controlling the phase difference, which is a phase delay of the second drive signal with respect to the first drive signal, to discretely change;
The step size of the control target value consisting of at least one of the frequency controlled in the frequency control step and the phase difference controlled in the phase difference control step is determined according to the resonance characteristic estimated in the estimation step. A decision step to decide;
A control method for a non-contact power feeding apparatus, comprising: A non-contact power feeding device according to any one of claims 1 to 4 and a non-contact power receiving device having the secondary side coil,
The non-contact power receiving apparatus is configured to be supplied with the output power in a non-contact manner from the non-contact power supply apparatus.

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