JP6675093B2 - Non-contact power supply device, program, non-contact power supply device control method, and non-contact power transmission system - Google Patents

Description

  The present invention generally relates to a non-contact power supply device, a program, a control method of a non-contact power supply device, and a non-contact power transmission system, and more particularly, to a non-contact power supply device that supplies power to a load in a non-contact manner, a program, and a non-contact power supply. The present invention relates to a device control method and a wireless power transmission system.

  2. Description of the Related Art Conventionally, a non-contact power supply device that supplies power to a load in a non-contact manner using electromagnetic induction has been proposed (for example, see Patent Document 1).

  The non-contact power supply device described in Patent Literature 1 includes a primary side coil (power supply coil) that supplies electric power by generating a magnetic field, and is used for power supply to a moving body such as an electric vehicle. Electric vehicles include a non-contact power receiving device. The non-contact power receiving device includes a secondary coil (power receiving coil) and a storage battery, and stores power supplied from the primary coil of the non-contact power feeding device to the secondary coil in the storage battery.

JP 2013-243929 A

  By the way, in the non-contact power feeding device as described above, the primary coil and the secondary coil are connected depending on the relative positional relationship between the primary coil of the non-contact power feeding device and the secondary coil of the load (moving body). The coupling coefficient changes between. Therefore, depending on the relative positional relationship between the primary coil and the secondary coil, the output power output from the non-contact power supply device fluctuates, and stable power supply may not be performed.

  The present invention has been made in view of the above circumstances, and has a non-contact power supply device, a program, and a non-contact power supply device capable of supplying stable power irrespective of a relative positional relationship between a primary coil and a secondary coil. And a non-contact power transmission system.

  The contactless power supply device according to one embodiment of the present invention includes an inverter circuit, a primary coil, a power correction circuit, an element control unit, a frequency control unit, a phase difference control unit, an acquisition unit, and a determination unit. , Is provided. The inverter circuit has a plurality of conversion switch elements electrically connected between a pair of input points and a pair of output points. The inverter circuit converts a DC voltage applied to the pair of input points into an AC voltage and outputs the AC voltage from the pair of output points by switching of the plurality of conversion switch elements. The primary side coil is electrically connected between the pair of output points, and supplies the secondary side coil in a non-contact manner by applying the AC voltage. The power correction circuit is electrically connected between at least one of the pair of output points and the primary coil, and includes a correction capacitor and a plurality of correction switch elements. The power correction circuit charges and discharges the correction capacitor by switching the plurality of correction switch elements. The element control unit controls the plurality of conversion switch elements using a first drive signal, and controls the plurality of correction switch elements using a second drive signal. The frequency controller adjusts the magnitude of the output power by controlling the frequency of the first drive signal to be discretely changed. The phase difference control unit adjusts the magnitude of the output power by controlling so as to discretely change a phase difference that is a delay of a phase of the second drive signal with respect to the first drive signal. The acquisition unit acquires a phase difference between an output voltage and an output current of the inverter circuit as a voltage-current phase difference. The determining unit determines a step width of a control target value including at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit according to the voltage-current phase difference. decide.

  A program according to one embodiment of the present invention causes a computer used for a wireless power supply device to function as an element control unit, a frequency control unit, a phase difference control unit, an acquisition unit, and a determination unit. The non-contact power supply device includes an inverter circuit, a primary coil, and a power correction circuit. The inverter circuit has a plurality of conversion switch elements electrically connected between a pair of input points and a pair of output points. The inverter circuit converts a DC voltage applied to the pair of input points into an AC voltage and outputs the AC voltage from the pair of output points by switching of the plurality of conversion switch elements. The primary side coil is electrically connected between the pair of output points, and supplies the secondary side coil in a non-contact manner by applying the AC voltage. The power correction circuit is electrically connected between at least one of the pair of output points and the primary coil. The power correction circuit has a correction capacitor and a plurality of correction switch elements. The power correction circuit charges and discharges the correction capacitor by switching the plurality of correction switch elements. The element control unit controls the plurality of conversion switch elements using a first drive signal, and controls the plurality of correction switch elements using a second drive signal. The frequency controller adjusts the magnitude of the output power by controlling the frequency of the first drive signal to be discretely changed. The phase difference control unit adjusts the magnitude of the output power by controlling so as to discretely change a phase difference that is a delay of a phase of the second drive signal with respect to the first drive signal. The acquisition unit acquires a phase difference between an output voltage and an output current of the inverter circuit as a voltage-current phase difference. The determining unit determines a step width of a control target value including at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit according to the voltage-current phase difference. decide.

  A control method for a wireless power supply device according to one embodiment of the present invention includes an element control step, a frequency control step, a phase difference control step, an acquisition step, and a determination step. The non-contact power supply device includes an inverter circuit, a primary coil, and a power correction circuit. The inverter circuit has a plurality of conversion switch elements electrically connected between a pair of input points and a pair of output points. The inverter circuit converts a DC voltage applied to the pair of input points into an AC voltage and outputs the AC voltage from the pair of output points by switching of the plurality of conversion switch elements. The primary side coil is electrically connected between the pair of output points, and supplies the secondary side coil in a non-contact manner by applying the AC voltage. The power correction circuit is electrically connected between at least one of the pair of output points and the primary coil. The power correction circuit has a correction capacitor and a plurality of correction switch elements. The power correction circuit charges and discharges the correction capacitor by switching the plurality of correction switch elements. In the element control step, the plurality of conversion switch elements are controlled by a first drive signal, and the plurality of correction switch elements are controlled by a second drive signal. In the frequency control step, the magnitude of the output power is adjusted by controlling the frequency of the first drive signal to change discretely. In the phase difference control step, the magnitude of the output power is adjusted by controlling so as to discretely change a phase difference which is a delay of a phase of the second drive signal with respect to the first drive signal. In the obtaining step, a phase difference between an output voltage and an output current of the inverter circuit is obtained as a voltage-current phase difference. In the determining step, the step width of a control target value consisting of at least one of the frequency controlled in the frequency control step and the phase difference controlled in the phase difference control step is determined according to the voltage / current phase difference. decide.

  A non-contact power transmission system according to one aspect of the present invention includes the non-contact power supply device described above and a non-contact power receiving device having the secondary coil. The non-contact power receiving device is configured to be supplied with the output power from the non-contact power supply device in a non-contact manner.

  The present invention has an advantage that stable power supply is possible regardless of the relative positional relationship between the primary coil and the secondary coil.

FIG. 1 is a circuit diagram showing a wireless power transmission system according to one embodiment of the present invention. FIG. 2 is a waveform diagram illustrating a first drive signal and a second drive signal of the wireless power supply device according to the embodiment of the present invention. FIG. 3 is a graph showing an example of a resonance characteristic of the wireless power supply device. FIGS. 4A and 4B are graphs showing examples of resonance characteristics in the wireless power feeding device of the above. 5A is a graph showing an example of a phase difference characteristic in the case of the initial phase advance in the wireless power supply device of the above, and FIG. 5B is a graph showing an example of the phase difference characteristic of the initial phase delay in the wireless power supply device of the above. is there. FIG. 6A is an explanatory diagram illustrating a first charging mode of a power correction circuit in the wireless power feeding device according to the first embodiment. FIG. 6B is an explanatory diagram illustrating a first discharge mode of the power correction circuit in the wireless power supply device. FIG. 6C is an explanatory diagram showing a second charging mode of the power correction circuit in the wireless power supply device. FIG. 6D is an explanatory diagram illustrating a second discharge mode of the power correction circuit in the wireless power supply device. FIG. 7 is a waveform diagram of the first drive signal, the primary-side current, and the second drive signal when the voltage-current phase difference in the wireless power supply device is 90 degrees. FIG. 8 is a waveform diagram of the first drive signal, the primary current, and the second drive signal when the voltage-current phase difference in the wireless power supply device is 45 degrees. FIG. 9 is a flowchart showing output power control in the wireless power supply device of the above. FIG. 10 is a flowchart showing a process for determining a frequency step size in the wireless power supply device according to the first embodiment. FIG. 11 is a graph showing frequency characteristics of output power at the time of frequency control in the wireless power supply device of the above. FIG. 12 is a circuit diagram illustrating a configuration of a power correction circuit according to a modification of the embodiment of the present invention.

  The non-contact power supply device according to the present embodiment supplies power to a load in a non-contact manner. The non-contact power supply device is configured such that the primary coil provided in the non-contact power supply device and the secondary coil provided in the load are electromagnetically coupled (at least one of electric field coupling and magnetic field coupling) in a state where the primary coil is connected to the primary coil. Power is transmitted to the load by transmitting power from the coil to the secondary coil. This type of non-contact power supply device forms a non-contact power transmission system together with a non-contact power receiving device provided in a load.

(1) Overview of Non-Contact Power Transmission System First, an overview of a non-contact power transmission system will be described with reference to FIG.

  The non-contact power transmission system 1 includes a non-contact power supply device 2 having a primary coil L1 and a non-contact power receiving device 3 having a secondary coil L2. The non-contact power receiving device 3 is configured to be supplied with output power from the non-contact power feeding device 2 in a non-contact manner. The “output power” here is the power output from the non-contact power supply device 2 and is applied from the primary coil L1 to the secondary coil L2 in a non-contact manner when an AC voltage is applied to the primary coil L1. Power to be supplied.

  In the present embodiment, a case where the non-contact power receiving device 3 is mounted on an electric vehicle as a load will be described as an example. The electric vehicle is a vehicle that includes the storage battery 4 and runs using the electric energy stored in the storage battery 4. The non-contact power receiving device 3 mounted on the electric vehicle is used as a charging device for the storage battery 4. Here, an electric vehicle that travels by a driving force generated by an electric motor will be described as an example of an electric vehicle. However, the electric vehicle is not limited to an electric vehicle, and may be, for example, a motorcycle (electric motorcycle) or an electric bicycle.

  The non-contact power supply device 2 charges the storage battery 4 of the electric vehicle by supplying the non-contact power receiving device 3 with electric power supplied from a commercial power supply (system power supply) or a power generation facility such as a solar power generation facility. The power supplied to the wireless power supply 2 may be either AC power or DC power, but in the present embodiment, the wireless power supply 2 is electrically connected to the DC power supply 5 and A case where DC power is supplied to the power supply device 2 will be described as an example. When AC power is supplied to the non-contact power supply device 2, the non-contact power supply device 2 is provided with an AC / DC converter for converting alternating current into direct current.

  The non-contact power supply device 2 is installed in, for example, a commercial facility, a public facility, or a parking lot such as an apartment house. The non-contact power supply device 2 has at least the primary coil L1 installed on the floor or the ground, and supplies electric power to the non-contact power receiving device 3 of the electric vehicle parked on the primary coil L1 in a non-contact manner. At this time, the secondary coil L2 of the non-contact power receiving device 3 is electromagnetically coupled to the primary coil L1 by being located above the primary coil L1. Therefore, the output power from the primary side coil L1 is transmitted (power transmission) to the secondary side coil L2. The primary side coil L1 is not limited to the configuration installed to be exposed from the floor or the ground, and may be installed so as to be embedded in the floor or the ground.

  The non-contact power receiving device 3 includes a secondary coil L2, a pair of secondary capacitors C21 and C22, a rectifier circuit 31, and a smoothing capacitor C2. The rectifier circuit 31 includes a diode bridge having a pair of AC input points and a pair of DC output points. One end of the secondary coil L2 is electrically connected to one AC input point of the rectifier circuit 31 via the first secondary capacitor C21, and the other end of the secondary coil L2 is connected to the second secondary capacitor C21. It is electrically connected to the other AC input point of the rectifier circuit 31 via the secondary capacitor C22. The smoothing capacitor C2 is electrically connected between a pair of DC output points of the rectifier circuit 31. Further, both ends of the smoothing capacitor C2 are electrically connected to a pair of output terminals T21 and T22. The storage battery 4 is electrically connected to the pair of output terminals T21 and T22.

  Thereby, the non-contact power receiving device 3 rectifies the AC voltage generated between both ends of the secondary coil L2 by receiving the output power from the primary coil L1 of the non-contact power feeding device 2 with the secondary coil L2. The DC voltage is obtained by rectification by the circuit 31 and smoothing by the smoothing capacitor C2. The non-contact power receiving device 3 outputs (applies) the DC voltage thus obtained from the pair of output terminals T21 and T22 to the storage battery 4.

  In the present embodiment, the non-contact power feeding device 2 includes a power correction circuit 22 that forms a resonance circuit (hereinafter, referred to as a “primary resonance circuit”) with the primary coil L1, and a pair of primary capacitors C11 and C12. ing. That is, the primary resonance circuit includes the primary coil L1, the power correction circuit 22, and the pair of primary capacitors C11 and C12. In the non-contact power receiving device 3, the secondary coil L2 forms a resonance circuit (hereinafter, referred to as a “secondary resonance circuit”) together with the pair of secondary capacitors C21 and C22. That is, the secondary-side resonance circuit includes the secondary-side coil L2 and the pair of secondary-side capacitors C21 and C22. The non-contact power transmission system 1 according to the present embodiment employs a magnetic field resonance method (magnetic resonance method) in which power is transmitted by resonating a primary side resonance circuit and a secondary side resonance circuit. That is, in the non-contact power transmission system 1, the resonance frequency of the primary side resonance circuit and the resonance frequency of the secondary side resonance circuit are matched, so that the primary side coil L1 and the secondary side coil L2 are relatively separated from each other. The output power of the contact power supply device 2 can be transmitted with high efficiency.

  The primary coil L1 and the secondary coil L2 in the present embodiment may be solenoid type coils in which a conductor is spirally wound around a core, but the conductor is spirally wound on a plane. It is preferably a spiral coil. Spiral coils (circular coils) have the advantage that unwanted radiation noise is less likely to occur than solenoid coils. In addition, the use of the spiral type coil has an advantage that unnecessary radiation noise is reduced, and as a result, the operating frequency range usable in the inverter circuit is expanded.

(2) Outline of Non-Contact Power Supply Device Next, an outline of the non-contact power supply device will be described with reference to FIG.

  The contactless power supply device 2 according to the present embodiment includes an inverter circuit 21, a power correction circuit 22, a control circuit 23, a measurement unit 24, a primary coil L1, and a pair of primary capacitors C11 and C12. ing.

  The inverter circuit 21 has a plurality of (here, four) conversion switch elements Q1 to Q4 electrically connected between the pair of input points 211 and 212 and the pair of output points 213 and 214. . The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 into an AC voltage by switching the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214.

  The primary coil L1 is electrically connected between the pair of output points 213 and 214. The primary-side coil L1 supplies output power to the secondary-side coil L2 in a non-contact manner when an AC voltage is applied.

  The power correction circuit 22 is electrically connected between the output point 213 and the primary coil L1, and includes a correction capacitor C1 and a plurality (four in this case) of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. In other words, the power correction circuit 22 adjusts the magnitude of the capacitance component of the primary resonance circuit between the output point 213 and the primary coil L1 by switching the plurality of correction switch elements Q5 to Q8. Thereby, the magnitude of the output power transmitted from the primary resonance circuit of the non-contact power feeding device 2 including the primary coil L1 to the secondary resonance circuit of the non-contact power receiving device 3 including the secondary coil L2 is corrected. Is done.

  The control circuit 23 has functions as an element control unit 231, a frequency control unit 232, a phase difference control unit 233, an acquisition unit 234, and a determination unit 235.

  The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8.

  The frequency controller 232 adjusts the magnitude of the output power by controlling the frequencies of the first drive signals G1 to G4 to discretely change. The phase difference control unit 233 adjusts the magnitude of the output power by controlling the phase difference, which is the delay of the phase of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to be discretely changed. I do.

  The “phase difference” here is a delay of the phase of the second drive signals G6 and G7 with respect to the first drive signals G1 and G4, or a delay of the phase of the second drive signals G5 and G8 with respect to the first drive signals G2 and G3. is there. This point will be described later in detail in the section “(4.2) With power correction circuit” of “(4) Basic operation”.

  The acquisition unit 234 acquires a phase difference between the output voltage and the output current of the inverter circuit 21 as a voltage-current phase difference. Since the output current of the inverter circuit 21 is a current flowing through the primary coil L1, it is also referred to as "primary current I1". In the present embodiment, the obtaining unit 234 obtains the voltage-current phase difference based on the output voltage of the inverter circuit 21 and the primary current I1 measured by the measuring unit 24. The measuring unit 24 receives the output of the current sensor 25 that measures the current flowing through the primary coil L1, and measures the primary current I1.

  The determining unit 235 determines a step size of a control target value including at least one of the frequency controlled by the frequency control unit 232 and the phase difference controlled by the phase difference control unit 233 according to the voltage-current phase difference. That is, when controlling the control target value (at least one of the frequency and the phase difference), if the voltage / current phase difference acquired by the acquisition unit 234 changes, the determination unit 235 changes the step size of the control target value accordingly. Is changed, the step size of the control target value is determined according to the voltage-current phase difference.

  The “step width” here means the minimum change amount of the control target value (at least one of the frequency and the phase difference) that changes discretely. For example, the frequencies of the first drive signals G1 to G4 are controlled by the frequency control unit 232 so as to discretely change. Therefore, when the magnitude of the output power is adjusted by the frequency control unit 232, the control target value (frequency) to be controlled by the frequency control unit 232 changes with the “step size” as the minimum unit. Will do. Similarly, a phase difference, which is a delay of the phase of the second drive signals G6, G7 (G5, G8) with respect to the first drive signals G1, G4 (G2, G3), discretely changes by the phase difference controller 233. Is controlled to be. Therefore, when the magnitude of the output power is adjusted by the phase difference control unit 233, the control target value (phase difference) to be controlled by the phase difference control unit 233 has a minimum unit of “step size”. Will change.

  According to the above configuration, the contactless power supply device 2 according to the present embodiment can control the magnitude of the output power by at least one of the control of the frequency by the frequency control unit 232 and the control of the phase difference by the phase difference control unit 233. Can be adjusted. Therefore, even if the relative positional relationship between the primary coil L1 and the secondary coil L2 changes and the coupling coefficient between the primary coil L1 and the secondary coil L2 changes, the wireless power supply device In the case of 2, the power can be supplied stably by adjusting the magnitude of the output power.

  Moreover, in the contactless power supply device 2, the step size of the control target value (at least one of the frequency and the phase difference) to be controlled when adjusting the output power is determined by the determination unit 235 to the voltage / current phase difference. Determined accordingly. Although described in detail later, there is a correlation between the voltage-current phase difference (the phase difference between the output voltage and the output current of the inverter circuit 21) and the output power of the wireless power supply device 2, and the voltage-current As the phase difference decreases, the output power of the non-contact power feeding device 2 increases. In short, the voltage-current phase difference and the output power have a negative proportional relationship, that is, a proportional relationship having a negative proportional constant. Therefore, for example, when the voltage-current phase difference is relatively large, that is, when the output power is far from the target value determined by the non-contact power receiving device 3, the step width of the control target value is made relatively large, so that the output power is increased. Can be reduced in the time required for the process to bring the value closer to the target value. When the voltage-current phase difference is relatively small, that is, when the output power approaches the target value, the step width of the control target value is made relatively small, so that the output power fluctuates near the target value, thereby causing ripple. It is possible to suppress the occurrence and the output power from greatly exceeding the target value. In other words, the contactless power supply device 2 determines the resolution of the control target value, which is the target of control when adjusting the output power, instead of making the resolution constant, according to the voltage-current phase difference. The output power can be stabilized while shortening the time required for adjusting the power. Therefore, the contactless power supply device 2 has an advantage that stable power supply is possible regardless of the relative positional relationship between the primary coil L1 and the secondary coil L2.

  Here, in the present embodiment, it is assumed that the inverter circuit 21 operates in the delayed mode in which the phase of the output current is delayed with respect to the phase of the output voltage. Therefore, in the present embodiment, the “voltage / current phase difference” is a delay of the phase of the output current with respect to the output voltage of the inverter circuit 21 unless otherwise specified. That is, since the phase delay φ1 of the output current with respect to the output voltage and the phase delay φ2 of the output voltage with respect to the output current have a relationship of “φ1 + φ2 = 360 (degrees)”, the output current is delayed with respect to the output voltage. The value of the voltage-current phase difference differs depending on whether or not it is advanced. In the present embodiment, the delay of the phase of the output current with respect to the output voltage is defined as a voltage-current phase difference because the inverter circuit 21 operates in the delay mode. The “slow mode” will be described in detail in the section “(5) Fast mode and slow mode”.

(3) Circuit Configuration Next, a specific circuit configuration of the contactless power supply device 2 according to the present embodiment will be described with reference to FIG.

  The contactless power supply device 2 according to the present embodiment includes a pair of input terminals T11 and T12. A DC power supply 5 is electrically connected to the pair of input terminals T11 and T12.

  The inverter circuit 21 is a full-bridge inverter circuit in which four conversion switch elements Q1 to Q4 are connected in full-bridge. That is, the inverter circuit 21 has a first arm and a second arm electrically connected in parallel between the pair of input points 211 and 212, and the first arm and the second arm each include four conversion switches. It is composed of elements Q1 to Q4. The first arm comprises a series circuit of a (first) conversion switch element Q1 and a (second) conversion switch element Q2, and the second arm has a (third) conversion switch element Q3 and a (fourth) conversion switch element Q3. 1) and a series circuit with a conversion switch element Q4. The midpoint of the first arm (the connection point of the conversion switch elements Q1 and Q2) and the midpoint of the second arm (the connection point of the conversion switch elements Q3 and Q4) are a pair of output points 213 and 214. In the present embodiment, each of the four conversion switch elements Q1 to Q4 is an n-channel depletion-type MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor).

  More specifically, a pair of input points 211 and 212 are arranged such that the first input point 211 is on the positive side of the DC power supply 5 and the second input point 212 is on the negative side of the DC power supply 5. It is electrically connected to terminals T11 and T12. The drains of the conversion switching elements Q1 and Q3 are electrically connected to the first input point 211. The sources of the conversion switching elements Q2 and Q4 are electrically connected to the second input point 212. The connection point between the source of the conversion switch element Q1 and the drain of the conversion switch element Q2 becomes the first output point 213 of the inverter circuit 21. The connection point between the source of the conversion switch element Q3 and the drain of the conversion switch element Q4 becomes the second output point 214 of the inverter circuit 21.

  The “input point” and the “output point” in the present embodiment may not have any substance as a component (terminal) for connecting an electric wire or the like, for example, a lead of an electronic component or a conductor included in a circuit board. May be a part of.

  Four diodes D1 to D4 are electrically connected between the drains and the sources of the four conversion switch elements Q1 to Q4 so as to correspond to the four conversion switch elements Q1 to Q4 one-to-one. . The diodes D1 to D4 are connected in such a manner that the drain sides of the conversion switching elements Q1 to Q4 are used as cathodes. Here, the diodes D1 to D4 are parasitic diodes of the conversion switching elements Q1 to Q4.

  The power correction circuit 22 has a correction capacitor C1 and four correction switch elements Q5 to Q8. The power correction circuit 22 has a third arm and a fourth arm electrically connected in parallel between a pair of output points 213 and 214 of the inverter circuit 21. It is composed of two correction switch elements Q5 to Q8. The third arm includes a series circuit of a (first) correction switch element Q5 and a (third) correction switch element Q7, and the fourth arm includes a (second) correction switch element Q6 and a (fourth) (A) in series with a correction switch element Q8. The correction capacitor C1 is electrically connected between the middle point of the third arm (the connection point of the correction switch elements Q5 and Q7) and the middle point of the fourth arm (the connection point of the correction switch elements Q6 and Q8). Connected. In the present embodiment, each of the four correction switch elements Q5 to Q8 is an n-channel depletion type MOSFET.

  More specifically, the source of the correction switch element Q5 and the drain of the correction switch element Q6 are electrically connected to the first output point 213 of the inverter circuit 21 via the first primary side capacitor C11. ing. The source of the correction switch element Q7 and the drain of the correction switch element Q8 are electrically connected to the second output point 214 via the second primary capacitor C12 and the primary coil L1. . One end of the correction capacitor C1 is electrically connected to a connection point between the drain of the correction switch element Q5 and the drain of the correction switch element Q7. The other end of the correction capacitor C1 is electrically connected to a connection point between the source of the correction switch element Q6 and the source of the correction switch element Q8. The capacity of the correction capacitor C1 is sufficiently larger than the capacity of the primary capacitors C11 and C12 in the primary resonance circuit. For example, if the capacity of the primary capacitors C11 and C12 is on the order of [nF], the correction capacitor C1 is used. Are of the order of [μF].

  The four diodes D5 to D8 are electrically connected between the drains and the sources of the four correction switch elements Q5 to Q8 so as to correspond one-to-one with the four correction switch elements Q5 to Q8. . The diodes D5 to D8 are connected in such a manner that the drain sides of the correction switch elements Q5 to Q8 are used as cathodes. Here, the diodes D5 to D8 are parasitic diodes of the correction switch elements Q5 to Q8.

  As described above, the measurement unit 24 has a function of measuring the output current of the inverter circuit 21, that is, the primary current I1, which is the current flowing through the primary coil L1. A current sensor 25 including, for example, a current transformer is provided between the primary coil L1 and the second primary capacitor C12. The measuring unit 24 is configured to receive the output of the current sensor 25, measure the primary current I1 as a measured value, and output the measured value to the control circuit 23.

  The control circuit 23 has functions as an element control unit 231, a frequency control unit 232, a phase difference control unit 233, an acquisition unit 234, and a determination unit 235. The control circuit 23 has, for example, a microcomputer as a main configuration. The microcomputer realizes a function as the control circuit 23 by executing a program recorded in a memory of the microcomputer by a CPU (Central Processing Unit). That is, the functions of the element control unit 231, the frequency control unit 232, the phase difference control unit 233, the acquisition unit 234, and the determination unit 235 are realized by the execution of the program by the processor of the microcomputer. The program may be recorded in a memory of a microcomputer in advance, may be provided by being recorded in a recording medium such as a memory card, or may be provided through an electric communication line.

  The element control section 231 outputs first drive signals G1 to G4 for switching on and off the conversion switch elements Q1 to Q4 of the inverter circuit 21. The four first drive signals G1 to G4 correspond one-to-one to the four conversion switch elements Q1 to Q4. Here, the element controller 231 controls the corresponding conversion switch elements Q1 to Q4 by outputting the first drive signals G1 to G4 to the gates of the corresponding conversion switch elements Q1 to Q4, respectively. I have.

  The element control unit 231 outputs second drive signals G5 to G8 for switching on and off of the four correction switch elements Q5 to Q8 of the power correction circuit 22. The four second drive signals G5 to G8 correspond one-to-one to the four correction switch elements Q5 to Q8. Here, the element controller 231 controls the corresponding correction switch elements Q5 to Q8 by outputting the second drive signals G5 to G8 to the gates of the corresponding correction switch elements Q5 to Q8. I have.

  In the present embodiment, the control circuit 23 (element control unit 231) supplies the first drive signals G1 to G4 and the second drive signal to the gates of the conversion switch elements Q1 to Q4 and the correction switch elements Q5 to Q8. Although G5 to G8 are directly output, the present invention is not limited to this configuration. For example, the non-contact power supply device 2 further includes a drive circuit, and the drive circuit receives the first drive signals G1 to G4 and the second drive signals G5 to G8 from the control circuit 23 (the element control unit 231) and performs conversion. The switch elements Q1 to Q4 and the correction switch elements Q5 to Q8 may be driven.

  The acquisition unit 234 acquires the phase difference between the output voltage and the output current of the inverter circuit 21 as a voltage-current phase difference, as described above. Since the phase of the output voltage of the inverter circuit 21 is equivalent to the phase of the first drive signals G1 to G4 output by the element control unit 231, the acquisition unit 234 determines the phase of the first drive signals G1 to G4 and the measurement unit From the phase of the primary current I1 measured at 24, a voltage-current phase difference is obtained. That is, the acquisition unit 234 is configured to acquire the delay of the phase of the primary current I1 with respect to the phases of the first drive signals G1 to G4 as a voltage / current phase difference.

  Functions (frequency control unit 232, phase difference control unit 233, and determination unit 235) of the control circuit 23 other than the element control unit 231 and the acquisition unit 234 will be described in the section of “(6) Output power control”.

  The primary coil L1 is electrically connected in series with the pair of primary capacitors C11 and C12 and the power correction circuit 22 between the pair of output points 213 and 214 of the inverter circuit 21. One end of the primary coil L1 is electrically connected to the first output point 213 of the inverter circuit 21 via the power correction circuit 22 and the first primary capacitor C11. The other end of the primary coil L1 is electrically connected to a second output point 214 of the inverter circuit 21 via a second primary capacitor C12.

(4) Basic Operation Next, a basic operation of the non-contact power supply device 2 of the present embodiment will be described with reference to FIGS. FIG. 2 shows signal waveforms of the first drive signals “G1, G4”, “G2, G3”, and the second drive signals “G5, G8”, “G6, G7” in order from the top, with the horizontal axis as the time axis. ing. Note that “ON” and “OFF” in FIG. 2 represent ON and OFF of corresponding switch elements (conversion switch elements and correction switch elements).

(4.1) No Power Correction Circuit First, when there is no power correction circuit 22, that is, only the primary coil L1 and the pair of primary capacitors C11 and C12 are electrically connected between the pair of output points 213 and 214. The operation of the non-contact power supply device 2 will be described assuming that it is connected. In this case, the operation of the non-contact power supply device 2 is such that in the circuit configuration of FIG. 1, when the power correction circuit 22 stops operating, that is, all the correction switch elements Q5 to Q8 of the power correction circuit 22 are turned on. The operation is equivalent to the operation of the non-contact power feeding device 2 when it is fixed.

  As shown in FIG. 2, the element control unit 231 generates first drive signals G1 and G4 corresponding to the switch elements Q1 and Q4 for conversion and first drive signals G2 and G3 corresponding to the switch elements Q2 and Q3 for conversion. , Signals having phases opposite to each other (a phase difference of 180 degrees). Thereby, in the inverter circuit 21, the pair of the first conversion switch element Q1 and the fourth conversion switch element Q4, and the pair of the second conversion switch element Q2 and the third conversion switch element Q3 are Are turned on alternately.

  As a result, a voltage (AC voltage) whose polarity (positive / negative) is periodically inverted is generated between the pair of output points 213 and 214 of the inverter circuit 21. In short, the inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 into an AC voltage by switching of the plurality of conversion switch elements Q1 to Q4 and outputs the AC voltage from the pair of output points 213 and 214. . Hereinafter, with respect to the output voltage of the inverter circuit 21, the voltage at which the first output point 213 of the pair of output points 213 and 214 has a high potential is referred to as "positive polarity", and the second output point 214 has the high potential. Is called “negative polarity”. That is, the output voltage of the inverter circuit 21 has a positive polarity when the conversion switching elements Q1 and Q4 are on, and has a negative polarity when the conversion switching elements Q2 and Q3 are on.

  As described above, the inverter circuit 21 outputs an AC voltage from the pair of output points 213 and 214, so that an AC current flows through the primary coil L1 electrically connected between the pair of output points 213 and 214, The side coil L1 generates a magnetic field. Thereby, the non-contact power supply device 2 can supply the output power to the secondary coil L2 of the non-contact power receiving device 3 from the primary coil L1 in a non-contact manner.

  By the way, in the case where the power correction circuit 22 is not provided, in the non-contact power supply device 2, the primary coil L1 forms a primary resonance circuit together with the pair of primary capacitors C11 and C12. Therefore, the magnitude of the output power output from the primary coil L1 changes according to the operating frequency of the inverter circuit 21 (that is, the frequency of the first drive signals G1 to G4), and the operating frequency of the inverter circuit 21 changes to the primary side. The peak is reached when the resonance frequency matches the resonance frequency of the resonance circuit.

  Here, when the relative positional relationship between the primary coil L1 and the secondary coil L2 changes and the coupling coefficient between the primary coil L1 and the secondary coil L2 changes, the non-contact power supply device 2 , The frequency characteristic of the output power (hereinafter, referred to as “resonance characteristic”) changes. FIG. 3 shows a change in the resonance characteristic of the non-contact power supply device 2 when the relative positional relationship between the primary coil L1 and the secondary coil L2 changes. In FIG. 3, the horizontal axis represents the frequency (the operating frequency of the inverter circuit 21), and the vertical axis represents the output power of the non-contact power feeding device 2, where the relative positional relationship between the primary coil L1 and the secondary coil L2 is different. Are represented by “X1” and “X2”.

  Here, as shown in FIG. 3, it is assumed that a frequency band usable as an operating frequency of the inverter circuit 21 (hereinafter, referred to as “permitted frequency band F1”) is limited. The permitted frequency band F1 is defined by laws such as the Radio Law. In this case, frequencies below the lower limit value fmin and above the upper limit value fmax of the permitted frequency band F1 cannot be used as the operating frequency of the inverter circuit 21. In such a case, if the resonance characteristic of the contactless power supply device 2 is in a state indicated by, for example, “X1” in FIG. There is a possibility that the power does not reach the required level (hereinafter referred to as “target value”).

  For example, as shown in FIG. 4A, when the resonance frequency fr0 of the primary side resonance circuit is out of the permitted frequency band F1, the magnitude of the output power of the non-contact power feeding device 2 does not reach the peak, and as a result, the target value The output power may be insufficient for P1. Further, for example, as shown in FIG. 4B, even when the resonance frequency fr0 of the primary side resonance circuit is within the permitted frequency band F1, the peak of the output power of the non-contact power feeding device 2 does not reach the target value P1, and as a result, , The output power may be insufficient with respect to the target value P1. That is, in the examples of FIGS. 4A and 4B, the power in the hatched (hatched) portion is insufficient for the target value P1.

  Therefore, the contactless power supply device 2 according to the present embodiment includes the power correction circuit 22 and has a function of correcting the magnitude of the output power so as to satisfy the target value P1.

(4.2) With Power Correction Circuit Next, as shown in FIG. 1, when the power correction circuit 22 is provided, that is, between the pair of output points 213 and 214, the primary coil L1 and the pair of primary capacitors C11 and C12 are provided. And the operation of the non-contact power supply device 2 when the power correction circuit 22 is electrically connected.

  As shown in FIG. 2, the element control unit 231 generates second drive signals G6 and G7 corresponding to the correction switch elements Q6 and Q7 and second drive signals G5 and G8 corresponding to the correction switch elements Q5 and Q8. , Signals having phases opposite to each other (a phase difference of 180 degrees). Thus, in the power correction circuit 22, a pair of the second correction switch element Q6 and the third correction switch element Q7 and a pair of the first correction switch element Q5 and the fourth correction switch element Q8 Are turned on alternately. In the present embodiment, the element controller 231 sets the frequencies of the first drive signals G1 to G4 and the second drive signals G5 to G8 to the same frequency.

  If the output voltage of the inverter circuit 21 is positive during the period in which the correction switch elements Q5 and Q8 are on and the correction switch elements Q6 and Q7 are off, the correction switch elements Q5 and Q8 are used. A voltage is applied to the capacitor C1. That is, the primary coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 via the correction capacitor C1 (hereinafter, also referred to as "first state").

  On the other hand, if the output voltage of the inverter circuit 21 is negative during a period in which the correction switch elements Q5 and Q8 are on and the correction switch elements Q6 and Q7 are off, a current flows through the diode D7 and the correction switch element Q5. . That is, the primary coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 without the intervention of the correction capacitor C1 (hereinafter, also referred to as "second state"). . In other words, both ends of the correction capacitor C1 are bypassed by the diode D7 and the correction switch element Q5.

  If the output voltage of the inverter circuit 21 is positive during the period when the correction switch elements Q6 and Q7 are on and the correction switch elements Q5 and Q8 are off, a current flows through the correction switch element Q6 and the diode D8. . That is, the primary side coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 without the intervention of the correction capacitor C1 (hereinafter, also referred to as “third state”). . In other words, both ends of the correction capacitor C1 are bypassed by the correction switch element Q6 and the diode D8.

  On the other hand, if the output voltage of the inverter circuit 21 is negative during the period in which the correction switch elements Q6 and Q7 are on and the correction switch elements Q5 and Q8 are off, the correction switch elements Q6 and Q7 are used. A voltage is applied to the capacitor C1. That is, the primary coil L1 is electrically connected between the pair of output points 213 and 214 of the inverter circuit 21 via the correction capacitor C1 (hereinafter, also referred to as “fourth state”). The polarity of the voltage applied to the correction capacitor C1 is the same between the first state and the fourth state. That is, in any of the first state and the fourth state, the voltage at which the connection point between the drain of the correction switch element Q5 and the drain of the correction switch element Q7 has a high potential is the voltage for correction. Applied to the capacitor C1.

  As described above, the power correction circuit 22 determines whether the primary coil L1 is electrically connected via the correction capacitor C1 between the pair of output points 213 and 214, and whether the primary coil L1 is electrically connected via the correction capacitor C1. Is switched to the connected state. As a result, the magnitude of the capacitance component of the primary resonance circuit between the pair of output points 213 and 214 and the primary coil L1 apparently changes.

  Therefore, the contactless power supply device 2 of the present embodiment can change the magnitude of the output power by adjusting the magnitude of the capacitance component of the primary-side resonance circuit by the power correction circuit 22. As a result, as described above, when the output power of the non-contact power supply device 2 becomes insufficient with respect to the target value P1, the power correction circuit 22 corrects the magnitude of the output power so as to satisfy the target value P1. It is possible. In other words, the power correction circuit 22 can charge and discharge the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8 to correct the magnitude of the output power.

  In the present embodiment, as described above, the first drive signals G1 and G4 corresponding to the conversion switch elements Q1 and Q4 and the first drive signals G2 and G3 corresponding to the conversion switch elements Q2 and Q3 are different from each other. The signals have phases opposite to each other (a phase difference of 180 degrees). Similarly, the second drive signals G6, G7 corresponding to the correction switch elements Q6, Q7 and the second drive signals G5, G8 corresponding to the correction switch elements Q5, Q8 have opposite phases (a phase difference of 180 degrees). Signal.

  Here, the phase difference θ between the first drive signal and the second drive signal in the present embodiment is the delay of the phase of the second drive signals G6 and G7 with respect to the first drive signals G1 and G4 or the first drive signal G2 , G3 with respect to the phase of the second drive signals G5, G8 (see FIG. 2). That is, the phase delay of the second drive signals G6 and G7 with respect to the first drive signals G1 and G4 and the phase delay of the second drive signals G5 and G8 with respect to the first drive signals G1 and G4 have a difference of 180 degrees. Therefore, the value of the phase difference θ differs depending on which phase delay is the phase difference θ. Therefore, in the present embodiment, the phase delay of the second drive signals G6 and G7 with respect to the first drive signals G1 and G4 or the phase delay of the second drive signals G5 and G8 with respect to the first drive signals G2 and G3 is determined by the phase difference. Defined as θ.

  Here, if the first drive signals G1, G4 and the second drive signals G6, G7 are all “ON”, in the power correction circuit 22, the primary side coil L1 is located between the pair of output points 213, 214. It is in the third state where it is electrically connected without the intervention of the correction capacitor C1. If the first drive signals G2 and G3 and the second drive signals G5 and G8 are all “ON”, the power correction circuit 22 corrects the primary coil L1 between the pair of output points 213 and 214. It is in the second state in which it is electrically connected without using the capacitor C1. That is, in the present embodiment, the first drive signal and the second drive signal are such that the primary coil L1 is electrically connected between the pair of output points 213 and 214 without the intervention of the correction capacitor C1. The phase difference for a combination with a signal is defined as a phase difference θ.

(5) Fast mode and slow mode Next, the fast mode and the slow mode will be described.

(5.1) No power correction circuit First, as in the section of “(4) Basic operation”, when there is no power correction circuit 22, that is, between the pair of output points 213 and 214, the primary side coil L1 and The case where only the pair of primary side capacitors C11 and C12 are electrically connected will be described.

  In this case, the inverter circuit 21 operates in one of the slow mode and the fast mode depending on, for example, the relationship between the operating frequency of the inverter circuit 21 and the resonance frequency of the primary resonance circuit.

  The phase advance mode is a mode in which the inverter circuit 21 operates in a state where the phase of the output current of the inverter circuit 21 (the current flowing through the primary coil L1) is ahead of the phase of the output voltage of the inverter circuit 21. In the phase advance mode, the switching operation of the inverter circuit 21 is hard switching. Therefore, in the phase advance mode, the switching loss of the conversion switching elements Q1 to Q4 tends to increase, and stress is easily applied to the conversion switching elements Q1 to Q4.

  On the other hand, the lag mode is a mode in which the inverter circuit 21 operates in a state where the phase of the output current of the inverter circuit 21 (the current flowing through the primary coil L1) is delayed from the phase of the output voltage of the inverter circuit 21. . In the slow mode, the switching operation of the inverter circuit 21 is soft switching. Therefore, in the slow mode, the switching loss of the conversion switching elements Q1 to Q4 can be reduced, and stress is less likely to be applied to the conversion switching elements Q1 to Q4. For this reason, it is preferable that the inverter circuit 21 operate in the late mode rather than the fast mode.

(5.2) With Power Correction Circuit Next, when there is the power correction circuit 22, that is, between the pair of output points 213 and 214, the primary coil L1, the pair of primary capacitors C11 and C12, and the power correction circuit 22 Are electrically connected.

  In this case, similarly to the inverter circuit 21, the power correction circuit 22 operates in one of the operation modes of the leading phase mode and the lagging mode. It is preferable that the power correction circuit 22 also operates in the lag mode instead of the lag mode.

  Further, when the power correction circuit 22 is provided, the operation modes (slow mode, fast mode) of the inverter circuit 21 and the power correction circuit 22 are the first drive signals G1 to G4 and the second drive signals G5 to G8. Has been confirmed to change according to the phase difference θ. Furthermore, the relationship between the operation mode of the inverter circuit 21 and the phase difference θ is such that the operation of the inverter circuit 21 is performed in a state where the power correction circuit 22 is not provided, that is, under the conditions described in “(5.1) No power correction circuit”. Varies depending on the mode. In other words, depending on whether the operation mode of the inverter circuit 21 determined by the relationship between the operation frequency of the inverter circuit 21 and the resonance frequency of the primary resonance circuit is the slow mode or the fast mode, the operation mode of the inverter circuit 21 and the phase difference θ The relationship changes.

  FIGS. 5A and 5B show the characteristics of the output power of the contactless power supply device 2 with respect to the phase difference θ when the inverter circuit 21 is in the leading mode and in the lagging mode without the power correction circuit 22, respectively. (Phase difference characteristics). 5A and 5B, the horizontal axis represents the phase difference θ between the first drive signals G1 to G4 and the second drive signals G5 to G8, and the vertical axis represents the output power of the contactless power supply device 2.

  That is, when the inverter circuit 21 is in the phase advance mode without the power correction circuit 22 (hereinafter, referred to as “initial phase advance”), the output power of the non-contact power supply device 2 is, for example, as shown in FIG. It changes according to the phase difference θ. In the example of FIG. 5A, the output power of the contactless power supply device 2 is maximized and maximized when the phase difference θ is 90 degrees, and is minimized and minimized when the phase difference θ is 180 degrees. Change. The principle that the output power of the non-contact power supply device 2 changes according to the phase difference θ will be described in the section “(6.2) Principle of output power control by phase difference control” below. Here, the phase difference θ (0 to 360 degrees) is divided into four sections, 0 to 90 degrees is a first section Z1, 90 to 180 degrees is a second section Z2, and 180 to 270 degrees is a third section Z1. The section Z3, 270 degrees to 360 degrees is defined as a fourth section Z4. Then, the relationship between the operation modes (slow mode, fast mode) of the inverter circuit 21 and each section of the phase difference θ is as shown in Table 1.

  In short, in the case of the “initial leading phase”, the inverter circuit 21 operates in the retarding mode when the phase difference θ is 90 degrees to 180 degrees among the first section Z1 to the fourth section Z4. Only the second section Z2.

  On the other hand, when the inverter circuit 21 is in the lagging mode without the power correction circuit 22 (hereinafter, referred to as “initial lagging”), the output power of the wireless power supply device 2 is, for example, as shown in FIG. It changes according to the phase difference θ. In the example of FIG. 5B, the output power of the non-contact power supply device 2 is maximized and maximized when the phase difference θ is 270 degrees, and is minimized and minimized when the phase difference θ is 180 degrees. Change. In this case, the relationship between the respective operation modes (slow mode, fast mode) of the inverter circuit 21 and the power correction circuit 22 and each section of the phase difference θ is as shown in Table 2.

  In short, in the case of the “initial delay”, the inverter circuit 21 operates in the delay mode only when the phase difference θ is 0 degree to 180 degrees in the first section Z1 to the fourth section Z4. There are three sections of a first section Z1, a second section Z2, and a fourth section Z4 which are degrees to 360 degrees. In addition, in the case of “initial delay”, the power correction circuit 22 operates in the delay mode because the phase difference θ is 0 degree to 90 degrees and 180 degrees to 180 degrees among the first section Z1 to the fourth section Z4. There are three sections of a first section Z1, a third section Z3, and a fourth section Z4 which are 360 degrees. That is, in the case of “initial delay”, both the inverter circuit 21 and the power correction circuit 22 operate in the delay mode because the phase difference θ is 0 among the first section Z1 to the fourth section Z4. There are two sections, a first section Z <b> 1 having a degree of 90 degrees and 270 degrees and 360 degrees, and a fourth section Z <b> 4.

(6) Output Power Control Next, the operation of “output power control” for adjusting the magnitude of the output power in the contactless power supply device 2 of the present embodiment will be described.

(6.1) Frequency Control and Phase Difference Control The control circuit 23 outputs the data by two methods, “frequency control” performed by the frequency control unit 232 and “phase difference control” performed by the phase difference control unit 233. It is configured to adjust the magnitude of the power. The frequency control is control for discretely changing the frequency of the first drive signals G1 to G4, that is, the operating frequency f1 of the inverter circuit 21. In contrast, the phase difference control discretely changes the phase difference θ, which is the delay of the phase of the second drive signals G6, G7 (G5, G8) with respect to the first drive signals G1, G4 (G2, G3). Control.

  That is, the control circuit 23 sets the operating frequency f1 in the frequency control and the phase difference θ in the phase difference control as the control target values, and controls the control target values so as to discretely change the output power. Adjust the size of. In the frequency control, the frequency control unit 232 discretely changes the operation frequency f1, which is the control target value, using the step size for the operation frequency f1 as a minimum unit. In the phase difference control, the phase difference control unit 233 discretely changes the phase difference θ, which is the control target value, using the step size for the phase difference θ as the minimum unit. Hereinafter, the step width of the control target value (operating frequency f1) in the frequency control, that is, the minimum change amount of the operating frequency f1 is also referred to as “frequency step width Δf”. Further, the step width of the control target value (phase difference θ) in the phase difference control, that is, the minimum change amount of the phase difference θ is also referred to as “phase difference step width Δθ”.

  In the present embodiment, the control circuit 23 first controls the frequency control unit 232 to change the frequency of the first drive signals G1 to G4 (operating frequency f1), thereby adjusting the frequency of the output power. Perform control. That is, as described in the column of “(4.1) No power correction circuit” in “(4) Basic operation”, the magnitude of the output power output from the primary coil L1 depends on the operation of the inverter circuit 21. It changes according to the frequency f1 (see FIG. 3). Therefore, the frequency control unit 232 adjusts the operating frequency f1 of the inverter circuit 21 by adjusting the frequency of the first drive signals G1 to G4, and adjusts the magnitude of the output voltage.

  Here, when the frequency band (permitted frequency band F1) usable as the operating frequency f1 of the inverter circuit 21 is limited, the operating frequency f1 that can be adjusted by the frequency control is limited to within the permitted frequency band F1. Is done. Then, when the magnitude of the output power after the frequency control is less than the predetermined target value P1, the control circuit 23 performs the phase difference control by the phase difference control unit 233. In other words, the phase difference control unit 233 performs the phase difference control for controlling the phase difference θ when the magnitude of the output power after the adjustment by the frequency control unit 232 is less than the predetermined target value P1. The power level is configured to approach the target value P1. As described above, when the output power is insufficient with respect to the target value P1 only by the frequency control (see FIGS. 4A and 4B), the control circuit 23 compensates for the insufficiency by the phase difference control.

  The phase difference control unit 233 changes the phase difference θ, which is the phase delay of the second drive signals G6, G7 (G5, G8) with respect to the first drive signals G1, G4 (G2, G3), within a specified range. By controlling, the magnitude of the output power of the wireless power feeding device 2 is adjusted. The “specified range” here is a control range of the phase difference θ in the phase difference control, that is, a range in which the phase difference θ is changed in the phase difference control. In short, the phase difference control unit 233 limits the movable range of the phase difference θ to a specified range, and controls so as to discretely change the phase difference θ within the specified range. Here, the specified range includes at least one of 270 degrees to 360 degrees and 90 degrees to 180 degrees.

  That is, as is clear from FIGS. 5A and 5B, the output power of the non-contact power feeding device 2 changes according to the phase difference θ, and the phase difference control unit 233 changes the phase difference θ to increase the output power. Adjustment of the length is possible. However, the phase difference θ between the first drive signals G1 to G4 and the second drive signals G5 to G8 depends on the operation mode (slow mode, fast mode) of the inverter circuit 21 and the power correction circuit 22 as described above. Affect. Therefore, in order to operate the inverter circuit 21 and the power correction circuit 22 in the slow mode, it is necessary to limit the range of the phase difference θ.

  First, the case of “initial advance” will be described with reference to FIG. 5A. In this case, as described above, the inverter circuit 21 operates in the delayed mode only in the second section Z2 in which the phase difference θ is 90 degrees to 180 degrees among the first section Z1 to the fourth section Z4. . Therefore, in the case of “initial phase advance”, the control range of the phase difference θ in the phase difference control is preferably in a range of 90 degrees to 180 degrees. Accordingly, when the phase difference control unit 233 controls the phase difference θ to be changed within the specified range and adjusts the magnitude of the output power of the wireless power supply device 2, the inverter circuit 21 operates in the slow mode. Can work.

  Next, the case of “initial delay” will be described with reference to FIG. 5B. In this case, as described above, the inverter circuit 21 operates in the slow mode in the three sections of the first section Z1, the second section Z2, and the fourth section Z4. However, in the first section Z1 in which the phase difference θ is 0 to 90 degrees, the magnitude of the output power hardly changes even if the phase difference θ changes. Therefore, in order to adjust the magnitude of the output power by adjusting the phase difference θ, the second section Z2 in which the phase difference θ is 90 degrees to 180 degrees and the fourth section Z2 in which the phase difference θ is 270 degrees to 360 degrees It is necessary to adjust the phase difference θ within two sections of section Z4. Therefore, if the specified ranges are 270 degrees to 360 degrees and 90 degrees to 180 degrees, the phase difference control unit 233 controls the phase difference θ to be changed within the specified range to reduce the magnitude of the output power. When adjusted, the inverter circuit 21 can operate in the slow mode.

  Furthermore, in the case of “initial delay”, both the inverter circuit 21 and the power correction circuit 22 operate in the delay mode because the phase difference θ is 0 to 90 degrees and 270 to 360 degrees as described above. There are two sections, a first section Z1 and a fourth section Z4. As described above, in the first section Z1, even if the phase difference θ changes, the magnitude of the output power hardly changes. Therefore, of the two sections of the first section Z1 and the fourth section Z4, the phase difference θ It is only in the fourth section Z4 that the magnitude of the output power can be adjusted by the adjustment. Therefore, in the case of “initial delay”, the control range of the phase difference θ in the phase difference control is more preferably in the range of 270 degrees to 360 degrees. Accordingly, when the phase difference control unit 233 controls the phase difference θ to be changed within the specified range and adjusts the magnitude of the output power of the wireless power supply device 2, the inverter circuit 21 and the power correction circuit 22 Can operate in the slow mode.

  In short, in the case of “early phase advance”, the prescribed range is preferably in a range of 90 degrees to 180 degrees (second section Z2). On the other hand, in the case of “initial delay”, the prescribed range is preferably in a range of 90 degrees to 180 degrees (second section Z2) or in a range of 270 degrees to 360 degrees (fourth section Z4). In the case of "initial delay", the specified range is more preferably in the range of 270 degrees to 360 degrees (fourth section Z4).

  In a section (at least one of the second section Z2 and the fourth section Z4) in which the inverter circuit 21 operates in the delayed mode, as shown in FIGS. 5A and 5B, as the phase difference θ becomes smaller, the output power becomes smaller. Becomes larger. Thus, for example, if the specified range is the second section (90 degrees to 180 degrees) Z2, the phase difference θ gradually decreases from the upper limit (180 degrees) to the lower limit (90 degrees) of the specified range. It is preferable that the phase difference controller 233 gradually reduces the phase difference θ within a specified range. Similarly, if the specified range is the fourth section (270 degrees to 360 degrees) Z4, the phase difference θ gradually decreases from the upper limit (360 degrees) to the lower limit (270 degrees) of the specified range. It is preferable that the phase difference controller 233 gradually reduces the phase difference θ within a specified range. Thereby, the output power of the non-contact power supply device 2 gradually increases as the phase difference control unit 233 gradually changes (decreases) the phase difference θ within the specified range.

  By the way, it has been confirmed that the voltage-current phase difference (the delay of the phase of the output current with respect to the output voltage of the inverter circuit 21) changes during the frequency control and the phase difference control, and the output power changes. . That is, in the frequency control, when the frequency (operating frequency f1) of the first drive signals G1 to G4 changes, the voltage-current phase difference changes according to the resonance characteristics, and the voltage-current phase difference changes with the change in the voltage-current phase difference. Output power changes. In the phase difference control, a change in the phase difference θ changes the voltage / current phase difference, and the output power changes in accordance with the change in the voltage / current phase difference. That is, there is a correlation between the voltage-current phase difference and the output power of the non-contact power supply device 2, and the voltage-current phase difference is essentially changed in any of the frequency control and the phase difference control. By doing so, the output power is adjusted.

  Specifically, the voltage-current phase difference and the output power have a negative proportional relationship, that is, a proportional relationship having a negative proportional constant. In short, the smaller the voltage-current phase difference, the larger the output power of the non-contact power feeding device 2. Conversely, as the voltage-current phase difference increases, the output power of the contactless power supply device 2 decreases. Therefore, by monitoring the voltage-current phase difference acquired by the acquisition unit 234, it is possible to monitor an increase or decrease in the output power of the wireless power supply device 2.

(6.2) Principle of Output Power Control by Phase Difference Control Hereinafter, the magnitude of the output power of the non-contact power feeding device 2 is controlled by the phase difference controller 233 controlling the phase difference θ to be changed within a specified range. The principle by which is adjusted will be described with reference to FIGS.

  The output power of the non-contact power feeding device 2 changes according to the voltage applied to the primary coil L1 of the primary resonance circuit. The voltage applied to the primary coil L1 is a combination of the output voltage of the inverter circuit 21 and the voltage across the power correction circuit 22 (the voltage between the source of the correction switch element Q5 and the source of the correction switch element Q7). Voltage. Therefore, when the output voltage of the inverter circuit 21 and the voltage between both ends of the power correction circuit 22 have the same polarity and strengthen each other, the voltage applied to the primary coil L1 increases, and the output of the non-contact power supply device 2 The power increases. In this case, as the voltage across the correction capacitor C1 increases, the voltage across the power correction circuit 22 increases, and the voltage applied to the primary coil L1 increases. Becomes larger. Therefore, the phase difference control unit 233 adjusts the phase difference θ to change the balance between charging and discharging of the correction capacitor C1, change the voltage across the correction capacitor C1, and change Change the output power.

  Here, whether the correction capacitor C1 is charged or discharged is determined by ON / OFF of the plurality of correction switch elements Q5 to Q8 and the direction of the output current (primary current I1) of the inverter circuit 21. The direction of the primary current I1 flowing from the first output point 213 to the second output point 214 through the primary side capacitor C11, the power correction circuit 22, the primary side coil L1, and the primary side capacitor C12, that is, in FIG. The direction of the primary current I1 indicated by the arrow is referred to as “positive direction”. The direction of the primary side current I1 flowing from the second output point 214 to the first output point 213 through the primary side capacitor C12, the primary side coil L1, the power correction circuit 22, and the primary side capacitor C11, that is, FIG. The direction opposite to the primary current I1 indicated by the arrow is called "negative direction".

  6A to 6D show a combination pattern of the on / off of the plurality of correction switch elements Q5 to Q8 and the direction of the primary current I1. 6A to 6D, the bold arrow indicates a current path, and the switch element for correction with a dotted circle indicates an element in an ON state.

  FIG. 6A shows the state of the power correction circuit 22 in which the correction switch elements Q6 and Q7 are on, the correction switch elements Q5 and Q8 are off, and the primary current I1 in the negative direction is flowing (hereinafter, referred to as “the current state”). "First charging mode"). FIG. 6B shows a state of the power correction circuit 22 in which the correction switch elements Q6 and Q7 are on, the correction switch elements Q5 and Q8 are off, and the primary current I1 in the positive direction is flowing (hereinafter, referred to as “the current state”). "First discharge mode"). FIG. 6C shows the state of the power correction circuit 22 in which the correction switch elements Q5 and Q8 are on, the correction switch elements Q6 and Q7 are off, and the primary current I1 in the positive direction is flowing (hereinafter, referred to as the following). "Second charging mode"). FIG. 6D shows the state of the power correction circuit 22 in which the correction switch elements Q5 and Q8 are on, the correction switch elements Q6 and Q7 are off, and the primary current I1 in the negative direction is flowing (hereinafter, referred to as “current”). "Second discharge mode"). In the first charging mode shown in FIG. 6A and the second charging mode shown in FIG. 6C, the correction capacitor C1 is charged. On the other hand, in the first discharge mode shown in FIG. 6B and the second discharge mode shown in FIG. 6D, the correction capacitor C1 is discharged.

  Next, the relationship between the phase difference θ and the balance between charging and discharging of the correction capacitor C1 will be described with reference to FIGS. 7 and 8, the horizontal axis is the time axis, and the waveforms of the first drive signal “G1, G4”, the primary current “I1”, and the two types of second drive signals “G6, G7” are arranged in order from the top. Is represented. Here, the two types of second drive signals have different phase differences θ. Note that “ON” and “OFF” in FIGS. 7 and 8 indicate ON and OFF of corresponding switch elements (conversion switch elements and correction switch elements).

  FIG. 7 illustrates the case of “initial phase delay”, in which the phase delay (voltage / current phase difference) φ of the primary current I1 with respect to the output voltage of the inverter circuit 21 is 90 degrees. FIG. 7 shows, as the waveforms of the two types of second drive signals “G6, G7”, a waveform when the phase difference θ is 360 degrees and a waveform when the phase difference θ is 320 degrees in order from the top. Further, in FIG. 7, when the phase difference θ is 360 degrees, the period of the first charging mode is “Tca1”, the period of the first discharging mode is “Tda1”, the period of the second charging mode is “Tca2”, The period of the two-discharge mode is represented by “Tda2”. Similarly, when the phase difference θ is 320 degrees, the period of the first charging mode is “Tcb1”, the period of the first discharging mode is “Tdb1”, the period of the second charging mode is “Tcb2”, and the second discharging mode is “Tcb2”. Is represented by “Tdb2”.

  As is clear from FIG. 7, when the phase difference θ is 360 degrees, the time during which the correction capacitor C1 is charged in one cycle of the second drive signal (hereinafter, referred to as “charge time”) and the correction capacitor The time when C1 is discharged (hereinafter, referred to as “discharge time”) is substantially balanced. That is, if the phase difference θ is 360 degrees, the sum of “Tca1” and “Tca2” and the sum of “Tda1” and “Tda2” have substantially the same length. On the other hand, if the phase difference θ is 320 degrees, the charging time exceeds the discharging time in one cycle of the second drive signal. That is, if the phase difference θ is 320 degrees, the sum of “Tcb1” and “Tcb2” is longer than the sum of “Tdb1” and “Tdb2”.

  As described above, when the phase difference θ changes from 360 degrees to approach 270 degrees, the balance between the charging time and the discharging time is broken in one cycle of the second drive signal, and the proportion of the charging time gradually increases. . In FIG. 7, for convenience of explanation, the description of the change in the voltage-current phase difference φ is omitted, but in practice, when the phase difference θ changes, the voltage-current phase difference φ also changes with the initial value as the phase difference θ changes. (Here, 90 degrees). That is, if the charging time exceeds the discharging time, a current flows into the correction capacitor C1, and the current phase advances by 90 degrees with respect to the phase of the voltage across the correction capacitor C1. That is, since the correction capacitor C1 functions as a phase advance capacitor, the voltage-current phase difference φ, which is a delay of the phase of the primary current I1 with respect to the output voltage of the inverter circuit 21, decreases, and the output power increases. As a result, as the phase difference θ approaches 270 degrees from 360 degrees, the voltage-current phase difference φ decreases, and the output power of the non-contact power feeding device 2 gradually increases.

  In other words, the smaller the phase difference θ, the smaller the voltage-current phase difference φ in order to maintain the balance between the charging time and the discharging time. That is, if the state where the charging time exceeds the discharging time continues, the voltage across the correction capacitor C1 diverges. To prevent such divergence from occurring, in practice, when the phase difference θ decreases, the voltage-current phase difference φ decreases, and the balance between the charging time and the discharging time is maintained.

  FIG. 8 illustrates the case of “initial delay”, in which the voltage-current phase difference φ is 45 degrees. FIG. 8 shows, as the waveforms of the two types of second drive signals “G6, G7”, a waveform when the phase difference θ is 315 degrees and a waveform when the phase difference θ is 290 degrees from the top. 8, when the phase difference θ is 315 degrees, the period of the first charging mode is “Tca1”, the period of the first discharging mode is “Tda1”, the period of the second charging mode is “Tca2”, The period of the two-discharge mode is represented by “Tda2”. Similarly, when the phase difference θ is 290 degrees, the period of the first charging mode is “Tcb1”, the period of the first discharging mode is “Tdb1”, the period of the second charging mode is “Tcb2”, and the second discharging mode is “Tcb2”. Is represented by “Tdb2”.

  If the voltage-current phase difference φ is 45 degrees, as is clear from FIG. 8, even if the phase difference θ is 315 degrees, the charging time and the discharging time are substantially balanced in one cycle of the second drive signal. . That is, even if the phase difference θ is 315 degrees, the total of “Tca1” and “Tca2” and the total of “Tda1” and “Tda2” have substantially the same length. On the other hand, if the phase difference θ is 290 degrees, the charging time exceeds the discharging time in one cycle of the second drive signal. That is, if the phase difference θ is 290 degrees, the sum of “Tcb1” and “Tcb2” is longer than the sum of “Tdb1” and “Tdb2”.

  From the above, not only when the voltage-current phase difference φ is 90 degrees, but in the case of “initial delay”, when the phase difference θ changes from 360 degrees to approach 270 degrees, one cycle of the second drive signal In, the balance between the charging time and the discharging time is broken, and the proportion of the charging time gradually increases. However, when the voltage-current phase difference φ is 90 degrees, the charging time exceeds the discharging time when the phase difference θ is 320 degrees, whereas when the voltage-current phase difference φ is 45 degrees, the phase difference θ is 315 degrees. The charging time and the discharging time are balanced even at the same time. As described above, when the phase difference θ is gradually reduced from 360 degrees, the phase difference θ (hereinafter, “change start”) corresponding to an inflection point where the balance between the charging time and the discharging time is broken and the output power starts to increase. Point) differs depending on the voltage-current phase difference φ. The change start point shifts closer to 270 degrees when the voltage-current phase difference φ is 45 degrees than when it is 90 degrees, that is, as the voltage-current phase difference φ is smaller.

  That is, in the case of the “initial delay”, a change start point exists within a specified range (for example, 270 degrees to 360 degrees) even though there is a difference due to the voltage / current phase difference φ. Therefore, if the phase difference control unit 233 gradually reduces the phase difference θ from the upper limit (360 degrees) to the lower limit (270 degrees) of the specified range, the phase difference control unit 233 does not operate after the phase difference θ reaches the change start point. The output power of the contact power supply device 2 gradually increases. Actually, as described above, the voltage-current phase difference φ also changes with the change of the phase difference θ, and after the phase difference θ reaches the change start point, the voltage-current phase difference φ decreases. Thus, the balance between the charging time and the discharging time is maintained.

  7 and 8 illustrate the case of “initial lag”, but in the case of “initial lag”, the phase of the primary current I1 is shifted by 180 degrees based on the example of “initial lag”. Is equivalent to That is, in the examples of FIGS. 7 and 8, if the phase of the primary current I1 is shifted by 180 degrees, it is an example of “initial advance”. Therefore, even in the case of the “initial phase advance”, the change start point exists within the specified range (180 degrees to 90 degrees) even if there is a difference due to the voltage / current phase difference φ. Therefore, if the phase difference controller 233 gradually reduces the phase difference θ from the upper limit value (180 degrees) to the lower limit value (90 degrees) of the specified range, the phase difference control unit 233 does not operate after the phase difference θ reaches the change start point. The output power of the contact power supply device 2 gradually increases.

  By the principle described above, the phase difference control unit 233 controls the phase difference θ to change within the specified range in any of the “early phase delay” and “early phase advance”. The magnitude of the output power of the power supply device 2 is adjusted.

(6.3) Overall Flow of Output Power Control Hereinafter, the overall flow of “output power control” of the present embodiment will be described with reference to the flowchart of FIG.

  When the output power control starts, the control circuit 23 first compares a predetermined target value P1 with the magnitude of the output power (S1). If the output power is within the allowable error range (± several%) of the target value P1 (S1: rated), the output power control ends.

  On the other hand, if the output power is below the lower limit of the allowable error range of the target value P1 (S1: insufficient), the control circuit 23 first adjusts the output power using the frequency control unit 232. Specifically, the frequency control unit 232 compares the operating frequency f1 of the inverter circuit 21 (that is, the frequency of the first drive signals G1 to G4) with the lower limit fmin of the permitted frequency band F1 (S11). If the operating frequency f1 is higher than the lower limit value fmin (S11: Yes), the frequency control unit 232 controls the operating frequency f1 of the inverter circuit 21 to decrease. At this time, the frequency step width Δf, which is the step width of the control target value (operating frequency f1) controlled by the frequency control unit 232, is determined by the determination unit 235 (S12). The determining unit 235 basically changes the frequency step width Δf when the voltage-current phase difference φ satisfies a predetermined condition. The process S12 of determining the frequency step width Δf will be described in detail in the section “(6.4) Resolution”. The frequency control unit 232 lowers the operating frequency f1 of the inverter circuit 21 by the frequency step width Δf determined in the processing S12 (S13), and returns to the operation of the processing S1. By repeating these processes (S11 to S13), the control circuit 23 can decrease the operating frequency f1 by the frequency step width Δf, and gradually increase the output power, that is, approach the target value P1. The initial value of the operating frequency f1 is, for example, the upper limit value fmax of the permitted frequency band F1.

  When the operating frequency f1 becomes equal to or lower than the lower limit value fmin (S11: No), the control circuit 23 adjusts the output power in the phase difference controller 233. Specifically, the phase difference controller 233 compares the phase difference θ with the lower limit of the specified range (S14). Here, if the specified range is between 270 degrees and 360 degrees, the initial value of the phase difference θ is 360 (degrees), and the lower limit of the specified range is 270 (degrees). If the specified range is between 90 degrees and 180 degrees, the initial value of the phase difference θ is 180 (degrees), and the lower limit of the specified range is 90 (degrees). If the phase difference θ is equal to or larger than the lower limit (S14: No), the phase difference controller 233 controls the phase difference θ to be small. At this time, the phase difference step width Δθ, which is the step width of the control target value (phase difference θ) controlled by the phase difference control unit 233, is determined by the determination unit 235 (S15). The determining unit 235 basically changes the phase difference step width Δθ when the voltage / current phase difference φ satisfies a predetermined condition. The process S15 for determining the phase difference step width Δθ will be described in detail in the section “(6.4) Resolution”. The phase difference controller 233 reduces the phase difference θ by the phase difference step width Δθ determined in step S15 (S16), and returns to the operation of step S1. By repeating these processes (S14 to S16), the control circuit 23 can reduce the phase difference θ by the phase difference step width Δθ and gradually increase the output power, that is, approach the target value P1.

  When the phase difference θ becomes smaller than the lower limit in the phase difference control (S14: Yes), the control circuit 23 determines that an error has occurred (S17) and ends the output power control.

  If the output power exceeds the upper limit of the allowable error range of the target value P1 (S1: excess), the control circuit 23 first adjusts the output power by the phase difference control unit 233. Specifically, the phase difference controller 233 compares the phase difference θ with the upper limit of the specified range (S21). Here, if the specified range is 270 degrees to 360 degrees, the upper limit of the specified range is 360 (degrees), and if the specified range is 90 degrees to 180 degrees, the upper limit of the specified range is 180 (degrees). Becomes If the phase difference θ is equal to or less than the upper limit (S21: No), the phase difference control unit 233 controls the phase difference θ to be large. At this time, the phase difference step width Δθ, which is the step width of the control target value (phase difference θ) controlled by the phase difference control unit 233, is determined by the determination unit 235 (S22). The process S22 for determining the phase difference step width Δθ will be described in detail in the section “(6.4) Resolution”. The phase difference control unit 233 increases the phase difference θ by the phase difference step width Δθ determined in step S22 (S23), and returns to the operation of step S1. By repeating these processes (S21 to S23), the control circuit 23 can gradually increase the phase difference θ by the phase difference step width Δθ to gradually reduce the output power, that is, approach the target value P1.

  When the phase difference θ exceeds the upper limit (S21: Yes), the control circuit 23 adjusts the output power by the frequency control unit 232 next. Specifically, the frequency control unit 232 compares the operating frequency f1 of the inverter circuit 21 (that is, the frequency of the first drive signals G1 to G4) with the upper limit value fmax of the permitted frequency band F1 (S24). If the operating frequency f1 is lower than the upper limit value fmax (S24: Yes), the frequency control unit 232 controls to increase the operating frequency f1 of the inverter circuit 21. At this time, the frequency step width Δf, which is the step width of the control target value (operating frequency f1) controlled by the frequency control unit 232, is determined by the determination unit 235 (S25). The process S25 of determining the frequency step width Δf will be described in detail in the section “(6.4) Resolution”. The frequency control unit 232 increases the operating frequency f1 of the inverter circuit 21 by the frequency step width Δf determined in the processing S25 (S26), and returns to the operation of the processing S1. By repeating these processes (S24 to S26), the control circuit 23 can gradually increase the operating frequency f1 by the frequency step width Δf to make the output power gradually smaller, that is, closer to the target value P1.

  When the operating frequency f1 becomes equal to or higher than the upper limit value fmax in the frequency control (S24: No), the control circuit 23 determines that an error has occurred (S27) and ends the output power control.

  By the way, during the period when the control circuit 23 adjusts the output power by the frequency control unit 232, that is, during the period when the frequency control is performed, all the correction switch elements Q5 to Q8 of the power correction circuit 22 are fixed to ON. Preferably, the function of the power correction circuit 22 is disabled. As a result, the non-contact power supply device 2 is in a state equivalent to “(4.1) No power correction circuit” (see the column “(4) Basic operation”). In this case, the control circuit 23 sets the phase difference θ to the initial value of 360 (degrees) before starting the adjustment of the output power by the phase difference control, and starts the operation of the power correction circuit 22. According to this configuration, when the magnitude of the output power reaches the target value P1 only by the frequency control, the control circuit 23 does not operate the power correction circuit 22, so that the efficiency by the power correction circuit 22 (power conversion efficiency) Can be avoided.

  When the control circuit 23 adjusts the output power by the phase difference control, the phase difference θ is gradually reduced from the initial value (the upper limit of the specified range) using the upper limit of the specified range as an initial value. This is not an essential configuration for the contactless power supply device 2. For example, when the specified range is 270 degrees to 360 degrees, the control circuit 23 sets the value exceeding the upper limit of the specified range (for example, 370 degrees) as an initial value, and gradually reduces the phase difference θ from this initial value. May be. Alternatively, if the specified range is in the range of 270 degrees to 360 degrees, the control circuit 23 sets the phase difference θ gradually from this initial value with a value (for example, 315 degrees) smaller than the upper limit of the specified range as an initial value. It may be smaller. In any case, in a region between the change start point existing in the specified range and the lower limit of the specified range, the output power of the non-contact power feeding device 2 changes according to the change in the phase difference θ.

(6.4) Resolution Assuming that the step width (frequency step width Δf, phase difference step width Δθ) of the control target value to be controlled when adjusting the output power is constant, the following problem occurs. There is.

  As a first problem, when the output power of the non-contact power supply device 2 is far from the target value P1, the time required for the process for bringing the output power close to the target value P1 may be long. That is, if the step size of the control target value is constant, even if the output power of the non-contact power supply device 2 is far from the target value P1, the control target value is changed by the step size and the output power is gradually increased to the target value P1. , The process may take time. In particular, when the non-contact power receiving device 3 charges the storage battery 4 after the output power reaches the target value P1, the time required to start charging the storage battery 4 increases.

  As a second problem, in order to solve the first problem, when the step width of the control target value is set to a relatively large value, ripple occurs due to the output power fluctuating near the target value P1. And the output power may greatly exceed the target value P1. In other words, if the step size of the control target value is a relatively large value, it is possible to reduce the time required for the process of bringing the output power close to the target value P1, but it is possible to reduce the occurrence of ripples and the target value P1 in the output power. May lead to a large excess of

  In short, when the step size of the control target value is relatively large, the amount of change in the output power by one frequency control or phase difference control becomes relatively large. If the change amount of the output power at this time is larger than the width of the allowable error range (± several%) of the target value P1, the output power exceeds the upper limit of the allowable error range and falls below the lower limit of the allowable error range. The state may be repeated. As a result, the output power continues to fluctuate near the target value P1, and “ripple” may occur in the power received by the non-contact power receiving device 3. In particular, when the non-contact power receiving device 3 applies a DC voltage to the storage battery 4, generation of ripples in the non-contact power receiving device 3 is considered from the viewpoint of the influence on the charge control of the storage battery 4 and the life of the storage battery 4. Is desired to be suppressed as much as possible.

  Similarly, if the step size of the control target value is relatively large, the amount of change in output power due to one frequency control or phase difference control becomes relatively large, so that when the output power is increased, the output power fluctuates rapidly. As a result, the output power may greatly exceed the target value P1. In particular, in the case where the non-contact power receiving device 3 applies a DC voltage to the storage battery 4, from the viewpoint of stress on the charging circuit of the storage battery 4, a large excess of the target value P1 in the output power is suppressed as much as possible. It is desired.

  Therefore, in the non-contact power supply device 2 according to the present embodiment, in order to solve these problems, the step width (frequency step width Δf, phase difference step width) of the control target value to be controlled when adjusting the output power is set. Δθ) is determined according to the voltage-current phase difference φ. That is, the frequency step Δf and the phase difference step Δθ are determined by the determination unit 235 as described in the section “(6.3) Overall Flow of Output Power Control”. Therefore, the frequency step width Δf and the phase difference step width Δθ are not constant, but change appropriately. In other words, in the non-contact power supply device 2, it is possible to appropriately change the resolution (resolution) of the control target value instead of making it constant. The change in resolution can be realized by, for example, high resolution (High Resolution) processing. As the resolution of the control target value increases, the step width (frequency step width Δf, phase difference step width Δθ) of the control target value decreases.

  Hereinafter, the processing (S12, S15, S22, S25 in FIG. 9) for determining the step size (frequency step Δf, phase difference step Δθ) of the control target value will be described in detail with reference to FIG. FIG. 10 is a flowchart illustrating a process when the control target value is the operating frequency f1 as an example, that is, when the frequency step width Δf is determined.

  First, the determination unit 235 compares the voltage-current phase difference φ acquired by the acquisition unit 234 with the first threshold Th1 (S31). That is, at the time of frequency control, the frequency control unit 232 changes the operating frequency f1 as the control target value by the frequency step width Δf. At this time, as the operating frequency f1 actually changes by the frequency step width Δf, the output power and the voltage / current phase difference φ that has a correlation with the output power change as needed. The acquisition unit 234 thus acquires the voltage / current phase difference φ that changes as needed with the frequency control, and the determination unit 235 determines whether the voltage / current phase difference φ is equal to or smaller than the first threshold Th1. (S31).

  If the voltage-current phase difference φ is not equal to or smaller than the first threshold Th1 (S31: No), the determination unit 235 applies the first value α1 as the frequency step width Δf (S32). In other words, if the condition that the voltage-current phase difference φ is larger than the first threshold value Th1 is satisfied, the determination unit 235 sets the frequency step width Δf to the first value α1.

  On the other hand, if the voltage-current phase difference φ is equal to or smaller than the first threshold Th1 (S31: Yes), the determining unit 235 determines a second threshold Th2 smaller than the first threshold Th1 and the voltage-current phase difference φ. A comparison is made (S33). That is, the determination unit 235 determines whether the voltage-current phase difference φ is equal to or smaller than the second threshold Th2 (<Th1) (S33). If the voltage-current phase difference φ is not equal to or smaller than the second threshold value Th2 (S33: No), the determination unit 235 applies the second value α2 smaller than the first value α1 as the frequency step width Δf (S34). . In other words, if the condition that the voltage-current phase difference φ is equal to or smaller than the first threshold value Th1 and larger than the second threshold value Th2 is satisfied, the determination unit 235 sets the frequency step width Δf to the second value α2 (<α1 ).

  If the voltage-current phase difference φ is equal to or smaller than the second threshold value Th2 (S33: Yes), the determination unit 235 applies a third value α3 smaller than the second value α2 as the frequency step width Δf (S35). . In other words, if the condition that the voltage-current phase difference φ is equal to or smaller than the second threshold value Th2 is satisfied, the determination unit 235 sets the frequency step width Δf to the third value α3 (<α2).

  That is, the determination unit 235 determines the frequency step width Δf according to the voltage / current phase difference φ such that the smaller the voltage / current phase difference φ, the smaller the frequency step width Δf. In other words, since the present embodiment is based on the premise that the inverter circuit 21 operates in the slow mode, as the voltage-current phase difference φ increases, that is, as the operation mode of the inverter circuit 21 becomes farther from the fast mode, The determination unit 235 increases the frequency step width Δf. Thus, at the time of frequency control, the frequency step width Δf switches between at least two (here, three) frequency step widths Δf.

  Here, the first threshold value Th1, the second threshold value Th2, the first value α1, the second value α2, and the third value α3 are stored in the memory of the control circuit 23 (microcomputer). Further, when the voltage / current phase difference φ is not determined, for example, immediately after the start of the frequency control, the determining unit 235 sets the voltage / current phase difference φ to a predetermined value (for example, 360 degrees) and determines the step size. Alternatively, the step size may be set to a predetermined value (for example, a first value α1).

  FIG. 11 shows an example in which the frequency step width Δf switches from the first value α1 to the second value α2 during frequency control. In FIG. 11, the horizontal axis represents the frequency (the operating frequency f1 of the inverter circuit 21), and the vertical axis represents the output power of the contactless power supply device 2, and represents the relationship between the operating frequency f1 and the output power during frequency control. In the example of FIG. 11, in the frequency band Fa equal to or higher than the switching frequency fc1, the voltage-current phase difference φ is larger than the first threshold Th1. On the other hand, in the frequency band Fb equal to or lower than the switching frequency fc1, the voltage-current phase difference φ is equal to or smaller than the first threshold Th1 and larger than the second threshold Th2. Therefore, in gradually lowering the operating frequency f1, the frequency control unit 232 applies the first value α1 as the frequency step width Δf up to the switching frequency fc1, and is smaller than the first value α1 below the switching frequency fc1. The second value α2 is applied as the frequency step width Δf. In short, the frequency control unit 232 lowers the operating frequency f1 by the first value α1 at points p1 to p8 in FIG. 11 and decreases the operating frequency f1 by the second value α2 (< α1). As a result, in the frequency band Fb equal to or lower than the switching frequency fc1, the frequency step width Δf is smaller than that in the frequency band Fa equal to or higher than the switching frequency fc1, and the rate of change of the magnitude of the output power (ΔP / A rapid change in Δf) is suppressed.

  For example, the control circuit 23 (microcomputer) has a function of a PWM (Pulse Width Modulation) timer, and has a PWM signal of an operating frequency f1 corresponding to a value (cycle register value) set in a cycle register of the PWM timer. Is output. At this time, if the step width of the cycle register value is “n / 255” (n is an integer), by adjusting “n” in the range of “1” to “255”, the step width of the operating frequency f1 (frequency The step width Δf) changes. As an example, it is assumed that the clock frequency of the CPU of the control circuit 23 (microcomputer) is 80 [MHz] and the initial value of the operating frequency f1 is 90 [kHz]. In this case, assuming that the cycle register value changes from “444” in steps of “1” (= 255/255), the operating frequency f1 becomes 89.9776 [kHz] when the cycle register value becomes “445”. At this time, the frequency step width Δf is 202.4 [Hz], and the resolution is the lowest. On the other hand, assuming that the cycle register value changes from "444" in steps of "1/255", when the cycle register value becomes "444 + 1/255", the operating frequency f1 becomes 89.9992 [kHz]. At this time, the frequency step width Δf is 0.8 [Hz], and the resolution is the highest. That is, the frequency step width Δf (0.8 [Hz]) when the resolution is the highest is 1/255 of the frequency step width Δf (202.4 [Hz]) when the resolution is the lowest.

  Up to this point, the case where the control target value is the operating frequency f1, that is, the operation of the determination unit 235 at the time of frequency control has been described, but the same applies to the case where the control target value is the phase difference θ, that is, at the time of the phase difference control. , The determination unit 235 determines the step width of the control target value (phase difference step width Δθ). That is, the determination unit 235 determines the phase difference step width Δθ according to the voltage / current phase difference φ such that the smaller the voltage / current phase difference φ, the smaller the phase difference step width Δθ. As a result, even during the phase difference control, the phase difference step width Δθ is switched between at least the binary phase difference step widths Δθ.

  As an example, when the voltage-current phase difference φ is larger than the threshold value, the determination unit 235 sets the phase difference step width Δθ to 0.4054 [degrees] and minimizes the resolution. In this case, the phase difference θ changes in steps of 0.4054 [degrees]. On the other hand, when the voltage-current phase difference φ is equal to or smaller than the threshold value, the determination unit 235 sets the phase difference step width Δθ to 0.0016 [degrees] and maximizes the resolution. In this case, the phase difference θ changes in steps of 0.0016 [degrees]. That is, the phase difference step width Δθ (0.0016 [degree]) when the resolution is the highest is 1/255 of the phase difference step width Δθ (0.4054 [degree]) when the resolution is the lowest.

(7) Startup Process The contactless power supply device 2 of the present embodiment soft-starts the inverter circuit 21 at the time of startup when the inverter circuit 21 and the power correction circuit 22 start operating, as described below.

  The control circuit 23 changes the duty ratio of the first drive signals G1 to G4 for controlling the conversion switching elements Q1 to Q4 from 0 (zero) to a predetermined value (for example, 0.5) when the inverter circuit 21 is started. By gradually raising, the soft start of the inverter circuit 21 is realized. As a result, a sudden change in the voltage or current input to the non-contact power supply device 2 is suppressed, and the stress applied to the circuit element can be reduced. Hereinafter, the process in which the control circuit 23 changes the duty ratio of the first drive signals G1 to G4 to soft start the inverter circuit 21 in this manner is referred to as “startup process”.

  In the contactless power supply device 2 of the present embodiment, while the control circuit 23 performs the start-up process, all the correction switch elements Q5 to Q8 are fixed to be ON for the power correction circuit 22, and the function of the power correction circuit 22 Disable. As a result, the non-contact power supply device 2 is in a state equivalent to “(4.1) No power correction circuit” (see the column “(4) Basic operation”).

  When the start-up processing ends, that is, when the duty ratio of the first drive signals G1 to G4 reaches a predetermined value (for example, 0.5), the control circuit 23 also starts the operation of the power correction circuit 22. In short, the control circuit 23 starts controlling the correction switch elements Q5 to Q8 with the second drive signals G5 to G8 after the end of the startup processing. Thereby, the non-contact power supply device 2 is in a state equivalent to “(4.2) With power correction circuit” (see the column of “(4) Basic operation”). At this time, if the prescribed range is in the range of 270 degrees to 360 degrees, the control circuit 23 sets the phase difference θ to the initial value of 360 (degrees), and starts the operation of the power correction circuit 22.

(8) Modifications The above embodiment is merely an example of the present invention, and the present invention is not limited to the above embodiment, and departs from the technical idea of the present invention even in other embodiments. Various changes can be made depending on the design and the like as long as they are not within the range. Hereinafter, modified examples of the above embodiment will be listed.

  The power correction circuit 22 is not limited to the configuration using the four correction switch elements Q5 to Q8 as in the above-described embodiment, but is configured using two correction switch elements Q9 and Q10, for example, as shown in FIG. It may be. In the power correction circuit 22 shown in FIG. 12, each of the correction switch elements Q9 and Q10 is a semiconductor switch element having a double gate structure having two gates. The first correction switch element Q9 is electrically connected in series with the correction capacitor C1. The second correction switch element Q10 is electrically connected in parallel to a series circuit of the correction switch element Q9 and the correction capacitor C1. A second drive signal G7 and a second drive signal G8 are input to the two gates of the correction switch element Q9, respectively. The second drive signal G5 and the second drive signal G6 are input to the two gates of the correction switch element Q10, respectively. The power correction circuit 22 illustrated in FIG. 12 functions equivalently to the power correction circuit 22 illustrated in FIG. 1 in which two correction switch elements Q9 and Q10 are controlled by the second drive signals G5 to G8.

  The load to which the output power is supplied (that is, supplied) from the non-contact power supply device 2 in a non-contact manner is not limited to the electric vehicle, but includes an electric device including a storage battery such as a mobile phone or a smartphone, or a storage battery. There may be no electrical equipment such as lighting equipment.

  Further, the transmission system of the output power from the non-contact power supply device 2 to the non-contact power receiving device 3 is not limited to the above-described magnetic field resonance system, and may be, for example, an electromagnetic induction system, a microwave transmission system, or the like.

  Further, each of the conversion switch elements Q1 to Q4 and each of the correction switch elements Q5 to Q8 may be formed of another semiconductor switching element such as a bipolar transistor or an IGBT (Insulated Gate Bipolar Transistor).

  Further, each of the diodes D1 to D4 is not limited to a parasitic diode of each of the conversion switching elements Q1 to Q4, and may be externally attached to each of the conversion switching elements Q1 to Q4. Similarly, the diodes D5 to D8 are not limited to the parasitic diodes of the correction switch elements Q5 to Q8, and may be externally connected to the correction switch elements Q5 to Q8.

  Further, in the above embodiment, it is assumed that the inverter circuit 21 operates in the delay mode. However, the present invention is not limited to this example. It may operate in mode. In this case, the acquisition unit 234 acquires the delay of the phase of the output voltage with respect to the output current of the inverter circuit 21 as a voltage-current phase difference. Also in this case, it is preferable that the determination unit 235 determines the step size according to the voltage / current phase difference so that the smaller the voltage / current phase difference, the smaller the step size of the control target value.

  Further, the measuring unit 24 is not limited to the configuration provided separately from the control circuit 23, and may be provided integrally with the control circuit 23. Further, the measuring unit 24 only needs to be able to measure the current flowing through the primary coil L1. Therefore, for example, the current sensor 25 is not limited to between the primary coil L1 and the second primary capacitor C12, and may be on the path of the current flowing through the primary coil L1. Furthermore, the measurement unit 24 is not limited to a configuration in which the current (primary current I1) flowing through the primary coil L1 is directly measured. For example, the current (secondary current) measured by the non-contact power receiving device 3 Current). That is, the secondary-side current corresponding to the current flowing through the primary-side coil L1 flows through the secondary-side resonance circuit of the non-contact power receiving device 3. Therefore, for example, the non-contact power receiving device 3 measures the secondary current, transmits the value from the non-contact power receiving device 3 to the non-contact power feeding device 2, and the measuring unit 24 outputs the current ( The primary current I1) may be measured indirectly.

  Inverter circuit 21 may be a voltage-type inverter capable of converting a DC voltage to an AC voltage and outputting the converted voltage, and is not limited to a full-bridge inverter circuit in which four conversion switch elements Q1 to Q4 are connected in a full-bridge manner. The inverter circuit 21 may be, for example, a half-bridge inverter circuit.

  Further, the control circuit 23 performs a search mode for estimating a coupling coefficient between the primary coil L1 and the secondary coil L2 in addition to the normal mode (including the start-up process) for performing the output power control as described above. You may have. In the search mode, the control circuit 23 is configured to gradually change the phase difference θ within a specified range and to estimate a coupling coefficient based on a change in the magnitude of the coil current accompanying a change in the phase difference θ. ing. The “coil current” here is a current flowing through the primary coil L1 (that is, the primary current I1), and is measured by the measurement unit 24.

  That is, it has been confirmed that the relationship between the phase difference θ and the coil current changes depending on the coupling coefficient between the primary coil L1 and the secondary coil L2. Specifically, when the phase difference θ is gradually reduced from the upper limit of the specified range (here, 360 degrees), the phase difference θ corresponding to the inflection point at which the coil current starts to increase differs depending on the coupling coefficient. I have. As the coupling coefficient between the primary coil L1 and the secondary coil L2 increases, the phase difference θ at which the coil current starts to increase decreases. Therefore, if the relationship between the phase difference θ and the coil current is used, the control circuit 23 can estimate the coupling coefficient from the change in the measurement value (the magnitude of the coil current) accompanying the change in the phase difference θ. .

  Further, the control circuit 23 can further estimate the resonance characteristics from the coupling coefficient. As a result, the control circuit 23 can estimate the frequency range in which the inverter circuit 21 operates in the slow mode (that is, does not enter the fast mode) for the operating frequency f1 of the inverter circuit 21, for example. Thereby, the control circuit 23 can set the initial value of the operating frequency f1 when starting the operation in the normal mode within the frequency range in which the inverter circuit 21 operates in the slow mode.

  Further, the power correction circuit 22 may be operating during a period in which the control circuit 23 adjusts the output power in the frequency control unit 232, that is, a period in which the frequency control is performed. In this case, the phase difference θ is set to 360 (degree), which is the initial value. In this case, the frequency control unit 232 changes the frequencies of the second drive signals G5 to G8 together with the frequencies of the first drive signals G1 to G4.

  In addition, the power correction circuit 22 only needs to be electrically connected between at least one of the pair of output points 213 and 214 and the primary side coil L1, and may be connected between the output point 213 and the primary side coil L1. Thus, it may be electrically connected between the output point 214 and the primary coil L1.

  In addition, the functions of the element control unit 231, the frequency control unit 232, the phase difference control unit 233, the acquisition unit 234, and the determination unit 235 are not limited to the configuration in which all of the functions are implemented in the control circuit 23. It may be provided separately from the control circuit 23.

  The determining unit 235 is configured to determine the step size only for one of the operating frequency f1 and the phase difference θ that are the control target values that are the control targets when adjusting the output power. Is also good. That is, the determination unit 235 determines the step size of the control target value including at least one of the frequency controlled by the frequency control unit 232 and the phase difference θ controlled by the phase difference control unit 233 according to the voltage / current phase difference φ. May be determined. For example, when the determination unit 235 determines only the step width (frequency step width Δf) for the operating frequency f1, the step width (phase difference step width Δθ) for the phase difference θ is a fixed value.

  Further, the determining unit 235 is not limited to the configuration in which the step width is switched between the binary step widths, and may switch between the three or more step widths. Further, the determining unit 235 may set, for example, a value obtained by multiplying the voltage / current phase difference φ by a predetermined coefficient as the step size. In this case, the step width is adjusted steplessly, that is, continuously.

  Furthermore, when determining the step width (phase difference step width Δθ) for the phase difference θ, the determination unit 235 may directly determine (specify) the phase difference step width Δθ or indirectly determine the phase difference step width Δθ. The phase difference step width Δθ may be determined. That is, the phase difference θ is a delay of the phase of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4. Therefore, for example, if any one of the frequency of the first drive signals G1 to G4 (operating frequency f1) and the delay time of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4 changes, the phase difference The step width Δθ changes. Therefore, the determination unit 235 may directly change the phase difference step width Δθ, but is not limited thereto, and for example, changes the delay time of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4. Thus, the phase difference step width Δθ may be indirectly changed.

(9) Summary As described above, the non-contact power supply device 2 includes the inverter circuit 21, the primary coil L1, the power correction circuit 22, the element control unit 231, the frequency control unit 232, and the phase difference control unit. 233, an acquisition unit 234, and a determination unit 235. The inverter circuit 21 has a plurality of conversion switch elements Q1 to Q4 electrically connected between a pair of input points 211 and 212 and a pair of output points 213 and 214. The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 into an AC voltage by switching the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214. The primary coil L1 is electrically connected between the pair of output points 213 and 214, and supplies an output power to the secondary coil L2 in a non-contact manner when an AC voltage is applied. The power correction circuit 22 is electrically connected between at least one of the pair of output points 213 and 214 and the primary coil L1, and has a correction capacitor C1 and a plurality of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8. The frequency controller 232 adjusts the magnitude of the output power by controlling the frequencies of the first drive signals G1 to G4 to discretely change. The phase difference control unit 233 controls the phase difference θ, which is the phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to discretely change the phase difference θ, thereby reducing the magnitude of the output power. Adjust. The acquisition unit 234 acquires a phase difference between the output voltage and the output current of the inverter circuit 21 as a voltage / current phase difference φ. The determination unit 235 determines the step size of the control target value including at least one of the frequency controlled by the frequency control unit 232 and the phase difference θ controlled by the phase difference control unit 233 according to the voltage-current phase difference φ. I do.

  According to this configuration, the contactless power supply device 2 adjusts the magnitude of the output power by at least one of the control of the frequency by the frequency control unit 232 and the control of the phase difference θ by the phase difference control unit 233. Is possible. Therefore, even if the relative positional relationship between the primary coil L1 and the secondary coil L2 changes and the coupling coefficient between the primary coil L1 and the secondary coil L2 changes, the wireless power supply device In the case of 2, the power can be supplied stably by adjusting the magnitude of the output power. Moreover, in the contactless power supply device 2, the step size of the control target value (at least one of the frequency and the phase difference) to be controlled when adjusting the output power is determined by the determining unit 235 by the voltage / current phase difference φ. Is determined according to. Therefore, for example, when the voltage-current phase difference φ is relatively large, that is, when the output power is far from the target value P1 determined by the non-contact power receiving device 3, the step size of the control target value is made relatively large, The time required for the process for bringing the output power closer to the target value P1 can be reduced. When the voltage-current phase difference φ is relatively small, that is, when the output power approaches the target value P1, the output power fluctuates around the target value P1 by making the step size of the control target value relatively small. And the output power greatly exceeds the target value P1. Therefore, the contactless power supply device 2 has an advantage that stable power supply is possible regardless of the relative positional relationship between the primary coil L1 and the secondary coil L2.

  Further, the inverter circuit 21 operates in the delay mode in which the phase of the output current is delayed with respect to the phase of the output voltage, and the acquisition unit 234 determines that the delay of the phase of the output current with respect to the output voltage of the inverter circuit 21 is a voltage current. It is preferable to be configured to obtain the phase difference φ. According to this configuration, the switching operation of the inverter circuit 21 is soft switching, the switching loss of the conversion switching elements Q1 to Q4 can be reduced, and stress is not easily applied to the conversion switching elements Q1 to Q4. However, this configuration is not indispensable to the non-contact power supply device 2, and for example, the inverter circuit 21 may operate in a phase advance mode in which the phase of the output current is ahead of the phase of the output voltage.

  Further, it is preferable that the determination unit 235 is configured to determine the step size according to the voltage / current phase difference φ so that the step size becomes smaller as the voltage / current phase difference φ becomes smaller. According to this configuration, when the voltage-current phase difference φ is relatively large, the step size of the control target value is relatively large, and the time required for processing for bringing the output power close to the target value P1 is reduced. Is possible. On the other hand, when the voltage-current phase difference φ is relatively small, the step size of the control target value is relatively small, and it is possible to suppress the occurrence of ripples and the output power from greatly exceeding the target value P1. However, this configuration is not indispensable to the non-contact power supply device 2. For example, the determining unit 235 may reduce the step size of the control target value as the voltage-current phase difference φ increases.

  Further, the phase difference control unit 233 controls the phase difference θ when the magnitude of the output power adjusted by the frequency control unit 232 is less than the predetermined target value P1, thereby reducing the magnitude of the output power. It is preferable that the configuration is such that the value approaches the target value P1. According to this configuration, the contactless power supply device 2 can adjust the output power in two stages of the frequency control and the phase difference control, so that the output power adjustment range can be widened. However, this configuration is not essential to the non-contact power supply device 2. For example, the phase difference control unit 233 controls the phase difference θ before the frequency control unit 232 adjusts the magnitude of the output power. The magnitude of the output power may be adjusted.

  The non-contact power transmission system 1 includes a non-contact power supply device 2 and a non-contact power receiving device 3 having a secondary coil L2. The non-contact power receiving device 3 is configured to be supplied with output power from the non-contact power feeding device 2 in a non-contact manner. According to this configuration, there is an advantage that stable power can be supplied regardless of the relative positional relationship between the primary coil L1 and the secondary coil L2.

  When the control circuit 23 has a microcomputer as a main configuration as in the present embodiment, a program recorded in a memory of the microcomputer causes the control circuit 23 to function as the control circuit 23. That is, the program is a program for causing a computer used for the non-contact power supply device 2 to function as the element control unit 231, the frequency control unit 232, the phase difference control unit 233, the acquisition unit 234, and the determination unit 235. The non-contact power supply device 2 includes an inverter circuit 21, a primary coil L1, and a power correction circuit 22.

  The inverter circuit 21 has a plurality of conversion switch elements Q1 to Q4 electrically connected between a pair of input points 211 and 212 and a pair of output points 213 and 214. The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 into an AC voltage by switching the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214. The primary coil L1 is electrically connected between the pair of output points 213 and 214, and supplies an output power to the secondary coil L2 in a non-contact manner when an AC voltage is applied. The power correction circuit 22 is electrically connected between at least one of the pair of output points 213 and 214 and the primary coil L1, and has a correction capacitor C1 and a plurality of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8. The frequency controller 232 adjusts the magnitude of the output power by controlling the frequencies of the first drive signals G1 to G4 to discretely change. The phase difference control unit 233 controls the phase difference θ, which is the phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to discretely change the phase difference θ, thereby reducing the magnitude of the output power. Adjust. The acquisition unit 234 acquires a phase difference between the output voltage and the output current of the inverter circuit 21 as a voltage / current phase difference φ. The determination unit 235 determines the step size of the control target value including at least one of the frequency controlled by the frequency control unit 232 and the phase difference θ controlled by the phase difference control unit 233 according to the voltage-current phase difference φ. I do. According to this program, a function equivalent to that of the non-contact power supply device 2 can be realized without using the dedicated control circuit 23, and the relative position relationship between the primary coil L1 and the secondary coil L2 depends on the relative position. Therefore, there is an advantage that stable power supply is possible.

  In addition, by controlling the non-contact power supply device 2 including the inverter circuit 21, the primary coil L1, and the power correction circuit 22 by the following control method, the non-contact power supply device 2 can be used without using the dedicated control circuit 23. A function equivalent to that of the contact power supply device 2 can be realized. That is, the control method of the wireless power supply device 2 includes an element control step, a frequency control step, a phase difference control step, an acquisition step, and a determination step. The non-contact power supply device 2 includes an inverter circuit 21, a primary coil L1, and a power correction circuit 22.

  The inverter circuit 21 has a plurality of conversion switch elements Q1 to Q4 electrically connected between a pair of input points 211 and 212 and a pair of output points 213 and 214. The inverter circuit 21 converts a DC voltage applied to the pair of input points 211 and 212 into an AC voltage by switching the plurality of conversion switch elements Q1 to Q4, and outputs the AC voltage from the pair of output points 213 and 214. The primary coil L1 is electrically connected between the pair of output points 213 and 214, and supplies an output power to the secondary coil L2 in a non-contact manner when an AC voltage is applied. The power correction circuit 22 is electrically connected between at least one of the pair of output points 213 and 214 and the primary coil L1, and has a correction capacitor C1 and a plurality of correction switch elements Q5 to Q8. The power correction circuit 22 charges and discharges the correction capacitor C1 by switching the plurality of correction switch elements Q5 to Q8. The element control unit 231 controls the plurality of conversion switch elements Q1 to Q4 with the first drive signals G1 to G4, and controls the plurality of correction switch elements Q5 to Q8 with the second drive signals G5 to G8. In the frequency control step, the magnitude of the output power is adjusted by controlling the frequencies of the first drive signals G1 to G4 to discretely change. In the phase difference control step, the magnitude of the output power is adjusted by controlling the phase difference θ, which is the phase delay of the second drive signals G5 to G8 with respect to the first drive signals G1 to G4, to be discretely changed. I do. In the obtaining step, a phase difference between the output voltage and the output current of the inverter circuit 21 is obtained as a voltage-current phase difference φ. In the determination step, the step width of the control target value, which is at least one of the frequency controlled in the frequency control step and the phase difference θ controlled in the phase difference control step, is determined according to the voltage-current phase difference φ. According to this control method, there is an advantage that stable power supply is possible regardless of the relative positional relationship between the primary coil L1 and the secondary coil L2.

REFERENCE SIGNS LIST 1 contactless power transmission system 2 contactless power supply device 3 contactless power receiving device 21 inverter circuit 211, 212 pair of input points 213, 214 pair of output points 22 power correction circuit 231 element controller 232 frequency controller 233 phase difference controller 234 Acquisition unit 235 Determination unit C1 Correction capacitor G1 to G4 First drive signal G5 to G8 Second drive signal L1 Primary coil L2 Secondary coil Q1 to Q4 Switch element for conversion Q5 to Q8 Switch element for correction Q9, Q10 Correction switch element f1 (operation) frequency θ phase difference φ voltage-current phase difference Δf (frequency) step width Δθ (phase difference) step width

Claims (7)

A plurality of conversion switch elements electrically connected between the pair of input points and the pair of output points, and a direct current applied to the pair of input points by switching of the plurality of conversion switch elements; An inverter circuit that converts a voltage into an AC voltage and outputs the AC voltage from the pair of output points;
A primary coil electrically connected between the pair of output points and supplying output power to the secondary coil in a non-contact manner by applying the AC voltage;
A correction capacitor and a plurality of correction switch elements electrically connected between at least one of the pair of output points and the primary side coil, wherein the correction is performed by switching the plurality of correction switch elements. Power correction circuit for charging and discharging the capacitor for
An element control unit that controls the plurality of conversion switch elements with a first drive signal and controls the plurality of correction switch elements with a second drive signal;
A frequency control unit that adjusts the magnitude of the output power by controlling the frequency of the first drive signal to change discretely;
A phase difference control unit that adjusts the magnitude of the output power by controlling the phase difference, which is the delay of the phase of the second drive signal with respect to the first drive signal, to change discretely;
An acquisition unit that acquires a phase difference between an output voltage and an output current of the inverter circuit as a voltage-current phase difference,
A determining unit that determines a step width of a control target value including at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit in accordance with the voltage-current phase difference; A non-contact power supply device comprising: The inverter circuit is operating in a delayed mode in which the phase of the output current is delayed with respect to the phase of the output voltage,
The wireless power supply device according to claim 1, wherein the acquisition unit is configured to acquire a phase delay of an output current with respect to an output voltage of the inverter circuit as the voltage / current phase difference. The determining unit includes:
The configuration is such that the step size is determined according to the voltage / current phase difference so that the step size becomes smaller as the voltage / current phase difference becomes smaller. A non-contact power supply device according to claim 1. The phase difference controller,
When the magnitude of the output power after adjustment by the frequency control unit is less than a predetermined target value, by controlling the phase difference, the magnitude of the output power is made to approach the target value. The non-contact power supply device according to any one of claims 1 to 3, wherein: A plurality of conversion switch elements electrically connected between the pair of input points and the pair of output points, and a direct current applied to the pair of input points by switching of the plurality of conversion switch elements; An inverter circuit that converts a voltage into an AC voltage and outputs the AC voltage from the pair of output points;
A primary coil electrically connected between the pair of output points and supplying output power to the secondary coil in a non-contact manner by applying the AC voltage;
A correction capacitor and a plurality of correction switch elements electrically connected between at least one of the pair of output points and the primary side coil, wherein the correction is performed by switching the plurality of correction switch elements. Computer used for a non-contact power supply device having a power correction circuit for charging and discharging the capacitor for
An element control unit that controls the plurality of conversion switch elements by a first drive signal and controls the plurality of correction switch elements by a second drive signal;
A frequency control unit that adjusts the magnitude of the output power by controlling the frequency of the first drive signal to change discretely;
A phase difference controller that adjusts the magnitude of the output power by controlling the phase difference, which is a phase delay of the second drive signal with respect to the first drive signal, to be discretely changed;
An acquisition unit that acquires a phase difference between an output voltage and an output current of the inverter circuit as a voltage-current phase difference,
And determining the step width of a control target value including at least one of the frequency controlled by the frequency control unit and the phase difference controlled by the phase difference control unit in accordance with the voltage / current phase difference. Department,
Program to function as A plurality of conversion switch elements electrically connected between the pair of input points and the pair of output points, and a direct current applied to the pair of input points by switching of the plurality of conversion switch elements; An inverter circuit that converts a voltage into an AC voltage and outputs the AC voltage from the pair of output points;
A primary coil electrically connected between the pair of output points and supplying output power to the secondary coil in a non-contact manner by applying the AC voltage;
A correction capacitor and a plurality of correction switch elements electrically connected between at least one of the pair of output points and the primary side coil, wherein the correction is performed by switching the plurality of correction switch elements. A power correction circuit for charging and discharging the capacitor for the non-contact power supply device, comprising:
An element control step of controlling the plurality of conversion switch elements by a first drive signal and controlling the plurality of correction switch elements by a second drive signal;
A frequency control step of adjusting the magnitude of the output power by controlling the frequency of the first drive signal to change discretely;
A phase difference control step of adjusting the magnitude of the output power by controlling so as to discretely change a phase difference that is a delay of the phase of the second drive signal with respect to the first drive signal;
An obtaining step of obtaining a phase difference between an output voltage and an output current of the inverter circuit as a voltage-current phase difference;
A determining step of determining a step size of a control target value including at least one of the frequency controlled in the frequency control step and the phase difference controlled in the phase difference control step, according to the voltage-current phase difference; ,
A method for controlling a non-contact power supply device, comprising: A non-contact power feeding device according to any one of claims 1 to 4, and a non-contact power receiving device having the secondary coil,
The non-contact power transmission system, wherein the non-contact power receiving device is configured to supply the output power from the non-contact power supply device in a non-contact manner.

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