JP2017112751A - Motor driving device and washing machine or washing and drying machine in which the same is incorporated - Google Patents

Abstract

PROBLEM TO BE SOLVED: To perform control of limiting input current to any value in a motor driving device used for a washing machine or the like.SOLUTION: Control means 6 decomposes motor current of a motor 4 into a current component corresponding to a magnetic flux and a current component corresponding to a torque, performs vector control on the current components so as to obtain desired values respectively, and also performs speed control so that the motor rotation speed by rotor position detection means 4a is equal to an instructed rotation speed. In accordance with the current motor rotation speed, variation in the instructed rotation speed is moderated when the motor current exceeds a limit value of a predetermined table, and the input current can be prevented from being excessively large without limiting the motor current more than necessary.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a motor driving device used in a washing machine or the like that drives a motor by an inverter circuit, and a washing machine or a washing / drying machine using the same.

  In a motor drive device for driving a washing machine or the like, it is necessary to limit the AC input current so that it does not become excessive when an abnormality occurs. Conventionally, a microcomputer that performs drive control of an inverter circuit is insulated from an AC input current line and often has a different GND potential (for example, see Patent Document 1).

  In Patent Document 1, as an inexpensive method for detecting an overcurrent state in a circuit insulated from a microcomputer, a current detection resistor is inserted in series in a line where current is to be detected, and a voltage generated at both ends of the inserted resistor. Discloses a motor driving device and a washing machine including an overcurrent detection device for driving a photocoupler.

JP 2008-122226 A

  However, the above-described conventional overcurrent detection device of the motor drive device requires circuit components such as a current detection resistor and a DC voltage source, and also requires expensive components such as a photocoupler because the GND potential is different. There is a problem that the heat generation of the resistance portion and the loss at the detection resistance portion may increase, and a more inexpensive motor drive device cannot be configured.

  In order to solve the above-described problems, a motor drive device of the present invention includes a rectifier circuit connected to a power source, an inverter circuit that converts DC power of the rectifier circuit into AC power, and a washing and unloading that is driven by the inverter circuit. A brushless motor for driving a load such as a water tank, rotor position detecting means for detecting a rotor position of the brushless motor, current detecting means for detecting a motor current of the brushless motor, and control means for controlling the inverter circuit. The control means performs a vector control for controlling the motor current of the brushless motor into a current component corresponding to a magnetic flux and a current component corresponding to a torque and controlling each of them to a desired value; Speed control is performed so that the motor rotation speed by the position detection means becomes the command rotation speed, and is detected by the current detection means. Primary current determination means for referring to a reference value of the motor current, wherein the primary current determination means is a table in which a reference value of the motor current detected by the current detection means is set in advance according to the motor rotation speed. When the limit value is exceeded, the change in the command rotational speed is made gradual.

  The motor drive device of the present invention can limit the primary current to an arbitrary desired range without using an expensive primary current detector, and can prevent the input current from becoming excessive. Further, since the motor current limit value is changed according to the command rotation speed, the motor current is not limited more than necessary at low rotation, so that the rotation speed is not reduced due to insufficient torque even at low speed. In addition, since a primary current detector that requires more expensive parts than in the past is not used, a low-cost motor drive device can be provided.

FIG. 2 is a partial block circuit diagram of the motor drive device according to the first and second embodiments of the present invention. Operation time chart of the motor drive unit Flowchart of motor current limit control of the motor drive device Flowchart of motor drive subroutine of the motor drive device Flowchart of carrier signal interrupt subroutine of the motor drive device Flowchart of position signal interrupt subroutine of the motor drive device Characteristic diagram showing the function of the Iu limit table of the motor drive device Flow chart for creating a speed command for the motor drive device Circuit diagram in which the motor driving apparatus according to the third embodiment of the present invention is partly blocked. Flow chart at the time of table switching control by detecting the rectified voltage value of the motor drive device The figure which shows the example of a low electric potential state at the time of the rectification voltage value detection of the motor drive device The figure which shows the example of a standard electric potential state at the time of the rectification voltage value detection of the motor drive device The figure which shows the example of a high potential state at the time of the rectification voltage value detection of the motor drive device Characteristic diagram showing the function of the Iu limit table of the motor drive device

  A motor drive device according to a first aspect of the present invention is a rectifier circuit connected to a power source, an inverter circuit for converting DC power of the rectifier circuit into AC power, and driving a load such as a washing / dehydrating tub driven by the inverter circuit. A brushless motor, a rotor position detecting means for detecting a rotor position of the brushless motor, a current detecting means for detecting a motor current of the brushless motor, and a control means for controlling the inverter circuit. The motor control of the brushless motor is divided into a current component corresponding to the magnetic flux and a current component corresponding to the torque, and the vector control is performed to control the motor current to a desired value. The speed is controlled so that the number becomes the command rotation speed, and the reference value of the motor current detected by the current detection means is A primary current determination means for illuminating the reference current, and the primary current determination means is configured such that a reference value of the motor current detected by the current detection means exceeds a preset table limit value according to the motor speed. Is characterized in that the change in the command rotational speed is moderated.

  The motor drive device according to a second aspect of the present invention is the motor drive device according to the first aspect of the invention, wherein, in the primary current determination means, the limit value of the table has a term proportional to the square of the motor rotational speed and monotonously decreases. It is defined by a function.

  With this configuration, it is possible to provide a low-cost motor driving device that can prevent the input current from becoming excessive without limiting the motor current more than necessary without using an expensive primary current detection unit. .

  The motor drive device according to a third aspect of the invention is particularly the first or second aspect of the invention in which the motor current detected by the current detection means and the rotor position detected by the rotor position detection means are compared to induce the motor current. Current phase detection means for detecting the phase of the voltage, wherein the primary current determination means adds the motor current and the induced voltage detected by the current phase detection means to the current value of the motor current detected by the current detection means; A value obtained by multiplying the cosine of the phase is a reference value of the motor current.

With this configuration, even when the phase of the current is changed with respect to the induced voltage of the brushless motor, such as field-weakening control used when the brushless motor is operated at high speed, the current of the brushless motor is not limited more than necessary. You can drive.

  A motor drive device according to a fourth aspect of the present invention is the motor drive device according to any one of the first to third aspects, further comprising voltage detection means for detecting a DC voltage of the rectifier circuit and a plurality of the tables, wherein the control means is configured to detect the voltage. The voltage detected by the means is compared with a predetermined standard value, and the limit value of the table is switched by selectively applying the plurality of tables according to the magnitude relationship.

  With this configuration, even when the power supply voltage fluctuates, a low-cost motor that can prevent the input current from becoming excessive without limiting the motor current more than necessary without using an expensive primary current detector. A drive device can be provided.

  The motor drive device of the fifth invention is the motor drive device of the fifth invention, in particular, in the fourth invention, when the voltage detected by the voltage detection means is compared with a predetermined standard value and determined to be higher than that, The limit value of the table has a term proportional to the square of the motor rotation speed, and increases the coefficient of a monotonically decreasing function.

  With this configuration, even when the power supply voltage fluctuates high, it is possible to prevent the input current from becoming excessive without limiting the motor current more than necessary without using an expensive primary current detection unit. A motor drive device can be provided.

  The motor drive device of the sixth invention is the motor drive device of the sixth invention, in particular, in the fourth invention, when the voltage detected by the voltage detection means is compared with a predetermined standard value and determined to be a lower potential than that, The limit value of the table has a term that is proportional to the square of the motor rotation number, and the coefficient of the monotonically decreasing function is reduced.

  With this configuration, even when the power supply voltage fluctuates low, it is possible to prevent the input current from becoming excessive without limiting the motor current more than necessary without using an expensive primary current detection unit. A motor drive device can be provided.

  7th invention uses the motor drive device of any one of 1st-6th invention for a washing machine or a washing dryer.

  Hereinafter, embodiments for carrying out the present invention will be described with reference to the drawings. Note that the present invention is not limited to the embodiments.

  FIG. 1 is a circuit diagram in which the motor drive apparatus according to the first embodiment of the present invention is partly blocked. As shown in FIG. 1, an AC power source 1 (power source) applies an AC voltage to a rectifier circuit 2, and the rectifier circuit 2 converts the DC voltage into a DC voltage by a rectifier 20 and a capacitor 21 and applies the DC voltage to the inverter circuit 3.

  The inverter circuit 3 is constituted by a three-phase full-bridge inverter circuit composed of six power switching semiconductors and an antiparallel diode, and normally includes an insulated gate bipolar transistor (IGBT), an antiparallel diode, its driving circuit, and a protection circuit. It consists of an intelligent power module (IPM). The power switching semiconductor may be composed of a metal oxide semiconductor field effect transistor (MOSFET) or the like in addition to the IGBT. Since the configuration of the inverter circuit 3 is the same as a well-known one, detailed description thereof is omitted.

The motor 4 is connected to the output terminal of the inverter circuit 3 to drive a load such as a washing blade or a stirring blade (not shown) or a washing / dehydrating tub (not shown) of a washing machine / dryer. The motor 4 is constituted by a brushless motor, and the relative position (rotor position) between the permanent magnet and the stator constituting the rotor (rotor) is detected by the rotor position detecting means 4a. The rotor position detection means 4a is normally composed of three Hall ICs, and detects the output reference signals H1 to H3 at positions at every electrical angle of 60 degrees.

  The current detection means 5 detects motor currents Iu, Iv, and Iw of the motor 4, and normally uses shunt resistors 5a and 5b. It can also be detected by a DC current transformer that can be measured from a low frequency including a DC current, or an AC current transformer. In the case of a three-phase motor, a general method is to obtain a two-phase current (for example, Iu, Iv) and obtain the remaining one-phase current (Iw) from Kirchhoff's law (Iu + Iv + Iw = 0).

  The rotor position detection means 4a detects the rotor position based on the output reference signals H1 to H3, but calculates the rotor position from the motor phase current and the three-phase motor drive control voltage without using the Hall IC. It is also possible to use a detection method (not shown).

  The control means 6 comprises a microcomputer and an inverter control timer (PWM timer) built in the microcomputer, a high-speed A / D conversion circuit, a memory circuit (ROM, RAM), and the like, and from an output signal of the rotor position detection means 4a An electrical angle detector 60 for detecting an electrical angle, an output signal from the current detector 5 and a signal from the electrical angle detector 60, a current component Id (d-axis current) corresponding to magnetic flux and a current component (torque current) corresponding to torque. ) Three-phase / two-phase dq conversion means 61 that decomposes into Iq (q-axis current) and sine wave data (sin, cos data) necessary for conversion from the stationary coordinate system to the rotation coordinate system or reverse conversion Storage means 62, and two-phase for converting voltage component Vd corresponding to magnetic flux and voltage component Vq corresponding to torque into three-phase motor drive control voltages Vu, Vv and Vw. A three-phase dq inverse conversion unit 63, a 3-phase motor drive control voltage Vu, Vv, and PWM control means 64 for controlling the IGBT switching of the inverter circuit 3 according to Vw.

  Further, setting change means 65 for controlling the start, stop, rotation speed, braking, etc. of the motor 4 according to the stroke, rotation speed detection means 66 for detecting the rotation speed from the output signal of the rotor position detection means 4a, and rotation Torque for determining q-axis current command value Iqs of torque current Iq, which is a current component corresponding to torque, with reference to detected rotation speed N detected by number detecting means 66 and speed command Ns set by setting changing means 65 Current control means 67, d-axis (direct-axis) current command value Ids from setting change means 65, q-axis (quadture-axis) current command value Iqs from torque current control means 67, and three-phase / 2-phase dq The Id and Iq calculated by the conversion means 61 are respectively compared, and the voltage component Vd corresponding to the magnetic flux for controlling the motor current and the torque are supported. And a motor current control means 68 for calculating the pressure component Vq.

  According to the input (three-phase motor drive control voltages Vu, Vv, Vw) from the two-phase / three-phase dq reverse conversion means 63 obtained by reverse conversion from the voltage component Vd corresponding to the magnetic flux and the voltage component Vq corresponding to the torque. Thus, the PWM control means 64 outputs a control signal to the inverter circuit 3.

  Constant torque control is possible by performing feedback control so that the q-axis current Iq corresponding to the torque becomes the q-axis current command value Iqs. However, since the motor-induced voltage increases and the torque current Iq does not increase as the rotational speed increases, the q-axis current Iq can also be increased by so-called weakening magnetic flux control that increases the d-axis current Id according to the rotational speed. And torque can be increased.

FIG. 2 shows the waveform relationship of each part during operation of the motor driving device. The edge signals of the output reference signals H1, H2, and H3 of the rotor position detecting means 4a change every 60 degrees, and 360 degrees from each part state signal becomes 6 degrees. Divided angles can be identified. A high edge where the output reference signal H1 changes from low to high is shown as a reference electrical angle of 0 degree, and the U-phase winding induced voltage Ec of the motor 4 has a waveform delayed by 30 degrees from the output reference signal H1. Maximum efficiency is obtained when the phases of the U-phase motor current Iu and the U-phase winding induced voltage Ec are the same. The U-phase winding induced voltage Ec is the same axis as the q axis, and the d axis is delayed by 90 degrees. Since the q-axis current is in phase with the motor induced voltage phase, it is called torque current.

  In FIG. 2, the U-phase motor current Iu is slightly advanced from the U-phase winding induced voltage Ec, and the motor applied voltage Vu shows a waveform advanced 30 degrees from the U-phase winding induced voltage Ec. Vc is a sawtooth (or triangular wave) waveform carrier signal generated in the PWM control means 64, and Vu is a sinusoidal U-phase control voltage, which is a PWM signal U that compares the carrier signal Vc and the U-phase control voltage Vu. It is generated in the control means 64 and added as a control signal for the U-phase upper arm transistor of the inverter circuit 3. ck is a synchronizing signal of the carrier signal Vc, and is an interrupt signal when the carrier counter counts up and overflows.

  Coordinate conversion from the stationary coordinate system to the rotating coordinate system, that is, dq conversion, is performed with the electrical angle at which the rotor magnet axis of the motor 4 and the magnetic flux axis of the stator coincided as the d axis, and a reference electrical angle of 0 degrees. 60 detects electrical angles such as 30 degrees, 90 degrees, and 150 degrees from the output reference signals H1, H2, and H3 of the rotor position detecting means 4a, and obtains the electrical angle θ by estimation except every 60 degrees.

  In general, the current component corresponding to the direction of the magnetic flux of the rotor magnet is called a d-axis current Id. Since the permanent magnet magnetic flux and the field magnetic flux are coaxial and the permanent magnet is attracted to the field, the torque is It becomes zero. Also, the axis that has the same phase as the induced voltage phase and the maximum torque at an electrical angle of 90 degrees from the d axis is called the q axis, and is called the q axis current Iq because it is a current component corresponding to the torque. Further, increasing the d-axis current Id in the negative direction is equivalent to weakening the field magnetic flux on the d-axis, and hence this is called field weakening control or field weakening control (or flux weakening control). Also, since it is divided into d-axis current Id and q-axis current Iq and controlled independently, it is called vector control.

  The three-phase / two-phase dq conversion means 61 converts the motor currents Iu, Iv, and Iw into the d-axis current Id and the q-axis current Iq using Equation 1, and the motor current instantaneous value detected corresponding to the electrical angle θ. Then, Id and Iq are calculated.

記憶手段６２には、ｓｉｎθとｃｏｓθのデータを記憶しているので、電気角データに
対応したデータを呼び出して積和演算を行うことにより、ｄ軸電流Ｉｄとｑ軸電流Ｉｑに分解できる。電気角θの検知とモータ電流瞬時値の検出はキャリヤ信号に同期して行うもので、後述するフローチャートに従い、詳細な説明を行う。

Since the storage unit 62 stores the data of sin θ and cos θ, it can be decomposed into the d-axis current Id and the q-axis current Iq by calling the data corresponding to the electrical angle data and performing the product-sum operation. The detection of the electrical angle θ and the detection of the instantaneous motor current value are performed in synchronization with the carrier signal, and will be described in detail according to the flowchart described later.

  The rotation speed detection means 66 detects the motor rotation speed from the output reference signal H3 of the rotor position detection means 4a, and applies the rotation speed signal to the setting change means 65 and the torque current control means 67.

  The setting change means 65 (primary current determination means) sets the rotation speed of the motor 4, sets the d-axis current command value Ids according to the rotation speed, and sets the speed command Ns which is the set rotation speed to the torque current control means 67. And the d-axis current command value Ids is added to the motor current control means 68. Note that H1 and H2 may be used for the output reference signal, respectively, or an average value of rotation speeds obtained from the signals H1 to H3 may be used (not shown).

  The torque current control means 67 corresponds to the rotation speed comparison means 67a for comparing the detected rotation speed N with the speed command Ns, the error signal ΔN between the detected rotation speed N and the speed command Ns, and the change rate (acceleration) of the rotation speed. The torque current setting means 67b for controlling the q-axis current command value Iqs.

  The torque current setting unit 67b performs so-called rotation speed control current minor loop control in which the q-axis current command value Iqs is PI-controlled according to the error signal ΔN. The setting change such as the gain in the PI control is received from the setting change means 65, and will be described according to the flowchart described later.

  The motor current control means 68 compares the output signals Iq and Id of the three-phase / two-phase dq conversion means 61 with the current command values Iqs and Ids and outputs control voltage signals Vq and Vd, respectively. It comprises means 68a, q-axis voltage setting means 68b, d-axis current comparison means 68c, and d-axis voltage setting means 68d, and generates voltage signals Vq and Vd for controlling q-axis current Iq and d-axis current Id, respectively.

  The d-axis current command value Ids is a signal applied from the setting change means 65 to the motor current control means 68. In the case of an embedded magnet motor, the d-axis current command value Ids is increased in accordance with the rotational speed to control the field weakening. I do. In the case of a surface magnet motor, the d-axis current command value Ids is normally set to zero, and the d-axis current command value Ids is increased in the case of high speed driving. The first embodiment is an explanation for limiting the current flowing from the AC power supply 1 to the rectifier circuit 2. In the surface magnet motor, the d-axis current command value Ids is set to a predetermined value close to zero. The d-axis current command value Ids is not particularly described. A case where the d-axis current is actively used will be described in a second embodiment described later.

  The two-phase / three-phase dq reverse conversion means 63 calculates the three-phase motor drive control voltages Vu, Vv, Vw from the voltage signals Vq, Vd according to Formula 2, and synchronizes with the carrier signal to detect the electrical angle detection means 60. A sinusoidal signal corresponding to the electrical angle θ detected by the above is applied to the PWM control means 64. The method of product-sum calculation performed by calling the data of sin θ and cos θ stored in the storage means 62 is almost the same as the calculation of the three-phase / 2-phase dq conversion means 61.

上記構成のモータ駆動装置について、図３から図８を参照しながら動作、作用を説明する。図３は本実施の形態におけるモータ制御の切り替わりを示すフローチャートで、ステップ１００により洗濯機や洗濯乾燥機のモータ４の速度制御を行うためのモータ制御を開始する。

The operation and action of the motor driving apparatus having the above configuration will be described with reference to FIGS. FIG. 3 is a flowchart showing switching of motor control in the present embodiment. At step 100, motor control for controlling the speed of the motor 4 of the washing machine or the washing / drying machine is started.

  In step 101, a speed command Ns is determined, and in step 102, the current rotational speed N is detected. A method for determining the speed command Ns will be described later. Thereafter, in step 103, the motor current is detected. At the same time, in step 104, a limit value (Iu limit value) of the motor current (here, Iu will be described as an example) is acquired by a method described later.

  In step 105, the effective value of the motor winding current obtained from the output signal of the current detection means 5 is set as the motor current reference value (described as Iu reference value in the figure), and the motor current limit value (Iu limit value in the figure as If the current reference value is higher than the limit value, the process proceeds to step 107, and if it is lower, the process proceeds to step 106.

  In step 106, the Iu limit flag is stored as 0 to indicate that the motor current reference value does not exceed the motor current limit value. In step 107, the Iu limit flag is stored as 1 to indicate that the reference value of the motor current has exceeded the limit value of the motor current.

  In step 108, a subroutine process for motor control is performed in accordance with the current command value. Thereafter, in step 109, if the motor control is continuing, the process returns to step 101 and the speed control is continued. When the motor control is finished, the speed control is finished at step 110.

The control of the motor drive subroutine (motor control subroutine) performed in step 108 will be described with reference to FIG. When the motor drive subroutine starts from step 600, the routine proceeds to step 601 where it is determined whether or not there is a carrier signal interrupt. The carrier signal interrupt is executed by an interrupt signal ck that is generated when the carrier counter of the PWM control means 64 overflows. Run.

  FIG. 5 shows a detailed flowchart of the carrier signal interrupt subroutine. The carrier signal interrupt subroutine is started from step 700, and the interrupt signal ck is counted in step 701.

  Next, the routine proceeds to step 702, where the electrical angle detection means 60 calculates the electrical angle θ at the rotor position. The electrical angle θ at the rotor position is obtained by multiplying the separately obtained electrical angle Δθ per carrier signal period by the count value k of the carrier counter k · Δθ every 60 degrees that can be detected by the rotor position detecting means 4a. Estimate by adding φ.

  If the motor 4 has 8 poles, the carrier frequency is 15.6 kHz, and the rotational speed is 900 r / min, the motor drive frequency is 60 Hz, and the count value k of the carrier counter within an electrical angle of 60 degrees is about 43. Therefore, Δθ is about 1.4 degrees. As the motor rotational speed is lower, the count value k within an electrical angle of 60 degrees is higher, and the electrical angle detection resolution in calculation is improved. Therefore, it can be understood that there is no problem even when the rotational speed is low and accuracy is required.

  Next, motor currents Iu and Iv are detected. Proceeding to step 703, the first motor current detection is performed to obtain Iu1 and Iv1. Since there is a possibility that noise is included in one current detection, the process proceeds to step 704 to detect again to obtain Iu2 and Iv2. In step 705, an average value of these two detection values is obtained, noise is removed, motor currents Iu and Iv are calculated, and motor current Iw is calculated from Iw = − (Iu + Iv).

  Here, the motor currents Iu and Iv are set to simple average values for noise removal, but the present invention is not limited to this method. For example, the amount of change between the current detected at the previous carrier signal interrupt and the current detected at the current carrier signal interrupt is calculated, and the amount of change is reduced by a fixed ratio and added to the previous detected current. Noise reduction may be performed by configuring a low-pass filter function that matches.

  Next, proceeding to step 706, the three-phase / two-phase dq conversion means 61 performs the calculation shown in Equation 1 from the electrical angle θ and the motor currents Iu, Iv, Iw, and performs the three-phase / 2-phase dq conversion. The d-axis current Id and the q-axis current Iq are obtained. Next, the routine proceeds to step 707, where the obtained current values Id and Iq are memorized and separately used as vector control data.

  Next, the process proceeds to step 708 to call up the d-axis control voltage Vd and the q-axis control voltage Vq, and the process proceeds to step 709 to perform the 2-phase / 3-phase dq reverse conversion by the 2-phase / 3-phase dq reverse conversion means 63 according to the equation 2. To obtain the three-phase control voltages Vu, Vv, and Vw. This inverse transformation uses sin θ and cos θ data corresponding to the electrical angle θ of the storage means 62 as in step 706 and performs a product-sum operation at high speed.

  Next, the process proceeds to step 710, where the PWM control means 64 performs PWM control corresponding to the three-phase control voltages Vu, Vv, Vw, and the process proceeds to step 711 to complete the subroutine and return.

  As described in FIG. 2, the PWM control is performed by comparing the sawtooth wave (or triangular wave) carrier signal with the control voltages Vu, Vv, and Vw in correspondence with the U phase, V phase, and W phase. The IGBT on / off control signal of the circuit 3 is generated and the motor 4 is driven in a sine wave. The signals of the upper arm transistor and the lower arm transistor are reversed waveforms. When the conduction ratio of the upper arm transistor is increased, the output voltage is When the positive voltage is increased and the conduction ratio of the lower arm transistor is increased, the output voltage is negative.

  When the conduction ratio is 50%, the output voltage becomes zero. When the control voltage is changed in a sine wave shape corresponding to the electrical angle θ, a sine wave current flows. In the case of sine wave drive, when the transistor conduction ratio is set to the maximum value of 100%, the output voltage is maximum and the modulation degree Am is 100%. When the maximum value of the conduction ratio is 50%, the output voltage is minimum and the modulation is performed. The degree Am is called 0%.

  Since 3-phase / 2-phase dq conversion and 2-phase / 3-phase dq reverse conversion for vector control of motor current are executed at high speed for each carrier signal, high-speed current control is possible, and further, vector for each carrier signal. By controlling the torque, the agitating blade and the washing / dehydrating tub can be appropriately torque-driven in response to load fluctuations.

  Returning to FIG. 4, after executing the carrier signal interrupt subroutine (step 602), the process proceeds to step 603 to determine whether or not there is a position signal interrupt. When any of the output reference signals H1, H2, and H3 of the rotor position detecting means 4a changes, an interrupt signal is generated, and the process proceeds to step 604 to execute the position signal interrupt subroutine shown in FIG. As shown in FIG. 2, an interrupt signal is generated every 60 electrical angles.

  Here, the position signal interruption subroutine will be described with reference to FIG. The position signal interruption subroutine is started from step 800, the process proceeds to step 801, and the output reference signals H1, H2, H3 are input to the electrical angle detection means 60 to perform position detection, and then the process proceeds to step 802 from the position signal. The rotor electrical angle θc is detected. Next, the process proceeds to step 803 where the count value k counted in the carrier signal interrupt subroutine is stored in kc, the process proceeds to step 804, the count value k is cleared, the process proceeds to step 805, and an electrical angle between 60 degrees is obtained. The electrical angle Δθ of one carrier is calculated from the count value kc of the carrier counter.

  Next, the routine proceeds to step 806, where the rotational speed detection means 66 determines whether or not it is an interrupt signal based on the output reference signal H3. If it is a reference position signal interrupt, the routine proceeds to step 807 and the count value T of the rotation period measurement timer is reached. Is stored as the cycle To, the process proceeds to step 808 to clear the count value T, and the process proceeds to step 809 to calculate the motor rotation speed N. Next, the routine proceeds to step 810 to start counting of the rotation period measurement timer, and the routine proceeds to step 811 to end the subroutine and return.

  If the detection resolution of the rotation period measurement timer is set to 8-bit accuracy, the clock becomes 64 μs and the carrier signal can be used as the clock. However, in order to improve the rotation control performance, it is necessary to improve the rotation period detection resolution. It is necessary to set to 1 to 10 μs. In this case, the microcomputer system clock is divided and used as the clock.

  Although the rotation speed detection method described above has shown the method of calculating | requiring from the period of the output reference signal H3, you may use all the output reference signals H1, H2, and H3. If the carrier signal is a triangular wave, the carrier counter timer has a period twice that of the sawtooth wave. Therefore, if the triangular wave overflow signal is used as a clock, the resolution is improved. Therefore, the triangular wave timer overflow signal may be used as a clock.

  Next, the Iu limit table for obtaining the motor current limit value (Iu limit value) in step 104 in FIG. 3 and the speed command generation method in step 101 will be described below.

FIG. 7 shows an example of an Iu limit table for obtaining a motor current limit value (Iu limit value) of the motor drive device of the present invention. FIG. 6 is a characteristic diagram showing the relationship between the motor rotation speed and the effective value of the motor winding current when the effective value of the current flowing to the rectifier circuit 2 is 7 A when the voltage of the AC power supply 1 is 100 V AC. 7 shows that the effective value of the current flowing to the rectifier circuit 2 can always be controlled to 7 A or less by holding the effective value of the motor winding current on the x-axis side indicating the motor rotation speed from the characteristic curve of FIG. Expression 3 represents the curve of FIG.

数式３は２次曲線で表記しているが、モータ巻線電流の実効値がモータ回転数に対してべき乗で単調減少する特性であれば、これに限られるものではなく、数式３であらわされる曲線からｘ軸、ｙ軸方向の領域にモータ巻線電流の実効値を制限することで、１次電流の実効値を検出することなく、任意の値に制限することが可能となる。

Equation 3 is expressed by a quadratic curve, but is not limited to this as long as the effective value of the motor winding current monotonously decreases with the power of the motor rotation speed, and is expressed by Equation 3. By limiting the effective value of the motor winding current to the region in the x-axis and y-axis directions from the curve, it is possible to limit to an arbitrary value without detecting the effective value of the primary current.

  For example, when the speed of the motor rotation is controlled to 700 rpm, the effective value of the current flowing to the rectifier circuit 2 is controlled to 7 A or less by controlling the effective value of the motor winding current so as not to exceed 1.71 A. It becomes possible.

  Similarly, when the motor rotation speed is controlled to 400 rpm, the effective value of the current flowing to the rectifier circuit 2 is controlled to 7 A or less by controlling the effective value of the motor winding current so as not to exceed 2.12 A. It becomes possible to do.

  As described above, the motor current limit value (Iu limit value) is acquired in correspondence with the motor rotation speed, and used as a reference when generating the Iu limit flag.

  Next, creation of the speed command Ns performed in association with the value of the above Iu limit flag will be described.

  The speed command Ns is created by a method of controlling the acceleration so that the motor current does not exceed the limit value (Iu limit value) of the motor current acquired according to the motor rotation speed by the above-described Iu limit table. That is, when the motor current exceeds a preset table limit value (motor current limit value), the command rotational speed (speed command Ns) is gradually changed.

  FIG. 8 is a flowchart showing a method of creating the speed command Ns. The flowchart at the time of acceleration of a brushless motor is shown for explanation.

  The previous value of the sampled cycle (control cycle) of the speed command Ns is represented by the subscript n-1, and the current value is represented by the subscript n. The current speed command is Ns (n), the previous speed command is Ns (n-1), the current acceleration command is ΔNs (n), the previous acceleration command is ΔNs (n-1), and after the acceleration is finished The target value of speed is Nss, the target value of acceleration is ΔNss, and the increase / decrease value of acceleration is α.

  The speed command Ns is created as follows. In step 300, the generation of the speed command Ns is started. In step 301, the Iu limit flag set in FIG.

  If the Iu limit flag is 0, that is, if the effective value of the motor winding current does not exceed the setting value of the aforementioned Iu limit table, the routine proceeds to step 302. In steps 302 to 305, the acceleration increase / decrease amount α is increased within a range not exceeding the acceleration target value ΔNss with respect to the previous acceleration command ΔNs (n−1). Thereafter, in step 307, a speed command is created.

  If the Iu limit flag is 1 in step 301, that is, if the effective value of the motor winding current exceeds the set value (Iu limit value) in the Iu limit table, the process proceeds to step 306. In step 306, a value obtained by subtracting the acceleration / deceleration increase / decrease value α from the previous acceleration command ΔNs (n−1) is set as the current acceleration command ΔNs (n). Thereafter, in step 307, a speed command is created.

  In Step 307, the current speed command Ns (n) is determined by adding the current acceleration command ΔNs (n) obtained according to the flow in Steps 301 to 306 to the previous speed command Ns (n−1).

  The current speed command Ns (n) and the current acceleration command ΔNs (n) are stored and used as the speed command Ns (n−1) and the acceleration command ΔNs (n−1) when the next speed command is created. The The speed command Ns (n) obtained in this way is used as the speed command Ns input from the setting change means 65 to the torque current control means 67 described above.

  Although not shown in the figure, the current speed command Ns (n) is determined in step 307, but it is necessary to increase the speed command within a range not exceeding the target value Nss of the speed after the acceleration is completed.

  In this embodiment, the acceleration increase / decrease value α is used as an example of a method for controlling the acceleration so that the effective value of the motor winding current does not exceed the setting value (Iu limit value) of the aforementioned Iu limit table. Although the method of performing is described, it does not stick to the method of creating the increase / decrease of acceleration. Similarly, it is also effective to increase or decrease the speed command so that the effective value of the motor winding current does not exceed the setting value (Iu limit value) of the aforementioned Iu limit table after shifting to a constant speed.

  The limit control of the effective value of the motor winding current described above makes it possible to control the current to a desired value without detecting the current flowing to the rectifier circuit 2, and more than necessary at the time of low rotation. Since the effective value of the current is not limited and the limit value can be appropriately changed according to the motor rotation speed, it is possible to prevent a problem due to insufficient torque at the time of low rotation.

(Embodiment 2)
In the first embodiment, the limit value characteristic of the effective value of the motor winding current has been described. However, the case where the induced voltage of the brushless motor and the current phase of the winding are extremely different, such as field weakening control, is considered. Absent. When the brushless motor is controlled at high speed, the induced voltage may approach or exceed the power supply voltage. In such a case, field-weakening control is performed by increasing the d-axis current command value Ids according to the rotational speed. When the field weakening control is performed, the winding current of the motor becomes larger than the primary side current. On the other hand, when the brushless motor is driven at a low speed, the d-axis current command Ids may be set to 0 (zero) as described above. In the case of a surface magnet type motor, the current with respect to the torque generated by the motor. Is minimal.

  In the present embodiment, even when the induced voltage of the brushless motor and the current phase of the winding are extremely different, the primary current can be limited to a desired value without reducing the effective value of the motor winding current more than necessary. explain. Hereinafter, the parts different from the first embodiment will be described, and other configurations are the same as those of the first embodiment, and the description thereof will be omitted.

  The electric angle detection means 60 in the circuit diagram in which the motor driving device shown in FIG. 1 is partially blocked is 30 degrees, 90 degrees, 150 degrees, etc. from the output reference signals H1, H2, H3 of the rotor position detection means 4a. The electrical angle is detected, and the electrical angle θ is obtained by estimation except for every 60 degrees. The three-phase / two-phase dq conversion means 61 also has a current phase detection means.

The current phase β is the phase of the induced voltage and the phase current of the motor 4, and the current phase detection means combined with the three-phase / 2-phase dq conversion means 61 described above at the timing of zero crossing of the current of the winding of the motor 4. It is obtained by detecting the electrical angle θ. The electrical angle θ at the time of zero-crossing of the U-phase current Iu becomes the current phase, and when using the detection values for the V-phase and W-phase currents, the values are shifted by 120 degrees. Alternatively, the current phase β can also be obtained by calculating tan ⁻¹ (Id / Iq) using the motor currents Id and Iq on the dq axis converted by the three-phase / 2-phase dq conversion means 61 described above.

  The current phase β thus obtained is input from the current phase detection means to the setting change means 65, and the value obtained by multiplying the effective value of the motor winding current by the cosine cos β is used as a reference value of the motor current to determine the current limit value in the Iu limit table. By comparing with the set value (Iu limit value), the motor 4 can be favorably operated without limiting the current more than necessary.

(Embodiment 3)
In the first embodiment, the limit limiting characteristic of the effective value of the single motor winding current when the power supply voltage is constant has been described. However, since the case where the power supply voltage fluctuates is not considered, the present embodiment In this embodiment, a configuration that can limit the primary current to a desired value without reducing the effective value of the motor winding current more than necessary even when the power supply voltage fluctuates will be described with reference to the drawings.

  FIG. 9 is a partial block diagram of the motor driving apparatus according to the third embodiment. In addition to the configuration of the first embodiment, the motor drive device of the present embodiment includes voltage detection means 69 that detects the voltage of the rectifier circuit 2, and a limiter 67c that limits the effective value of the motor winding current. Other configurations are the same as those of the first embodiment. Hereinafter, a different part from Embodiment 1 is demonstrated and description is abbreviate | omitted about another structure.

  The limiter 67c constitutes a part of the torque current control means 67, and compares the value of the q-axis current command value Iqs with the value of the maximum Iqs derived from the function with the current detected rotational speed N, and the limiter 67c is limited. .

  FIG. 10 illustrates the switching process of the Iu limit table of the effective value of the motor winding current by the rectified voltage detection performed by the voltage detection means 69 for each step. In step 200, the voltage Vp of the capacitor 21 of the rectifier circuit 2 is detected, and in step 201, it is compared with a predetermined high-level specified value. In step 201, if the Vp voltage is higher than the high-level specified value, the process proceeds to step 203, and the Iu limit table corresponding to the high potential is selected, and if it is low, the process proceeds to step 202. In step 202, the lower specified value is further compared with the Vp voltage. If the Vp voltage is lower than the lower specified value, the process proceeds to step 204, and the Iu limit table corresponding to the low potential is selected. Iu limit table of effective value of motor winding current is selected. In step 206, the table selected in steps 203 to 205 is set as the limit table of the limiter 67c of the effective value of the motor winding current.

  The specified value to be compared with the Vp voltage value in steps 201 and 202 will be described with reference to FIGS.

FIG. 11 shows the voltage waveform (calculated waveform) of the rectified voltage detector by the voltage detector 69 under the condition of the power supply voltage of 85V. In FIG. 11A, the state of the motor 4 is energized (rotating), the voltage value pulsates from 210V to 225V, and the average value is about 215V. In FIG. 11B, the state of the motor 4 is stopped, the pulsation of the voltage value is small, and the average value is 235.
About V. If the power supply voltage is 85V, the effective value of the appropriate motor winding current against power supply voltage fluctuations is set by setting a table with an average value of 235V or less when the motor is rotating and 255V or less when the motor is stopped. The limit table can be selected.

  Similarly, FIG. 12 shows the voltage waveform (calculated waveform) of the rectified voltage detection unit by the voltage detection means 69 under the condition that the power supply voltage is 100 V, which is a standard voltage. In FIG. 12A, the state of the motor 4 is energized (rotating), the voltage value pulsates from 235V to 265V, and the average value is about 255V. In FIG. 12B, the state of the motor 4 is stopped, the pulsation of the voltage value is small, and the average value is about 275V. When the power supply voltage is 100V, the effective value of the appropriate motor winding current against the power supply voltage fluctuation is set by setting a standard table with an average value of 235V to less than 275V when the motor is rotating and 255V to less than 295V when stopped. The limit table can be selected.

  Similarly, FIG. 13 shows the voltage waveform (calculated waveform) of the rectified voltage detection unit by the voltage detection means 69 under the condition that the power supply voltage is 115V. In FIG. 13A, the state of the motor 4 is energized (rotating), the voltage value pulsates from 285V to 305V, and the average value is about 295V. In FIG. 13B, the state of the motor 4 is stopped, the pulsation of the voltage value is small, and the average value is about 315V. When the power supply voltage is 115 V, the effective value of the appropriate motor winding current against the power supply voltage fluctuation is set by setting the table as an average value of 275 V or higher when the motor is rotating and 295 V or higher when the motor is stopped. A value limit table can be selected.

  FIG. 14 shows the case where the effective value of the current flowing to the rectifier circuit 2 is 7 A when the voltage of the AC power source 1 is in a high voltage condition (AC 115 V), a standard voltage (AC 100 V), and a low voltage condition (AC 85 V), respectively. It is a characteristic view which shows the relationship between the motor rotation speed and the effective value of motor winding current. By holding the effective value of the motor winding current on the x-axis side indicating the number of rotations from each characteristic curve of FIG. 14, the effective value of the current flowing to the rectifier circuit 2 is always 7 A or less according to the power supply voltage. It shows that it can be controlled.

  In the curve of FIG. 14, the curve of the high voltage condition (the effective value limit table of the motor winding current at the high potential) is expressed by the equation 4 and the low voltage condition (the effective value of the motor winding current at the low potential). (5) is a mathematical expression of the curve of (limit table). A curve representing the curve at the time of the standard voltage (the effective value limit table of the motor winding current at the standard potential) is expressed by the same equation 3 as in the first embodiment.

数式４、数式５は２次曲線で表記しているが、モータ巻線電流の実効値がモータ回転数に対してべき乗で単調減少する特性であれば、これに限られるものではなく、数式４、数式５であらわされる曲線からｘ軸、ｙ軸方向の領域にモータ巻線電流の実効値を制限することで、１次側電流の実効値を検出することなく、任意の値に制限することが可能となる。

Expressions 4 and 5 are expressed by quadratic curves, but the present invention is not limited to this as long as the effective value of the motor winding current monotonously decreases with the power of the motor speed. By limiting the effective value of the motor winding current to the x-axis and y-axis direction regions from the curve expressed by Equation 5, the effective value of the primary current is limited to an arbitrary value without being detected. Is possible.

  In addition, although the above-mentioned limit table of the effective value of the motor winding current has been described in three ways, the reference value corresponding to the power supply voltage fluctuation may be set in four or more ways. Furthermore, the coefficient of the function that monotonously decreases in proportion to the square of the motor rotation speed may be continuously changed with respect to the power supply voltage fluctuation range.

  The motor driving device and the washing machine and washing / drying machine using the motor driving device are limited so that the AC input current does not become excessive in the event of an abnormality, and the capacity and bias of the laundry in the drum driven by the motor. As a result, the motor can be driven by the optimum torque control according to the load that varies depending on the power consumption, and the energy saving performance can be improved. In addition, since the number of drum rotations can be controlled finely and efficiently according to various washing, rinsing, dehydration and drying conditions, it is possible to provide a multifunctional and easy-to-use washing machine and washing dryer. . In addition, since the circuit configuration of the motor drive device can be simplified and expensive parts such as a photocoupler can be eliminated, an inexpensive washing machine and washing dryer can be provided.

  The motor drive device according to the present invention can provide a low-cost motor drive device that can prevent the input current from becoming excessive without restricting the motor current more than necessary. It can be suitably used for a motor drive device such as a washing machine or a washing / drying machine that needs to be limited to an arbitrary value.

1 AC power supply
2 Rectifier circuit 3 Inverter circuit 4 Motor (brushless motor)
4a Rotor position detection means 5 Current detection means 6 Control means 60 Electrical angle detection means 61 3 phase / 2 phase dq conversion means 62 Storage means 63 2 phase / 3 phase dq reverse conversion means 64 PWM control means 65 Setting change means (primary Current judgment means)
66 Rotational speed detection means 67 Torque current control means 67c Limiter 68 Motor current control means 69 Voltage detection means

Claims (7)

A rectifier circuit connected to a power source; an inverter circuit that converts DC power of the rectifier circuit into AC power; a brushless motor that is driven by the inverter circuit and drives a load such as a washing and dewatering tub; and a rotor of the brushless motor Rotor position detecting means for detecting position, current detecting means for detecting motor current of the brushless motor, and control means for controlling the inverter circuit, wherein the control means uses the motor current of the brushless motor as magnetic flux. Vector control is performed so that the current component corresponding to the current component and the current component corresponding to the torque are decomposed and controlled so as to have respective desired values, and the motor rotational speed by the rotor position detecting means is set to the command rotational speed. A primary current determination unit that performs control and refers to a reference value of the motor current detected by the current detection unit; The primary current determination means moderates the change in the command rotational speed when the reference value of the motor current detected by the current detection means exceeds a preset table limit value according to the motor rotational speed. A motor drive device characterized by that. 2. The motor drive according to claim 1, wherein, in the primary current determination means, the limit value of the table is defined by a monotonically decreasing function having a term proportional to the square of the motor rotation speed. apparatus. Comparing the motor current detected by the current detection means with the rotor position detected by the rotor position detection means, the current detection means has a current phase detection means for detecting the phase of the motor current and the induced voltage. A value obtained by multiplying the current value of the motor current detected by the current detection means by the cosine of the phase of the motor current and the induced voltage detected by the current phase detection means is used as the reference value of the motor current. The motor drive device according to claim 1, wherein: A voltage detection means for detecting a DC voltage of the rectifier circuit and a plurality of the tables are provided, and the control means compares the voltage detected by the voltage detection means with a predetermined standard value, and the magnitude relationship thereof. The motor driving device according to claim 1, wherein the limit value of the table is switched by selectively applying the plurality of tables. When the voltage detected by the voltage detection means is compared with a predetermined standard value and determined to be higher than that, the limit value of the table is a term proportional to the square of the motor rotation speed. The motor drive device according to claim 4, wherein the coefficient of the monotonically decreasing function is increased. The voltage detected by the voltage detection means is compared with a predetermined standard value, and when it is determined that the potential is lower than that, the limit value of the table is a term proportional to the square of the motor rotation speed. The motor driving apparatus according to claim 4, wherein a coefficient of a monotonically decreasing function is reduced. A washing machine or a washing dryer using the motor driving device according to any one of claims 1 to 6.

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