JP2017017878A - Motor controller and power generator controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor controller capable of facilitating an inverter to execute superior performance even when operating in a modulation region.SOLUTION: A motor controller 103 has a magnitude identification section 140 that identifies command magnitude as the magnitude of command magnetic flux vector. In a first case that the magnitude of an estimated motor voltage vector is larger than a voltage limitation value, the magnitude identification section 140 identifies the command magnitude through performing a control including a voltage feedback control that converges a difference between the magnitude of an estimated motor voltage vector and a voltage limitation value to zero.SELECTED DRAWING: Figure 3

Description

  The present disclosure relates to a motor control device and a generator control device.

  Conventionally, various methods are known as a driving method of a three-phase motor. An example of a driving method of the three-phase motor is direct torque control (DTC). Direct torque control can be realized using a motor control device and an inverter. With direct torque control, position sensors such as encoders and resolvers can be omitted.

  Patent Document 1 and Non-Patent Documents 1 and 2 describe techniques related to direct torque control.

Japanese Patent No. 4972135

Masanori Inoue, Shigeo Morimoto, Masayuki Sanada, “A reference value calculation scheme for torque and flux and an anti-windup implementation of torque controller for direct torque control of permanent magnet synchronous motor), IEEJ Transactions D, 130, 6, p. 777-784 (2010) Yuichiro Maeda, Masanori Inoue, Shigeo Morimoto, Masayuki Sanada, “Operating Characteristics of Direct Torque Control System for Interior Permanent Magnet Synchronous Motors in Overmodulation Region) ”2012 IEEJ Industrial Application Division Conference, Vol. 3, pp 243-246 (2012-8)

  The present inventors have sought to provide a motor control device that easily exhibits excellent performance even when the inverter operates in a region where the degree of modulation is large. The present disclosure relates to a technique for realizing such a motor control device.

That is, this disclosure
A command magnetic flux vector to be followed by the motor magnetic flux vector of the three-phase motor and a first command voltage vector to be followed by the motor voltage vector of the three-phase motor are specified, and the three-phase motor is configured using a PWM inverter. A motor control device for controlling,
A motor voltage estimation unit for estimating the motor voltage vector;
A motor magnetic flux estimator for estimating the motor magnetic flux vector;
An amplitude specifying unit for specifying a command amplitude to be an amplitude of the command magnetic flux vector;
A command magnetic flux specifying unit for specifying the command magnetic flux vector using the command amplitude;
A command voltage specifying unit for specifying the first command voltage vector using the command magnetic flux vector and the estimated motor magnetic flux vector;
The DC voltage converted into three-phase AC by the PWM inverter is defined as a DC link voltage, and each phase voltage of the first command voltage vector when expressed in three-phase AC coordinates with respect to a half value of the DC link voltage Is defined as a modulation degree, and when a threshold value larger than the magnitude of the first command voltage vector when the modulation degree is 1 is defined as a voltage limit value, the amplitude specifying unit estimates Feedback control for converging a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value to zero in a first case where the magnitude of the motor voltage vector thus determined is larger than the voltage limit value A motor control device that specifies the command amplitude through control including:

  The motor control device described above easily exhibits excellent performance even when the inverter operates in a region where the degree of modulation is large.

Block diagram of motor controller Diagram for explaining dq coordinate system and αβ coordinate system Block diagram of the motor control unit of the first embodiment Inverter configuration diagram The block diagram of the correction amplitude specific | specification part of 1st Embodiment The flowchart for demonstrating operation | movement of the switching part of 1st Embodiment. Explains the waveform of the U-phase component of the first command voltage vector, the waveform of the U-phase component of the second command voltage vector, and the waveform of the U-phase component of the motor voltage vector when the inverter is operating in the overmodulation region Illustration to do Diagram for explaining the relationship between the modulation factor and the amplitude compensation coefficient Graph showing simulation results Block diagram of motor control section of modification 1-1A Block diagram of motor control unit of modification 1-1B Block diagram of modified amplitude specifying unit of modification 1-1A Block diagram of modified amplitude identification unit of modification 1-1B Block diagram of motor control unit of modification 1-2 Block diagram of motor control unit of second embodiment The block diagram of the position specific | specification part of 2nd Embodiment Block diagram of motor control unit of modification 2-1A Block diagram of motor control unit of modification 2-1B Block diagram of motor control unit of modification 2-1C Block diagram of motor control unit of modification 2-1D Block diagram of position specifying unit of modification 2-1A Block diagram of position specifying part of modification 2-1B Block diagram of position specifying unit of modification 2-1C Block diagram of position specifying unit of modification 2-1D Block diagram of a conventional motor control unit

The inventors examined a motor control unit 903 as shown in FIG. 17 in improving the motor control device. The motor control unit 903 has a configuration for realizing direct torque control in the speed control system. In the motor control unit 903, the phase currents i _u and i _w are converted into axial currents i _α and i _β by the u, w / α, β conversion unit 906. The motor voltage estimation unit 922 estimates the motor voltage vector applied to the three-phase motor from the second command voltage vectors v _2u ^* , v _2v ^* , v _2w ^* (phase voltage limit vectors v _{u_lim} , v _{v_lim} and v _{w_lim} are obtained). u, v, w / alpha, the β conversion unit 923, phase voltage limiting vector _v u_lim, v v_lim, v w_lim is converted restricted axis voltage _v α_lim, v the _{Beta_lim.} A motor magnetic flux estimation unit 908 estimates a motor magnetic flux vector (estimated magnetic flux Ψ _s is obtained) from the limited shaft voltages v _{α_lim} and v _{β_lim} and the shaft currents i _α and i _β . The α-axis component and the β-axis component of the estimated magnetic flux Ψ _s are described as an estimated magnetic flux Ψ _α and an estimated magnetic flux Ψ _β , respectively. The speed and the position specifying unit 911, the estimated magnetic flux [psi _s, speed and phase are estimated motor flux vector of the three-phase motor (phases theta _s estimated velocity omega _r and the estimated magnetic flux [psi _s is determined). The torque estimation unit 914 estimates the motor torque (estimated torque _Te is obtained) from the estimated magnetic flux Ψ _s and the shaft currents i _α and i _β . The command torque specifying unit 925 generates the command torque T _e ^* so that the current estimated speed ω _r matches the command speed ω _ref ^* . A command amplitude | Ψ _s ^* | is generated from the command torque T _e ^* by the amplitude specifying unit 915. The torque deviation calculation unit 921 obtains a deviation (torque deviation) ΔT between the estimated torque _Te and the command torque _Te ^* . The position specifying unit 990 specifies the phase θ _s ^{* of the} command magnetic flux vector Ψ _s ^* from the phase θ _s and the torque deviation ΔT. The command magnetic flux specifying unit 912 obtains the command magnetic flux vector ψ _s ^* from the command amplitude | ψ _s ^* | and the phase θ _s ^* . The α-axis component and β-axis component of the command magnetic flux vector ψ _s ^* are described as α-axis command magnetic flux ψ _α ^* and β-axis command magnetic flux ψ _β ^* , respectively. the alpha-axis magnetic flux deviation calculation unit 913a, alpha axis command flux [psi _alpha ^* and the estimated flux [psi _alpha and deviation (the magnetic flux deviation) [Delta] [Psi] _alpha is obtained. the beta-axis magnetic flux deviation calculation unit 913b, beta axis command flux [psi _beta ^* and the estimated flux [psi _beta and deviation (the magnetic flux deviation) [Delta] [Psi] _beta is obtained. The command voltage specifying unit 907 obtains first command voltage vectors (axis command voltages) v _α ^* and v _β ^* from the magnetic flux deviations ΔΨ _α and ΔΨ _β and the shaft currents i _α and i _β . alpha, the beta / u, v, w conversion unit 909, the first command voltage vector _{v α ^*,} v β ^* the first command voltage vector _{v 1u} ^*, v _1v ^*, is converted to v _{1 w} ^*. The amplitude compensation unit 910, the first command voltage vector ^{_{v 1u *, v 1v *,}} v 1w * from the second command voltage vector ^{_{v 2u *, v 2v *,}} v 2w * is determined. The inverter is switched based on the second command voltage vectors v _2u ^* , v _2v ^* , v _2w ^* . Note that the first command voltage vectors v _1u ^* , v _1v ^* , and v _1w ^* represent vectors that the motor voltage vector should follow. The second command voltage vectors v _2u ^* , v _2v ^* , v _2w ^* are vectors for eliminating an error in the motor voltage vector derived from the gain of the inverter.

  By the way, Non-Patent Document 2 introduces a technique for performing control that combines direct torque control and flux-weakening control while operating an inverter in a region where the degree of modulation is greater than one. The flux weakening control is a control for suppressing the magnitude of the motor voltage vector to a voltage limit value or less. In the technique of Non-Patent Document 2, a term of a voltage limit value is included in a mathematical expression that gives a command amplitude for flux-weakening control (see Expression (2) of Non-Patent Document 2). That is, the technique of Non-Patent Document 2 attempts to realize flux-weakening control by feedforward control.

  In a situation where the flux weakening control is being performed (a situation where the increase in the magnitude of the motor voltage vector is suppressed based on the flux weakening control), the magnitude of the motor voltage vector may be maintained at the voltage limit value. However, according to the study by the present inventors, in the flux weakening control of Non-Patent Document 2, the magnitude of the motor voltage vector is kept at a value deviated from the voltage limit value due to various errors. As far as the present inventors know, no weak flux control method has been proposed so far that can ensure high accuracy when the inverter is operated in a region where the degree of modulation is greater than one.

  In view of the above, an object of the present disclosure is to provide a motor control device suitable for performing magnetic flux control with high accuracy while operating an inverter in a region where the degree of modulation is greater than 1.

That is, the first aspect of the present disclosure is:
A command magnetic flux vector to be followed by the motor magnetic flux vector of the three-phase motor and a first command voltage vector to be followed by the motor voltage vector of the three-phase motor are specified, and the three-phase motor is configured using a PWM inverter. A motor control device for controlling,
A motor voltage estimation unit for estimating the motor voltage vector;
A motor magnetic flux estimator for estimating the motor magnetic flux vector;
An amplitude specifying unit for specifying a command amplitude to be an amplitude of the command magnetic flux vector;
A command magnetic flux specifying unit for specifying the command magnetic flux vector using the command amplitude;
A command voltage specifying unit for specifying the first command voltage vector using the command magnetic flux vector and the estimated motor magnetic flux vector;
The DC voltage converted into three-phase AC by the PWM inverter is defined as a DC link voltage, and each phase voltage of the first command voltage vector when expressed in three-phase AC coordinates with respect to a half value of the DC link voltage Is defined as a modulation degree, and when a threshold value larger than the magnitude of the first command voltage vector when the modulation degree is 1 is defined as a voltage limit value, the amplitude specifying unit estimates Feedback control for converging a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value to zero in a first case where the magnitude of the motor voltage vector thus determined is larger than the voltage limit value The motor control apparatus which specifies the said command amplitude through control including is provided.

  The voltage limit value of the first aspect is larger than the first command voltage vector when the modulation degree is 1. That is, according to this voltage limit value, the flux-weakening control can be executed while operating the PWM inverter in a region where the degree of modulation is larger than 1. Specifically, in the first aspect, in the first case where the magnitude of the motor voltage vector is larger than the voltage limit value, the voltage feedback control that converges the deviation between the magnitude of the motor voltage vector and the voltage limit value to zero. Execute. In this way, the magnitude of the motor voltage vector matches the voltage limit value with higher accuracy than when the feedforward control is executed. That is, according to the first aspect, it is possible to execute the flux-weakening control with high accuracy while operating the PWM inverter in a region where the modulation degree is larger than 1.

The second aspect of the present disclosure includes, in addition to the first aspect,
The amplitude specifying unit includes:
A temporary setting unit for setting a temporary amplitude of the command magnetic flux vector;
In the first case, a corrected amplitude specifying unit that specifies a corrected amplitude through control including the voltage feedback control;
There is provided a motor control device including a correction unit that identifies the command amplitude by adding the correction amplitude to the temporary amplitude.

  The motor control device of the second aspect is simple.

The third aspect of the present disclosure includes, in addition to the second aspect,
In the first case, the correction amplitude specifying unit provides a motor control device that specifies the correction amplitude smaller than zero.

  According to the third aspect, the temporary amplitude is corrected so as to obtain a demagnetizing action.

In addition to the second aspect or the third aspect, the fourth aspect of the present disclosure includes:
The modified amplitude specifying unit has a voltage deviation compensating unit,
In the first case, the voltage deviation compensator uses the corrected amplitude as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. A motor control device that executes feedback control is provided.

  The configuration of the fourth aspect is simpler than the configuration of the sixth aspect.

The fifth aspect of the present disclosure includes, in addition to the fourth aspect,
The modified amplitude specifying unit includes a switching unit that switches between the first mode and the second mode,
In the first mode, the voltage deviation compensation unit performs the voltage feedback control using the corrected amplitude as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. Is the mode to run,
The second mode is a mode in which a constant is the corrected amplitude,
The switching unit selects the first mode in the first case, and selects the second mode in the second case where the estimated magnitude of the motor voltage vector is equal to or less than the voltage limit value. A motor control device is provided.

  The configuration of the fifth aspect is simpler than the configuration of the seventh aspect. In the motor control device according to the fifth aspect, when the magnitude of the motor voltage vector is equal to or less than the voltage limit value, control other than the magnetic flux weakening control can be performed. Also, according to the fifth aspect, it is possible to smoothly switch between the flux-weakening control and the other control at an appropriate timing.

In the sixth aspect of the present disclosure, in addition to the second aspect or the third aspect,
The modified amplitude specifying unit includes a voltage deviation compensation unit and an inner product compensation unit,
In the first case, the voltage deviation compensator uses the first command error parameter as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. Executing the voltage feedback control,
In the first case, the inner product compensation unit is configured to (a) a first inner product of the estimated motor magnetic flux vector and a motor current vector of the three-phase motor, or (b) an estimation of the three-phase motor. First feedback with the corrected amplitude as the manipulated variable so that a deviation between an error parameter, which is a second inner product of the magnet magnetic flux vector and the motor current vector, and the first command error parameter converge to zero. A motor control device that executes control is provided.

The seventh aspect of the present disclosure includes, in addition to the sixth aspect,
The modified amplitude specifying unit includes a switching unit that switches between the first mode and the second mode,
In the first mode, the voltage feedback control is performed using the first command error parameter as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. A voltage deviation compensation unit executes the first feedback control using the corrected amplitude as an operation amount so that a deviation between the first command error parameter and the error parameter converges to zero. Is the mode executed by
In the second mode, the inner product compensation unit performs second feedback control using the corrected amplitude as an operation amount so that a deviation between a constant second command error parameter and the error parameter converges to zero. The mode to run,
The switching unit selects the first mode in the first case, and selects the second mode in the second case where the estimated magnitude of the motor voltage vector is equal to or less than the voltage limit value. A motor control device is provided.

  In the motor control devices of the sixth aspect and the seventh aspect, the degree of freedom of control is high in the second case (second mode) in which the magnitude of the motor voltage vector is equal to or less than the voltage limit value. Specifically, various controls can be performed by appropriately setting a constant at which the first inner product or the second inner product should converge. Since the first inner product or the second inner product is an effective index for correcting the provisional amplitude, feedback control according to the purpose can be performed in the second case. Moreover, according to the 7th aspect, it becomes possible to switch weakly magnetic flux control and other control smoothly at an appropriate timing.

The eighth aspect of the present disclosure includes, in addition to the seventh aspect,
The error parameter is the second inner product;
A motor controller is provided in which the constant is zero.

  In the motor control apparatus according to the eighth aspect, the flux-weakening control can be executed in the first case, and the maximum torque / current control (MTPA) can be executed in the second case.

The ninth aspect of the present disclosure is in addition to any one of the first aspect to the eighth aspect,
When the theoretical value of the modulation factor when the operation region of the PWM inverter transitions from a linear region to an overmodulation region is defined as a boundary value, the voltage limit value is the first value when the modulation factor is the boundary value. Provided is a motor control device having a magnitude greater than one command voltage vector.

  The voltage limit value of the ninth aspect is appropriate for performing the flux weakening control while operating the PWM inverter in the overmodulation region.

In a tenth aspect of the present disclosure, in addition to any one of the first aspect to the ninth aspect,
Each phase voltage of the motor voltage vector that the PWM inverter outputs to the three-phase motor has an upper limit that can be output and a lower limit that is smaller than the upper limit by the DC link voltage,
The motor control device specifies a reference voltage vector used to generate a PWM duty that defines the operation of the PWM inverter,
The motor voltage estimator replaces the voltage of the phase whose voltage is larger than the upper limit with the upper limit in the reference voltage vector expressed by the three-phase AC coordinates, and the voltage of the phase whose voltage is smaller than the lower limit. Provided is a motor control device that estimates a value replaced with the lower limit as the motor voltage vector.

  According to the tenth aspect, the motor voltage vector can be estimated easily and accurately even when the PWM inverter operates in the overmodulation region.

In an eleventh aspect of the present disclosure, in addition to any one of the first aspect to the tenth aspect,
The motor control device includes an amplitude compensator that specifies a reference voltage vector used to generate a PWM duty that defines an operation of the PWM inverter from the first command voltage vector,
When the theoretical value of the modulation factor when the operation region of the PWM inverter transitions from a linear region to an overmodulation region is defined as a boundary value, the reference voltage vector is when the modulation factor is equal to or less than the boundary value , A second command voltage vector that is the same as the first command voltage vector,
The reference voltage vector is an amplitude compensation coefficient larger than 1 when the degree of modulation is greater than the boundary value, and an amplitude compensation coefficient that increases as the degree of modulation increases is the first command voltage vector. A motor control device that is a second command voltage vector obtained by multiplying by is provided.

  When the PWM inverter is operating in the overmodulation region, the gain of the PWM inverter decreases as the modulation degree increases. In the eleventh aspect, the reference voltage vector (second command voltage vector) corresponding to the amplitude of each phase voltage of the first command voltage vector increases as the degree of modulation increases when the PWM inverter operates in the overmodulation region. The ratio of the amplitude of each phase voltage increases. As a result, the deviation between the first command voltage vector and the motor voltage vector is suppressed when the PWM inverter is operating in the overmodulation region.

The twelfth aspect of the present disclosure includes, in addition to the eleventh aspect,
The amplitude compensation coefficient is the product of the gain of the PWM inverter and the amplitude of the fundamental wave component of each phase voltage of the second command voltage vector, and the amplitude of the fundamental wave component of each phase voltage of the first command voltage vector A motor control device having a coefficient determined to match

  According to the twelfth aspect, the amplitude of the fundamental wave component of each phase voltage of the first command voltage vector matches the amplitude of the fundamental wave component of each phase voltage of the motor voltage vector with high accuracy.

The thirteenth aspect of the present disclosure is in addition to any one of the first aspect to the twelfth aspect,
The motor magnetic flux estimating unit estimates the motor magnetic flux vector using a motor current vector of the three-phase motor and the estimated motor voltage vector,
The motor control device
A torque estimator for estimating a motor torque of the three-phase motor from the motor current vector and the estimated motor magnetic flux vector;
Using the torque deviation between the command torque to be followed by the motor torque and the estimated motor torque, the movement amount for each control cycle in which the phase of the motor magnetic flux vector should move is specified, and the specified movement And a position specifying unit that specifies the phase of the command magnetic flux vector using the estimated amount and the phase of the estimated motor magnetic flux vector.

  In the motor control device of the thirteenth aspect, the motor torque is estimated. The estimated motor torque as well as the estimated motor magnetic flux vector phase is reflected in the motor magnetic flux vector phase. That is, since feedback information regarding the motor torque is used for specifying the phase of the motor magnetic flux vector, the motor control device according to the thirteenth aspect is excellent in response to the motor torque.

The fourteenth aspect of the present disclosure is in addition to any one of the first aspect to the twelfth aspect,
The motor control device specifies a movement amount for each control cycle in which the phase of the motor magnetic flux vector should move using a command speed that the speed of the three-phase motor should follow, and uses the specified movement amount A position specifying unit for specifying the phase of the command magnetic flux vector;
The command magnetic flux specifying unit provides a motor control device that specifies the command magnetic flux vector using the phase specified by the position specifying unit and the command amplitude specified by the amplitude specifying unit.

  According to the fourteenth aspect, the phase of the command magnetic flux vector can be specified simply.

A fifteenth aspect of the present disclosure includes
A command magnetic flux vector to be followed by a generator magnetic flux vector of a three-phase generator and a first command voltage vector to be followed by a generator voltage vector of the three-phase generator are specified, and the PWM converter is used to specify the command magnetic flux vector A generator control device for controlling a three-phase generator,
A generator voltage estimator for estimating the generator voltage vector;
A generator magnetic flux estimator for estimating the generator magnetic flux vector;
An amplitude specifying unit for specifying a command amplitude to be an amplitude of the command magnetic flux vector;
A command magnetic flux specifying unit for specifying the command magnetic flux vector using the command amplitude;
A command voltage specifying unit for specifying the first command voltage vector using the command magnetic flux vector and the estimated generator magnetic flux vector;
The DC voltage converted into three-phase AC by the PWM converter is defined as a DC link voltage, and each phase voltage of the first command voltage vector when expressed in three-phase AC coordinates with respect to a half value of the DC link voltage Is defined as a modulation degree, and when a threshold value larger than the magnitude of the first command voltage vector when the modulation degree is 1 is defined as a voltage limit value, the amplitude specifying unit estimates In the first case where the magnitude of the generated generator voltage vector is larger than the voltage limit value, the voltage that converges the deviation between the estimated generator voltage vector magnitude and the voltage limit value to zero Provided is a generator control device that specifies the command amplitude through control including feedback control.

  According to the 15th aspect, the generator control apparatus which has the same effect as the motor control apparatus of the 1st aspect can be provided. It is also possible to provide an electric motor that combines the features of the first aspect and the fifteenth aspect. That is, it is also possible to provide an electric motor that performs both power running drive and regenerative drive, and operates in the same manner as the first and fifteenth aspects and exhibits the same effects as those aspects.

The sixteenth aspect of the present disclosure includes
A command magnetic flux vector to be followed by the motor magnetic flux vector of the three-phase motor and a first command voltage vector to be followed by the motor voltage vector of the three-phase motor are specified, and the three-phase motor is configured using a PWM inverter. A motor control method for controlling,
Estimating the motor voltage vector;
Estimating the motor flux vector;
Specifying a command amplitude to be an amplitude of the command magnetic flux vector;
Identifying the command magnetic flux vector using the command amplitude;
Identifying the first command voltage vector using the command magnetic flux vector and the estimated motor magnetic flux vector,
The DC voltage converted into three-phase AC by the PWM inverter is defined as a DC link voltage, and each phase voltage of the first command voltage vector when expressed in three-phase AC coordinates with respect to a half value of the DC link voltage Defining the command amplitude when a ratio greater than the magnitude of the first command voltage vector when the modulation factor is 1 is defined as a voltage limit value. Then, in the first case where the estimated magnitude of the motor voltage vector is larger than the voltage limit value, the deviation between the estimated magnitude of the motor voltage vector and the voltage limit value is converged to zero. Provided is a motor control method for specifying the command amplitude through control including voltage feedback control.

A seventeenth aspect of the present disclosure includes
A command magnetic flux vector to be followed by a generator magnetic flux vector of a three-phase generator and a first command voltage vector to be followed by a generator voltage vector of the three-phase generator are specified, and the PWM converter is used to specify the command magnetic flux vector A generator control method for controlling a three-phase generator,
Estimating the generator voltage vector;
Estimating the generator flux vector;
Specifying a command amplitude to be an amplitude of the command magnetic flux vector;
Identifying the command magnetic flux vector using the command amplitude;
Identifying the first command voltage vector using the command magnetic flux vector and the estimated generator magnetic flux vector,
The DC voltage converted into three-phase AC by the PWM converter is defined as a DC link voltage, and each phase voltage of the first command voltage vector when expressed in three-phase AC coordinates with respect to a half value of the DC link voltage Defining the command amplitude when a ratio greater than the magnitude of the first command voltage vector when the modulation factor is 1 is defined as a voltage limit value. Then, in the first case where the estimated generator voltage vector magnitude is larger than the voltage limit value, the deviation between the estimated generator voltage vector magnitude and the voltage limit value is zero. Provided is a generator control method for specifying the command amplitude through control including voltage feedback control for convergence.

  According to the sixteenth aspect and the seventeenth aspect, effects similar to the effects of the first aspect and the fifteenth aspect are obtained.

  The technology related to the motor control device can be applied to the generator control device. The technology related to the generator control device can be applied to the motor control device. This is because the control mode is very similar in both cases. There is a difference between the motor and the generator such that the phase of the current flowing through the motor / generator is reversed, but those skilled in the art can configure both control devices in consideration of these differences.

  The technology related to the motor control device and the generator control device can be applied to the motor control method and the generator control method. The technology related to the motor control method and the generator control method can be applied to the motor control device and the generator control device.

  In this specification, vectors relating to current, voltage, magnetic flux, etc. are expressed using three-phase alternating current coordinates (UVV coordinates), α-β coordinates, dq coordinates, dm-qm coordinates, and the like. . However, the technique of this specification can also be described using other coordinates such as MT coordinates. That is, in the following, a control example using specific coordinates will be described for convenience, but it should be considered that this specification discloses a control example using other coordinates as well as the control example. The motor magnetic flux vector is a concept including both an armature linkage magnetic flux on a three-phase AC coordinate applied to a three-phase motor and a magnetic flux obtained by coordinate conversion of the armature linkage flux. The same applies to the command magnetic flux vector, the motor voltage vector, the first command voltage vector, the second command voltage vector, the motor current vector, and the like. In this specification, “amplitude” may simply refer to magnitude (absolute value). Further, unless otherwise specified, “speed” represents the angular speed (unit: rad / s) of the rotor of the three-phase motor.

  Hereinafter, an embodiment according to the present disclosure will be described with reference to the drawings.

(First embodiment)
As shown in FIG. 1, the motor control apparatus 100 includes a first current sensor 105a, a second current sensor 105b, a motor control unit 103, and a duty generation unit 104. Motor control device 100 is connected to inverter 101 and three-phase motor 102.

  The motor control unit 103 is configured to execute position sensorless operation of the three-phase motor 102. The position sensorless operation is an operation that does not use a position sensor such as an encoder or a resolver.

  The motor control device 100 may include elements provided by a control application executed in a DSP (Digital Signal Processor) or a microcomputer. The DSP or the microcomputer may include peripheral devices such as a core, a memory, an A / D conversion circuit, and a communication port. Further, the motor control device 100 may include an element configured by a logic circuit.

(Outline of control using the motor control device 100)
An overview of control using the motor control device 100 will be described with reference to FIG. The phase sensors (motor current vectors) i _u and i _w are detected by the current sensors 105a and 105b. The motor control unit 103 generates reference voltage vectors v _u ^* , v _v ^* , v _w ^* from the command speed ω _ref ^* and the phase currents i _u , i _w . Each component of the reference voltage vectors v _u ^* , v _v ^* , and v _w ^* corresponds to the U-phase voltage v _u ^* , the V-phase voltage v _v ^*, and the W-phase voltage v _w ^* on the three-phase AC coordinates, respectively. The duty generation unit 104, the reference voltage vector _{v u} ^*, v v ^*, the v _w ^*, the duty D _u, D _v, D _w is generated. The inverter 101 generates motor voltage vectors v _u , v _v , v _w to be applied to the three-phase motor 102 from the duties D _u , D _v , D _w . The command speed ω _ref ^* is given to the motor control device 100 from the host control device. The command speed ω _ref ^* represents the speed that the speed of the three-phase motor 102 should follow. With such control, the three-phase motor 102 is controlled such that the speed follows the command speed ω _ref ^* .

  Hereinafter, the motor control device 100 may be described based on α-β coordinates (two-phase coordinates). FIG. 2 shows α-β coordinates, UVW coordinates, and dq coordinates. The α-β coordinates are fixed coordinates. The α-β coordinates are also referred to as stationary coordinates and AC coordinates. The α axis is set as an axis extending in the same direction as the U axis. The d axis is defined as an axis that coincides with the rotor of the rotor, and its position is defined by θ. A position advanced 90 degrees from the d axis is defined as the q axis.

(Motor controller 103)
As shown in FIG. 3, the motor control unit 103 includes a u, w / α, β conversion unit (three-phase two-phase coordinate conversion unit) 106, a motor voltage estimation unit (voltage limiting unit) 122, u, v, w / α, β conversion unit (three-phase two-phase coordinate conversion unit) 123, motor magnetic flux estimation unit 108, torque estimation unit (torque estimator) 114, speed / position specifying unit 111, command torque specifying unit 125, torque deviation calculating unit 121 , Position specifying unit 190, amplitude specifying unit 140, command magnetic flux specifying unit 112, α-axis magnetic flux deviation calculating unit 113a, β-axis magnetic flux deviation calculating unit 113b, command voltage specifying unit 107, α, β / u, v, w converting unit (Two-phase three-phase coordinate conversion unit) 109 and amplitude compensation unit 110 are included. The amplitude specifying unit 140 includes a temporary setting unit 126, a corrected amplitude specifying unit 124, and an adding unit (correcting unit) 127.

In the motor control unit 103, the u, w / α, β conversion unit 106 converts the phase currents i _u , i _w into shaft currents i _α , i _β . The axial currents i _α and i _β are collectively described as the α-axis current i _α and the β-axis current i _β on the α-β coordinate of the three-phase motor 102. By the motor voltage estimation unit 122, the reference voltage vector ^{_{v u *, v v *,}} v w * from the motor voltage vector that is applied to the 3-phase motor 102 is estimated (phase voltage limiting vector _v u_lim, v v_lim, v _{w_lim} is required). The reference voltage vectors v _u ^* , v _v ^* , and v _w ^* are voltage vectors used for generating the PWM duties D _u , D _v , and D _w that define the operation of the inverter 101. In the present embodiment, the reference voltage vector ^{_{v u *, v v *,}} v w * , the second command voltage vector v _2u ^*, v _2v ^*, a v _2w ^*. u, v, w / alpha, the β conversion unit 123, phase voltage limiting vector _v u_lim, v v_lim, v w_lim is converted restricted axis voltage _v α_lim, v the _{Beta_lim.} The limit axis voltages v _{α_lim} and v _{β_lim} collectively describe the α axis limit command voltage v _{α_lim} and the β axis limit command voltage v _{β_lim} on the α-β coordinate of the three-phase motor 102. The motor magnetic flux estimator 108 estimates the motor magnetic flux vector (the estimated magnetic flux Ψ _s is obtained) from the limited shaft voltages v _{α_lim} and v _{β_lim} and the shaft currents i _α and i _β . The α-axis component and the β-axis component of the estimated magnetic flux Ψ _s are described as an estimated magnetic flux Ψ _α and an estimated magnetic flux Ψ _β , respectively. The torque estimation unit 114, the estimated magnetic flux [psi _s and the axial current i _alpha, i _beta from the estimated torque T _e is identified. The speed / position specifying unit 111 specifies the phase θ _s and the estimated speed ω _{r of the} estimated magnetic flux Ψ _s from the estimated magnetic flux Ψ _s . The command torque specifying unit 125 specifies the command torque T _e ^* from the command speed ω _ref ^* and the estimated speed ω _r . By the torque deviation calculation unit 121, the deviation (torque deviation) of the command torque T _e ^* and the estimated torque T _e [Delta] T is calculated. The position specifying unit 190 specifies the phase θ _s ^{* of the} command magnetic flux vector Ψ _s ^* from the phase θ _s and the torque deviation ΔT. By modifying the amplitude determination unit 124, the voltage limit value V _am and limitations axis voltage v _{Arufa_lim,} from v _{Beta_lim,} modified amplitude ΔΨ is obtained. The temporary setting unit 126 specifies the temporary amplitude | ψ _{s_MTPA} ^* | from the command torque T _e ^* . The adder 127 generates the amplitude | Ψ _s ^* | (command amplitude | Ψ _s ^* |) of the command magnetic flux vector Ψ _s ^* from the temporary amplitude | Ψ _{s_MTPA} ^* | and the modified amplitude ΔΨ. The command magnetic flux specifying unit 112 obtains the command magnetic flux vector ψ _s ^* from the command amplitude | ψ _s ^* | and the phase θ _s ^* . The α-axis component and β-axis component of the command magnetic flux vector ψ _s ^* are described as α-axis command magnetic flux ψ _α ^* and β-axis command magnetic flux ψ _β ^* , respectively. the alpha-axis magnetic flux deviation calculation unit 113a, alpha axis command flux [psi _alpha ^* and the estimated flux [psi _alpha and deviation (the magnetic flux deviation) [Delta] [Psi] _alpha is obtained. the beta-axis magnetic flux deviation calculation unit 113b, beta axis command flux [psi _beta ^* and the estimated flux [psi _beta and deviation (the magnetic flux deviation) [Delta] [Psi] _beta is obtained. The command voltage specifying unit 107 obtains first command voltage vectors (axis command voltages) v _α ^* and v _β ^* from the magnetic flux deviations ΔΨ _α and ΔΨ _β and the shaft currents i _α and i _β . The axis command voltages v _α ^* and v _β ^* are a summary of the α axis command voltage v _α ^* and the β axis command voltage v _β ^* on the α-β coordinate of the three-phase motor 102. alpha, beta / u, v, by w converter 109, the first command voltage vector v α ^_*, v β ^* is converted first command voltage vector ^{_{v 1u *, v 1v *,}} v to _{1 w} ^* . The amplitude compensation unit 110, the first command voltage vector ^{_{v 1u *, v 1v *,}} v 1w * from the second command voltage vector ^{_{v 2u *, v 2v *,}} v 2w * is determined. Although expressed in different coordinate systems, the first command voltage vectors v _α ^* , v _β ^* and the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* are substantially the same. It is. In the following description, the first command voltage vectors v _α ^* and v _β ^* may be referred to as axis command voltages v _α ^* and v _β ^* for the purpose of distinguishing these vectors.

With such control, a voltage vector is applied to the three-phase motor 102 via the inverter 101. The speed of the three-phase motor 102 follows the command speed ω _ref ^* . The motor magnetic flux vector and the motor torque of the three-phase motor 102 follow the command magnetic flux vector ψ _s ^* and the command torque _Te ^* . In addition, it is possible to execute highly accurate flux-weakening control while operating the inverter 101 in the overmodulation region. That is, when the flux-weakening control is being executed, the magnitudes of the first command voltage vectors v _α ^* , v _β ^* (voltage command value = √ (v _α ^* ^ 2, v _β ^* ^ 2)) and It is possible to make the magnitude of the motor voltage vector coincide with the desired voltage limit value _Vam with high accuracy.

In the present specification, the shaft currents i _α and i _β mean current values transmitted as information, not currents actually flowing through the three-phase motor 102. Phase voltage limiting vector _{v u_lim, v v_lim, v w_lim} , limited-axis voltage _v α_lim, v β_lim, estimated magnetic flux [psi _s, corrected amplitude [Delta] [Psi], the estimated torque T _e, the voltage limit value V _am, the command speed omega _ref ^*, the command torque T _e ^* , command magnetic flux vector Ψ _s ^* , first command voltage vector v _α ^* , v _β ^* (v _1u ^* , v _1v ^* , v _1w ^* ), second command voltage vector v _2u ^* , v _2v ^The same applies to ^* , v _2w ^{*, and the} like.

  Each component regarding control of this embodiment is explained below.

(Duty generator 104)
Duty generation unit 104 shown in FIG. 1, the reference voltage vector _{^{v u *, v v *,}} v w * ( in the present embodiment the second command voltage vector _{^{v 2u *, v 2v *,}} v 2w *) from the duty D _u , D _v and D _w are generated. The duty is also called PWM duty. Specifically, the duty generation unit 104 converts each component (each phase voltage) of the reference voltage vectors v _u ^* , v _v ^* , and v _w ^* into the duty _Du , D _v , and D _w of each phase. As a method for generating the duties D _u , D _v and D _w, a method used for a general voltage source inverter can be used. For example, the duty D _u, D _v, D _w is the reference voltage vector _{v u} ^*, v v ^*, and v _w ^*, determined by dividing in half the value of the voltage V _dc of the DC power supply 118 (FIG. 4) be able to. In this case, the duty _Du is 2 ^* _vu ^* / _Vdc . The duty D _v is 2 × v _v ^* / V _dc . The duty D _w is 2 × v _w ^* / V _dc . Duty generation unit 104, duty D _u, D _v, and outputs the D _w.

However, duty generation unit 104, duty D _u, D _v, each phase component of D _w is configured so as not to exceed 1. This is because the duty represents the ratio of the ON period to one switching cycle in the inverter 101 and should be defined in the range of 0 to 1. Specifically, the duty D _u generated by the method described above, D _v, component exceeds one of the phases components of D _w is replaced by 1.

(Inverter 101)
The inverter 101 is a PWM inverter. As shown in FIG. 4, the inverter 101 of this embodiment includes a conversion circuit in which switching elements 119a, 119b, 119c, 119d, 119e, and 119f and free-wheeling diodes 120a, 120b, 120c, 120d, 120e, and 120f are paired. A base driver 116, a smoothing capacitor 117, and a DC power supply 118 are included. The DC power supply 118 represents an output rectified by a diode bridge or the like. In this specification, the voltage V _dc of the DC power supply 118 may be referred to as a DC link voltage V _dc . In the present specification, a configuration in which the conversion circuit and the smoothing capacitor 117 are combined is described as an inverter.

The inverter 101 applies a voltage vector to the three-phase motor 102 by PWM control. Specifically, power is supplied to the three-phase motor 102 from the DC power supply 118 via the switching elements 119a to 119f. More specifically, first, the duty D _u, D _v, D _w is input to the base driver 116. Then, the duty D _u, D _v, D _w is converted into a drive signal for electrically driving the switching elements 119A～119f. Next, each of the switching elements 119a to 119f operates according to the drive signal.

  In the present embodiment, the inverter 101 is a three-phase switching circuit using switching elements 119a to 119f. Examples of the switching elements 119a to 119f include a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) and an IGBT (Insulated Gate Bipolar Transistor).

Herein, the first command voltage vector v _1u for half the value of the DC link voltage V _dc ^*, _{v 1v} ^*, is defined as v _{1 w} ^* amplitude modulation factor m _i the ratio of values of each phase voltage . In the present embodiment, the amplitude of the voltage v _1u ^* , the amplitude of the voltage v _1v ^* , and the amplitude of the voltage v _1w ^* are the same (the same applies to the following embodiments and modifications).

As is well known, there are several types of PWM inverter modulation schemes. The inverter 101 of this embodiment operates based on a sine wave PWM modulation method. In the present embodiment, in accordance with the customary case of using the sine wave PWM modulation method, a region where the modulation factor is 1 or less is defined as a linear region, and a region where the modulation factor is greater than 1 is defined as an overmodulation region. The linear region indicates the amplitude of each phase voltage of the motor voltage vectors v _u , v _v , v _w ^* with respect to the change in the amplitude of each phase voltage of the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^*. Is a region that changes linearly. The overmodulation region is obtained by changing each phase voltage of the motor voltage vectors v _u , v _v , v _w ^* with respect to a change in amplitude of each phase voltage of the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* . This is a region where the amplitude changes nonlinearly. For details on the sine wave PWM modulation method, see Non-Patent Document 2, Principle and Design Method of Energy Saving Motor (Shigeo Morimoto, Masayuki Sanada, Scientific Information Publishing Co., Ltd., issued on July 19, 2013), PWM Power Conversion System (Taniguchi) See Katsunori, Kyoritsu Publishing Co., Ltd., August 2007). Needless to say, the present disclosure can also be applied to a case where the sine wave PWM modulation method is not employed. In this case, the modulation degree that becomes the boundary between the linear region and the overmodulation region may be other than 1. is there.

(Three-phase motor 102)
A three-phase motor 102 illustrated in FIG. 1 is a control target of the motor control device 100. A motor voltage vector is applied to the three-phase motor 102 by the inverter 101. “A motor voltage vector is applied to the three-phase motor 102” means that a voltage is applied to each of the three phases (U phase, V phase, W phase) on the three-phase AC coordinate in the three-phase motor 102. Point to. In the present embodiment, each of the three phases (U phase, V phase, W phase) is selected from two types: a high voltage phase having a relatively high voltage and a low voltage phase having a relatively low voltage. The three-phase motor 102 is controlled so as to be one of the following.

The three-phase motor 102 in this embodiment is a synchronous motor. Specifically, the three-phase motor 102 in the present embodiment is a permanent magnet synchronous motor. The three-phase motor 102 may be an SPMSM (Surface Permanent Magnet Synchronous Motor) or an IPMSM (Interior Permanent Magnet Synchronous Motor). In SPMSM, the d-axis inductance L _d and the q-axis inductance L _q are the same. The IPMSM has a saliency in which the d-axis inductance L _d and the q-axis inductance L _q are different (generally, a reverse saliency such that L _q > L _d ). IPMSM can use reluctance torque in addition to magnet torque. For this reason, the driving efficiency of the IPMSM is extremely high.

(First current sensor 105a, second current sensor 105b)
As the first current sensor 105a and the second current sensor 105b shown in FIG. 1, known current sensors can be used. In the present embodiment, the first current sensor 105a is provided to measure the phase current i _u flowing through the u phase. The second current sensor 105b is provided to measure the phase current i _w flowing through the w phase. However, the first current sensor 105a and the second current sensor 105b may be provided so as to measure a two-phase current of a combination other than the u-phase and w-phase two phases.

(U, w / α, β converter 106)
The u, w / α, β converter 106 shown in FIG. 3 converts the phase currents i _u , i _w into axial currents i _α , i _β . Specifically, the u, w / α, β conversion unit 106 converts the phase currents i _u , i _w into the axial currents i _α , i _β by the equations (1) and (2), and the axial current i _α. , I _β .

(Motor voltage estimation unit 122)
Motor voltage estimation unit 122 of the present embodiment, the reference voltage vector _{^{v u *, v v *,}} v w * ( in the present embodiment the second command voltage vector _{^{v 2u *, v 2v *,}} v 2w *) from 3 estimates the motor voltage vector that is applied to the phase-motor 102 (limit voltage vector _v u_lim, v v_lim, seek _v w_lim). The motor voltage estimation unit 122 is configured to accurately estimate the motor voltage vector even when the inverter 101 is operating in the overmodulation region. Details of the motor voltage estimation unit 122 will be described later.

(U, v, w / α, β converter 123)
u, v, w / α, β conversion unit 123 limits the voltage vector _v u_lim, v v_lim, from v _{W_lim,} determined limit shaft voltage v _{Arufa_lim,} the _v β_lim. Specifically, u, v, w / α , β conversion unit 123, by Equation (3), limiting the voltage vector _v u_lim, v v_lim, the v _{W_lim,} converts restricted axis voltage _v α_lim, v in _{Beta_lim,} limit The shaft voltages v _{α_lim} and v _{β_lim} are output.

(Motor magnetic flux estimation unit 108)
The motor magnetic flux estimation unit 108 estimates the motor magnetic flux vector using the motor current vector of the three-phase motor 102 and the estimated motor voltage vector (determined magnetic flux Ψ _s (estimated magnetic flux Ψ _α , Ψ _β )). ). Specifically, the shaft currents i _α and i _β representing the motor current vector by two-phase coordinates and the shaft voltage (restricted shaft voltage) representing the estimated motor voltage vector by two-phase coordinates. v seek _{α_lim,} v and _{β_lim,} the estimated magnetic flux Ψ _s using. More specifically, the motor magnetic flux estimation unit 108 obtains the estimated magnetic fluxes Ψ _α and Ψ _β using the equations (4) and (5). In equations (4) and (5), Ψ _{α | t = 0} and Ψ _{β | t = 0} are initial values of the estimated magnetic fluxes Ψ _α and Ψ _β , respectively. R in the equations (4) and (5) is the winding resistance of the three-phase motor 102. When the motor magnetic flux estimation unit 108 is incorporated in a digital control device such as a DSP or a microcomputer, the integrator required for the calculations in the equations (4) and (5) can be configured as a discrete system. In a typical example in this case, a value derived from the current control cycle is added to or subtracted from the estimated magnetic fluxes ψ _α and ψ _β before one control cycle (one time step).

(Torque estimation unit 114)
Torque estimating unit 114, the axial current i _alpha, i _beta and the estimated magnetic flux [psi _s (estimated magnetic flux Ψ _α, Ψ _β) from estimating the motor torque (obtaining the estimated torque T _e). Specifically, the torque estimation portion 114, using equation (6), determine the estimated torque T _e. P _n in Equation (6) is the _number of pole pairs of the three-phase motor 102.

(Speed / position identification unit 111)
The speed / position specifying unit 111 obtains the phase θ _s of the estimated magnetic flux ψ _s from the estimated magnetic flux ψ _s (estimated magnetic flux ψ _α , ψ _β ). Specifically, the speed / position specifying unit 111 obtains the phase θ _s of the estimated magnetic flux Ψ _s by the equation (7). Further, the speed / position specifying unit 111 uses the phase θ _s (n) obtained in the current control cycle and the phase θ _s (n−1) obtained in the previous control cycle according to the equation (8). The estimated speed ω _r is obtained. The speed / position specifying unit 111 is a known phase estimator. Here, T _s means a control cycle (sampling cycle) in this control. n is a time step.

(Command torque specifying unit 125)
The command torque specifying unit 125 obtains the command torque T _e ^* from the command speed ω _ref ^* and the estimated speed ω _r . Specifically, the command torque specifying unit 125 obtains the command torque _Te ^{* according} to the equation (9). K _sP in equation (9) is a proportional gain. K _sI is an integral gain. s is a Laplace operator. The command torque specifying unit 125 is a known PI compensator.

(Torque deviation calculator 121)
Torque deviation calculation unit 121, the command torque T _e ^* and the estimated torque T _e and the deviation (torque deviation ΔT: T _e ^* -T _e) Request. A known operator can be used as the torque deviation calculation unit 121.

(Position specifying unit 190)
The position specifying unit 190 specifies the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* from the phase θ _s and the torque deviation ΔT. Specifically, using the torque deviation ΔT, a movement amount (rotation amount) Δθ _s for each control period in which the phase of the motor magnetic flux vector should move is specified, and the motor magnetic flux vector estimated as the specified movement amount Δθ _s The phase θ _s ^* of the command magnetic flux vector is specified using the phase θ _{s of} More specifically, the movement amount Δθ _s of the motor magnetic flux vector is obtained by the equation (10). Using the equation (11), the phase θ _s ^* is obtained. K _θP in equation (10) is a proportional gain. K _θI is an integral gain. The position specifying unit 190 brings the torque deviation ΔT close to zero. In this respect, it can be said that the position specifying unit 190 constitutes a torque compensation mechanism. When the position specifying unit 190 is incorporated in a digital control device such as a DSP or a microcomputer, the integrator required for the calculation in Expression (10) can be configured as a discrete system.

(Modified amplitude specifying unit 124)
Modifying amplitude determination unit 124, the voltage limit value V _am and limitations axis voltage v _{Arufa_lim,} obtains a correction amplitude ΔΨ from _v β_lim. As shown in FIG. 5, the corrected amplitude specifying unit 124 includes an amplitude calculation unit 128, a voltage deviation calculation unit 129, a voltage deviation compensation unit 130, a constant unit 131, and a switching unit 132.

The amplitude calculation unit 128 _obtains the amplitude (magnitude) V _a of the limit axis voltage from the limit axis voltages v _{α_lim} and v _{β_lim} . Specifically, the amplitude calculation unit 128 calculates the amplitude V _{a according} to Expression (12). As understood from the equation (12), in this embodiment, the amplitude V _a is calculated as the square sum of squares of the components (v _{α_lim} and v _{β_lim} ) of the limit axis voltage.

The voltage deviation calculation unit 129 acquires the voltage limit value V _am and the amplitude V _a of the limit axis voltage, and obtains the deviation (voltage deviation ΔV: V _am −V _a ). A known operator can be used as the voltage deviation calculation unit 129. The voltage limit value V _am is a threshold value larger than the magnitudes (= √6 / 4 × V _dc ) of the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* when the modulation degree is 1. .

The voltage deviation compensation unit 130 obtains the first corrected amplitude ΔΨ ₁ from the voltage deviation ΔV. Specifically, the voltage deviation compensator 130 calculates the _first corrected amplitude ΔΨ _{1 according} to equations (13) and (14). K _1P in equation (14) is a proportional gain. K _1I is an integral gain. The voltage deviation compensation unit 130 is a known PI compensator. As the voltage deviation compensation unit 130, a compensator other than the PI compensator can be used.

  The constant unit 131 has constants that can be acquired by the switching unit 132. In this embodiment, the constant is zero.

The switching unit 132 operates according to the flowchart of FIG. Step S101 in FIG. 6 is a step in which the switching unit 132 determines whether or not the amplitude V _a of the limit axis voltage is larger than the voltage limit value V _am . If the amplitude V _a is determined to be greater than the voltage limit value V _am, the process proceeds to step S102. If the amplitude V _a is determined to be below the voltage limit value V _am, the process proceeds to step S103. In step S102, the switching unit 132 selects the first corrected amplitude ΔΨ1 as the corrected amplitude ΔΨ. In step S103, the switching unit 132 selects zero as the correction amplitude ΔΨ.

(Temporary setting unit 126)
The temporary setting unit 126 specifies a temporary amplitude from the command torque _Te ^* . The temporary setting unit 126 can be configured using a lookup table, an operator storing a calculation formula (approximation formula), or the like. When using a lookup table, it is possible to prepare in advance a lookup table that represents the correspondence between the command torque _Te ^* and the temporary amplitude. Formulas for operators can also be prepared in advance. Such a lookup table and calculation formula can be set based on measurement data or theory performed in advance.

In the present embodiment, the temporary setting unit 126 is configured so that the maximum torque / current control (MTPA) is executed when the flux-weakening control is not performed (when the switching unit 132 selects zero as the correction amplitude ΔΨ). ing. MTPA is control for generating maximum torque with minimum current. In this embodiment, the provisional amplitude for MTPA may be referred to as provisional amplitude | Ψ _{s_MTPA} ^* | or MTPA command amplitude | Ψ _{s_MTPA} ^* |. The provisional amplitude | Ψ _{s_MTPA} ^* | is set based on the following concept. When a motor having no magnetic saliency is used as the three-phase motor 102, the amplitude | Ψ _s | of the motor magnetic flux vector and the motor torque T are approximated by equations (15A) and (15B). | Ψ _a | is a magnetic flux parameter. The magnetic flux parameter | Ψ _a | is a constant given as the amplitude of a magnet magnetic flux vector (also called a field magnetic flux vector) created by a permanent magnet in the three-phase motor 102. L is an inductance per phase of the armature winding of the three-phase motor 102. i _q is a q-axis current. P _n is the number of pole pairs of the motor. Expression (15C) is derived from Expressions (15A) and (15B). By substituting T for the command torque T _e ^* and | Ψ _s | for the temporary amplitude | Ψ _{s_MTPA} ^* |, respectively, a relational expression between the command torque T _e ^* and the temporary amplitude | Ψ _{s_MTPA} ^* | is derived. If this relational expression is used, the temporary amplitude | Ψ _{s_MTPA} ^* | can be obtained from the command torque T _e ^* . Of course, a conversion table can also be created. When a motor having magnetic saliency is used as the three-phase motor 102, the amplitude | ψ _s | of the motor magnetic flux vector and the motor torque T are approximated by equations (15D) and (15E). L _d is the d-axis inductance. L _q is a q-axis inductance. i _d is a d-axis current. The d-axis current i _d and the q-axis current i _q generally satisfy the relationship of Expression (15F). Expressions (15D), (15E), and (15F) provide a conversion table that can specify the amplitude | Ψ _s | of the motor magnetic flux vector from the motor torque T without using the variables i _d and i _q . A T command torque T _e ^*, | ^Ψ _s | a temporary amplitude _| Ψ s_MTPA ^* | to by replacing each temporary amplitude from the command torque T _{e ^*} | ^{Ψ s_MTPA *} | can be specified. Regarding MTPA, reference is made to known documents such as Yoji Takeda, Shigeo Morimoto, Nobuyuki Matsui, Yukio Honda, “Design and Control of Embedded Magnet Synchronous Motor”, Ohm Co., Ltd., issued on October 25, 2001, etc. Become. As a matter of course, the temporary setting unit 126 can also specify a temporary amplitude that is not for MTPA.

(Adder 127)
Adding unit (correction unit) 127, temporary amplitude _| Ψ s_MTPA ^* | to adds the corrected amplitude [Delta] [Psi], instruction amplitude | Ψ _s ^* | to identify. As the adder 127, a known operator can be used.

As described above, the amplitude specifying unit 140 is configured by the corrected amplitude specifying unit 124, the temporary setting unit 126, and the adding unit (correcting unit) 127. The operation of the amplitude specifying unit 140 can be summarized as follows. That is, the amplitude determination unit 140, when the amplitude V _a (magnitude of the estimated motor voltage vector) is first greater than the voltage limit value V _am, between the amplitude V _a and the voltage limit value V _am The command amplitude | Ψ _s ^* | is specified through control including voltage feedback control for converging the deviation to zero. Specifically, the amplitude specifying unit 140 sets the temporary amplitude | Ψ _{s_MTPA} ^* | of the command magnetic flux vector using the temporary setting unit 126, and _performs voltage feedback control using the modified amplitude specifying unit 124 in the first case. The correction amplitude ΔΨ is specified through the control including the correction, and the correction amplitude 127 is added to the temporary amplitude | Ψ _{s_MTPA} ^* | using the correction unit 127 to specify the command amplitude | Ψ _s ^* |. In more specifically, the first case, the voltage deviation compensation unit 130 of the correction amplitude determination unit 124, a modified amplitude ΔΨ so that the deviation between the amplitude V _a and the voltage limit value V _am converges to zero Execute voltage feedback control as the manipulated variable. In the first case, the corrected amplitude specifying unit 124 specifies a corrected amplitude ΔΨ smaller than zero. Needless to say, voltage feedback control is feedback control using voltage. The feedback control is a control for comparing the control amount with a target value by feedback and generating an operation amount so as to match them as defined in Japanese Industrial Standard JIS Z8116 (1994). In the voltage feedback control of this embodiment, the control amount is the magnitude (amplitude V _a ) of the motor voltage vector, the target value is the voltage limit value V _am , and the operation amount is the corrected amplitude ΔΨ.

The switching unit 132 can be described as follows. That is, the switching unit 132 is configured to switch between the first mode and the second mode. The first mode is a mode in which the voltage deviation compensator 130 executes voltage feedback control using the modified amplitude ΔΨ as an operation amount so that the deviation between the amplitude V _a and the voltage limit value V _am converges to zero. The second mode is a mode in which the constant is a modified amplitude ΔΨ. Then, the switching unit 132, the first case the first mode is selected, in the case of the second amplitude V _a is less than or equal to the voltage limit value V _am selects the second mode.

As described above, the voltage limit value _Vam is a threshold value that is larger than the magnitude of the first command voltage vector when the modulation degree is 1. According to the switching unit 132 of the present embodiment, it is possible to switch between the flux-weakening control (first mode) and another control (second mode) in a situation where the inverter 101 is operating in the overmodulation region. Specifically, the other control is MTPA. In the present embodiment, since zero is stored in the constant portion 131, the command amplitude | Ψ _s ^* | becomes | Ψ _s ^* | = | Ψ _{s_MTPA} ^* | + 0 = | Ψ _{s_MTPA} ^* | in the second mode. It is. That is, according to the switching unit 132 of the present embodiment, it is possible to switch between the flux-weakening control and the MTPA in a situation where the inverter 101 is operating in the overmodulation region. In the present embodiment, the amplitude V _a is when the following voltage limit value V _am, regardless of whether the inverter 101 is operating in either linear region operating in the overmodulation region, MTPA is performed.

(Command magnetic flux specifying unit 112)
The command magnetic flux specifying unit 112 obtains a command magnetic flux vector ψ _s ^* (command magnetic flux ψ _α ^* , ψ _β ^* ) from the command amplitude | Ψ _s ^* | and the phase θ _s ^* . Specifically, the command magnetic fluxes Ψ _α ^* and Ψ _β ^* are obtained using the equations (16) and (17).

(Α-axis magnetic flux deviation calculator 113a, β-axis magnetic flux deviation calculator 113b)
alpha -axis magnetic flux deviation calculation unit 113a obtains the command flux [psi _alpha ^* and the estimated flux [psi _alpha, these deviations (the magnetic flux deviation ^{_{ΔΨ α: Ψ α * -Ψ α}} ) obtained. The β-axis magnetic flux deviation calculation unit 113b acquires the command magnetic flux Ψ _β ^* and the estimated magnetic flux Ψ _β and obtains the deviation (magnetic flux deviation ΔΨ _β : Ψ _β ^* -Ψ _β ). As the magnetic flux deviation calculators 113a and 113b, known operators can be used.

(Command voltage specifying unit 107)
The command voltage specifying unit 107 specifies the first command voltage vectors (axis command voltages) v _α ^* and v _β ^* from the magnetic flux deviations ΔΨ _α and ΔΨ _β and the shaft currents i _α and i _β . Specifically, the command voltage specifying unit 107 obtains the α-axis command voltage v _α ^* using Expression (18). Moreover, the command voltage specific | specification part 107 calculates | requires (beta) axis command voltage v ( _beta) ^* using Formula (19).

(Α, β / u, v, w converter 109)
The α, β / u, v, w conversion unit 109 converts the first command voltage vector (axis command voltage) v _α ^* , v _β ^* into the first command voltage vectors v _1u ^* , v _1v ^* , v _1w. Convert to ^* . Specifically, the α, β / u, v, w conversion unit 109 converts the first command voltage vectors v _α ^* , v _β ^* into the first command voltage vectors v _1u ^* , v _1v according to equation (20). ^* it is converted into v _{1 w} ^*, the first command voltage vector v _1u ^*, v _{1 v} ^*, v and outputs a _{1 w} ^*.

(Amplitude compensation unit 110)
Amplitude compensation unit 110, the first command voltage vector _{^{v 1u *, v 1v *,}} v 1w * from the second command voltage vector _{^{v 2u *, v 2v *,}} v seek _2w ^*. Even when the inverter 101 is operating in the overmodulation region, the amplitude compensation unit 110 suppresses the difference between the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* and the motor voltage vector. The Hereinafter, details of the amplitude compensation unit 110 will be described together with details of the voltage limiting unit 122.

(Details of Amplitude Compensator 110 and Motor Voltage Estimator 122)
The amplitude compensation unit 110 and the motor voltage estimation unit 122 operate effectively when the inverter 101 operates in the overmodulation region. Details of the amplitude compensation unit 110 and the motor voltage estimation unit 122 will be described with reference to FIGS. 7 and 8 are incorporated from Non-Patent Document 2. FIG.

Assuming that the reference voltage vectors v _u ^* , v _v ^* , and v _w ^* used for generating the PWM duties D _u , D _v , and D _w that define the operation of the PWM inverter 101 are the first command voltage vector v _1u ^*. , V _1v ^* , v _1w ^* are used. Even in this case, if the modulation degree _mi is 1 or less, the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* and the motor voltage vectors v _u , v _v , v _w are the same. It becomes. However, when the modulation factor m _i is greater than 1, the first command voltage vector ^{_{v 1u *, v 1v *,}} v 1w * and the voltage vector v _{u, v} v, v do not match exactly the _w. This is because the motor voltage vector that can be output from the inverter 101 is restricted based on the DC link voltage _Vdc . Expressed quantitatively, the gain G (m of the inverter 101 when the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* are used as the reference voltage vectors v _u ^* , v _v ^* , v _w ^* . _i), in the overmodulation region is greater than the degree of modulation m _i is 1 changes as shown in equation (21). Equation (21), the gain G (m _i) is 1 when modulation factor m _i is 1, it indicates that the degree of modulation m _i is decreased more you increase from 1.

In the present embodiment, the amplitude compensation unit 110 is provided in consideration of a decrease in the gain of the inverter 101 in the overmodulation region. When the inverter 101 is operating in the overmodulation region, the amplitude compensator 110 multiplies the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* by the amplitude compensation coefficient k to generate the second command The voltage vectors v _2u ^* , v _2v ^* , v _2w ^* are specified. Amplitude compensation coefficient k is larger coefficient as becomes the greater the degree of modulation m _i is a factor greater than 1. Since the amplitude compensation unit 110 uses such an amplitude compensation coefficient k, the waveform of the U-phase component v _1u ^* of the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* , the second command voltage vector v ^{_{2u *, v 2v *, v}} 2w * for U-phase component v _2u ^* waveforms and the motor voltage vector v _{u, v} v, v waveforms of U-phase component v _u and _w has a waveform shown in FIG. 7 . As can be understood from FIG. 7, the amplitude compensation unit 110 increases the amplitude of the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* in anticipation of a decrease in the gain of the inverter 101 in the overmodulation region. The second command voltage vectors v _2u ^* , v _2v ^* , v _2w ^* are generated. Specifically, the amplitude compensation coefficient k is associated with the modulation index m _i as shown in FIG. If the amplitude compensation coefficient k is defined as shown in FIG. 8, the fundamental wave component of each phase voltage of the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* and the motor voltage vectors v _u , v _v , v a fundamental component of the phase voltages of _w can be matched. That is, the fundamental wave component of the U-phase voltage v _1u ^{* and} the fundamental wave component of the U-phase voltage v _u are matched, and the fundamental wave component of the V-phase voltage v _1v ^* is matched with the fundamental wave component of the V-phase voltage v _v. Thus, the fundamental wave component of the W-phase voltage v _1w ^{* and} the fundamental wave component of the W-phase voltage v _w can be matched.

The operation of the amplitude compensation unit 110 can be summarized as follows. That reference, the amplitude compensation unit 110, the first command voltage vector _{^{v 1u *, v 1v *,}} v 1w * PWM duty for defining the operation of the PWM inverter 101 from D _u, D _v, used for generating D _w The voltage vectors v _u ^* , v _v ^* , v _w ^* are specified. Specifically, the operation of the amplitude compensation unit 110 can be described in association with the operation region of the PWM inverter 101. That is, when the operation region of the PWM inverter 101 and the theoretical value of the modulation factor m _i of the transition from the linear region to the overmodulation region is defined as a boundary value, the reference voltage vector _{^{v u *, v v *,}} v w * is If the degree of modulation m _i is less than the boundary value, the first command voltage vector _{^{v 1u *, v 1v *,}} v 1w * which is the same as the second command voltage vector _{^{v 2u *, v 2v *,}} v _2w ^* . Reference voltage vector _{^{v u *, v v *,}} v w * , if modulation factor m _i is greater than the boundary value is larger as becomes the greater the degree of modulation m _i is a large amplitude compensation coefficient k than 1 comprising amplitude compensation coefficient k the first command voltage vector _{^{v 1u *, v 1v *,}} v 1w * to multiply the second command voltage vector v obtained by _2u ^*, v _2v ^*, a v _2w ^*. In this embodiment, since the sine wave PWM method is adopted, the boundary value is 1 (when the third harmonic injection modulation method is adopted, the boundary value is 2 / √3). More specifically, the amplitude compensation coefficient k is the product of the gain of the PWM inverter 101 and the amplitude of the fundamental component of each phase voltage of the second command voltage vectors v _2u ^* , v _2v ^* , v _2w ^* . 1 is a coefficient determined to coincide with the amplitude of the fundamental component of each phase voltage of the command voltage vectors v _1u ^* , v _1v ^* , and v _1w ^* . Although Mind just in case, the gain here, the second command voltage vector _{^{v 2u *, v 2v *,}} v 2w * and the motor voltage vector v _u, v _v, represents the relationship between v _w Thus, it is different from G (m _i ) mentioned in the description of the tentative case.

The motor voltage estimation unit 122 can be described as follows. That is, as described above, the motor control device 100 uses the reference voltage vectors v _u ^* , v _v ^* , v _w ^* used to generate the PWM duties D _u , D _v , D _w that define the operation of the PWM inverter 101 ^. (in this embodiment, the second command voltage vector ^{_{v 2u *, v 2v *,}} v 2w *) is specified. In the present embodiment, the motor voltage estimation unit 122 uses the relationship between the waveforms of the reference voltage vectors v _u ^* , v _v ^* , and v _w ^{* and} the waveforms of the motor voltage vectors v _u , v _v , and v _w. , motor voltage vector v _{u, v} v, identifies a v _w (limit voltage vector _v u_lim, v v_lim, seek _v w_lim). That is, as shown in FIG. 7, an upper limit (1 in this embodiment) that can be output for each phase voltage of the motor voltage vectors v _u , v _v , and v _w that the inverter 101 outputs to the three-phase motor 102. / 2V _dc ) and a lower limit (−1 / 2V _dc in this embodiment) that is smaller than the upper limit by the DC link voltage V _dc . The motor voltage estimation unit 122 replaces the reference voltage vectors v _u ^* , v _v ^* , and v _w ^* with the voltage of the phase whose voltage is higher than the upper limit, and the voltage of the phase whose voltage is lower than the lower limit with the lower limit. _Are estimated as motor voltage vectors v _u , v _v and v _w . As can be understood from FIG. 7, by doing this, it is possible to ensure the estimation accuracy of the motor voltage vectors v _u , v _v , and v _w even when the inverter 101 is operating in the overmodulation region.

As described above, the inverter 101 of this embodiment operates based on the sine wave PWM method. The linear region of this embodiment is a region where the modulation degree _mi is 1 or less. The overmodulation region of this embodiment, the degree of modulation m _i is larger region than 1. The inverter 101 of another example operates based on the third harmonic injection modulation method. In this alternative example, when the modulation degree m _i is 2 / √3 (≒ 1.154) or less, the first command voltage vector ^{_{v 1u *, v 1v *,}} v of _{1 w} ^* of the respective phase voltage amplitude The amplitude of each phase voltage of the motor voltage vectors v _u , v _v and v _w ^* changes linearly with respect to the change. When the modulation degree is larger than 2 / √3, the first command voltage vector ^{_{v 1u *, v 1v *,}} v 1w * motor voltage vector v _u relative change in amplitude of each phase voltage, v _v, The amplitude of each phase voltage of v _w ^* changes nonlinearly. Therefore, in this alternative example, the area modulation m _i is 2 / √3 or less can be defined as the linear region, a region larger than the modulation factor m _i is 2 / √3 can be defined as the overmodulation region. In short, the degree of modulation serving as the boundary between the linear region and the overmodulation region is a value of 1 or more corresponding to the modulation method. Further, as understood from the above description, when the sinusoidal PWM method is adopted, the voltage limit value for performing the flux weakening control in the overmodulation region is the first limit when the modulation degree _mi is 1. When the third harmonic injection modulation method is adopted, the voltage limit value for performing the flux weakening control in the overmodulation region can be set to a modulation factor of 2 / √3. Can be set to a value larger than the magnitude of the first command voltage vector. Regardless of which modulation method is employed, the voltage limit value is a threshold value that is larger than the magnitude of the first command voltage vector when the degree of modulation _mi is 1. Therefore, the modulation method adopted by the inverter according to the present disclosure is limited to the sine wave PWM method with the expression “threshold value larger than the magnitude of the first command voltage vector when the modulation degree is 1”. Should not. Other modulation schemes can be applied not only to this embodiment but also to embodiments and modifications described later.

(Effect of this embodiment)
As described above, the operation region of the PWM inverter 101 includes an overmodulation region and a linear region. When the operation region of the PWM inverter 101 is defined as a boundary value theoretical value of the modulation index m _i of the transition from the linear region to the overmodulation region, the voltage limit value V _am, the time modulation factor m _i is the boundary value It can be said that it is larger than the magnitude of the first command voltage vector. In short, in the present embodiment, the mode is smoothly switched at an appropriate timing when the inverter 101 is operating in the overmodulation region. That is, in the first case where the amplitude V _a of the limit axis voltage is larger than the voltage limit value V _{am, the} flux weakening control is performed (first mode). If the amplitude V _a of the second or less voltage limit value V _am is, MTPA is carried out (second mode).

In the technique described in Non-Patent Document 2, the magnetic flux weakening control is executed by feedforward by giving a command amplitude by a mathematical expression. As described above, the flux-weakening control described in Non-Patent Document 2 cannot be said to have high accuracy. In contrast, in the present embodiment, the voltage feedback control for specifying a correction amplitude ΔΨ so that the deviation is converged to zero between the amplitude V _a and the voltage limit value V _am are executed. Due to the feedback control, even if there is an error in the resistance parameter (note that the resistance parameter is included in Equation (2) of Non-Patent Document 2), an error in the amplitude compensation coefficient k, etc., The deviation between the amplitude V _a and the voltage limit value V _am converges to zero. Therefore, according to the voltage feedback control of the present embodiment, it is possible to execute the flux weakening control with high accuracy while operating the inverter 101 in the overmodulation region. That is, in the situation where the flux-weakening control is performed, the amplitude V _a of the limit axis voltage coincides with the voltage limit value _Vam with high accuracy.

According to the motor control apparatus 100 of the first embodiment, the speed of the three-phase motor 102 matches the command speed ω _ref ^* with high accuracy, including when the inverter 101 is operating in the overmodulation region. That is, the motor control device 100 can be suitably used at the time of driving the compressor in the refrigeration air conditioner.

As described above, in the present embodiment, the fundamental wave component of each phase voltage of the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* and each phase voltage of the motor voltage vectors v _u , v _v , v _w . The amplitude compensator 110 is provided for the purpose of matching the fundamental wave component. However, the amplitude compensation unit 110 can be omitted. In that case, the second command voltage vector ^{_{v 2u *, v 2v *,}} v 2w * instead first command voltage vector ^{_{v 1u *, v 1v *,}} v 1w * reference voltage vector v _u ^*, v _v ^*, the v _w ^*. The motor voltage estimation unit 122 estimates the motor voltage vector applied to the three-phase motor 102 using the first command voltage vectors v _1u ^* , v _1v ^* , v _1w ^* (phase voltage limit vector). _v u_lim, v v_lim, will be determined _v w_lim). Duty generation unit 104, first command voltage vector ^{_{v 1u *, v 1v *,}} v 1w * from the duty D _u, D _v, will produce a D _w.

(simulation)
FIG. 9 is a graph showing a simulation result. In this simulation, there is a model that can analyze various voltages when the same control as in the present embodiment is performed except that the inverter operates based on the third harmonic injection modulation method instead of the sine wave PWM modulation method. It has been adopted. In the simulation, the DC link voltage was 20V. When the inverter is operating in the linear region, the maximum value of the magnitude of the motor voltage vector output from the inverter is 2 / √3 × (√6 / 4 × 1 × 20) ≈14.14V. √6 / 4 × 1 × 20 represents the magnitude of the motor voltage vector output from the inverter operating based on the sine wave PWM modulation method when the modulation degree is 1 and the DC link voltage is 20V. 2 / √3 is a coefficient for improving the voltage utilization factor in the third harmonic injection modulation method. In the simulation, the voltage limit value was set to 15V. That is, this simulation is set so that the degree of modulation is maintained at 15 / (√6 / 4 × 1 × 20) ≈1.22 in the situation where the flux-weakening control is performed. However, in this simulation, an error is given to the amplitude compensation coefficient. Specifically, the amplitude compensation coefficient should be larger than 1 in the overmodulation region, but is always 1 in this simulation.

The upper graph in FIG. 9 represents the time change of the rotation speed of the three-phase motor. The horizontal axis of this graph is time (unit: second). The vertical axis of this graph represents the rotation speed (unit: rad / sec) of the three-phase motor. The lower graph of FIG. 9 represents the time change of the magnitude or amplitude of various voltages. The horizontal axis of this graph is time (unit: second). The vertical axis of this graph represents the magnitude or amplitude (unit: volts) of the voltage. V _a is the amplitude of the limit axis voltage. V _ref1 is the magnitude of the first command voltage vector. V _LLmax is the maximum value of the magnitude of the motor voltage vector output from the inverter when the inverter is operating in the linear region. V _LLmax is approximately 14.14 V as calculated above. V _am is a voltage limit value, and is 15 V as described above. As shown in Figure 9, during the period in which the inverter is operating in the overmodulation region, the magnitude V _ref1 of the first command voltage vector does not coincide exactly with the voltage limit V _am. This is because there is an error in the amplitude compensation coefficient. However, the amplitude V _a of the limit axis voltage coincides with the voltage limit value V _am with high accuracy. This is because doing deviation voltage feedback control to converge to zero between the amplitude V _a and the voltage limit value V _am. From the simulation result, according to the voltage feedback control, even if the amplitude compensation coefficient is the error, it can be seen that it is possible to match the amplitude V _a of the limit-axis voltage to the voltage limit value V _am with high accuracy. In reality, there are errors derived from other than the amplitude compensation coefficient. However, it will be apparent to those skilled in the art that even if there is an error other than the error of the amplitude compensation coefficient, these match with high accuracy. That is, it can be seen from the simulation results that the magnetic flux control can be performed with high accuracy according to the voltage feedback control, and the accuracy does not greatly decrease even in the region where the modulation degree is high, particularly in the overmodulation region.

(Modification 1-1A)
Hereinafter, the motor control device of Modification 1-1A will be described. In the modified example 1-1A, the same parts as those in the first embodiment are denoted by the same reference numerals, and the description thereof may be omitted.

  As illustrated in FIG. 10A, the motor control unit 203a of the modified example 1-1A includes an amplitude specifying unit 240a instead of the amplitude specifying unit 140. The amplitude specifying unit 240a includes a modified amplitude specifying unit 224a instead of the corrected amplitude specifying unit 124.

(Modified amplitude identification unit 224a)
The corrected amplitude specifying unit 224a corrects the corrected amplitude from the motor current vector (axis current i _α , i _β ) of the three-phase motor 102, the estimated magnetic fluxes ψ _α , ψ _β , the voltage limit value V _am and the limit axis voltages v _{α_lim} , v _{β_lim.} Find ΔΨ. As shown in FIG. 11A, the corrected amplitude specifying unit 224a includes an error parameter calculating unit 231a, an error parameter deviation calculating unit 232, and an inner product compensating unit 233 in addition to the components of the corrected amplitude specifying unit 124. .

The corrected amplitude specifying unit 224a performs the same operation regardless of whether the d-axis inductance L _d and the q-axis inductance L _q are different from each other. Specifically, when there is an inductance difference, a value between the d-axis inductance and the q-axis inductance can be used as the inductance L. Further, if the magnetic saliency is not large, there is no harm in handling the L = L _d. That is, as the inductance value, a d-axis inductance value, a value larger than the d-axis inductance and smaller than the q-axis inductance, or a value smaller than the d-axis inductance and larger than the q-axis inductance can be used.

Japanese Patent No. 4972135 is helpful in setting the inductance L as described above. Patent Document 1 describes a technique related to a dm-qm coordinate system. The dm-qm coordinate system makes it possible to treat a motor having magnetic saliency such as a permanent magnet synchronous motor having an embedded magnet structure in the same manner as a permanent magnet synchronous motor having no magnetic saliency. Maximum torque control (maximum torque / current control) can be performed by using the dm-qm coordinate system and setting the dm-axis current (γ-axis current in the control coordinate system) to zero. When the three-phase motor 102 has magnetic saliency, the d-axis current can be replaced with the dm-axis current, the magnet magnetic flux vector ψ _a can be replaced with the extended linkage magnetic flux vector Φ _exm , and the inductance L can be replaced with the virtual inductance L _m. it can. For details of the dm-axis current, the extended flux linkage vector Φ _exm, and the virtual inductance L _m , Patent Document 1 (Formula 36 and paragraphs 0182 to 0183) can be referred to. Note that L _m satisfies L _d ≦ L _m <L _q . When there is no inductance difference, the dm-qm coordinate system and the general dq coordinate system coincide with each other, and L _m = L _d = L _q may be set. That is, the idea about the case where there is an inductance difference includes the idea about the case where there is no inductance difference.

(Error parameter calculator 231a)
The error parameter calculation unit 231a calculates an error parameter ε from the virtual inductance (inductance of the three-phase motor 102) L _m , the shaft currents i _α , i _β, and the estimated magnetic flux ψ _s (estimated magnetic flux ψ _α , ψ _β ). . Specifically, first, estimate the armature reaction magnetic flux (obtain the estimated armature reaction flux L _m i _a). The alpha-axis component and beta-axis component of the estimated armature reaction flux L _m i _a, respectively estimated armature reaction flux L _m i _α, referred to as estimated armature reaction flux L _m i _β. The estimated armature reaction magnetic flux L _m i _α and the estimated armature reaction magnetic flux L _m i _β are the products of the virtual inductance L _m and the axial currents i _α and i _β . Next, a magnetic flux vector is estimated from the estimated magnetic flux Ψ _s (estimated magnetic flux Ψ _α , Ψ _β ) and the estimated armature reaction magnetic flux L _m i _a (estimated armature reaction magnetic flux L _m i _α , L _m i _β ). ( _Determine estimated magnetic flux Ψ ′ _ae ). The α-axis component and β-axis component of the estimated magnet magnetic flux Ψ ′ _ae are described as estimated magnet magnetic flux Ψ ′ _aeα and Ψ ′ _aeβ , respectively. Specifically, as shown in equations (22) and (23), the estimated magnetic flux Ψ ′ _{aeα is} obtained by subtracting the estimated armature reaction magnetic fluxes L _m i _α and L _m i _β from the estimated magnetic fluxes Ψ _α and Ψ _β. , Ψ ′ _aeβ . Next, an error parameter ε is calculated from the estimated magnet magnetic fluxes ψ ′ _aeα , ψ ′ _aeβ and the axial currents i _α , i _β as shown in Expression (24).

(Error parameter deviation calculator 232)
The error parameter deviation calculation unit 232 acquires the command error parameter ε ^* and the error parameter ε, and obtains a deviation (error parameter deviation Δε: ε ^* −ε). A known operator can be used as the error parameter deviation calculation unit 232. The command error parameter ε ^* is the same as the corrected amplitude ΔΨ of the first embodiment.

(Inner product compensation unit 233)
The inner product compensation unit 233 acquires the error parameter deviation Δε and specifies the corrected amplitude ΔΨ so that it becomes zero. The inner product compensation unit 233 of Modification 1-1A is a PI compensation unit. Therefore, the inner product compensation unit 233 obtains the corrected amplitude ΔΨ by performing a proportional / integral calculation with the error parameter deviation Δε as an input, as shown in Expression (25).

The operations of the amplitude specifying unit 240a (the corrected amplitude specifying unit 224a, the temporary setting unit 126, and the addition unit (correction unit) 127) can be summarized as follows. That is, the amplitude determination section 240a, in the case where the amplitude V _a (magnitude of the estimated motor voltage vector) is first greater than the voltage limit value V _am, between the amplitude V _a and the voltage limit value V _am The command amplitude is specified through control including voltage feedback control for converging the deviation to zero. Specifically, the amplitude specifying unit 240a sets the temporary amplitude of the command magnetic flux vector using the temporary setting unit 126, and in the first case, the corrected amplitude is controlled through control including voltage feedback control using the corrected amplitude specifying unit 224a. identify [Delta] [Psi], temporary amplitude using the correction unit ₁₂₇ | Ψ s_MTPA ^* | corrected amplitude [Delta] [Psi] an addition to instruction amplitude | Ψ _s ^* | to identify. More specifically, in the first case, the voltage deviation compensator 130, a first command error parameters epsilon ^* operations so that the deviation is converged to zero between the amplitude V _a and the voltage limit value V _am Voltage feedback control is executed. In the first case, the inner product compensator includes estimated magnet magnetic fluxes Ψ ′ _aeα and Ψ ′ _aeβ (estimated magnetic flux vectors of the three-phase motor 102) and axial currents i _α and i _β (motor current vectors). The first feedback control is executed using the corrected amplitude ΔΨ as the manipulated variable so that the deviation between the error parameter ε and the first command error parameter ε ^* , which is the inner product (second inner product) of, converges to zero.

The switching unit 132 can be described as follows. That is, the switching unit 132 is configured to switch between the first mode and the second mode. In the first mode, the voltage deviation compensator 130 executes voltage feedback control using the first command error parameter ε ^* as an operation amount so that the deviation between the amplitude V _a and the voltage limit value V _am converges to zero. In this mode, the inner product compensator 233 executes the first feedback control using the modified amplitude ΔΨ as the manipulated variable so that the deviation between the first command error parameter ε ^* and the error parameter ε converges to zero. . In the second mode, the inner product compensator 233 performs the second feedback control using the modified amplitude ΔΨ as the manipulated variable so that the deviation between the constant second command error parameter ε ^* and the error parameter ε converges to zero. Is the mode to execute. Then, the switching unit 132, the first case the first mode is selected, in the case of the second amplitude V _a is less than or equal to the voltage limit value V _am selects the second mode.

(Effect of Modification 1-1A)
Also in the modified example 1-1A, as in the first embodiment, when the inverter 101 is operating in the overmodulation region, the mode is switched at an appropriate timing. That is, in the first case where the amplitude V _a of the limit axis voltage is larger than the voltage limit value V _{am, the} flux weakening control is performed (first mode). Specifically, in the first case, a corrected amplitude ΔΨ suitable for the flux-weakening control is generated. If the amplitude V _a of the second or less voltage limit value V _am, it fixes the amplitude ΔΨ adapted to MTPA produced (second mode).

Furthermore, in Modification 1-1A, MTPA can be executed with very high accuracy in the second case. Hereinafter, this point will be described. In order to establish MTPA, it is necessary to make the estimated magnetic flux Ψ ′ _ae and the axial currents i _α and i _β orthogonal to each other. Therefore, in the modified example 1-1A, the command error parameter ε ^* is set to zero in order to make the inner product of the estimated magnet magnetic flux ψ ′ _ae and the shaft currents i _α and i _β zero in the second case. . That is, in the modified example 1-1A, the provisional amplitude is the provisional amplitude | Ψ _{s_MTPA} ^* | suitable for MTPA. In the second case, the provisional amplitude is corrected by being corrected by the correction amplitude ΔΨ suitable for MTPA. An amplitude ΔΨ is generated. Therefore, according to Modification 1-1A, MTPA can be performed with very high accuracy in the second case. That is, if the motor control unit 203a of the modified example 1-1A is used, the three-phase motor 102 can be driven more efficiently than when the motor control unit 103 is used.

(Modification 1-1B)
Hereinafter, the motor control device of Modification 1-1B will be described. In addition, in the modified example 1-1B, the same parts as those in the modified example 1-1A are denoted by the same reference numerals, and description thereof may be omitted.

  As illustrated in FIG. 10B, the motor control unit 203b of the modification 1-1B includes an amplitude specifying unit 240b instead of the amplitude specifying unit 240a. The amplitude specifying unit 240b includes a modified amplitude specifying unit 224b instead of the corrected amplitude specifying unit 224a.

(Modified amplitude identification unit 224b)
As illustrated in FIG. 11B, the modified amplitude specifying unit 224b includes an error parameter calculation unit 231b instead of the error parameter calculation unit 231a in the modified example 1-1A.

(Error parameter calculation unit 231b)
The error parameter calculator 231b calculates an error parameter ε from the virtual inductance L _m (inductance of the three-phase motor 102), the shaft currents i _α , i _β and the estimated magnetic flux ψ _s (estimated magnetic flux ψ _α , ψ _β ). . Specifically, the inner product Ψ _α i _α + Ψ _β i _β of the estimated magnetic flux Ψ _s (estimated magnetic flux Ψ _α , Ψ _β ) and the axial currents i _α , i _β is specified. A product L _m (i _α ² + i _β ² ) of the virtual inductance and the square of the amplitude of the current is specified from the virtual inductance L _m and the axial currents i _α and i _β . The product L _m (i _α ² + i _β ² ) is subtracted from the inner product Ψ _α i _α + Ψ _β i _β to determine the difference Ψ _α i _α + Ψ _β i _β −L _m (i _α ² + i _β ² ), and the error parameter Get ε. The calculation performed by the error parameter calculation unit 231b is expressed by Expression (26).

The modified amplitude specifying unit 224b of the modified example 1-1B uses an expression different from the formula used in the modified amplitude specifying unit 224a of the modified example 1-1A, but the same inner product (estimated magnet magnetic flux as in the modified example 1-1A). (Second inner product of ψ ′ _aeα , ψ ′ _aeβ and axial currents i _α , i _β ) is specified as an error parameter ε, and the same corrected amplitude ΔΨ is specified. Therefore, if the motor control unit 203b of the modified example 1-1B is used, the same effect as that obtained when the motor control unit 203a is used can be obtained. In this specification, “the inner product of the estimated magnet magnetic fluxes Ψ ′ _aeα and Ψ ′ _aeβ and the axial currents i _α and i _β ” is specified as “the estimated magnet magnetic fluxes Ψ ′ _aeα and Ψ ′ _aeβ and the axial current i _α. , I _β is an expression indicating that the inner product with the result is specified, and is not an expression indicating the calculation process.

  In the following, examples using the corrected amplitude specifying unit 224a may be described, but in these examples, the corrected amplitude specifying unit 224b can be provided instead of the corrected amplitude specifying unit 224a.

(Modification 1-2)
Hereinafter, the motor control device of Modification 1-2 will be described. In the modified example 1-2, the same reference numerals are given to the same parts as the modified example 1-1A, and the description may be omitted.

  As shown in FIG. 12, the motor control unit 203c of the modified example 1-2 includes an amplitude specifying unit 240c instead of the amplitude specifying unit 240a. The amplitude specifying unit 240 c includes a temporary setting unit 226 instead of the temporary setting unit 126.

(Temporary setting unit 226)
The temporary setting unit 226 sets (specifies) the temporary amplitude of the command magnetic flux vector Ψ _s ^* . In the modified example 1-2, a temporary amplitude that is a constant is stored in the temporary setting unit 226 in advance. In the modified example 1-2, since the modified amplitude specifying unit 224a is used, MTPA can be realized in the second case even if the temporary amplitude is a constant. The temporary amplitude of the modified example 1-2 is the magnetic flux parameter Ψ _a . However, the provisional amplitude is not limited to the magnetic flux parameter Ψ _a . The temporary amplitude of the modified example 1-2 is expressed as a temporary amplitude | Ψ _s0 ^* |.

  According to the modified example 1-2, the flux-weakening control and the MTPA can be realized with sufficient accuracy. Further, according to the modified example 1-2, it is possible to control the three-phase motor 102 with a small amount of calculation compared to the modified example 1-1A.

In the modified example 1-1A, the modified example 1-1B, and the modified example 1-2, the control using the inner product of the estimated magnet magnetic fluxes Ψ ′ _aeα and Ψ ′ _aeβ and the axial currents i _α and i _β is performed. Specifically, in the second case where the amplitude V _a is equal to or less than the voltage limit value V _am , feedback control is performed so that the deviation between the inner product and a predetermined constant (zero in the above modification) is zero. ing. More generally, in the above modification, in the second case, a physical quantity correlated with the reactive power of the three-phase motor 102 is obtained from the estimated magnetic fluxes ψ _α , ψ _β and the shaft currents i _α , i _β , The physical quantity is controlled to a predetermined value. That is, in the above modification, the inner product of the estimated magnet magnetic fluxes Ψ ′ _aeα and Ψ ′ _aeβ and the axial currents i _α and i _β is used as a physical quantity correlated with the reactive power.

Another example of the physical quantity correlated with reactive power is the first inner product of the estimated magnetic flux ψ _s (estimated motor magnetic flux vector) and the axial currents i _α , i _β (motor current vector of the three-phase motor 102). . Even when such a first inner product is used as the error parameter ε, the same control as in the modified examples 1-1A and 1-1B can be realized. That is, in the first case, the flux-weakening control can be performed. In the second case, various controls can be performed if appropriate constants are stored in the constant unit 131.

Further, in the second case, the corrected amplitude specifying unit may obtain reactive power of the three-phase motor 102 and generate an amplitude correction amount ΔΨ for controlling the reactive power to a predetermined value. In this case, the reactive power can be obtained by multiplying the inner product Ψ _α i _α + Ψ _β i _β by the command speed ω _ref ^* . The reactive power can be obtained by differentiating the phase of the estimated magnetic flux Ψ _s to estimate the speed of the three-phase motor 102 and multiplying the speed by the inner product Ψ _α i _α + Ψ _β i _β . The reactive power can also be obtained from the shaft currents i _α and i _β and the shaft voltages v _α ^* and v _β ^* . Specifically, the error parameter calculation unit is in charge of specifying the reactive power, the constant to be taken by the reactive power is stored in the constant unit, and the deviation between the reactive power and the constant is set to zero instead of the inner product compensation unit. Compensation unit (PI compensation unit or the like) can be provided. Even in such a configuration, the magnetic flux weakening control is performed in the first case.

(Second Embodiment)
Hereinafter, the motor control device of the second embodiment will be described. In addition, in 2nd Embodiment, the same code | symbol may be attached | subjected about the part similar to the modification 1-2, and description may be abbreviate | omitted.

  As illustrated in FIG. 13, the motor control unit 303 according to the second embodiment does not include the torque estimation unit 114, the command torque identification unit 125, the torque deviation calculation unit 121, and the speed / position identification unit 111. On the other hand, the motor control unit 303 includes a position estimation unit 335 and a position specification unit 336.

(Position estimation unit 335)
The position estimation unit 335 identifies the phase θ _s of the estimated magnetic flux ψ _s from the estimated magnetic flux ψ _s (estimated magnetic flux ψ _α , ψ _β ). Specifically, the position estimation unit 335 specifies the phase θ _s using the same equation as the equation (7) of the speed / position specifying unit 111.

(Position specifying unit 336)
The position specifying unit 336 obtains the phase θ _s ^* of the command magnetic flux vector ψ _s ^* from the command speed ω _ref ^* and the phase θ _s of the estimated magnetic flux ψ _s . As illustrated in FIG. 14, the position specifying unit 336 includes a multiplying unit 337 and a phase addition calculating unit 338. The multiplication unit 337 identifies the movement amount Δθ by multiplying the command speed ω _ref ^* by the control period T _s . The phase addition computing unit 338 obtains the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* by adding the movement amount Δθ and the phase θ _s of the estimated magnetic flux Ψ _s . In short, the position specifying unit 336 uses the command speed ω _ref ^* to specify the movement amount Δθ for each control cycle in which the phase of the motor magnetic flux vector should move, and uses the specified movement amount Δθ to specify the command magnetic flux vector Ψ _s. ^Specify the phase θ _s ^* of ^* .

The operation of the motor control unit 303 of the second embodiment is summarized as follows. The u, w / α, β converter 106 converts the phase currents i _u , i _w into axial currents i _α , i _β . Motor voltage estimation unit 122, the reference voltage vector ^{_{v u *, v v *,}} v w * ( in the present embodiment the second command voltage vector ^{_{v 2u *, v 2v *,}} v 2w *) from the 3-phase motor 102 It estimates the motor voltage vector that is applied (limit voltage vector _v u_lim, v v_lim, seek _v w_lim). u, v, w / α, β conversion unit 123, the phase voltage limiting vector _v u_lim, v v_lim, the v _{W_lim,} converts limiting axis voltage _v α_lim, v the _{Beta_lim.} The motor magnetic flux estimating unit 108 estimates the motor magnetic flux vector applied to the three-phase motor 102 using the shaft currents i _α and i _β and the limited shaft voltages v _{α_lim} and v _{β_lim} (determining the estimated magnetic flux Ψ _s) . ). The position estimation unit 335 estimates the phase θ _s of the motor magnetic flux vector using the estimated magnetic flux ψ _s . The position specifying unit 336 specifies the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* from the command speed ω _ref ^* and the estimated magnetic flux phase θ _s . The corrected amplitude specifying unit 224a obtains a corrected amplitude ΔΨ from the voltage limit value V _am , the limit axis voltages v _{α_lim} , v _{β_lim} , the axis currents i _α , i _β, and the estimated magnetic flux ψ _s . The temporary setting unit 226 sets (specifies) the temporary amplitude | Ψ _s0 ^* |. The adder 127 generates a command amplitude | Ψ _s ^* | from the temporary amplitude | Ψ _s0 ^* | and the modified amplitude ΔΨ. The command magnetic flux specifying unit 112 specifies the phase θ _s ^* specified by the position specifying unit 336 and the command specified by the amplitude specifying unit 240c (the corrected amplitude specifying unit 223a, the temporary setting unit 226, and the adding unit (correcting unit) 127). The command magnetic flux vector Ψ _s ^* is obtained using the amplitude | Ψ _s ^* |. The α-axis magnetic flux deviation calculating unit 113a obtains a deviation (magnetic flux deviation) ΔΨ _α between the α-axis command magnetic flux ψ _α ^* and the estimated magnetic flux ψ _α . The β-axis magnetic flux deviation calculator 113b obtains a deviation (magnetic flux deviation) ΔΨ _β between the β-axis command magnetic flux Ψ _β ^* and the estimated magnetic flux Ψ _β . The command voltage specifying unit 107 obtains first command voltage vectors (axis command voltages) v _α ^* and v _β ^* from the magnetic flux deviations ΔΨ _α and ΔΨ _β and the shaft currents i _α and i _β . α, β / u, v, w conversion unit 109, the first command voltage vector v _alpha ^*, the v _beta ^*, the first command voltage vector ^{_{v 1u *, v 1v *,}} v is converted to _{1 w} ^*. Amplitude compensation unit 110, the first command voltage vector ^{_{v 1u *, v 1v *,}} v 1w * from the second command voltage vector ^{_{v 2u *, v 2v *,}} v seek _2w ^*.

According to the motor control unit 303 of the second embodiment, the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* can be easily specified. Further, a motor magnetic flux vector estimated for specifying the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* is used, and axial currents i _α and i _β are used for the estimation. This means that the motor current vector actually flowing in the three-phase motor 102 is reflected in the specification of the phase θ _s ^* of the command magnetic flux vector ψ _s ^* . That is, this contributes to appropriately setting the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* , and helps to stabilize the driving of the three-phase motor 102.

(Modification 2-1A)
Hereinafter, the motor control device of Modification 2-1A will be described. Note that in the modified example 2-1A, the same portions as those in the second embodiment are denoted by the same reference numerals, and description thereof may be omitted.

  As illustrated in FIG. 15A, the motor control unit 403a of Modification 2-1A includes a position specifying unit 436a instead of the position specifying unit 336 of the second embodiment. In addition, the motor control unit 403a includes the torque estimation unit 114 described in the first embodiment.

(Position identifying unit 436a)
Position specifying unit 436a includes a command velocity omega _ref ^*, the estimated torque T _e, and a phase theta _s estimated flux [psi _s, identifies the command flux vector [psi _s ^* phase theta _s ^*. As illustrated in FIG. 16A, the position specifying unit 436a includes a high-pass filter 439, a gain multiplication unit 440, a speed deviation calculation unit 451, a multiplication unit 452, and a phase addition calculation unit 453.

(High pass filter 439)
High-pass filter 439, the vibration component of the estimated torque T _e identify only (torque vibration component) T _H (extraction).

(Gain multiplier 440)
Gain multiplication section 440 identifies the speed vibration component K ₁ T _H is multiplied by a gain K ₁ to the torque vibration component T _H.

  The operations of the high pass filter 439 and the gain multiplication unit 440 are expressed by Expression (27). g is a cut-off frequency, and its unit is [rad / s]. s is a Laplace operator.

(Speed deviation calculation unit 451)
Speed difference calculating unit 451 calculates a command speed omega _ref ^* and speed vibration component K ₁ T _H speed deviation (corrected command ^{_{speed) ω ref * -K 1 T H}} .

(Multiplier 452)
Multiplying unit 452 calculates a movement amount Δθ by multiplying the control period T _S of the speed deviation ω _ref ^* -K ₁ T _H.

(Phase addition operation unit 453)
The phase addition operation unit 453 obtains the phase θ _s ^* of the command magnetic flux vector ψ _s ^* by adding the movement amount Δθ to the phase θ _s of the estimated magnetic flux ψ _s .

Position specifying portion 436a of the modification 2-1A corrects the command speed omega _ref ^* using the estimated torque fluctuation component T _H, the moving amount using the corrected command speed ω _ref ^* -K ₁ T _H By specifying Δθ and using the specified movement amount Δθ (specifically, adding the phase θ _s of the motor magnetic flux Ψ _s estimated by the position estimation unit 335 and the movement amount Δθ), the command magnetic flux vector Ψ _s ^* of identifying the phase θ _s ^*. When the movement amount Δθ is specified using the torque fluctuation information as in the modified example 2-1A, the driving of the three-phase motor 102 is stabilized. That is, the motor control device of the modified example 2-1A has a higher stability than the motor control device of the second embodiment while having a simple configuration.

(Modification 2-1B)
Hereinafter, the motor control device of Modification 2-1B will be described. Note that in the modification 2-1B, the same parts as those in the modification 2-1A are denoted by the same reference numerals, and description thereof may be omitted.

  As illustrated in FIG. 15B, the motor control unit 403b of Modification 2-1B has a phase specifying unit 436b instead of the phase specifying unit 436a of Modification 2-1A.

(Phase identifying unit 436b)
The phase specifying unit 436b specifies the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* from the command speed ω _ref ^* , the estimated torque _Te, and the phase θ _s of the estimated magnetic flux Ψ _s . As illustrated in FIG. 16B, the phase specifying unit 436b includes a multiplying unit 460, a high pass filter 439, a sign inverting unit 462, a PI compensating unit 463, an adding unit 464, and a phase addition calculating unit 453. Yes.

(Multiplier 460)
The multiplication unit 460 obtains the movement amount ω _ref ^* T _S by multiplying the command speed ω _ref ^* by the control period T _S.

(Sign inversion unit 462)
Sign inversion unit 462 obtains a vibration component -T _H by multiplying -1 to torque vibration component T _H.

(PI compensation unit 463)
The PI compensation unit 463 acquires the vibration component −T _H and specifies the correction amount Δω _ref ^* T _S so that it becomes zero. Specifically, as shown in Expression (28), the correction amount Δω _ref ^* T _S is obtained by performing a proportional / integral calculation with the vibration component −T _H as an input.

(Adder 464)
The adding unit 464 corrects the movement amount ω _ref ^* T _S using the correction amount Δω _ref ^* T _S. Specifically, the movement amount Δθ is obtained by adding the correction amount Δω _ref ^* T _S to the movement amount ω _ref ^* T _S.

In the modified example 2-1B, the torque estimating unit 114 estimates the motor torque using the estimated motor magnetic flux vector (estimated magnetic flux Ψ _s ) and the axial currents i _α and i _β (the estimated torque _Te is Ask). In the phase identification unit 436b, to estimate the vibration component T _H. In the phase specifying unit 436b, the movement amount ω _ref ^* T _S is specified using the command speed ω _ref ^* , and the specified movement amount ω _ref ^* T _S is corrected using the estimated vibration component T _H. Command flux vector Ψ _s ^* using the travel amount Δθ thus obtained (specifically, the phase θ _s of the motor magnetic flux ψ _s estimated by the position estimation unit 335 and the corrected travel amount Δθ are added together) ^. The phase θ _s ^{* of} is specified.

As understood from FIGS. 16A and 16B, the operation of the phase specifying unit 436a of Modification 2-1A and the operation of the phase specifying unit 436b of Modification 2-1B are very similar. The gain multiplication unit 440 multiplies the torque vibration component T _H by the gain K ₁ , the speed deviation calculation unit 451 multiplies −1, and the multiplication unit 452 multiplies the control cycle T _S (FIG. 16A), multiplied by -1 to torque vibration component T _H at the inverting portion 462, the proportional control performed by are calculation results obtained as part of function in the PI compensator 463 (FIG. 16B), it is for the corresponding. However, that perform modification 2-1B the integrals as input vibration component -T _H Control (some functions of the PI compensator 463) is different from the modification 2-1A. Since the phase specifying unit 436b performs integration control, the phase θ _s ^* of the command magnetic flux vector Ψ _s ^* can be specified with higher accuracy than the phase specifying unit 436a. Further, instead of the PI compensation unit 463, a P compensation unit or an I compensation unit can be used.

(Modification 2-1C)
Hereinafter, the motor control device of Modification 2-1C will be described. Note that in the modification 2-1C, the same parts as those in the modification 2-1B are denoted by the same reference numerals, and description thereof may be omitted.

  As illustrated in FIG. 15C, the motor control unit 403c of Modification 2-1C has a phase specifying unit 436c instead of the phase specifying unit 436b of Modification 2-1B.

(Phase identifying unit 436c)
Phase identification unit 436c includes a command velocity omega _ref ^*, the estimated torque T _e, and a phase theta _s estimated flux [psi _s, identifies the command flux vector [psi _s ^* phase theta _s ^*. As illustrated in FIG. 16C, the phase specifying unit 436c includes a multiplying unit 460, a low-pass filter 471, a subtracting unit 472, a PI compensating unit 463, an adding unit 464, and a phase addition calculating unit 453. .

(Low-pass filter 471)
Low-pass filter 471 extracts the steady component T _L from the estimated torque T _e.

(Subtraction unit 472)
Subtraction unit 472, by subtracting the estimated torque T _e from the steady component T _L, determine the vibration components -T _H.

In Modification Example 2-1C, the phase identification unit 436c, to estimate the vibration component -T _H (multiplied by -1 to vibration component T _H of the motor torque). In the phase specifying unit 436c, the movement amount ω _ref ^* T _S is specified using the command speed ω _ref ^* , and the specified movement amount ω _ref ^* T _S is corrected using the estimated vibration component −T _H. Using the corrected movement amount Δθ (specifically, adding the phase θ _s of the motor magnetic flux Ψ _s estimated by the position estimation unit 335 and the movement amount Δθ), the phase of the command magnetic flux vector Ψ _s ^* Specify θ _s ^* . As understood from FIGS. 16B and 16C, the operation of the phase specifying unit 436b of the modification 2-1B and the operation of the phase specifying unit 436c of the modification 2-1C are the same. Therefore, according to the modified example 2-1C, the same effect as the modified example 2-1B can be obtained.

(Modification 2-1D)
Hereinafter, the motor control device of Modification 2-1D will be described. Note that in the modification 2-1D, the same parts as those in the modification 2-1C are denoted by the same reference numerals, and description thereof may be omitted.

  As illustrated in FIG. 15D, the motor control unit 403d of Modification 2-1D includes a phase specifying unit 436d instead of the phase specifying unit 436c of Modification 2-1C.

(Phase identifying unit 436d)
Phase identification unit 436d includes a command velocity omega _ref ^*, the estimated torque T _e, and a phase theta _s estimated flux [psi _s, identifies the command flux vector [psi _s ^* phase theta _s ^*. As illustrated in FIG. 16D, the phase specifying unit 436d is different from the phase specifying unit 436c in that a torque limiter 480 is provided between the low pass filter 471 and the subtracting unit 472.

(Torque limiter 480)
The torque limiter 480 specifies the limit torque T _lim from the steady component _TL . Specifically, the torque limiter 480 obtains the limit torque T _lim using the equations (29A) and (29B). Here, I _am means a limited current. See Non-Patent Document 1 for details of the limit torque T _lim .

In variation 2-1D, the limit torque T _lim is inputted to the subtraction unit 472 instead of the stationary component T _L of the variation 2-1C.

According to the modified example 2-1D, the configuration in which the motor torque does not exceed the limit torque T _lim can be easily realized.

  The present disclosure can be applied to synchronous motors such as SPMSM and IPMSM. Those synchronous motors are suitable for a heat pump refrigeration apparatus used in an air conditioner or a water heater.

DESCRIPTION OF SYMBOLS 100 Motor control apparatus 101 Inverter 102 Three-phase motor 103,203a, 203b, 203c, 303,403a, 403b, 403c, 403d, 903 Motor control part 104 Duty generation part 105a 1st current sensor 105b 2nd current sensor 106,906 u , W / α, β conversion unit 107,907 Command voltage specifying unit 108,908 Motor flux estimation unit 109,909 α, β / u, v, w conversion unit 110,910 Amplitude compensation unit 111,911 Speed / position specifying unit 112,912 Command magnetic flux specifying unit 113a, 913a α-axis magnetic flux deviation calculating unit 113b, 913b β-axis magnetic flux deviation calculating unit 114,914 Torque estimating unit 116 Base driver 117 Smoothing capacitor 118 DC power supply 119a-119f Switching element 120a-120f Free-wheeling diode 121 921 Torque deviation calculation unit 122, 922 Motor voltage estimation unit 123, 923 u, v, w / α, β conversion unit 124, 224a, 224b Modified amplitude specifying unit 125, 925 Command torque specifying unit 126, 226 Temporary setting unit 127 Addition Part (correction part)
128 Amplitude calculation unit 129 Voltage deviation calculation unit 130 Voltage deviation compensation unit 131 Constant unit 132 Switching unit 140, 240a, 240b, 240c, 915 Amplitude specifying unit 190, 336, 436a, 436b, 436c, 436d, 990 Position specifying unit 231a, 231b Error parameter calculation unit 232 Error parameter deviation calculation unit 233 Inner product compensation unit 335 Position estimation unit 337 Multiplication unit 338 Phase addition calculation unit 439 High-pass filter 440 Gain multiplication unit 451 Speed deviation calculation unit 452, 460 Multiplication unit 453 Phase addition calculation unit 462 Sign inversion unit 463 PI compensation unit 464 Addition unit 471 Low pass filter 472 Subtraction unit 480 Torque limiter

Claims (15)

A command magnetic flux vector to be followed by the motor magnetic flux vector of the three-phase motor and a first command voltage vector to be followed by the motor voltage vector of the three-phase motor are specified, and the three-phase motor is configured using a PWM inverter. A motor control device for controlling,
A motor voltage estimation unit for estimating the motor voltage vector;
A motor magnetic flux estimator for estimating the motor magnetic flux vector;
An amplitude specifying unit for specifying a command amplitude to be an amplitude of the command magnetic flux vector;
A command magnetic flux specifying unit for specifying the command magnetic flux vector using the command amplitude;
A command voltage specifying unit for specifying the first command voltage vector using the command magnetic flux vector and the estimated motor magnetic flux vector;
The DC voltage converted into three-phase AC by the PWM inverter is defined as a DC link voltage, and each phase voltage of the first command voltage vector when expressed in three-phase AC coordinates with respect to a half value of the DC link voltage Is defined as a modulation degree, and when a threshold value larger than the magnitude of the first command voltage vector when the modulation degree is 1 is defined as a voltage limit value, the amplitude specifying unit estimates Feedback control for converging a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value to zero in a first case where the magnitude of the motor voltage vector thus determined is larger than the voltage limit value A motor control device that identifies the command amplitude through control including: The amplitude specifying unit includes:
A temporary setting unit for setting a temporary amplitude of the command magnetic flux vector;
In the first case, a corrected amplitude specifying unit that specifies a corrected amplitude through control including the voltage feedback control;
The motor control device according to claim 1, further comprising: a correction unit that identifies the command amplitude by adding the correction amplitude to the temporary amplitude.   3. The motor control device according to claim 2, wherein, in the first case, the correction amplitude specifying unit specifies the correction amplitude smaller than zero. The modified amplitude specifying unit has a voltage deviation compensating unit,
In the first case, the voltage deviation compensation unit uses the corrected amplitude as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. The motor control device according to claim 2, wherein feedback control is executed. The modified amplitude specifying unit includes a switching unit that switches between the first mode and the second mode,
In the first mode, the voltage deviation compensation unit performs the voltage feedback control using the corrected amplitude as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. Is the mode to run,
The second mode is a mode in which a constant is the corrected amplitude,
The switching unit selects the first mode in the first case, and selects the second mode in the second case where the estimated magnitude of the motor voltage vector is equal to or less than the voltage limit value. The motor control device according to claim 4. The modified amplitude specifying unit includes a voltage deviation compensation unit and an inner product compensation unit,
In the first case, the voltage deviation compensator uses the first command error parameter as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. Executing the voltage feedback control,
In the first case, the inner product compensation unit is configured to (a) a first inner product of the estimated motor magnetic flux vector and a motor current vector of the three-phase motor, or (b) an estimation of the three-phase motor. First feedback with the corrected amplitude as the manipulated variable so that a deviation between an error parameter, which is a second inner product of the magnet magnetic flux vector and the motor current vector, and the first command error parameter converge to zero. The motor control device according to claim 2 or 3, wherein control is executed. The modified amplitude specifying unit includes a switching unit that switches between the first mode and the second mode,
In the first mode, the voltage feedback control is performed using the first command error parameter as an operation amount so that a deviation between the estimated magnitude of the motor voltage vector and the voltage limit value converges to zero. A voltage deviation compensation unit executes the first feedback control using the corrected amplitude as an operation amount so that a deviation between the first command error parameter and the error parameter converges to zero. Is the mode executed by
In the second mode, the inner product compensation unit performs second feedback control using the corrected amplitude as an operation amount so that a deviation between a constant second command error parameter and the error parameter converges to zero. The mode to run,
The switching unit selects the first mode in the first case, and selects the second mode in the second case where the estimated magnitude of the motor voltage vector is equal to or less than the voltage limit value. The motor control device according to claim 6. The error parameter is the second inner product;
The motor control device according to claim 7, wherein the constant is zero.   When the theoretical value of the modulation factor when the operation region of the PWM inverter transitions from a linear region to an overmodulation region is defined as a boundary value, the voltage limit value is the first value when the modulation factor is the boundary value. The motor control device according to claim 1, wherein the motor control device is larger than the magnitude of one command voltage vector. Each phase voltage of the motor voltage vector that the PWM inverter outputs to the three-phase motor has an upper limit that can be output and a lower limit that is smaller than the upper limit by the DC link voltage,
The motor control device specifies a reference voltage vector used to generate a PWM duty that defines the operation of the PWM inverter,
The motor voltage estimator replaces the voltage of the phase whose voltage is larger than the upper limit with the upper limit in the reference voltage vector expressed by the three-phase AC coordinates, and the voltage of the phase whose voltage is smaller than the lower limit. The motor control device according to claim 1, wherein a value replaced with the lower limit is estimated as the motor voltage vector. The motor control device includes an amplitude compensator that specifies a reference voltage vector used to generate a PWM duty that defines an operation of the PWM inverter from the first command voltage vector,
When the theoretical value of the modulation factor when the operation region of the PWM inverter transitions from a linear region to an overmodulation region is defined as a boundary value, the reference voltage vector is when the modulation factor is equal to or less than the boundary value , A second command voltage vector that is the same as the first command voltage vector,
The reference voltage vector is an amplitude compensation coefficient larger than 1 when the degree of modulation is greater than the boundary value, and an amplitude compensation coefficient that increases as the degree of modulation increases is the first command voltage vector. The motor control device according to claim 1, wherein the motor control device is a second command voltage vector obtained by multiplying by.   The amplitude compensation coefficient is the product of the gain of the PWM inverter and the amplitude of the fundamental wave component of each phase voltage of the second command voltage vector, and the amplitude of the fundamental wave component of each phase voltage of the first command voltage vector The motor control device according to claim 11, wherein the coefficient is a coefficient determined so as to match. The motor magnetic flux estimating unit estimates the motor magnetic flux vector using a motor current vector of the three-phase motor and the estimated motor voltage vector,
The motor control device
A torque estimator for estimating a motor torque of the three-phase motor from the motor current vector and the estimated motor magnetic flux vector;
Using the torque deviation between the command torque to be followed by the motor torque and the estimated motor torque, the amount of movement for each control cycle in which the phase of the motor magnetic flux vector should move is specified, and the specified movement The position control part which specifies the phase of the said command magnetic flux vector using the quantity and the phase of the estimated said motor magnetic flux vector, The motor control apparatus as described in any one of Claims 1-12. The motor control device specifies a movement amount for each control cycle in which the phase of the motor magnetic flux vector should move using a command speed that the speed of the three-phase motor should follow, and uses the specified movement amount A position specifying unit for specifying the phase of the command magnetic flux vector;
The command magnetic flux specifying unit specifies the command magnetic flux vector using the phase specified by the position specifying unit and the command amplitude specified by the amplitude specifying unit. The motor control device according to claim 1. A command magnetic flux vector to be followed by a generator magnetic flux vector of a three-phase generator and a first command voltage vector to be followed by a generator voltage vector of the three-phase generator are specified, and the PWM converter is used to specify the command magnetic flux vector A generator control device for controlling a three-phase generator,
A generator voltage estimator for estimating the generator voltage vector;
A generator magnetic flux estimator for estimating the generator magnetic flux vector;
An amplitude specifying unit for specifying a command amplitude to be an amplitude of the command magnetic flux vector;
A command magnetic flux specifying unit for specifying the command magnetic flux vector using the command amplitude;
A command voltage specifying unit for specifying the first command voltage vector using the command magnetic flux vector and the estimated generator magnetic flux vector;
The DC voltage converted into three-phase AC by the PWM converter is defined as a DC link voltage, and each phase voltage of the first command voltage vector when expressed in three-phase AC coordinates with respect to a half value of the DC link voltage Is defined as a modulation degree, and when a threshold value larger than the magnitude of the first command voltage vector when the modulation degree is 1 is defined as a voltage limit value, the amplitude specifying unit estimates In the first case where the magnitude of the generated generator voltage vector is larger than the voltage limit value, the voltage that converges the deviation between the estimated generator voltage vector magnitude and the voltage limit value to zero A generator control device that identifies the command amplitude through control including feedback control.

Images