JP2015201860A - Radio transmitter, radio receiver, radio communication system, elevator control system, and transformation facility control system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a technique that improves the reliability of digital radio communication getting interference of multiple waves under an environment in which there are a plurality of electromagnetic wave scattering matters.SOLUTION: A first transmission wave having a first carrier frequency f0+Δf obtained by a modulator 13-1 having modulated a carrier wave with an information signal having a frequency band f1 is transmitted from a transmission antenna 20-1. A second transmission wave having a second carrier frequency f0-Δf obtained by a modulator 13-2 having modulated the carrier wave with the information signal is transmitted from a transmission antenna 20-2. By controlling frequency difference Δf keeping the average frequency f0 of the first carrier frequency f0+Δf and the second carrier frequency f0-Δf constant, the first carrier frequency f0+Δf and the second carrier frequency f0-Δf are made variable.

Description

  The present invention relates to a long-lived wireless transmitter, wireless receiver, and wireless communication system that realize highly reliable wireless communication. In particular, when the installation environment of wireless transmitters and wireless receivers has obstacles that reflect and scatter radio waves, wireless transmission that can suppress sensitivity degradation due to interference of multiple waves generated by the obstacles The present invention relates to a machine, a radio receiver, a radio communication system, an elevator control system, and a substation control system.

  In recent years, wireless communication technology has made remarkable progress in the broadcasting and communication fields, and has overcome problems related to reliability, such as a disconnection peculiar to wireless communication. As a result, wireless communication technology is increasingly applied to the control field and the measurement field, which require higher reliability than the broadcasting field and the communication field.

  In particular, in the control field and the measurement field, equipment that constructs social infrastructure (hereinafter referred to as “social infrastructure equipment”) has higher communication quality and communication equipment than general consumer equipment in the broadcasting and communication fields. High reliability, that is, a long life is particularly required. The social infrastructure equipment is, for example, an elevator system shown in FIG. 12, a substation monitoring system shown in FIG.

  Social infrastructure equipment is overwhelmingly larger in size than general consumer equipment and is made of a solid metal member. This social infrastructure device itself becomes an electromagnetic wave scattering source. Therefore, wireless communication in social infrastructure equipment is often performed in an environment where multiple waves (multipath) generated by scattering interfere with each other. For this reason, it is desired to realize highly reliable wireless communication in an environment where interference due to multiple waves (multipath) occurs.

  When the difference in the distance from the transmission point to the reception point is an odd multiple of a half wavelength, the electromagnetic wave energy is canceled out by the interference of the multiple electromagnetic waves and becomes zero, and communication becomes impossible. Conventionally, this problem has been addressed by a spatial diversity technique in which a plurality of antennas are spatially separated by half a wavelength. Spatial diversity technology is based on the fact that even if the electromagnetic wave energy received by one antenna becomes zero due to interference, the electromagnetic wave energy received by other antennas located half a wavelength apart is strengthened by interference. It is a technology that enables reception with an antenna.

  In a wireless communication environment of a social infrastructure system, reflection is caused by the distribution of furniture that is an electromagnetic wave scatterer. When the average distance of this radio wave reflection is approximately the same as the antenna distance (half-wave distance of the electromagnetic wave) for realizing space diversity, the energy of the electromagnetic wave reaching the antenna due to interference due to another multiple reflection is zero. The possibility of becoming greater. Therefore, it becomes difficult to ensure the reliability of wireless communication.

  In social infrastructure equipment, electromagnetic waves generated by a wireless transmitter are reflected by the social infrastructure equipment itself to become multiple waves (multipath) and may arrive at the receiver from all directions. Therefore, when the space diversity technique is applied, many antennas are required. For example, it is necessary to prepare a plurality of arrayed antennas even if it is limited when multiple waves (multipath) come in the plane direction. Since the distance between adjacent antennas is a half wavelength of the received electromagnetic wave, there is a possibility that it exceeds the size that can be installed in this social infrastructure device.

  The summary of Patent Document 1 (Japanese Patent Laid-Open No. 10-135919) and FIG. 3 disclose a technique for rotating the polarization plane of a radio wave in order to suppress the effects of fading and noise in wireless communication. Further, paragraph 0006 of the specification of Patent Document 1 states, “On the transmitting side, two pairs of dipole antennas that are crossed at right angles and rotated at right angles to the transmission direction for rotating and transmitting the polarization plane of the radio wave, and this Comprising a transmitter having two sets of balanced modulated wave outputs for exciting the signal and comprising a receiver for detecting and receiving the rotation of the polarization plane of the incoming radio wave on the receiver side. .

  In Patent Document 2 (Japanese Patent Application Laid-Open No. 61-024339), a carrier having two different first frequencies is used without using the third frequency, and different information is used using the second frequency. And a method of transmitting these two carriers using different polarized waves and detecting a difference frequency from these two carriers in the receiver to obtain a third frequency.

Japanese Patent Laid-Open No. 10-135919 Japanese Patent Laid-Open No. 61-024339

  The invention of Patent Document 1 is effective in removing the effects of fading and noise that occur during transmission and reception of radio waves. However, the present invention describes nothing about realizing highly reliable wireless communication in an environment where interference due to multiple waves (multipath) occurs, and reducing the size of the transmitting antenna and the receiving antenna. Absent.

  The invention of Patent Document 2 does not require a pilot signal wave, and can be applied to an analog modulation method such as FM (Frequency Modulation). The configuration is simple and compensation for each communication signal wave is easy. However, this invention does not describe anything about digital processing of information signals after detecting a predetermined frequency component in the receiver.

  Accordingly, the present invention provides a wireless transmitter, a wireless receiver, a wireless communication system, and an elevator control system capable of improving the reliability of digital wireless communication due to multiple wave interference in an environment where a plurality of electromagnetic wave scatterers are present. It is another object of the present invention to provide a substation equipment control system.

In order to solve the above problems and achieve the object of the present invention, the present invention is configured as follows.
That is, the wireless transmitter of the present invention includes a first transmission wave having a first carrier frequency modulated by an information signal having a predetermined frequency band, and a second transmission wave modulated by the information signal. A wireless transmitter for transmitting a second transmission wave having a carrier frequency, wherein an average frequency of the first carrier frequency and the second carrier frequency is constant, and the first carrier frequency and the second carrier frequency The carrier frequency is variable.
Other means will be described in the embodiment for carrying out the invention.

  According to the present invention, a wireless transmitter, a wireless receiver, a wireless communication system, and an elevator control system capable of improving the reliability of digital wireless communication due to interference of multiple waves in an environment where a plurality of electromagnetic wave scatterers are present. And a substation equipment control system can be provided.

It is a figure which shows the structure of the radio | wireless communications system in 1st Embodiment. It is a figure which shows the structure of the radio | wireless communications system in 2nd Embodiment. It is a figure which shows operation | movement of the radio | wireless communications system in 2nd Embodiment. It is a figure which shows the channel example (the 1) of the radio | wireless communications system in 2nd Embodiment. It is a figure which shows the channel example (the 2) of the radio | wireless communications system in 2nd Embodiment. It is a figure which shows the structure of the radio | wireless receiver in 3rd Embodiment. It is a figure which shows the structure of the radio | wireless receiver in 4th Embodiment. It is a figure which shows the structure of the radio | wireless receiver in 5th Embodiment. It is a figure which shows the example of mounting of the radio | wireless receiver in 5th Embodiment. It is a figure which shows the structure of the radio | wireless communications system in 6th Embodiment. It is a figure which shows the structure of the radio | wireless communications system in 7th Embodiment. It is a figure which shows the structure of the elevator system in 8th Embodiment. It is a figure which shows the structure of the substation equipment monitoring system in 9th Embodiment.

  Hereinafter, a mode for carrying out the present invention (referred to as “the present embodiment”) will be described in detail with reference to the drawings.

(Configuration of the first embodiment)
FIGS. 1A to 1D are diagrams showing a configuration of a wireless communication system in the first embodiment. FIG. 1A shows a wireless transmitter 10 of the present embodiment. FIG. 1B shows the wireless receiver 30 of the present embodiment. FIG.1 (c) has shown the power spectrum of the transmission signal of the wireless transmitter 10 of this embodiment. FIG. 1D shows the power spectrum of the output signal of the low pass filter 35.
The wireless communication system according to this embodiment includes a wireless transmitter 10 and a wireless receiver 30.

  A radio transmitter 10 shown in FIG. 1A includes a transmission-side control unit 14, variable oscillators 12-1 and 12-2, modulators 13-1 and 13-2, an information generation circuit 11, and a baseband. A circuit 17 and transmission antennas 20-1 and 20-2 are included.

The output side of the transmission side control unit 14 is connected to the variable oscillators 12-1 and 12-2. The output side of the variable oscillator 12-1 is connected to the modulator 13-1. The output side of the variable oscillator 12-2 is connected to the modulator 13-2. The output side of the information generation circuit 11 is connected to the baseband circuit 17. The output side of the baseband circuit 17 is connected to the modulators 13-1 and 13-2, respectively. The output of the modulator 13-1 is connected to the transmission antenna 20-1. The output of the modulator 13-2 is connected to the transmission antenna 20-2.
The information generation circuit 11 generates an information signal. The baseband circuit 17 converts the input information signal into an information signal having a frequency band f1 that is a predetermined frequency.

The transmission side control unit 14 controls the signal frequency output from the variable oscillators 12-1 and 12-2. The variable oscillator 12-1 outputs a first carrier wave having a frequency (f0 + Δf) obtained by adding a frequency difference Δf corresponding to the output signal of the transmission side control unit 14 from the frequency f0.
The variable oscillator 12-2 outputs a second carrier wave having a frequency (f0−Δf) obtained by subtracting the frequency difference Δf corresponding to the output signal of the transmission side control unit 14 from the frequency f0.

  The modulator 13-1 modulates the information signal based on the input oscillation signal that is the first carrier wave. The modulator 13-2 modulates the information signal based on the input oscillation signal that is the second carrier wave. In the present embodiment, the frequency difference Δf is smaller than the frequency f0. Furthermore, the frequency f1, which is a predetermined frequency, is smaller than the frequency difference Δf. That is, f0> Δf> f1.

FIG. 1B shows the wireless receiver 30 of the present embodiment.
The radio receiver 30 illustrated in FIG. 1B includes a reception antenna 31, a mixer 32, an oscillator 33, a low-pass filter 35, an analog / digital converter 36, and a baseband circuit 51.

  The reception antenna 31 receives radio waves transmitted from the transmission antennas 20-1 and 20-2. The oscillator 33 outputs an oscillation signal having a frequency f0. The mixer 32 mixes two input signals and outputs them. Specifically, the oscillation signal having the frequency f0 and the signal received by the receiving antenna 31 are mixed. Thereby, product detection is performed. The low-pass filter 35 suppresses a spectral component exceeding a predetermined frequency in the input signal and transmits a low-frequency spectral component. The analog / digital converter 36 converts the input signal into a digital signal. The baseband circuit 51 converts the input digital signal into a baseband signal that is an original information signal.

  The output side of the receiving antenna 31 and the output side of the oscillator 33 are connected to the mixer 32. The output side of the mixer 32 is connected to an analog / digital converter 36 via a low-pass filter 35. The output side of the analog / digital converter 36 is connected to the baseband circuit 51.

  FIG. 1C shows the power spectrum of the wireless transmitter 10 of this embodiment. The horizontal axis of the figure indicates the frequency. The vertical axis in the figure indicates the spectral density corresponding to the frequency. In the drawings, the frequency may be abbreviated as freq., And the spectral density may be described as spectrum.

The power spectrum shown in FIG. 1C relates to electromagnetic waves radiated from the transmission antennas 20-1 and 20-2. The variable oscillator 12-1 outputs a signal having a frequency (f0 + Δf) around the frequency f0. The variable oscillator 12-2 outputs a signal having a frequency (f0−Δf) around the frequency f0.
The frequency difference Δb is a peak power spectrum difference between the electromagnetic waves radiated from the transmitting antennas 20-1 and 20-2 when the frequency Δf is the largest.
The frequency difference Δa is the difference between the power spectrum peaks of the electromagnetic waves radiated from the transmitting antennas 20-1 and 20-2 when the frequency Δf is the smallest.

FIG. 1D shows the power spectrum of the output signal of the low-pass filter 35 of the present embodiment. The horizontal axis of the figure indicates the frequency. The vertical axis in the figure indicates the spectral density corresponding to the frequency.
In this embodiment, the power spectrum of the signal output from the receiving antenna 31 to the mixer 32 is product-detected by the mixer 32 to a frequency corresponding to the peak frequency difference. Therefore, the power spectrum of the output signal of the low-pass filter 35 has a peak at the frequency Δb / 2 to the frequency Δa / 2.

(Operation of the first embodiment)
The operation of the wireless transmitter 10 of this embodiment will be described based on FIG.
The information generation circuit 11 generates an information signal and outputs it to the baseband circuit 17. The baseband circuit 17 converts the input information signal into an information signal having a frequency band f1, which is a predetermined frequency, and outputs the information signal to the modulators 13-1 and 13-2. The modulator 13-1 modulates the information signal having the frequency band f1 with the oscillation signal output from the variable oscillator 12-1, and transmits the modulated information signal from the transmission antenna 20-1. The modulator 13-2 modulates the information signal having the frequency band f1 with the oscillation signal output from the variable oscillator 12-2, and transmits the modulated information signal from the transmission antenna 20-2.

  The electromagnetic waves radiated from the transmission antennas 20-1 and 20-2 are reflected at different incident angles and different frequencies when passing through a space where a large number of non-specific reflectors exist. The electromagnetic wave has a polarization, and the phase shift angle of this polarization changes depending on the polarization vector for different incident angles. For example, if the polarization vector is orthogonal to the incident surface, the phase shift angle is 180 °. If the polarization vector is included with respect to the incident surface, the phase shift angle is 0 °. The electromagnetic waves of different frequencies radiated from the transmission antennas 20-1 and 20-2 are reflected various times by various incident paths and various incident angles by a plurality of reflectors. At this time, when the frequency corresponding to the wavelength equivalent to the average distribution distance of the reflector is about the same as the difference between the two transmission frequencies, a beat wave is formed on the time axis by this frequency difference. Are synthesized with different polarization vector directions and different phases. On average, the polarization vector rotates once per reflection.

  Therefore, when a specific multiple wave is synthesized at the reception point, the phase of the plurality of reflected waves arriving at the reception point changes between 0 ° and 180 ° in the rotation period due to the rotation of the polarization vector. Therefore, when the rotation period is divided on the time axis and the power of the received wave at each point after the division is observed, the set of each point after the division is combined with the reflected wave in reverse phase at the reception point. And a time point where reflected waves are combined in phase at the reception point and reception power is strengthened. By extracting the time points where reflected waves are combined in the same phase at the receiving point and the received power is strengthened by digital signal processing technology etc., it is easy to secure a wireless communication path even in a radio wave environment where there are many reflectors It becomes.

The operation of the wireless receiver 30 of this embodiment will be described based on FIG.
The linearly polarized waves transmitted by the transmission antennas 20-1 and 20-2 are received by the reception antenna 31. The received signal includes a signal having a frequency (f0 ± Δf). This received signal and the oscillation signal having the frequency f 0 output from the oscillator 33 are mixed by the mixer 32. Thereby, product detection is performed, and a signal having a frequency Δf corresponding to the frequency difference between the two is extracted. From the output signal of the mixer 32, a signal (noise) having a predetermined frequency or higher is suppressed by the low-pass filter 35, and a signal spectrum of the frequency Δf is extracted. The output signal of the low-pass filter 35 is converted into a digital signal via the analog / digital converter 36. This digital signal is converted into the original information signal by the baseband circuit 51.

  According to the wireless receiver 30 of the present embodiment, a radio wave emitted from the wireless transmitter 10 and subjected to multiple reflections by a plurality of reflectors and reaching the receiving antenna 31 is transmitted by the wireless receiver 30 into two transmission waves. Is converted into a signal having a frequency Δf corresponding to the difference between the carrier frequency (f0 ± Δf) of the current and the frequency wind f0 output from the oscillator 33. The maximum value of the frequency difference between the carrier frequencies (f0 ± Δf) of the two transmission waves is the frequency Δb, and the minimum value is the frequency Δa. As a result, in the frequency range (Δa / 2 to Δb / 2) that is significantly lower than the carrier frequency f0 of the transmission wave, it is possible to easily extract the time point at which the reflected wave is synthesized in phase at the reception point. It is easy to secure a wireless communication path of the communication system.

(Effects of the first embodiment)
The first embodiment described above has the following effects (A) and (B).
(A) The wireless transmitter 10 of the present embodiment transmits two electromagnetic waves having different frequencies from the transmission antennas 20-1 and 20-2. As a result, when the rotation cycle is divided on the time axis and the power of the received wave at each point after the division is observed, the set of each point after the division is combined with the reflected wave in reverse phase at the reception point. And a time point where reflected waves are combined in phase at the reception point and reception power is strengthened. Securing a wireless communication path even in a radio wave environment where there are many reflectors by extracting time points where reflected waves are combined in phase at this receiving point by digital signal processing technology and the received power is strengthened Becomes easy.

(B) The radio receiver 30 of the present embodiment converts the signal into a signal having a frequency Δf corresponding to the difference between the carrier frequencies of the two transmission waves. As a result, at the frequency Δa / 2 to the frequency Δb / 2 which is significantly lower than the carrier frequency f0 of the transmission wave, it is possible to easily extract the time point at which the reflected wave is synthesized in phase at the reception point. It is easy to secure a wireless communication path.

(Configuration of Second Embodiment)
FIGS. 2A to 2D are diagrams showing the configuration of a wireless communication system in the second embodiment. The same reference numerals are assigned to the same elements as those in the wireless communication system according to the first embodiment shown in FIGS.
The radio transmitter 10 shown in FIG. 2A has the same configuration as the radio transmitter 10 of the first embodiment shown in FIG.

  The radio receiver 30A shown in FIG. 2B has a variable bandpass filter 35A different from the lowpass filter 35 of the radio receiver 30 of the first embodiment shown in FIG. Other than having the part 34, it has the structure similar to the radio | wireless receiver 30 of 1st Embodiment shown in FIG.1 (b).

  The output side of the receiving side control unit 34 of this embodiment is connected to the variable bandpass filter 35A. The reception-side control unit 34 changes the pass frequency band of the variable bandpass filter 35A at the same time as the transmission wave frequency variable period used by the wireless transmitter 10.

  FIG. 2C shows the power spectrum of the wireless transmitter 10 of the present embodiment. The power spectrum of the wireless transmitter 10 of the present embodiment is the same as the power spectrum of the wireless transmitter 10 of the first embodiment shown in FIG.

  FIG. 2D shows the power spectrum of the output signal of the variable bandpass filter 35A of the present embodiment. The power spectrum of the output signal of the variable bandpass filter 35A of this embodiment is the same as the power spectrum of the output signal of the lowpass filter 35 of the first embodiment shown in FIG.

(Operation of Second Embodiment)
FIGS. 3A to 3H are diagrams illustrating the operation of the wireless communication system according to the second embodiment.
FIG. 3A shows a predetermined sequence in which the wireless transmitter 10 repeats the training mode and the communication mode. FIG. 3B shows a predetermined sequence in which the wireless receiver 30A repeats the training mode and the communication mode. The horizontal axes in FIGS. 3A and 3B indicate a common time t.

  When the wireless transmitter 10 starts communication, the wireless transmitter 10 operates in a training mode in which an optimal frequency is trained for a predetermined time, and then transitions to the communication mode to perform communication at an optimal frequency. Similarly, the radio receiver 30A operates in a training mode in which an optimal frequency is trained for a predetermined time, and then transitions to a communication mode to perform communication at an optimal frequency. Here, the optimum frequency is a frequency at which the interference of multiple waves is the smallest and therefore the component of the frequency f0 by the variable bandpass filter 35A is the highest.

  FIG. 3C shows a detailed sequence of the training mode of the wireless transmitter 10. FIG. 3D shows a detailed sequence of the training mode of the radio receiver 30A. The horizontal axes of FIGS. 3C and 3D indicate a common time t.

  The transmission-side control unit 14 of the wireless transmitter 10 is a variable oscillator at frequencies (f0 ± Δf1), (f0 ± Δf2), (f0 ± Δf3), and (f0 ± Δf4) every time T in the training mode. 12-1 and 12-2 are switched so as to oscillate.

  Similarly, in the training mode, the radio receiver 30A switches so that the variable bandpass filter 35A performs a filter operation that passes signals of frequencies Δf1, Δf2, Δf3, and Δf4 every time 4T. That is, the reception frequency of the radio receiver 30A is switched. By switching between these modes, training is performed so that the combination of the oscillation frequency of the wireless transmitter 10 and the reception frequency of the wireless receiver 30A is optimized.

FIG. 3E shows a detailed sequence of the communication mode of the wireless transmitter 10. FIG. 3F shows a detailed sequence of the communication mode of the radio receiver 30A. The horizontal axes in FIGS. 3E and 3F indicate the common time t.
In the communication mode, the wireless transmitter 10 and the wireless receiver 30A transmit and receive information with the optimal oscillation frequency Δfi and the optimal reception frequency Δfi.

  The vertical axis in FIG. 3G shows the power spectrum of the transmission signal of the wireless transmitter 10. The vertical axis in FIG. 3H indicates the power spectrum of the output signal of the variable bandpass filter 35A of the radio receiver 30A. The horizontal axes of FIGS. 3 (g) and 3 (h) indicate the frequency.

  FIG. 3G shows that the wireless transmitter 10 outputs a transmission signal having a peak at the frequency (f0 + Δfi) and a transmission signal having a peak at the frequency (f0−Δfi).

  FIG. 3H shows that the variable bandpass filter 35A of the radio receiver 30A outputs a signal having a peak at a frequency Δfi that is a difference between the peaks of two transmission signals by product detection and bandpass filter processing. Show.

  FIG. 4 is a diagram illustrating a channel example (part 1) of the wireless communication system according to the second embodiment. The horizontal axis indicates the frequency, and the rectangle indicates the respective wireless communication channels Ch-1 to Ch-n (n is a natural number).

  The light gray portion indicates a combination of the carrier frequency (f0−Δf1) of the first transmission wave and the carrier frequency (f0 + Δf1) of the second transmission wave. The frequency f0 indicates the average frequency of the combination of the carrier frequency (f0−Δf1) of the first transmission wave and the carrier frequency (f0 + Δf1) of the second transmission wave.

  The dark gray portion indicates a combination of the carrier frequency (f0−Δf2) of the first transmission wave and the carrier frequency (f0 + Δf2) of the second transmission wave. Similarly to the above, the frequency f0 indicates the average frequency of the combination of the carrier frequency (f0−Δf2) of the first transmission wave and the carrier frequency (f0 + Δf2) of the second transmission wave.

  The radio communication system according to the present embodiment divides a frequency band to be used into radio communication channels Ch-1 to Ch-n that are a plurality of narrow frequency bands. In the wireless communication channels Ch-1 to Ch-n, signal modulation is performed in the same manner as in the first embodiment. One center frequency (average frequency) f0 is set inside the frequency band to be used, two channels having the same frequency interval are selected on the left and right on the frequency axis of this frequency f0, and modulation is performed by the same signal. Then, it is emitted into the air by the transmitting antennas 20-1 and 20-2. The present embodiment can be applied to an existing wireless communication system that performs frequency division multiplexing in accordance with the Radio Law.

FIG. 5 is a diagram illustrating a channel example (part 2) of the wireless communication system according to the second embodiment.
The difference from FIG. 4 is that different center frequencies f0a and f0b are set in the frequency band. The wireless communication system according to the present embodiment further uses a communication channel having a frequency (f0a + Δf1) and a communication channel having a frequency (f0a−Δf1) located at equal intervals on the frequency axis from the center frequency f0a. At the same time, a communication channel having a frequency (f0b + Δf1) and a communication channel having a frequency (f0b−Δf1) located at equal intervals on the frequency axis from the center frequency f0b are used. In this way, a plurality of wireless channels can be realized simultaneously by selecting two communication channels located at equal intervals on the frequency axis from a plurality of center frequencies f0a and f0b without using the same channel at the same time. It is possible to increase the information communication capacity and improve the reliability of the wireless communication line.

(Effect of 2nd Embodiment)
The second embodiment described above has the following effects (C) and (D).
(C) The radio communication system of the present embodiment can be applied to an existing radio communication system that performs frequency division multiplexing in accordance with the Radio Law.
(D) By simultaneously selecting two communication channels located at equal intervals on the frequency axis from a plurality of center frequencies f0a and f0b without simultaneously using the same channel, a plurality of wireless lines can be realized simultaneously. Thus, there is an effect in increasing the information communication capacity and improving the reliability of the wireless communication line.

(Configuration of Third Embodiment)
FIGS. 6A and 6B are diagrams illustrating the configuration of the wireless receiver according to the third embodiment. The same components as those of the wireless receiver 30 of the first embodiment shown in FIG.

  Unlike the wireless receiver 30 of the first embodiment, the wireless receiver 30B of this embodiment shown in FIG. 6A is connected to the baseband circuit 51 from the receiving antenna 31 via the delta-sigma modulator 40. Other than that, the configuration is the same as that of the wireless receiver 30 of the first embodiment.

The delta-sigma modulator 40 provided in the wireless receiver 30B of this embodiment includes resonators 42-1, 42-2, an analog / digital converter 43, a digital / analog converter 45, and an oscillator 44. And a reverse phase synthesizer 41-1. The resonator 42-1 which is the first resonator resonates at a resonance frequency corresponding to the carrier frequency (f0−Δf) of the transmission wave. The resonator 42-2 which is the second resonator resonates at a resonance frequency corresponding to the carrier frequency (f0 + Δf) of the transmission wave.
The analog / digital converter 43 compares, for example, a predetermined threshold value with an input signal and converts it to 1-bit digital. The oscillator 44 outputs an oscillation signal having a frequency fs. The digital / analog converter 45 converts, for example, a predetermined analog value according to a 1-bit digital signal.

  FIG. 6B is a diagram showing a power spectrum of the delta-sigma modulator 40 of the wireless receiver 30B of the present embodiment. The horizontal axis indicates the frequency, and the vertical axis indicates the spectrum density of the output signal of the delta-sigma modulator 40.

  Due to the alias signal peculiar to the digital signal, the power spectrum of the output signal of the bandpass type delta sigma modulator 40 becomes 0 for every integer multiple of the sampling frequency fs. In the frequency region below the sampling frequency fs, there is a zeroth-order harmonic peak. There is a peak of the first harmonic in the frequency region of the sampling frequency fs to 2fs. Similarly, the peak of the nth-order harmonic exists in the frequency region of the sampling frequency (n × fs) to ((n + 1) × fs) (n is a natural number).

(Operation of Third Embodiment)
The operation of the wireless receiver 30B will be described based on FIG.
The digital / analog converter 45 outputs a feedback signal to the antiphase synthesizer 41-1. The feedback signal is subtracted from the reception signal output from the reception antenna 31 by the antiphase synthesizer 41-1. The output signal of the negative phase synthesizer 41-1 resonates at the respective resonance frequencies via the resonators 42-1 and 42-2 connected in parallel. Thereby, noise due to a frequency higher than the frequency f0 can be removed.

The output signals of the resonators 42-1 and 42-2 are input to the analog / digital converter 43 and converted into digital signals. The digital signal is output to the baseband circuit 51 and also to the digital / analog converter 45, and is converted into the feedback signal described above. The analog / digital converter 43 and the digital / analog converter 45 are sampled by the common oscillator 44 at the same sampling frequency fs. The sampling frequency fs is an integer multiple of the carrier frequency (f0−Δf) of the two transmission waves described above and the average frequency f0 of the carrier frequency (f0 + Δf), and satisfies the following Expression 1.
f0 = M × fs (Formula 1).
The delta sigma modulator 40 outputs a digital signal in which the time density of the bit “1” is increased according to the differential value (change amount) of the input signal.

  As shown in FIG. 6B, in this embodiment, since the center frequency f0 of the carrier wave of the received signal is M times the sampling frequency fs, the carrier frequency (f0−Δf) is obtained from the delta sigma modulator 40. The output component of the transmitted wave signal and the transmitted wave signal having the carrier frequency (f0 + Δf) is output as a digital signal.

(Effect of the third embodiment)
The third embodiment described above has the following effect (E).
(E) According to the delta-sigma modulator 40 of the present embodiment, a modulated signal wave whose center frequency is a frequency sufficiently lower than the transmission wave is digitally used without using the mixer 32 and the oscillator 33 which are analog nonlinear circuits. It can be taken out as a signal. As a result, it is possible to easily extract the time point at which the reflected wave is synthesized in phase at the reception point by the digital signal processing performed by the baseband circuit 51 at the subsequent stage, and the radio receiver 30B can be simplified and highly reliable. Can be realized.

(Configuration of Fourth Embodiment)
FIGS. 7A and 7B are diagrams illustrating the configuration of a wireless receiver according to the fourth embodiment. The same elements as those of the wireless receiver according to the third embodiment shown in FIG.

  The wireless receiver 30C according to the present embodiment has a delta-sigma modulator 40C different from the delta-sigma modulator 40 included in the wireless receiver 30B according to the third embodiment. The wireless receiver 30B has the same configuration.

  The delta sigma modulator 40C of this embodiment further includes a digital signal interpolator 46 in addition to the delta sigma modulator 40 of the third embodiment, and the oscillator 44C includes a digital / analog converter 45, The configuration is the same as that of the delta-sigma modulator 40B of the third embodiment except that an oscillation signal having a sampling frequency (M × fs) is output.

  The digital signal interpolator 46 receives a 1-bit signal having a predetermined period and outputs a signal at a designated sampling period. For example, the digital signal interpolator 46 outputs the input digital signal as it is at the timing that coincides with the predetermined cycle, and interpolates and outputs “0” at the timing that does not coincide with the predetermined cycle.

  The delta-sigma modulator 40C of this embodiment inputs the output signal of the analog / digital converter 43 to the digital / analog converter 45 via the digital signal interpolator 46.

  FIG. 7B is a diagram illustrating the operation of the wireless receiver 30C according to the present embodiment. The horizontal axis indicates the frequency, and the vertical axis indicates the power spectrum. The solid line indicates the power spectrum of the output signal of the delta-sigma modulator 40C, and for comparison, the power spectrum of the output signal of the delta-sigma modulator 40 shown in FIG. .

  Due to the alias signal specific to the digital signal, the power spectrum of the output signal of the delta-sigma modulator 40C becomes 0 for every integer multiple of the sampling frequency (M × fs) of the digital-analog converter 45. In the frequency region below the sampling frequency (M × fs), there is a peak of the 0th harmonic.

(Operation of Fourth Embodiment)
The radio receiver 30C of the fourth embodiment shown in FIG. 7 is different from the radio receiver 30B of the third embodiment shown in FIG. 6 in that the digital-analog converter 45 used by the delta-sigma modulator 40C is different. The sampling frequency fs is an integral multiple of the sampling frequency (M × fs) of the analog / digital converter 43, and when the digital output of the analog / digital converter 43 is returned to the feedback loop, the digital signal interpolator 46 performs digital processing. The frequency of the signal is an integral multiple.

  The digital-analog converter 45 exhibits a low-pass attenuation characteristic of the SINC function due to the zero-order hold effect. As the difference between the average frequency of the carrier frequencies of the two transmission waves and the sampling frequency of the clock generation circuit increases, the gain of the hood back loop with respect to the transmission frequency decreases.

  As shown in FIG. 7B, in this embodiment, since the center frequency f0 of the carrier wave of the received signal is M times the sampling frequency fs, the signal of the transmission wave having the frequency Δf is output from the delta-sigma modulator 40C. The output component of the transmission wave signal having the carrier frequency (f0−Δf) and the transmission wave signal having the carrier frequency (f0 + Δf) are output as digital signals.

  Due to the zero-order hold effect, the signal having the frequency Δf has the highest spectral density. Therefore, the information signal having the frequency f1 carried by the signal having the frequency Δf can be easily extracted.

(Effect of the fourth embodiment)
The fourth embodiment described above has the following effect (F).
(F) According to the present embodiment, the frequency of the digital signal input to the digital-to-analog converter 45 of the hood back loop can be raised in advance at a low frequency, so that the transmission wave due to the zero-order hold effect can be prevented. A decrease in the gain of the food bag can be suppressed. Thereby, compared with the delta-sigma modulator 40 of 3rd Embodiment shown in FIG. 6, the sampling frequency fs can be made low and the cost reduction of hardware and the reduction in power consumption are attained.

(Configuration of Fifth Embodiment)
FIG. 8 is a diagram illustrating a configuration of a wireless receiver according to the fifth embodiment. The same code | symbol is provided to the structure same as the radio | wireless receiver of 4th Embodiment shown in FIG.

  As shown in FIG. 8A, the radio receiver 30D of this embodiment has a delta-sigma modulator 40D different from the radio receiver 30C of the fourth embodiment shown in FIG.

In addition to the delta sigma modulator 40C of the fourth embodiment, the delta sigma modulator 40D of the present embodiment includes forward amplifiers 48-1, 48-2, 48-3, backward amplifiers 47-1, 47-2 and 47-3, and reverse phase synthesizers 41-2 and 41-3.
The output side of the receiving antenna 31 is connected to the forward amplifiers 48-1, 48-2, 48-3.
The output side of the digital / analog converter 45 is connected to the reverse amplifiers 47-1, 47-2, 47-3.

  The output side of the forward amplifier 48-1 is connected to the in-phase input point of the negative phase synthesizer 41-1. The output side of the reverse amplifier 47-1 is connected to the negative phase input point of the negative phase synthesizer 41-1.

  The negative phase synthesizer 41-2 is connected between the resonator 42-1 and the resonator 42-2. The output side of the resonator 42-1 and the output side of the forward amplifier 48-2 are connected to the in-phase input point of the negative phase synthesizer 41-2. The output side of the reverse amplifier 47-2 is connected to the negative phase input point of the negative phase synthesizer 41-2.

  The negative phase synthesizer 41-3 is connected between the resonator 42-2 and the analog / digital converter 43. The output side of the resonator 42-2 and the output side of the forward amplifier 48-3 are connected to the in-phase input point of the negative phase synthesizer 41-3. The output side of the reverse amplifier 47-3 is connected to the negative phase input point of the negative phase synthesizer 41-3.

  As described above, the received signals are input to the in-phase input points of the antiphase synthesizers 41-1, 41-2, and 41-3 via the forward amplifiers 48-1, 48-2, and 48-3. . The feedback output of the digital / analog converter 45 is fed to the negative phase input points of the negative phase synthesizers 41-1, 41-2, 41-3 via the reverse amplifiers 47-1, 47-2, 47-3. Added.

  Third-order feedforward control is performed by the forward amplifiers 48-1, 48-2, and 48-3, and third-order feedback control is performed by the backward amplifiers 47-1, 47-2, and 47-3. . Therefore, the signal transfer function of the delta sigma modulator 40D is expressed by a sixth order function.

  FIG. 8B is a diagram showing the configuration and operation of the radio receiver 30D in the fifth embodiment, and shows the phase characteristics of the signal transfer function of the delta-sigma modulator 40D. The horizontal axis indicates the phase ω, and the vertical axis indicates an example of the phase distortion of the signal transfer function STF (Signal Transfer Function).

  When the phase ω is 0, the phase distortion of the signal transfer function STF is a predetermined positive value. When the phase is (ω0−Δω), it becomes almost 0 and thereafter monotonously decreases to the phase π. The phase distortion of the signal transfer function STF increases when the phase exceeds π, and becomes almost zero again when the phase is (ω0 + Δω). The frequency (f0−Δf) corresponds to the phase (ω0−Δω). The frequency (f0 + Δf) corresponds to the phase (ω0 + Δω).

FIG. 9 is a diagram illustrating an implementation example of a wireless receiver according to the fifth embodiment.
A power element circuit 64, a high frequency connector 61, and a digital signal connector 62 are mounted on the multilayer printed circuit board 63, and functional element blocks to which the same symbols as those in FIG. Connected. The direct current generated in the power supply circuit 64 is supplied to the active element of the functional element block using a through hole or the like by a power supply line (not shown) provided in the inner layer of the multilayer printed board 63. A ground plane (not shown) for the analog signal line 65 and the digital signal line 66 is formed on the inner layer of the multilayer printed circuit board 63, and a strip line is formed by the ground plane and these signal lines to form a signal transmission path. .

The radio receiver 30D of the multilayer printed circuit board 63 is equipped with a high frequency connector 61 as an input end of a reception wave and a digital signal connector 62 as an output end of a digital signal. Thus, the delta-sigma modulator 40D shown in FIG. 8 is configured. The delta-sigma modulator 40D can be mass-produced using a printed circuit board process and an automatic surface mounting process for components, and is effective in reducing production costs.
(Operation of Fifth Embodiment)

  When the signal generated by the signal generation circuit is modulated using the phase information, the two carrier waves are modulated in opposite phases. In the case of converting a signal having two carrier frequencies into a signal having a frequency difference between the two carrier frequencies, it is desirable that the delta-sigma modulator 40D does not give phase distortion to the two carrier frequencies.

  The radio receiver 30D of the present embodiment makes the phase distortion of the signal transfer function STF of the delta-sigma modulator 40D zero at the carrier frequencies (f0 + Δf) and (f0−Δf) of the two transmission waves. Is possible. Therefore, it is possible to improve the phase modulation sensitivity of the radio receiver 30D.

(Effect of 5th Embodiment)
The fifth embodiment described above has the following effect (G).
(G) According to the delta-sigma modulator 40D of the present embodiment, the phase distortion is suppressed to be small at the carrier frequencies (f0 + Δf) and (f0−Δf) of the two transmission waves. It is possible to improve the phase modulation sensitivity.

(Configuration of the sixth embodiment)
FIGS. 10A and 10B are diagrams illustrating the configuration of a wireless communication system according to the sixth embodiment. Elements similar to those of the wireless communication system of the second embodiment shown in FIGS. 2A and 2B are assigned the same reference numerals.
FIG. 10A is a diagram illustrating a configuration of a wireless transmitter according to the sixth embodiment.

  In addition to the wireless transmitter 10 of the second embodiment, the wireless transmitter 10E of the present embodiment includes a combiner / distributor 19, a phase shifter 18, and linearly polarized transmission antennas 20b-1, 20b-2.

  The combiner / distributor 19 combines the two input signals and distributes them to the two output sides. The phase shifter 18 is, for example, a delay line, and delays the input signal by a time corresponding to a quarter wavelength of the carrier wave frequency f0 and outputs the delayed signal.

  The output sides of the modulators 13-1 and 13-2 are input to the combiner / distributor 19. The output side of the combiner / distributor 19 is connected to the transmission antenna 20b-1 and is also connected to the transmission antenna 20b-2 via the phase shifter 18.

FIG. 10B is a diagram illustrating the configuration of the wireless receiver according to the sixth embodiment.
The radio receiver 30E of this embodiment includes linearly polarized wave receiving antennas 31b-1 and 31b-2, a phase shifter 37, demodulators 40E-1 and 40E-2, and a receiving side control unit. 34A and a baseband circuit 51A.

  The phase shifter 37 delays the input signal by a time corresponding to a quarter wavelength of the carrier wave frequency f0 and outputs the delayed signal. Demodulators 40E-1 and 40E-2 demodulate the input signal at a predetermined carrier frequency. The baseband circuit 51A generates an information signal based on the two demodulated signals.

(Operation of the sixth embodiment)
Of the operation of the wireless communication system according to the sixth embodiment shown in FIGS. 10A and 10B, the description of the same parts as those of the wireless communication system according to the second embodiment shown in FIG. .

  In the wireless transmitter 10E, unlike the second embodiment, the output signal of the modulator 13-1 and the output signal of the modulator 13-2 are combined by the combiner / distributor 19 and then divided into two to shift the phase. The unit 18 adds a phase difference of 90 ° to the frequency f0, and transmits it to the air via two transmitting antennas 20b-1 and 20b-2 which are integral antennas of orthogonally polarized waves.

  In the radio receiver 30E, unlike the second embodiment, the received wave taken from the receiving antenna 31b-1 is demodulated by the demodulator 40E-1 and supplied to the baseband circuit 51A. The received wave captured from the receiving antenna 31b-2 is given a phase difference of 90 ° with respect to the frequency f0 by the phase shifter 37, demodulated by the demodulator 40E-2, and supplied to the baseband circuit 51A.

  According to the present embodiment, since transmission antennas 20b-1 and 20b-2, which are linearly polarized integrated antennas orthogonal to each other, are transmitted into the air, the polarization vector has a frequency difference between the two carrier frequencies. A rotating transmission wave can be emitted, and the rotation angle of the polarization vector of this transmission wave can be detected by the receiver. Thereby, compared with the radio | wireless communications system of 2nd Embodiment, size reduction is possible by integrating the transmitting antennas 20b-1 and 20b-2 and the receiving antennas 31b-1 and 31b-2, respectively. It is.

(Effect of 6th Embodiment)
The sixth embodiment described above has the following effect (H).
(H) Compared to the wireless communication system of the second embodiment, the transmission antennas 20b-1 and 20b-2 and the reception antennas 31b-1 and 31b-2 can be integrated to reduce the size. It is.

(Configuration of the seventh embodiment)
FIGS. 11A and 11B are diagrams illustrating the configuration of a wireless communication system according to the seventh embodiment. The same elements as those in the wireless communication system according to the sixth embodiment shown in FIG.

  Unlike the wireless transmitter 10E of the sixth embodiment, the wireless transmitter 10F illustrated in FIG. 11A includes transmission antennas 20c-1 and 20c-2 that are circularly polarized integrated antennas. Other than that, the configuration is the same as that of the wireless transmitter 10E of the sixth embodiment.

  Unlike the wireless receiver 30E of the sixth embodiment, the wireless receiver 30F illustrated in FIG. 11B includes receiving antennas 31c-1 and 31c-2 that are circularly polarized integrated antennas. Others have the same configuration as the wireless receiver 30E of the sixth embodiment.

(Manufacture of 7th Embodiment)
When producing two orthogonally polarized antennas such as the transmitting antennas 20b-1 and 20b-2 and the receiving antennas 31b-1 and 31b-2 of the sixth embodiment, strictly orthogonal straight lines It is necessary to physically realize the conductor. In manufacturing this linearly polarized antenna, it is practically difficult to maintain accuracy so as to be strictly orthogonal.
In this embodiment, the transmitting antennas 20b-1 and 20b-2 and the receiving antennas 31b-1 and 31b-2 are circularly polarized antennas having different rotation directions. In order to manufacture this, a right-hand circularly knitted wave antenna and a left-hand circularly polarized antenna need only be bonded together, and the relative position accuracy between the two antennas need not be considered. Therefore, it is not necessary to consider the pasting accuracy in the mass production process, so that the cost of the antenna can be reduced.

(Effect of 7th Embodiment)
The seventh embodiment described above has the following effect (I).
(I) In the mass production process of the transmitting antenna and the receiving antenna, it is not necessary to consider the pasting accuracy of the two antennas, so the cost of the antenna can be reduced.

(Configuration of Eighth Embodiment)
FIG. 12 is a diagram showing a configuration of an elevator system in the eighth embodiment.
The elevator system 100 includes a building 101 that is a vertically long rectangular parallelepiped and an elevator basket 111. Inside the building 101, a space in which the lifting cage 111 is raised and lowered is provided. The elevator cage 111 moves up and down the interior space of the building 101 by a rope and a drive mechanism (not shown).

  A base station radio 102-1 and an antenna 103-1 are installed on the ceiling of the internal space of the building 101, and a base station radio 102-2 and an antenna 103- are installed on the floor of the internal space of the building 101. 2 are installed. The base station radio devices 102-1 and 102-2 are radio devices having the same configuration as the radio receiver 30F shown in FIG. The antennas 103-1 and 103-2 are integrated reception antennas similar to the reception antennas 31b-1 and 31b-2 shown in FIG.

  An antenna 113-1 is provided on the upper surface of the lift cage 111, and an antenna 113-2 is provided on the lower surface. The terminal station radio 112 is a radio similar to the radio transmitter 10F shown in FIG. The antennas 113-1 and 113-2 are integrated transmission antennas similar to the transmission antennas 20c-1 and 20c-2 shown in FIG.

(Operation of Eighth Embodiment)
The radio wave transmitted from the terminal station radio 112 is transmitted via the antenna 113-1 and the antenna 113-2. The transmitted radio wave is subjected to multiple reflections by the inner wall of the building 101 and the outer wall of the elevating basket 111 because the internal space of the building 101 is used as a wireless transmission medium. That is, the internal space of the building 101 forms a multiwave interference environment. The radio waves that have received multiple reflections reach antennas 103-1 and 103-2, respectively.

  In the present embodiment, high-quality wireless transmission can be realized even in a multiwave interference environment. Since the elevator basket 111 can be controlled / monitored from the building 101 by wireless connection means, a space in which the elevator basket 111 is raised and lowered by wired connection means such as a cable is not wasted. Therefore, it is possible to improve the transportation capacity by making the building 101 a small volume or increasing the size of the elevating cage 111 with the same building 101 volume.

  In addition, the lifting cage 111 can be reduced in weight. This is because the weight of wired connection means such as a cable connected to the elevator cage 111 is a weight that cannot be ignored in high-rise buildings.

(Effect of 8th Embodiment)
The eighth embodiment described above has the following effect (J).
(J) Since the elevator basket 111 can be controlled / monitored from the building 101 by wireless connection means, the space in which the elevator basket 111 is raised and lowered by wired connection means such as cables is not wasted. Therefore, it is possible to improve the transport capability by setting the volume of the small building 101 or increasing the size of the lifting cage 111 with the same building 101 volume.

(Configuration of the ninth embodiment)
FIG. 13 is a diagram illustrating a configuration of a substation equipment monitoring system according to the ninth embodiment.
The substation equipment monitoring system 200 of the present embodiment includes a plurality of substations 201-1 to 201-12 and a plurality of radio base stations 211-1 to 211-4 set in the vicinity thereof. In the present embodiment, the number of transformers 201-1 to 201-12 is greater than the number of radio base stations 211-1 to 211-4.

  Each of the transformers 201-1 to 201-12 includes a terminal station radio 203 and an orthogonal polarization integrated antenna 202. The dimensions of the transformers 201-1 to 201-12 are on the order of several meters.

Each of the radio base stations 211-1 to 211-4 includes a base station radio 213 and an orthogonal polarization integrated antenna 212. The dimensions of the transformers 201-1 to 201-12 are overwhelmingly larger than the wavelength of electromagnetic waves having a frequency of several hundreds of MHz to several GHz used by the radio equipment.
(Operation of the ninth embodiment)

  In the substation equipment monitoring system 200 of the present embodiment, the electromagnetic waves are subjected to multiple reflections by the plurality of transformers 201-1 to 201-12. In the substation equipment monitoring system 200, a multi-wave interference environment is formed.

  The terminal station radio 203 and the base station radio 213 according to the present embodiment can realize high-quality radio transmission even in a multi-wave interference environment, and can be changed by a plurality of radio base stations 211-1 to 211-4. Remote control and remote monitoring of the electric machines 201-1 to 201-12 are possible. Therefore, it is possible to solve the problem of high-voltage induction power that becomes a problem when using a cable and the like, and the cost of laying the cable becomes unnecessary, improving the safety of the control / monitoring system of the transformers 201-1 to 201-12, and Cost reduction is possible.

(Effect of 9th Embodiment)
The ninth embodiment described above has the following effect (K).
(K) According to the wireless device of the present embodiment, high-quality wireless transmission can be realized even in a multi-wave interference environment. Control and monitoring of the substations 201-1 to 201-12 can be performed remotely by a plurality of radio base stations 211-1 to 211-4. Therefore, it is possible to solve the problem of high-voltage induced power that becomes a problem when the wired connection means such as a cable is used, and it is possible to eliminate the cable laying cost, and the safety of the control / monitoring system for the substations 201-1 to 201-12. Improvement and cost reduction are possible.

(Modification)
The present invention is not limited to the above embodiment, and can be modified without departing from the spirit of the present invention. As this usage pattern and modification, for example, there is the following (a).

(A) The transmission antennas 20-1 and 20-2 and the reception antenna 31 of the first to fifth embodiments are all linearly polarized antennas. However, the present invention is not limited to this, and a circularly polarized antenna may be used.

10, 10E, 10F Wireless transmitter 11 Information generation circuit 12-1, 12-2 Variable oscillator 13-1, 13-2 Modulator 17 Baseband circuit 20-1, 20-2 Transmitting antenna 30, 30A, 30B, 30C , 30D, 30E, 30F Wireless receiver 31 Reception antenna 32 Mixer 33 Oscillator 34 Reception-side control unit 35 Low-pass filter 35A Variable band-pass filter 36 Analog / digital converters 40, 40C, 40D Delta-sigma modulators 41-1, 41 -2, 41-3 Reverse phase synthesizers 42-1, 42-2 Resonator 43 Analog to digital converter 44 Oscillator 45 Digital to analog converter 46 Digital signal interpolator 47-1 to 47-3 Reverse amplifier 48- 1-48-3 Forward amplifier 51 Baseband circuit 100 Elevator system 200 Substation equipment monitoring system Temu

Claims (11)

A first transmission wave having a first carrier frequency modulated by an information signal having a band of a predetermined frequency;
A wireless transmitter for transmitting a second transmission wave having a second carrier frequency modulated by the information signal,
An average frequency that is an average of the first carrier frequency and the second carrier frequency is made constant, and a difference frequency that is a difference between the first carrier frequency and the second carrier frequency is made variable. A wireless transmitter. By selecting the first carrier frequency and the second carrier frequency from the frequency band divided into a plurality of channels, the first carrier frequency and the second carrier frequency are variable,
Transmitting the first transmission wave using a first transmission antenna;
The wireless transmitter according to claim 1, wherein the second transmission wave is transmitted using a second transmission antenna.   The wireless transmitter according to claim 2, wherein the information signals different from each other are transmitted by a plurality of combinations of the first transmission wave and the second transmission wave. A radio transmitter according to any one of claims 1 to 3 receives the first transmission wave and the second transmission wave transmitted to detect the difference frequency. And demodulating the information signal having a predetermined frequency band carried by the differential frequency. A first resonator that receives the reception signal of the wireless transmitter according to any one of claims 1 to 3 and uses the first carrier frequency as a resonance frequency;
A second resonator that receives the received signal and has the second carrier frequency as a resonance frequency;
An analog-to-digital converter that samples the output signal of the first resonator and the output signal of the second resonator at the frequency of the average frequency;
A digital signal interpolator for interpolating the signal sampled by the analog-to-digital converter;
A digital-to-analog converter that converts the output signal of the digital signal interpolator at a frequency that is an integral multiple of the average frequency;
A reverse phase synthesizer that synthesizes the output signal of the digital-analog converter from the received signal in reverse phase;
A radio transmitter characterized by detecting at the difference frequency by a delta-sigma modulator comprising:   The radio transmitter according to claim 5, wherein the phase of the signal transfer function of the delta-sigma modulator is substantially zero at the first carrier frequency and the second carrier frequency. . A printed circuit board on which the first resonator, the second resonator, the analog / digital converter, the digital signal interpolator, the digital / analog converter, and the antiphase synthesizer are mounted; The wireless transmitter according to claim 5, wherein: A wireless communication system comprising: the wireless transmitter according to claim 1; and the wireless transmitter according to claim 5.
The wireless transmitter varies the difference frequency according to a predetermined sequence,
The wireless receiver detects a combination of the detection frequency and the differential frequency having the highest conversion output by varying a detection frequency to be detected according to a predetermined sequence,
A radio communication system, wherein the information signal having the predetermined frequency band is demodulated. The predetermined sequence includes a training mode and a communication mode,
In the training mode, the wireless transmitter varies the differential frequency while selecting the differential frequency having the highest reception sensitivity in the wireless receiver,
The wireless transmitter and the wireless receiver perform communication using the first carrier frequency and the second carrier frequency corresponding to the difference frequency in the communication mode. The wireless communication system according to claim 8.   An elevator control system comprising the wireless communication system according to claim 8 or claim 9.   A substation equipment control system comprising the radio communication system according to claim 8 or claim 9.

Images