JP2015080353A - Motor driving device and motor driving method - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor driving device and a motor driving method capable of variably controlling currents flowing through a motor under a wide range of operating conditions, and stably driving the motor without individually compensating various control errors.SOLUTION: The driving device of a motor for controlling a voltage value to be applied to a motor (10), and for controlling the driving of the motor by allowing currents flowing through the motor to follow up a current command value includes: a current detector (5) for detecting currents flowing through the motor as a current detection value; a current deviation calculation unit for calculating a difference between the current command value and the current detection value as a current deviation; a magnetic flux estimation unit (1) for estimating a motor interlinkage magnetic flux interlinking the armature of the motor on the basis of the current deviation; and a current control unit (2) for determining a voltage value to be applied to the motor on the basis of the estimation value of the motor interlinkage magnetic flux, the current deviation and the rotational speed of the motor.

Description

  The present invention relates to an electric motor drive device that controls a current flowing through an electric motor by a voltage applied to the electric motor, and in particular, an electric motor drive device capable of variably controlling the electric current flowing through the electric motor over a wide range of operating conditions, and The present invention relates to a method for driving an electric motor.

  Various ideas have been made to drive an AC motor stably and with high performance in a wide range of operation. For example, when the rotor of the electric motor is rotating at high speed, non-interference control is performed in order to compensate for an induced voltage that increases substantially in proportion to the rotation speed.

  In the non-interference control for the electric motor, the voltage to be compensated is generally determined based on the current flowing through the electric motor (and the permanent magnet magnetic flux in an electric motor including a permanent magnet) and the rotation speed.

  Non-interference control in a conventional motor drive device is based on a current detection value obtained by detecting a current flowing in the motor (for example, see Patent Document 1), based on a current command value, or a combination of both. (For example, refer to Patent Document 2) and those based on current deviation (for example, refer to Patent Document 3 / Patent Document 4).

JP-A-6-153567 Japanese Patent No. 4128891 Japanese Patent No. 1641300 Japanese Patent No. 3321356 Japanese Patent No. 3683304 Japanese Patent No. 3527207

However, the prior art has the following problems.
In a conventional motor drive device, if the motor to be driven is driven in a wide operating range, the current flowing to the motor is limited during critical operation such as when the motor rotates at high speed or when the current flowing through the motor is large. There was a problem that proper control could not be achieved.

  For example, in the method of performing non-interference based on the current detection value and the rotation speed, if this is digitally mounted, it may vibrate at a frequency several times that of the control cycle due to delay, or remarkably current control There was a problem that became impossible. Although there are cases where such a problem can be solved by shortening the control cycle, there are cases where the control cycle cannot be shortened sufficiently due to the performance of the microcomputer used and restrictions due to PWM.

  Alternatively, in the method of performing non-interference based on the current command value and the rotation speed, the phase of the current command value is usually advanced as compared with the actual current. For this reason, the non-interacting voltage is not appropriate, and there is a case where current oscillation is observed in a frequency band near the design response of the current control system or in a frequency band higher than the design response. In addition, when current detection or voltage application includes a phase error, current pulsation may occur at a frequency lower than the above-described frequency band.

  Alternatively, in the method using the current detection value and the current command value in combination, some of the above-described problems may be solved by adjusting the weighting between the detection value and the command value. However, there are cases where all cannot be solved simultaneously. In particular, low frequency current pulsations caused by current detection and voltage application phase errors are often not solved.

  Note that a correction method for a current detection or voltage application phase error is known (for example, Patent Document 5 describes a voltage application error). However, when the error amount is unknown, an appropriate correction amount is also unknown. For this reason, another problem that correction cannot be performed properly occurs.

  Alternatively, in the method based on the current deviation and the rotation speed, the existing method has a problem that the non-interference compensation voltage cannot cope with the sudden change in the rotation speed and the current waveform is disturbed when the rotation speed changes suddenly.

  In the method disclosed in Patent Document 3, since the amount multiplied by the angular frequency is integrated, the integrator holds a large value during high-speed rotation. However, when the rotational speed is suddenly reduced, a value that is significantly larger than the appropriate amount during low-speed rotation remains in the integrator. As a result, there is a problem that the output voltage value becomes excessive and the current waveform is transiently disturbed.

  In the method disclosed in Patent Document 4, the value corresponding to the integral is not multiplied by the rotation speed equivalent amount. For this reason, at first glance, it seems that there is no influence of a sudden change in the rotational speed, but in reality it is affected by a sudden change in the rotational speed. This is because the necessary decoupling voltage is proportional to the rotation speed, but the integral value does not change instantaneously according to the change in the rotation speed, but includes a delay, so that the current waveform is transient. It is because there is no change in being disturbed.

  In the method disclosed in Patent Document 6, it is described that the integration gain multiplied by the integrated current deviation value has a high degree of freedom in setting a specific value thereof. However, no matter what value is set to the gain multiplied by the current deviation value, a transient disturbance occurs in the current waveform in accordance with the physical law, as described above.

  As described above, in the conventional motor drive device, if the motor is driven in a wide range of operating conditions, the pulsation is superimposed on the current at the limit operation such as at high speed rotation or at high current. In some cases, the current cannot be properly controlled and becomes unstable, for example, the control response of the current significantly decreases.

  The present invention has been made to solve the above-described problems, and can variably control the current flowing through the motor over a wide range of operating conditions, and can compensate the motor without individually compensating for various control errors. An object of the present invention is to obtain an electric motor drive device and an electric motor drive method capable of stably driving the motor.

  An electric motor drive apparatus according to the present invention is an electric motor drive apparatus that controls the voltage value applied to the electric motor and controls the electric motor by causing the current flowing through the electric motor to follow the current command value. A current detector that detects current as a current detection value, a current deviation calculator that calculates a difference between the current command value and the current detection value as a current deviation, and an armature that links the armatures of the motor based on the current deviation A magnetic flux estimator for estimating the linkage flux and a current controller for determining a voltage value to be applied to the motor based on the estimated value of the armature linkage flux, the current deviation, and the rotation speed of the motor.

  In addition, the motor driving method according to the present invention includes a control unit that controls a voltage value applied to the motor and controls the driving of the motor by causing the current flowing through the motor to follow the current command value. The motor drive method is executed by the control unit, wherein the control unit reads the rotation speed of the motor, a current detection step of detecting a current flowing through the motor as a current detection value by a current detector, and a current command A current deviation calculating step for calculating a difference between the current value and the detected current value as a current deviation, a magnetic flux estimating step for estimating an armature linkage flux that links the armatures of the motor based on the current deviation, and an armature linkage And a current control step for determining a voltage value to be applied to the electric motor based on the estimated value of the magnetic flux, the current deviation, and the rotational speed of the electric motor.

  According to the present invention, the armature linkage flux is estimated based on the current deviation that is the difference between the current command value and the current detection value, and the estimated value of the armature linkage flux, the current deviation, and the rotation speed of the motor are calculated. Based on this, the voltage value to be applied to the electric motor is determined. As a result, it is possible to variably control the current flowing through the motor over a wide range of operating conditions, and to stably drive the motor without individually compensating for various control errors. Can be obtained.

It is an illustration figure which shows the structure of the drive device of the electric motor which concerns on Embodiment 1 of this invention. It is an illustration figure which shows the structure of the magnetic flux estimator which concerns on Embodiment 1 of this invention. It is an illustration figure which shows the structure of the current controller which concerns on Embodiment 1 of this invention. It is an illustration figure which shows the structure of the drive device of the electric motor which concerns on Embodiment 3 of this invention. It is an illustration figure which shows the structure of the magnetic flux estimator which concerns on Embodiment 3 of this invention. It is an illustration figure which shows the structure of the current controller which concerns on Embodiment 3 of this invention.

  DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, preferred embodiments of an electric motor drive device and an electric motor drive method according to the present invention will be described with reference to the drawings. In addition, the same code | symbol is attached | subjected and demonstrated about the part which is the same or it corresponds in each figure.

Embodiment 1 FIG.
FIG. 1 is an exemplary diagram showing a configuration of a drive device for an electric motor according to Embodiment 1 of the present invention. 1 includes a magnetic flux estimator 1, a current controller 2, a dq / uvw converter 3, a voltage output unit 4, a current detector 5, a uvw / dq converter 6, and a differentiator 7. Configured.

  In addition to the electric motor driving device, FIG. 1 shows an electric motor 10 driven by the electric motor driving device and an electric angle detector 31 for detecting the electric angle of the rotor of the electric motor 10. The rotation angle detection value detected by the electrical angle detector 31 is illustrated as θ, and this rotation angle detection value θ is represented by the dq / uvw converter 3, the uvw / dq converter 6, and the differentiator 7. Given to.

  Hereinafter, the function of each component of the electric motor drive device according to Embodiment 1 of the present invention will be described with reference to FIG.

The current detector 5 detects a three-phase current flowing through the electric motor 10 to be driven. The detection values for these three phases are collectively referred to as a three-phase current detection value, and the detection values for each phase of uvw are respectively represented as u-phase current detection value (i _u ), v-phase current detection value (i _v ), The w-phase current detection value (i _w ) is shown in FIG.

The uvw / dq converter 6 converts the three-phase current detection value into a dq coordinate system synchronized with the rotation of the electric motor 10. In the first embodiment, the d-axis direction is the direction of the magnetic flux of the rotor of the electric motor 10, and the q-axis is the orthogonal direction of the d-axis (the direction advanced 90 degrees counterclockwise along the rotation axis of the electric motor). . These converted values by uvw / dq converter 6, shown in the figure as respectively d-axis current detection value _{(i d)} and the q-axis current detection value _{(i q).}

  Specifically, the uvw / dq converter 6 performs the conversion of the following expression (1).

D-axis current command value given from the outside _(i ^{d *)} and the d-axis current detection value _{(i d)} and the deviation of is calculated by the following equation (2), which with the d-axis current deviation _(i ^{d ~)} Call. Similarly, the deviation between the q-axis current command value (i _q ^* ) given from the outside and the q-axis current detection value (i _q ) is also calculated by the following equation (2), which is calculated as the q-axis current deviation (i _q ^~ )

FIG. 2 is an exemplary diagram showing a configuration of the magnetic flux estimator 1 according to the first embodiment of the present invention. The magnetic flux estimator 1 estimates the armature linkage magnetic flux of each of the d axis and the q axis from the d axis current deviation and the q axis current deviation. FIG. 1 and FIG. 2 show the d-axis component of this estimated value as the d-axis armature flux linkage estimated value (φ _d ^). Further, the q-axis component is shown in FIG. 1 and FIG. 2 as the q-axis armature flux linkage estimated value (φ _q ^).

  Specifically, the magnetic flux estimator 1 performs the calculation of the following expression (3).

Here, _Kd represents a d-axis armature linkage flux estimation gain, and _Kq represents a q-axis armature linkage flux estimation gain.

  The differentiator 7 receives and differentiates the rotation angle detection value θ from the electrical angle detector 31 and outputs it to the current controller 2 as the rotation speed (ω).

FIG. 3 is an exemplary diagram showing a configuration of the current controller 2 according to the first embodiment of the present invention. The current controller 2 calculates a voltage command value from the current deviation, the armature flux linkage estimated value, and the rotation speed (ω). In the first embodiment, the current deviation indicates the d-axis current deviation and the q-axis current deviation, and the armature linkage flux estimated value is the d-axis armature linkage flux estimate and the q-axis electrical machine. Refers to the estimated value of flux linkage. The rotation speed is a value output from the differentiator 7. The voltage command value means a voltage suitable for converging the current flowing through the armature toward the command value, and in the first embodiment, for each component of the d-axis component and the q-axis component. Calculated. These are respectively shown in FIG. 1 and FIG. 3 as a d-axis voltage command value (v _d ^* ) and a q-axis voltage command value (v _q ^* ).

  Specifically, the current controller 2 performs the calculation of the following expression (4).

Here, K _Pd is a d-axis proportional gain, K _Id is a d-axis integral gain, K _Pq is a q-axis proportional gain, and K _Iq is a q-axis integral gain.

  The place shown in FIG. 3 and the above equation (4) is as follows. The d-axis voltage command value means that the product of the q-axis armature flux linkage estimated value and the rotation speed is subtracted from the proportional-integral control based on the d-axis current deviation. Further, the q-axis voltage command value means that the product of the d-axis armature flux linkage estimated value and the rotational speed is determined by proportional integral control based on the q-axis current deviation.

The dq / uvw converter 3 converts the d-axis voltage command value and the q-axis voltage command value into voltage command values on three-phase coordinates. These command values for three phases are collectively referred to as a three-phase voltage command value. These three-phase voltage command values are shown in FIG. 1 as a u-phase voltage command value (v _u ^* ), a v-phase voltage command value (v _v ^* ), and a w-phase voltage command value (v _w ^* ), respectively.

  Specifically, the dq / uvw converter 3 performs the conversion of the following equation (5).

  The voltage output unit 4 applies a voltage that substantially matches the three-phase voltage command value to the electric motor 10 that is the drive target. The hardware configuration of this part is a configuration according to a three-phase voltage source inverter, and the voltage output unit 4 applies a pulse-width modulated voltage to the electric motor 10.

  An example of setting various gains in the first embodiment is collectively shown in the following formula (6).

Here, ω _{cc, d} is a d-axis current response design parameter, ω _{cc, q} is a q-axis current response design parameter, Ld is a d-axis inductance, Lq is a q-axis inductance, and R is a winding resistance value. .

  As described above, the motor drive apparatus according to the first embodiment reflects the estimated magnetic flux value based on the current deviation, which is the difference between the current command value and the detected current value, in the applied voltage to the motor 10. ing. As a result, adverse effects on the stability of the electric motor 10 can be suppressed, and the electric motor 10 can be stably driven in a wide range of operation without individually compensating for various errors. Furthermore, even when the rotational speed changes suddenly, it is possible to obtain an excellent effect that cannot be obtained by the conventional technique, such as no current disturbance.

  In addition, as shown in FIG. 1, by converting the three-phase coordinates to the dq axis and integrating the deviation between the current command value and the detected current value for each of the d axis and the q axis, the d axis can be obtained with a simple configuration. It becomes possible to estimate the armature linkage magnetic flux of each q axis.

  However, the present invention itself can be implemented on arbitrary coordinate axes, and the implementation is not limited only to the dq coordinate system.

  As described above, according to the first embodiment, the armature linkage flux is estimated based on the current deviation which is the difference between the current command value and the current detection value, and the estimated value of the armature linkage flux, the current deviation The voltage value to be applied to the motor is determined based on the rotation speed of the motor. As a result, it is possible to variably control the current flowing through the motor over a wide range of operating conditions, and to stably drive the motor without individually compensating for various control errors. Can be obtained.

Embodiment 2. FIG.
In the second embodiment, a mode suitable for a case where a permanent magnet type synchronous motor having a permanent magnet is targeted as the motor 10 to be driven by the driving device of the motor will be described. The configuration of the motor driving device according to the second embodiment is the same as the configuration of the motor driving device according to the first embodiment shown in FIG.

  When a permanent magnet type synchronous motor having a permanent magnet is used as the electric motor 10 to be driven in the first embodiment, the estimated value of the armature flux linkage and the actual value are calculated at the time of starting. Make a difference. This is because, in the magnetic flux estimator 1 in the first embodiment, the initial value of the integrator is zero, so that the armature flux linkage estimated value is also zero. On the other hand, when the electric motor 10 is a permanent magnet type synchronous motor 10, the magnetic flux generated by the permanent magnet is linked to the armature before starting driving (before the electric current flows through the electric motor). This is because the flux linkage is non-zero.

  Therefore, in the second embodiment, a value corresponding to the armature linkage component of the permanent magnet magnetic flux is set in advance as the initial value of the integrator in the magnetic flux estimator 1. As a specific setting example, the measured value of the torque constant or counter electromotive voltage constant divided by the number of pole pairs is set as the initial value of the integrator on the d-axis armature linkage component side, and the q-axis armature linkage is set. Zero is set as the initial value on the component side. Alternatively, the design value of the electric motor 10 may be used. In this case, the d-axis component is a non-zero finite value and the q-axis side is zero. Note that the initial value on the q-axis side of zero is not necessarily completely zero, and may be substantially zero.

  As described above, according to the second embodiment, when a permanent magnet type synchronous motor having a permanent magnet is targeted as an electric motor to be driven by an electric motor drive device, an integrator in the magnetic flux estimator 1 is used. As an initial value, an appropriate value is set in advance. As a result, it is possible to obtain an electric motor drive device and an electric motor drive method that provide good control characteristics when the electric motor is started.

Embodiment 3 FIG.
In the present third embodiment, a mode suitable for a case where the electric motor 10 to be driven by the electric motor drive device is a type of electric motor 10 called a rotary field type synchronous motor having field windings will be described. .

  FIG. 4 is an exemplary diagram showing a configuration of a motor drive device according to Embodiment 3 of the present invention. 4 includes a magnetic flux estimator 1a, a current controller 2a, a dq / uvw converter 3, a voltage output unit 4, a current detector 5, a uvw / dq converter 6, a differentiator 7, a field magnet. A winding voltage output device 8 and a field winding current detector 9 are provided.

  In addition to the electric motor driving device, FIG. 4 shows a rotating field type synchronous motor 10a driven by the electric motor driving device and an electric angle for detecting the electric angle of the rotor of the rotating field type synchronous motor 10a. An angle detector 31 is shown. The rotation angle detection value detected by the electrical angle detector 31 is illustrated as θ, and this rotation angle detection value θ is represented by the dq / uvw converter 3, the uvw / dq converter 6, and the differentiator 7. Given to.

  Hereinafter, the function of each component of the electric motor drive device according to Embodiment 3 of the present invention will be described with reference to FIG. 4, particularly focusing on the differences from the previous Embodiment 1.

  In the rotating field type synchronous motor 10a to be driven, in addition to the three-phase winding of the stator, the rotor also has a winding for the rotating field. The winding for the rotating field is usually connected to the outside of the rotating field type synchronous motor 10a via a slip ring or the like.

The field winding current detector 9 detects a current flowing in the rotating field winding. This detected value is a field winding current detected value ( _if ) and is shown in FIG. In addition, the coordinate axis corresponding to the field winding is referred to as f-axis for convenience.

F-axis current command value given from the outside and (i f ^_*), is calculated the deviation between the f-axis current detection value, called At the the f-axis current deviation (i f ^_~). This f-axis current deviation (i _f ^˜ ) is given to the magnetic flux estimator 1a and the current controller 2a.

FIG. 5 is an exemplary diagram showing a configuration of a magnetic flux estimator 1a according to the third embodiment of the present invention. The magnetic flux estimator 1a shown in FIG. 5 performs estimation including a component interlinked with the armature in the magnetic flux generated by the field winding in addition to the magnetic flux estimator 1 of the first embodiment. That is, the magnetic flux estimator 1a estimates the armature linkage magnetic flux from the d-axis current deviation, the q-axis current deviation, and the f-axis current deviation. Of the estimated values, the d-axis component is referred to as a d-axis armature flux linkage estimated value (φ _d ^), and the q-axis component is referred to as a q-axis armature flux linkage estimated value (φ _q ^) This is the same as in the first embodiment.

  Specifically, the magnetic flux estimator 1a estimates the magnetic flux by the calculation of the following equation (7).

Here, K _f denotes the f-axis armature flux linkage estimation gain.

The current controller 2a calculates a voltage command value from the current deviation, the armature flux linkage estimated value, and the rotation speed (ω). In the third embodiment, the current deviation indicates the d-axis current deviation, the q-axis current deviation, and the f-axis current deviation, and the armature linkage flux estimated value is the d-axis armature linkage flux estimation. Value and q-axis armature flux linkage estimated value. The rotation speed is a value corresponding to a differential value obtained by differentiating the rotation angle detection value θ with the differentiator 7. The voltage command value means a voltage suitable for converging the current flowing through the armature toward the command value. In the third embodiment, the d-axis component, the q-axis component, and the field winding are It is calculated for each component of the f-axis component shown. These are shown in FIG. 4 as a d-axis voltage command value (v _d ^* ), a q-axis voltage command value (v _q ^* ), and an f-axis voltage command value (v _f ^* ), respectively.

  Specifically, the current controller 2a performs the calculation of the following formula (8).

Here, K _Pf is the f-axis proportional gain, K _If is the f-axis integral gain, K _Pdf is the d-axis f-axis cross proportional gain, and K _Pfd is the f-axis d-axis cross proportional gain.

FIG. 6 is an exemplary diagram showing a configuration of a current controller 2a according to Embodiment 3 of the present invention. FIG. 6 illustrates the calculation of the current controller 2a. The P is proportional control in FIG. 6, P _I represents a proportional integral control.

  The place shown in FIG. 6 and the above equation (8) is as follows. The f-axis voltage command value means that it is determined by the sum of proportional-integral control based on f-axis current deviation and proportional control based on d-axis current deviation. The d-axis voltage command value is determined by subtracting the product of the q-axis armature flux linkage estimated value and the rotational speed from the sum of the proportional integral control based on the d-axis current deviation and the proportional control based on the f-axis current deviation. It means that. Further, the q-axis voltage command value means that the product of the d-axis armature flux linkage estimated value and the rotational speed is determined by proportional integral control based on the q-axis current deviation.

  As in the first embodiment, the voltage output device 4 applies a voltage that substantially matches the three-phase voltage command value to the rotating field synchronous motor 10a that is the drive target. The hardware configuration of this part is a configuration according to a three-phase voltage source inverter, and the voltage output unit 4 applies a pulse-width modulated voltage to the three phases of the rotating field type synchronous motor 10a. is there.

  On the other hand, the field winding voltage output device 8 which is a hardware configuration of a portion for applying a voltage to the field winding is preferably a single-phase inverter circuit. The field winding voltage output unit 8 applies a pulse-width modulated voltage to the field winding of the rotating field type synchronous motor 10a in the same manner as the voltage output unit 4.

  An example of setting various gains in the third embodiment is collectively shown in the following equation (9).

Here, ω _{cc, f} is a design parameter of the f-axis current response, M _df is a mutual inductance between the d-axis and the f-axis, L _f is an inductance of the f-axis, and R _f is a resistance of the f-axis. Other gains not described are the same settings as in the above equation (6) shown in the first embodiment.

  As described above, according to the third embodiment, when a rotating field type synchronous motor having a field winding is targeted as an electric motor to be driven by an electric motor drive device, the rotating field Further considering the winding, the voltage value to be applied to the field winding is determined based on the field winding current deviation. As a result, the current flowing through the rotating field synchronous motor can be variably controlled over a wide range of operating conditions, and the rotating field synchronous motor can be stably driven without individually compensating for various control errors. An electric motor drive device and an electric motor drive method can be obtained.

  1, 1a magnetic flux estimator, 2, 2a current controller, 3 dq / uvw converter, 4 voltage output device, 5 current detector, 6 uvw / dq converter, 7 differentiator, 8 field winding voltage output device , 9 Field winding current detector, 10 electric motor, 10a rotating field synchronous motor, 11 d-axis inductance multiplier, 12 d-axis current design response multiplier, 13 d-axis integrator, 14 q-axis inductance multiplier, 15 q-axis current design response multiplier, 16 q-axis integrator, 17 mutual inductance multiplier, 18 f-axis current design response multiplier, 21 d-axis current proportional integration controller, 22 rotational speed multiplier for q-axis flux estimate , 23 q-axis current proportional integration controller, 24 d-axis flux estimated rotation speed multiplier, 25 f-axis current proportional integration controller, 26 d-axis f-axis cross-proportional controller, 27 f-axis d-axis cross-proportional control , 31 Electrical angle detector.

The electric motor drive device according to the present invention employs a rotating field type synchronous motor having a field winding as the electric motor, controls the voltage value applied to the electric motor, and causes the current flowing through the electric motor to follow the current command value. An electric motor drive device that controls the electric motor by a current detector that detects a current flowing through the electric motor as a current detection value, and a current deviation calculation that calculates a difference between the current command value and the current detection value as a current deviation A magnetic flux estimator that estimates the armature interlinkage magnetic flux that links the armature of the motor based on the current deviation, and the motor based on the estimated value of the armature interlinkage magnetic flux, the current deviation, and the rotation speed of the motor. A current controller for determining a voltage value to be applied to the current, a uvw / dq converter for converting a current detection value output from the current detector from a three-phase coordinate system to a dq coordinate system, and a voltage value output from the current controller D Dq / uvw converter for converting from a coordinate system to a three-phase coordinate system, a field winding current detector for detecting an f-axis current detection value that is a current flowing in the field winding, and an f-axis current applied from the outside An f-axis current deviation calculator that calculates a difference between the command value and the detected f-axis current value as an f-axis current deviation. The current deviation calculator is a current detection value in the dq coordinate system output by the dq / uvw converter. And the current command value given as the dq coordinate system are calculated as a d-axis current deviation and a q-axis current deviation, and the magnetic flux estimator integrates the sum of the d-axis current deviation and the f-axis current deviation. A d-axis integrator for estimating the d-axis component of the interlinkage magnetic flux and a q-axis integrator for estimating the q-axis component of the armature interlinkage flux by integrating the q-axis current deviation; Are the estimated values of the d-axis component and q-axis component of the armature flux linkage, and the d-axis current deviation and and the q-axis current deviation, and f-axis current deviation, and determines the value of the voltage applied to the motor as a dq coordinate system on the basis of the rotational speed of the electric motor, is what determines the value of the voltage applied to the field winding .

The electric motor driving method according to the present invention applies a rotating field type synchronous motor having a field winding as the electric motor, controls the voltage value applied to the electric motor, and sets the current flowing through the electric motor to the current command value. An electric motor driving method executed by an electric motor drive device including a control unit that controls driving of the electric motor by following the electric motor. The control unit includes a rotation speed reading step for reading the rotation speed of the electric motor, and a current detector. A current detection step for detecting a current flowing through the motor as a current detection value, a current deviation calculation step for calculating a difference between the current command value and the current detection value as a current deviation, and linkage of the armature of the motor based on the current deviation. Voltage estimation step for estimating the armature interlinkage magnetic flux to be applied, and the voltage value to be applied to the motor based on the estimated value of the armature interlinkage magnetic flux, the current deviation, and the rotation speed of the motor A current control step determining, the current detection value output by the current detection step, the uvw / dq conversion step of converting the dq coordinate system from the three-phase coordinate system, the voltage value output from the current control step, dq coordinates A dq / uvw conversion step for converting the system into a three-phase coordinate system, a field winding current detection step for detecting an f-axis current detection value that is a current flowing in the field winding, and an f-axis current command given from the outside And an f-axis current deviation calculation step for calculating a difference between the value and the detected f-axis current value as an f-axis current deviation. The current deviation calculation step is a current detection in the dq coordinate system output in the dq / uvw conversion step. The difference between the value and the current command value given as the dq coordinate system is calculated as the d-axis current deviation and the q-axis current deviation, and the magnetic flux estimation step integrates the sum of the d-axis current deviation and the f-axis current deviation. The d-axis component of the armature linkage flux is estimated, and the q-axis component of the armature linkage flux is estimated by integrating the q-axis current deviation, and the current control step includes the d-axis component of the armature linkage flux and Based on the estimated value of the q-axis component, the d-axis current deviation and the q-axis current deviation, the f-axis current deviation, and the rotation speed of the motor, the voltage value applied to the motor is determined as the dq coordinate system, and the field The voltage value to be applied to the winding is determined .

Claims (5)

A motor drive device for controlling the voltage value applied to the electric motor and controlling the electric motor by causing the current flowing through the electric motor to follow the current command value;
A current detector for detecting a current flowing through the motor as a current detection value;
A current deviation calculator for calculating a difference between the current command value and the current detection value as a current deviation;
A magnetic flux estimator for estimating an armature interlinkage magnetic flux interlinking the armature of the electric motor based on the current deviation;
A motor drive device comprising: a current controller that determines the voltage value to be applied to the motor based on the estimated value of the armature flux linkage, the current deviation, and the rotation speed of the motor. A uvw / dq converter for converting the current detection value output by the current detector from a three-phase coordinate system to a dq coordinate system;
A dq / uvw converter that converts the voltage value output from the current controller from the dq coordinate system to the three-phase coordinate system;
The current deviation calculator calculates a difference between a current detection value of the dq coordinate system output from the dq / uvw converter and a current command value given as the dq coordinate system as a d-axis current deviation and a q-axis current deviation. And
The magnetic flux estimator integrates the d-axis current deviation to estimate a d-axis component of the armature linkage magnetic flux, and integrates the q-axis current deviation to integrate the armature linkage. A q-axis integrator for estimating the q-axis component of the magnetic flux,
The current controller controls the motor based on the estimated values of the d-axis component and the q-axis component of the armature flux linkage, the d-axis current deviation and the q-axis current deviation, and the rotation speed of the motor. The motor drive device according to claim 1, wherein the voltage value to be applied is determined as the dq coordinate system. The electric motor is a permanent magnet type synchronous motor having a permanent magnet,
The magnetic flux estimator is
The electric motor according to claim 2, wherein a non-zero finite value defined in advance according to a magnetic flux of the permanent magnet is set as an initial value of the d-axis integrator, and zero is set as an initial value of the q-axis integrator. Drive device. The electric motor is a rotating field type synchronous motor having a field winding,
A field winding current detector for detecting an f-axis current detection value that is a current flowing through the field winding;
An f-axis current deviation calculator that calculates a difference between the f-axis current command value given from the outside and the f-axis current detection value as an f-axis current deviation;
The d-axis integrator of the magnetic flux estimator estimates a d-axis component of the armature flux linkage by integrating the sum of the d-axis current deviation and the f-axis current deviation,
The current controller includes estimated values of the d-axis component and the q-axis component of the armature flux linkage, the d-axis current deviation and the q-axis current deviation, the f-axis current deviation, and the rotation speed of the motor. The motor drive device according to claim 2, wherein the voltage value to be applied to the electric motor is determined as the dq coordinate system based on and the voltage value to be applied to the field winding is determined. An electric motor driving method executed by an electric motor driving device including a control unit that controls a drive of the electric motor by controlling a voltage value applied to the electric motor and causing a current flowing through the electric motor to follow a current command value. And
The controller is
A rotation speed reading step for reading the rotation speed of the electric motor;
A current detection step of detecting a current flowing through the motor as a current detection value by a current detector;
A current deviation calculating step of calculating a difference between the current command value and the current detection value as a current deviation;
A magnetic flux estimation step for estimating an armature interlinkage magnetic flux interlinking the armature of the electric motor based on the current deviation;
And a current control step of determining the voltage value to be applied to the motor based on the estimated value of the armature flux linkage, the current deviation, and the rotational speed of the motor.

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