JP2015042080A - Switching power supply device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a zero voltage switching type switching power supply device that reduces an effect of resonance generated by an inductance on a primary side and a capacitance component on a secondary side.SOLUTION: In a switching power supply device 100, a first protection diode D1 is connected between a first junction of a first resonant inductor L1 on a primary side and one terminal of a primary winding of a transformer T1, and a power line on a high voltage side of a DC power supply E. A second protection diode D2 is connected between the first junction and a power line on a low voltage side of the DC power supply E. A third protection diode D3 is connected between a second junction of a second resonant inductor L2 and the other terminal of the primary winding, and the power line on the high voltage side of the DC power supply E. A fourth protection diode D4 is connected between the second junction and the power line on the low voltage side of the DC power supply E.

Description

  The present invention relates to a zero voltage switching (ZVS) switching power supply.

  Soft switching with reduced switching loss of switching elements has been put into practical use in switching power supplies. A typical example of a switching power supply realizing soft switching is a phase shift type full bridge circuit. In this circuit, the resonance between the parasitic capacitance formed in parallel with the switching element and the inductance connected in series with the primary winding of the transformer is used. By this resonance, zero voltage switching is realized by turning on the switching element when the voltage across the switching element becomes zero (see, for example, Patent Document 1).

  The switching loss occurs when the switching element is turned on while the charge is stored in the parasitic capacitance of the switching element, and the parasitic capacitance is short-circuited. In this regard, when the series resonance between the parasitic capacitance and the inductance is used, the parasitic capacitance can be made zero by causing a current in the reverse direction to flow through the parasitic capacitance by the energy stored in the inductance.

JP 2003-158873 A

  In order to realize the above-described zero voltage switching, it is necessary to provide an inductance between the switching element and the primary winding of the transformer. In this configuration, resonance also occurs between the inductance and the parasitic capacitance of the diode constituting the rectifier circuit on the secondary side of the transformer. Due to this resonance, voltage oscillation and surge voltage are generated at the connection point between the inductance and the parasitic capacitance. This voltage oscillation leads to power loss of the entire circuit, and the surge voltage becomes a factor that causes malfunctions in the components.

  The present invention has been made in view of such circumstances, and an object of the present invention is to reduce the influence of resonance generated by primary-side inductance and secondary-side capacitance components in a zero-voltage switching switching power supply device. Is to provide.

  In order to solve the above problems, a switching power supply device according to an aspect of the present invention includes a first arm including two switching elements, a full bridge circuit in which a second arm including two switching elements is connected in parallel, A transformer for transforming the output power of the bridge circuit; an output terminal of the first arm; a first inductor connected between one terminal of the primary winding of the transformer; an output terminal of the second arm; A second inductor connected between the other terminal of the primary winding, a rectifier circuit connected to the secondary winding of the transformer and rectifying the output power of the transformer, and the power rectified by the rectifier circuit being smoothed The anode terminal is connected to the first connection point of the smoothing circuit, the first inductor, and one terminal of the primary winding, and the power supply on the high voltage side of the power source supplying power to the full bridge circuit A first diode having a cathode terminal connected to the line; a second diode having a cathode terminal connected to the first connection point; and an anode terminal connected to the power line on the low voltage side of the power supply; a second inductor and a primary winding; A third diode having an anode terminal connected to the second connection point with the other terminal of the line, a cathode terminal connected to the power line on the high voltage side of the power supply, and a cathode terminal connected to the second connection point; And a fourth diode having an anode terminal connected to the low-voltage power line.

  Another embodiment of the present invention is also a switching power supply device. This device includes a first arm including two switching elements, a full bridge circuit in which a second arm including two switching elements is connected in parallel, a transformer for transforming output power of the full bridge circuit, A first inductor connected between the output terminal and one terminal of the primary winding of the transformer, a rectifier circuit connected to the secondary winding of the transformer and rectifying the output power of the transformer, and rectified by the rectifier circuit A smoothing circuit for smoothing the generated power, and a high voltage of a power source that supplies power to the full bridge circuit with the anode terminal connected to the first connection point of the first inductor and one terminal of the primary winding A first diode whose cathode terminal is connected to the power line on the side, a second diode whose cathode terminal is connected to the first connection point, and whose anode terminal is connected to the power line on the low voltage side of the power supply. It includes a diode, a.

  It should be noted that any combination of the above-described constituent elements and a representation obtained by converting the expression of the present invention between apparatuses, methods, systems, and the like are also effective as an aspect of the present invention.

  ADVANTAGE OF THE INVENTION According to this invention, the influence of the resonance which generate | occur | produces with the primary side inductance and the capacitance component of a secondary side can be reduced in the switching power supply device of a zero voltage switching system.

It is a figure which shows the basic structural example of the switching power supply device of a zero voltage switching system. It is a figure which shows the timing chart for demonstrating the operation example 1 of the switching power supply device of FIG. It is a figure which shows the timing chart for demonstrating the operation example 2 of the switching power supply device of FIG. It is a figure which shows the structural example of the switching power supply device which concerns on a comparative example. It is a figure which shows the structural example of the switching power supply device which concerns on embodiment of this invention. FIGS. 6A to 6B are diagrams illustrating waveforms of voltages applied to the primary winding of the transformer. It is a figure which shows the structural example of the switching power supply device which concerns on a modification.

  FIG. 1 is a diagram illustrating a basic configuration example of a zero-voltage switching switching power supply device 100. The switching power supply apparatus 100 is a phase shift type full bridge type DC-DC converter. The phase shift type full-bridge type DC-DC converter can convert large power with high efficiency.

  The switching power supply device 100 of FIG. 1 includes a full bridge circuit 10, a first resonant inductor L1, a transformer T1, a rectifier circuit 20, a smoothing circuit 30, and a control unit 40. The full bridge circuit 10 includes a first arm including a first switching element Qa on the upper side and a second switching element Qb on the lower side, a second arm including a third switching element Qc on the upper side and a fourth switching element Qd on the lower side. Is provided. The first arm and the second arm are connected to the DC power supply E in parallel. For the first switching element Qa to the fourth switching element Qd, semiconductor switching elements such as MOSFET (Metal Oxide Semiconductor Field Effect Transistor) and IGBT (Insulated Gate Bipolar Transistor) are used, for example. Hereinafter, it is assumed that N-channel MOSFETs are used for the first switching element Q1 to the fourth switching element Qd.

  A first parasitic diode Da and a first parasitic capacitance Ca are formed in parallel with the first switching element Qa. Since the N-channel MOSFET is used for the first switching element Qa, the first parasitic diode Da is formed in the direction opposite to the conduction direction of the first switching element Qa. That is, the direction from the source to the drain is formed in the forward direction. The first parasitic capacitance Ca is an output capacitance Coss formed when the drain and source of the first switching element Qa are short-circuited in an alternating manner.

  Similarly, a second parasitic diode Db and a second parasitic capacitor Cb are formed in parallel with the second switching element Qb. Similarly, a third parasitic diode Dc and a third parasitic capacitor Cc are formed in parallel with the third switching element Qc. Similarly, a fourth parasitic diode Dd and a fourth parasitic capacitor Cd are formed in parallel with the fourth switching element Qd.

  The first control pulse signal Pa to the fourth control pulse signal Pd generated by the control unit 40 are input to control terminals (gate terminals in the case of MOSFET) of the first switching element Qa to the fourth switching element Qd, respectively. The full bridge circuit 10 receives the first control pulse signal Pa to the fourth control pulse signal Pd, switches the DC power from the DC power source E, and converts it into AC power.

  The transformer T1 transforms the output power of the full bridge circuit 10 according to the winding ratio of the primary winding and the secondary winding, and insulates the primary side and the secondary side. The upper terminal of the primary winding of the transformer T1 is connected to the output terminal of the first arm of the full bridge circuit 10 via the first resonant inductor L1. The lower terminal of the primary winding of the transformer T1 is connected to the output terminal of the second arm of the full bridge circuit 10.

  The rectifier circuit 20 is connected to the secondary winding of the transformer T1, and rectifies the output power of the transformer T1. The rectifier circuit 20 includes a first rectifier diode Dr1 and a second rectifier diode Dr2. A first secondary side parasitic capacitor Cr1 is formed in parallel with the first rectifier diode Dr1, and a second secondary side parasitic capacitor Cr2 is formed in parallel with the second rectifier diode Dr2.

  The anode terminal of the first rectifier diode Dr1 is connected to the upper terminal of the secondary winding of the transformer T1. The anode terminal of the second rectifier diode Dr2 is connected to the lower terminal of the secondary winding of the transformer T1. The cathode terminal of the first rectifier diode Dr1 and the cathode terminal of the second rectifier diode Dr2 are connected and connected to the high voltage side input terminal of the smoothing circuit 30. The midpoint of the secondary winding of the transformer T1 is connected to the low voltage side input terminal of the smoothing circuit 30.

  The smoothing circuit 30 smoothes the power rectified by the rectifying circuit 20 and supplies the smoothed power to an external load Ro. The smoothing circuit 30 includes an output inductor Lo and an output capacitor Co, and smoothes the output power of the rectifier circuit 20.

  The control unit 40 generates the first control pulse signal Pa to the fourth control pulse signal Pd to be input to the control terminals of the first switching element Qa to the fourth switching element Qd, and drives the full bridge circuit 10 by the phase shift method. To do. The controller 40 turns on the first switching element Qa after the first parasitic capacitance Ca of the first switching element Qa is discharged by a current based on the energy of the first resonant inductor L1. The same applies to the second switching element Qb to the fourth switching element Qd.

  2 is a timing chart for explaining an operation example 1 of the switching power supply apparatus 100 of FIG. The operation example 1 is an operation example of a general phase shift method. In the full bridge circuit, a positive voltage is applied to the transformer T1 when the first switching element Qa and the fourth switching element Qd are on and the second switching element Qb and the third switching element Qc are off. Conversely, when the first switching element Qa and the fourth switching element Qd are off and the second switching element Qb and the third switching element Qc are on, a reverse voltage is applied to the transformer T1.

  In the hard switching method, the output voltage Vo is adjusted by adjusting the ON time with respect to the unit period T, that is, by adjusting the duty ratio. On the other hand, in the phase shift method according to the operation example 1, the duty ratio of the first control pulse signal Pa to the fourth control pulse signal Pd is fixed at 50%. The first switching element Qa and the second switching element Qb operate complementarily except for the dead time. Similarly, the third switching element Qc and the fourth switching element Qd operate complementarily except for the dead time.

  Based on the above, the pulse phase of the first arm configured by the first switching element Qa and the second switching element Qb and the pulse phase of the second arm configured by the third switching element Qc and the fourth switching element Qd are The output voltage Vo is adjusted by adjusting the phase difference. Specifically, the control unit 40 compares the output voltage Vo output to the load Ro with the target voltage, and sets the pulse phase of the second arm with respect to the pulse phase of the first arm so that both voltages approach each other. Shift adaptively. When the phase difference between the pulse phase of the first arm and the pulse phase of the second arm changes, the amount of current flowing through the primary winding of the transformer T1 can be adjusted, so that the output voltage Vo can be adjusted.

  In FIG. 2, in order from the top, the waveform of the first control pulse signal Pa input to the first switching element Qa, the waveform of the second control pulse signal Pb input to the second switching element Qb, and the third switching element Qc The waveform of the input third control pulse signal Pc and the waveform of the fourth control pulse signal Pd input to the fourth switching element Qd are shown. A dead time is provided when the first switching element Qa and the second switching element Qb are inverted in phase and when the third switching element Qc and the fourth switching element Qd are inverted.

Subsequently, the waveform of the voltage V _T1 applied to the primary winding of the transformer T1 and the waveform of the current i _L1 flowing through the first resonance inductor L1 are shown. Hereinafter, the current waveform is drawn with a thick line. Subsequently, the waveform of the voltage V _Qa applied across the first switching element Qa and the waveform of the current i _Qa flowing through the first switching element Qa are shown. Subsequently, the waveform of the voltage V _Qb applied across the second switching element Qb and the waveform of the current i _Qb flowing through the second switching element Qb are shown. Subsequently, the waveform of the voltage V _Qc applied across the third switching element Qc and the waveform of the current i _Qc flowing through the third switching element Qc are shown. Finally, the waveform of the voltage V _Qd applied across the fourth switching element Qd and the waveform of the current i _Qd flowing through the fourth switching element Qd are shown.

  The initial state of the timing chart of FIG. 2 is a state in which the first switching element Qa and the third switching element Qc are off and the second switching element Qb and the fourth switching element Qd are on. In this state, a negative current flows through the first resonant inductor L1, a positive current flows through the second switching element Qb, and a negative current flows through the fourth switching element Qd. The first parasitic capacitance Ca and the third parasitic capacitance Cc are in a charged state.

  From this state, the phase of the first arm is reversed. At that time, a dead time dt1 is provided. That is, the dead time dt1 is inserted after the third switching element Qc is turned off, and the first switching element Qa is turned on after the third switching element Qc is turned off. During the dead time dt1, the current is maintained by the energy stored in the first resonant inductor L1, and a negative current flows through the first switching element Qa. Due to this negative current, the charge stored in the first parasitic capacitance Ca is discharged. Thereafter, since the first switching element Qa is turned on, switching loss due to the electric charge stored in the first parasitic capacitance Ca does not occur.

Since the switching loss Pc is defined by the following formula (1), the switching loss Pc can be reduced as the output capacitance Coss of the switching element is reduced.
Pc = 1/2 · Coss · V ² · f Formula (1)
V represents a voltage applied to the switching element, and f represents a switching frequency.

The zero voltage resonance condition can be defined by the following formula (2).
L · It ² > Coss · V ² Formula (2)
L represents the inductance of the resonant inductor, and It represents the current flowing through the resonant inductor.

  If the relationship of said Formula (2) is satisfy | filled, zero voltage resonance will be attained. Therefore, it is necessary to provide a resonant inductor having a sufficient size with respect to the output capacitance Coss of the switching element. The first resonant inductor L1 in FIG. 1 may be configured with a leakage inductance of the primary winding of the transformer T1. However, when the relationship of the above equation (2) is not satisfied, complete zero voltage switching cannot be realized. Therefore, in an application where the output capacity Coss of the switching element is increased, it is preferable to provide an inductor element separately from the primary winding.

  Returning to FIG. 2, when the phase of the first arm is reversed, the first switching element Qa and the fourth switching element Qd are turned on, and the second switching element Qb and the third switching element Qd are turned off. In this state, a positive voltage is applied to the primary winding of the transformer T1, a positive current flows through the first resonant inductor L1, a positive current flows through the first switching element Qa, and a positive current flows through the fourth switching element Qd. Yes. The second parasitic capacitor Cb and the third parasitic capacitor Cc are in a charged state.

  From this state, the phase of the second arm is reversed. At that time, a dead time dt2 is provided. That is, the dead time dt2 is inserted after the fourth switching element Qd is turned off, and the third switching element Qc is turned on after the end. During the dead time dt2, the current is maintained by the energy stored in the first resonant inductor L1, and a negative current flows through the third switching element Qc. The electric charge stored in the third parasitic capacitance Cc is discharged by this negative current. Thereafter, since the third switching element Qc is turned on, a switching loss due to the charge stored in the third parasitic capacitance Cc does not occur. According to the same principle, no switching loss occurs when the second switching element Qb is turned on and when the fourth switching element Qd is turned on. As described above, zero voltage switching of the full bridge circuit 10 is realized.

FIG. 3 is a timing chart for explaining an operation example 2 of the switching power supply apparatus 100 of FIG. The operation example 2 is a modification in which the same power as the operation example 1 is supplied to the transformer T1 using the first control pulse signal Pa to the fourth control pulse signal Pd different from the operation example 1. In the phase shift method according to the operation example 2, the duty ratio of the first control pulse signal Pa and the third control pulse signal Pc is fixed at 50%. The duty ratio of the second control pulse signal Pb and the fourth control pulse signal Pd is fixed at about 25%. The first switching element Qa and the third switching element Qc operate complementarily except for the dead time. The second switching element Qb and the fourth switching element Qd operate with a phase difference of 180 °. The rising phase of the first switching element Qa and the rising phase of the fourth switching element Qd are synchronized, and the rising phases of the second switching element Qb and the third switching element Qc are synchronized. Thereby, the voltage V _T1 applied to the primary winding having the same waveform as that of the phase shift method according to the operation example 1 can be generated.

Waveforms of voltages V _{Qa to} V _Qd applied to both ends of each of the first switching element Qa to the fourth switching element Qd, and currents i _{Qa to} i _Qd flowing in the first switching element Qa to the fourth switching element Qd, respectively. These waveforms are as shown in FIG. 3 and will not be described in detail. Since all of the first switching element Qa to the fourth switching element Qd are turned on in a state where the first parasitic capacitance Ca to the fourth parasitic capacitance Cd are discharged and no electric charge is stored, the zero of the full bridge circuit 10 is obtained. Voltage switching is realized.

  Although the switching power supply device 100 of the zero voltage switching system of FIG. 1 is a highly efficient switching power supply device as described above, there are problems described below. That is, a surge voltage is generated by resonance between the first resonant inductor L1, the first secondary parasitic capacitance Cr1, and the second secondary parasitic capacitance Cr2. If this surge voltage exceeds the withstand voltage of various elements in the switching power supply apparatus 100, there is a possibility that problems may occur in the various elements. In addition, the voltage and current supplied to the primary winding of the transformer T1 vibrate due to resonance, and power loss occurs.

The first secondary side parasitic capacitance Cr1 and the second secondary side parasitic capacitance Cr2 are converted to the primary side capacitance C0 by the following equation (3).
C0 = N ² · Cr (3)
N = N2 / N1
N1 represents the number of turns of the primary winding of the transformer T1, N2 represents the number of turns of the secondary winding, and Cr represents one capacitance of the first secondary side parasitic capacitance Cr1 and the second secondary side parasitic capacitance Cr2.

  Hereinafter, a mechanism for suppressing the influence of resonance between the first resonant inductor L1, the first secondary side parasitic capacitance Cr1, and the second secondary side parasitic capacitance Cr2 will be considered.

  FIG. 4 is a diagram illustrating a configuration example of the switching power supply apparatus 100 according to the comparative example. Compared with the switching power supply device 100 of FIG. 1, the switching power supply device 100 has a configuration in which a CR absorber is connected in parallel to each of the first rectifier diode Dr1 and the second rectifier diode Dr2. Specifically, a series circuit of the first protection resistor Ra1 and the first protection capacitor Ca1 is connected in parallel to the first rectifier diode Dr1. Similarly, a series circuit of the second protection resistor Ra2 and the second protection capacitor Ca2 is connected in parallel to the second rectifier diode Dr2. The first protection capacitor Ca1 and the second protection capacitor Ca2 absorb a high-frequency surge voltage.

  However, providing a CR absorber increases the circuit size. Moreover, it is insufficient to suppress power loss due to vibration.

  FIG. 5 is a diagram illustrating a configuration example of the switching power supply device 100 according to the embodiment of the present invention. Compared with the switching power supply device 100 of FIG. 1, the switching power supply device 100 has a configuration in which a second resonant inductor L2 and a first protection diode D1 to a fourth protection diode D4 are added to the primary side.

  The second resonant inductor L2 is connected between the output terminal of the second arm of the full bridge circuit 10 and the lower terminal of the primary winding of the transformer. The first protection diode D1 is connected between a first connection point between the first resonant inductor L1 and the upper terminal of the primary winding of the transformer T1 and the high voltage side power line of the DC power supply E. Specifically, the anode terminal of the first protection diode D1 is connected to the first connection point, and the cathode terminal is connected to the power line. The second protection diode D2 is connected between the first connection point and the low voltage side power line of the DC power supply E. Specifically, the cathode terminal of the second protection diode D2 is connected to the first connection point, and the anode terminal is connected to the power line.

  The third protection diode D3 is connected between the second connection point between the second resonant inductor L2 and the lower terminal of the primary winding of the transformer T1 and the power line on the high voltage side of the DC power supply E. Specifically, the anode terminal of the third protection diode D3 is connected to the second connection point, and the cathode terminal is connected to the power line. The fourth protection diode D4 is connected between the second connection point and the low voltage side power line of the DC power supply E. Specifically, the cathode terminal of the fourth protection diode D4 is connected to the second connection point, and the anode terminal is connected to the power line.

  Thereby, the upper limit voltage and the lower limit voltage of the two connection points can be clamped to the high voltage side power line and the low voltage side power line of the DC power supply E. That is, the surge voltage generated at the two connection points can be fed back to the DC power source E via the first protection diode D1 to the fourth protection diode D4.

FIGS. 6A to 6B are diagrams illustrating the waveform of the voltage V _T1 applied to the primary winding of the transformer T1. 6A shows a waveform in the switching power supply device 100 in FIG. 1 where the resonance countermeasures are not taken, and FIG. 6B shows a waveform in the switching power supply device 100 in which the resonance measures are taken in FIG. Show.

In FIG. 6A, the voltage V _T1 applied to the primary winding is oscillating due to the resonance between the first resonant inductor L1 and the first secondary parasitic capacitance Cr1 and the second secondary parasitic capacitance Cr2. The peak voltage exceeds the voltage range −V to + V of the DC power supply E.

The frequency f of this vibration is defined by the following formula (4), and the period T is defined by the following formula (5).
f = 1 / (2π√ (L1 · C0)) (4)
T = 1 / f Expression (5)

In FIG. 6B, the voltage V _T1 applied to the primary winding by the first protection diode D1 to the fourth protection diode D4 becomes the voltage of the power line on the high voltage side and the power line on the low voltage side of the DC power supply E. Clamped and vibration can be suppressed.

  As described above, according to the embodiment of the present invention, the diode is connected in the opposite direction between the connection point between the primary side resonant inductor and the primary winding of the transformer and the power supply voltage line. Thereby, it is possible to reduce the influence of resonance generated by the primary side resonance inductor and the secondary side capacitance component. Since the surge voltage due to the resonance can be suppressed, various elements in the switching power supply device 100 can be protected. Since it is not necessary to increase the breakdown voltage of various elements, it is possible to suppress the increase in size and cost of the elements. Further, since vibration due to the resonance can be suppressed, power loss due to vibration can be suppressed, and a decrease in conversion efficiency can be suppressed.

  The present invention has been described based on the embodiments. This embodiment is an exemplification, and it will be understood by those skilled in the art that various modifications can be made to combinations of the respective constituent elements and processing processes, and such modifications are also within the scope of the present invention. is there.

  FIG. 7 is a diagram illustrating a configuration example of the switching power supply device 100 according to the modification. The switching power supply device 100 according to the modification has a configuration in which the second resonant inductor L2, the third protection diode D3, and the fourth protection diode D4 are omitted from the switching power supply device 100 of FIG. Even if a protective diode is connected to only one terminal of the primary winding of the transformer T1, a certain resonance suppression effect can be obtained. Moreover, if it uses together with the secondary side CR absorber shown with the switching power supply apparatus 100 of FIG. 4, the resonance suppression effect can be improved more.

  In the above-described embodiment, the example in which the smoothing circuit 30 includes the first rectifier diode Dr1 and the second rectifier diode Dr2 has been described. However, a switching element such as a MOSFET may be used instead. In this case as well, resonance occurs between the parasitic capacitance of the switching element and the primary resonance inductor, but the influence of the resonance can be reduced by the first protection diode D1 to the fourth protection diode D4.

  Further, although an example in which a parasitic capacitance and a parasitic diode are formed in parallel with the switching element has been described, a capacitance element and / or a diode element may be connected in parallel with the switching element.

  Further, instead of the DC power supply E that supplies the input voltage to the full bridge circuit 10, an AC power supply, a rectifier circuit, and a PFC (Power Factor Correction) circuit may be used.

  Qa first switching element, Da first parasitic diode, Ca first parasitic capacitance, L1 first resonant inductor, T1 transformer, E DC power supply, Dr1 first rectifier diode, Cr1 first secondary side parasitic capacitance, Ra1 first protection Resistor, Ca1 first protection capacitor, D1 first protection diode, Qb second switching element, Db second parasitic diode, Cb second parasitic capacitance, L2 second resonant inductor, Dr2 second rectifier diode, Cr2 second secondary side Parasitic capacitance, Ra2 second protection resistor, Ca2 second protection capacitance, D2 second protection diode, Qc third switching element, Dc third parasitic diode, Cc third parasitic capacitance, D3 third protection diode, Qd fourth switching element Dd 4th parasitic diode, Co Output capacitance , Lo output inductor, Ro load, Cd fourth parasitic capacitance, D4 fourth protection diode, 10 full bridge circuit, 20 rectifier circuit, 30 smoothing circuit, 40 control unit, 50 DC-DC converter, 100 switching power supply device.

Claims (5)

A full bridge circuit in which a first arm including two switching elements and a second arm including two switching elements are connected in parallel;
A transformer for transforming the output power of the full bridge circuit;
A first inductor connected between the output terminal of the first arm and one terminal of the primary winding of the transformer;
A second inductor connected between the output terminal of the second arm and the other terminal of the primary winding of the transformer;
A rectifier circuit connected to the secondary winding of the transformer and rectifying the output power of the transformer;
A smoothing circuit that smoothes the power rectified by the rectifier circuit;
An anode terminal is connected to a first connection point between the first inductor and one terminal of the primary winding, and a cathode terminal is connected to a power line on a high voltage side of a power supply that supplies power to the full bridge circuit. A first diode to be
A second diode having a cathode terminal connected to the first connection point and an anode terminal connected to a power line on a low voltage side of the power source;
A third diode having an anode terminal connected to a second connection point between the second inductor and the other terminal of the primary winding, and a cathode terminal connected to a power line on the high voltage side of the power source;
A fourth diode having a cathode terminal connected to the second connection point and an anode terminal connected to a power line on a low voltage side of the power source;
A switching power supply device comprising: A control unit for controlling the full bridge circuit;
2. The switching power supply according to claim 1, wherein the control unit turns on the switching element after the parasitic capacitance of the switching element is discharged by a current based on the first inductor and the second inductor. apparatus. A full bridge circuit in which a first arm including two switching elements and a second arm including two switching elements are connected in parallel;
A transformer for transforming the output power of the full bridge circuit;
An inductor connected between the output terminal of the first arm and one terminal of the primary winding of the transformer;
A rectifier circuit connected to the secondary winding of the transformer and rectifying the output power of the transformer; and a smoothing circuit for smoothing the power rectified by the rectifier circuit;
An anode terminal is connected to a first connection point between the inductor and one terminal of the primary winding, and a cathode terminal is connected to a power line on a high voltage side of a power supply that supplies power to the full bridge circuit. A first diode;
A second diode having a cathode terminal connected to the first connection point and an anode terminal connected to a power line on a low voltage side of the power source;
A switching power supply device comprising: A control unit for controlling the full bridge circuit;
4. The switching power supply device according to claim 3, wherein the control unit turns on the switching element after the parasitic capacitance of the switching element is discharged by a current based on the inductor.   5. The switching power supply device according to claim 1, wherein the switching element is a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) or an IGBT (Insulated Gate Bipolar Transistor).

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