JP2014090643A - Controller of permanent magnet synchronous motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To eliminate the need for amplitude adjustment of high frequency voltage superposed on a synchronous motor for magnetic pole position detection.SOLUTION: A controller for controlling with a power converter by estimating the magnetic pole position of a permanent magnet synchronous motor includes: an adder-subtracter 22 for generating a γ-axis voltage command value by adding a γ-axis high frequency voltage command value to a γ-axis basic wave voltage command value; bandpass filters 33a, 33b for detecting a γ-axis high frequency current amplitude and a δ-axis high frequency current amplitude, respectively, from a γ-axis current detection value and a δ-axis current detection value: a current adjuster 31 for operating a γ-axis high frequency voltage amplitude command value so as to match the γ-axis high frequency current amplitude with a γ-axis high frequency current amplitude command value: a rectangular wave oscillator 32 for operating a γ-axis high frequency current voltage command value from a γ-axis high frequency voltage amplitude command value; an angular difference computing unit 34 for computing a magnetic pole position from the δ-axis high frequency current amplitude; a speed computing unit 35; and an integrator 36.

Description

  The present invention relates to a control device for calculating a magnetic pole position of a rotor of a permanent magnet type synchronous motor and driving the synchronous motor by a power converter.

Conventionally, a so-called sensorless control technique for calculating a magnetic pole position using the saliency of a rotor of a permanent magnet type synchronous motor (hereinafter referred to as IPMSM) having an embedded magnet structure and performing control based on the calculated value of the magnetic pole position Has been developed.
For example, Patent Document 1 discloses a technique for calculating a magnetic pole position by detecting a high-frequency current that flows when a high-frequency alternating voltage is applied to an electric motor.

  In the magnetic pole position calculation method using the saliency of the rotor such as Patent Document 1, since the N pole and the S pole of the rotor magnetic pole cannot be distinguished in principle, the magnetic pole position calculation value has an error of 180 °. Sometimes. Therefore, a technique for correcting the calculation error using the magnetic saturation characteristics of the motor core and accurately discriminating the N pole and the S pole of the rotor magnetic pole (hereinafter, this technique is referred to as “NS discrimination”) has been proposed. ing.

For example, in Patent Document 2, as shown in FIG. 6, a high-frequency square wave voltage is superimposed in the direction of the straight axis (synonymous with the d axis), and the polarity of the direct current is determined by the direct current command value i _d ^*. the direct axis when controlled positive and negative high frequency current _{i dh} switch 409, via the memory 410 incorporated into the pole discriminator 411 generates a preset signal by comparing the amplitude of _{i DHP,} magnitude relation of _{i dhn,} speed A technique for correcting the magnetic pole position calculation value obtained by the estimator 412 and the integrator 413 is described.
In FIG. 6, 50 is a three-phase AC power source, 70 is a power converter such as an inverter, 81 is a permanent magnet synchronous motor, 401 is a direct-axis current regulator, 402 is a horizontal-axis current regulator, and 403 is an adder. , 404 and 407 are coordinate converters, 405 is a PWM circuit, 406 is a current detector, 408 is a high-frequency separation filter, and the switch 409 is configured to be turned on when i _d ^* is positive or negative.

Further, in Patent Document 3, as shown in FIG. 7, a predetermined high-frequency component I _h sin ω _h · t is applied as an observation command to the d-axis current command value i _d ^* by the observation command application unit 501, and the d-axis current command value i _d ^* It is applied to the positive-negative symmetrical DC bias component I _b sized to the motor a constant value to the current command value i _d ^* for a predetermined period of time has magnetic saturation as the observation command, feedback corresponding to the observed current command value based on the amplitude of the high frequency component of the d-axis response voltage V _d which is calculated on the basis of the current i _d, the polarity judging unit 507 in the rotation detecting section 508 is described a technique of performing NS discrimination.
In FIG. 7, 502 is a current control unit, 503 and 505 are coordinate conversion units, 504 is a modulation unit, 506 is a position determination unit, and 509 is a bandpass filter.

Japanese Patent No. 331472 (paragraphs [0014] to [0040], FIG. 1 etc.) JP 2002-171798 A (claim 4, paragraphs [0033] to [0036], FIG. 4 etc.) JP 2011-205832 A (paragraphs [0020] to [0038], FIG. 2 etc.)

In order to realize the magnetic pole position calculation described in Patent Document 1 and Patent Document 2, it is necessary to adjust the high-frequency current to an appropriate magnitude. For this purpose, it is necessary to adjust the high-frequency voltage superimposed on the voltage command value to an appropriate magnitude, and this adjustment work is generally complicated. In addition, when the motor core is magnetically saturated, such as during heavy load or NS determination, the inductance is reduced and the high-frequency current may be excessive.
Also, have been in the prior art described in Patent Document 3, due to control error due to the delay of the current control system, it is difficult to increase the frequency omega _h of the high frequency component I _h sinω _h · t as the observation command When this frequency is low, a delay in the magnetic pole position calculation and a decrease in calculation accuracy become a problem.

  Therefore, the problem to be solved by the present invention is that it is not necessary to adjust the amplitude of the high-frequency voltage superimposed on the synchronous motor, and the high-frequency current can be controlled to a constant value even when the inductance decreases due to the magnetic saturation of the motor core. It is an object of the present invention to provide a control device that prevents an excessive increase. Furthermore, another problem to be solved by the present invention is to provide a control device capable of calculating the magnetic pole position with high accuracy.

In order to solve the above problems, an invention according to claim 1 is directed to driving the electric motor by a power converter using a magnetic pole position calculation value obtained by utilizing the saliency of a rotor of a permanent magnet type synchronous motor. A control device,
The current and voltage are regarded as vectors, and the γ-δ axis as the orthogonal rotational coordinate axis for control is set inside with respect to the dq axis composed of the d axis in the magnetic flux axis direction of the rotor and the q axis orthogonal thereto. In the control device for driving the electric motor by giving each phase voltage command value obtained from the γ-axis voltage command value and the δ-axis voltage command value to the power converter,
means for adding the γ-axis high-frequency voltage command value to the γ-axis fundamental wave voltage command value to generate the γ-axis voltage command value; means for detecting the γ-axis high-frequency current amplitude from the γ-axis current detection value; Means for detecting the δ-axis high-frequency current amplitude from the detected value; means for calculating the γ-axis high-frequency voltage amplitude command value so that the γ-axis high-frequency current amplitude matches the γ-axis high-frequency current amplitude command value; Means for calculating the γ-axis high-frequency voltage command value from the voltage amplitude command value, and magnetic pole position calculating means for calculating the magnetic pole position from the δ-axis high-frequency current amplitude.
Here, as described in claim 2, the magnetic pole position calculation means includes an angle difference calculation means for obtaining an angle difference between the γ-axis and the d-axis from the δ-axis high frequency current amplitude, Speed calculating means for obtaining an angular velocity of the γ-δ axis from a magnetic pole position calculating error; and integrating means for obtaining a magnetic pole position from the angular velocity.
As a result, even when the frequency of the γ-axis high-frequency voltage applied to the synchronous motor is high, the γ-axis high-frequency current can be controlled according to the command value, and the adjustment work of the amplitude of the γ-axis high-frequency voltage is unnecessary. become. Further, since the γ-axis high-frequency current can be controlled to a constant value even if the inductance of the synchronous motor varies, it is possible to prevent the current flowing through the synchronous motor from becoming excessive.

According to a third aspect of the present invention, in the control device for the permanent magnet type synchronous motor according to the first or second aspect, the means for calculating the γ-axis high-frequency voltage amplitude command value includes the γ-axis high-frequency current amplitude command value and the Integral adjusting means for adjusting the deviation from the γ-axis high-frequency current amplitude to be zero is provided.
Thereby, the γ-axis high-frequency current can be controlled to the command value by a relatively simple calculation.

According to a fourth aspect of the present invention, in the control device for the permanent magnet type synchronous motor according to the first or second aspect, the means for calculating the γ-axis high-frequency voltage amplitude command value includes the γ-axis high-frequency voltage amplitude command value and the parameter. Means for calculating a γ-axis high-frequency current amplitude estimated value from the estimated value, means for amplifying a deviation between the γ-axis high-frequency current amplitude estimated value and the γ-axis high-frequency current amplitude, and calculating the parameter estimated value; means for calculating the γ-axis high-frequency voltage amplitude command value from the γ-axis high-frequency current amplitude command value and the parameter estimation value.
Thereby, even when the inductance of the synchronous motor changes, the γ-axis high-frequency current can be controlled with high response.

The invention according to claim 5 is the control device according to any one of claims 1 to 4, wherein the means for controlling the value of the γ-axis current to be positive and negative, and the value of the γ-axis current are positive Means for calculating the magnetic pole position from the γ-axis high-frequency voltage amplitude and the γ-axis high-frequency voltage amplitude when the value of the γ-axis current is negative.
For example, as described in claim 6, means for controlling the γ-axis current command value to be positive and negative, and the γ-axis high-frequency voltage amplitude command value and the γ-axis current when the γ-axis current command value is controlled to be positive According to the result of comparing the γ-axis high-frequency voltage amplitude command value when the command value is controlled to be negative, the magnetic angle position calculation value at the previous calculation timing or the magnetic angle position calculation value at the previous calculation timing is 180 ° And means for correcting the magnetic pole position calculation value at the current calculation timing based on the value obtained by adding.

The invention according to claim 7 is the control device according to claim 4, wherein means for controlling the value of the γ-axis current to be positive and negative, the parameter estimated value when the value of the γ-axis current is positive, and γ Means for calculating the magnetic pole position from the parameter estimated value when the value of the axial current is negative.
For example, as described in claim 8, the means for controlling the γ-axis current command value to be positive and negative, and the parameter estimated value and the γ-axis current command value when the γ-axis current command value is controlled to be negative Depending on the result of comparison with the parameter estimation value when the current control is performed, the current calculation is performed using the magnetic pole position calculation value at the previous calculation timing or the value obtained by adding the electrical angle 180 ° to the magnetic pole position calculation value at the previous calculation timing. And means for correcting the magnetic pole position calculation value at the timing.
According to these inventions described in claims 5 to 8, NS discrimination can be realized with high accuracy by utilizing the magnetic saturation characteristics of the motor iron core.

  According to the present invention, it is not necessary to adjust the high frequency voltage applied to the synchronous motor, and even if the inductance of the synchronous motor changes, the high frequency current is controlled to a constant value to prevent the motor current from becoming excessive. At the same time, the magnetic pole position can be calculated with high accuracy.

It is the block diagram which showed the control apparatus which concerns on embodiment of this invention with the main circuit. It is explanatory drawing of a dq axis coordinate and a (gamma) -delta axis coordinate. It is a block diagram which shows 1st Example of the (gamma) -axis high frequency current regulator in FIG. It is a block diagram which shows 2nd Example of the (gamma) -axis high frequency current regulator in FIG. It is explanatory drawing of the principle of NS discrimination | determination in embodiment of this invention. It is a block diagram of the prior art described in patent document 2. FIG. It is a block diagram of the prior art described in patent document 3. FIG.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is a block diagram showing a control device according to this embodiment together with a main circuit. First, in the main circuit, 50 is a three-phase AC power source, 60 is a rectifier circuit, 70 is a power converter such as a three-phase voltage source inverter, and 80 is a permanent magnet type synchronous motor (IPMSM) having an embedded magnet structure.

On the other hand, in the control device, 11u and 11w are current detectors, 12 is a voltage detector, and the u-phase current detection value i _u and the w-phase current detection value i _w of the synchronous motor 80 are input to the coordinate converter 14, The detected DC voltage value E _dc of the power converter 70 is input to the PWM circuit 13.
Further, a deviation between the angular velocity command value ω _r ^* and the angular velocity calculation value ω ₁ is obtained by the adder / subtractor 16, and the torque command value τ ^* is calculated by the speed regulator 17 so that this deviation becomes zero. The current command calculator 18 calculates a γ-axis current command value i _γ ^* and a δ-axis current command value i _δ ^* for generating torque according to the torque command value τ ^* .

The γ-axis current regulator 20a performs an adjustment operation so that the deviation between the γ-axis current command value i _γ ^* obtained by the adder / subtractor 19a and the detected γ-axis fundamental wave current value i _γf becomes zero, and the γ-axis fundamental wave voltage is adjusted. The command value v _γf ^* is calculated, and the δ-axis current controller 20b is set so that the deviation between the δ-axis current command value i _δ ^* obtained by the adder / subtractor 19b and the detected δ-axis fundamental wave current value i _δf becomes zero. The adjustment operation is performed to calculate the δ-axis fundamental wave voltage command value v _δf ^* .
Here, the γ-axis fundamental wave current detection value i _γf and the δ-axis fundamental wave current detection value i _δf are converted from the γ-axis current detection value i _γ and the δ-axis current detection value i _δ by the notch filters 21a and 21b. Each is required to be removed. The γ-axis current detection value i _γ and the δ-axis current detection value i _δ are obtained by inputting the u-phase and w-phase current detection values i _u and i _w to the coordinate converter 14 and using the magnetic pole position calculation value θ ₁ . Obtained by coordinate conversion.

The band pass filter 33a calculates the γ-axis high-frequency current amplitude I _γh having the same frequency component as the γ-axis high-frequency voltage from the γ-axis current detection value i _γ , and similarly, the band-pass filter 33b calculates the δ-axis current detection value i _δ. To δ-axis high-frequency current amplitude I _δh having the same frequency component as the γ-axis high-frequency voltage.
The γ-axis high-frequency current regulator 31 controls the γ-axis high-frequency voltage amplitude command value V _γh ^* so that the γ-axis high-frequency current amplitude I _γh matches the γ-axis high-frequency current amplitude command value I _γh ^* . A specific configuration of the γ-axis high-frequency current regulator 31 will be described later.
The rectangular wave oscillator 32 generates and outputs a rectangular wave γ-axis high-frequency voltage command value v _γh ^* whose amplitude is controlled by the γ-axis high-frequency voltage amplitude command value V _γh ^* .

Further, the adder / subtracter 22 calculates the γ-axis voltage command value v _γ ^* by superimposing the γ-axis high-frequency voltage command value v _γh ^* output from the rectangular wave oscillator 32 on the γ-axis fundamental wave voltage command value v _γf ^*. The γ-axis voltage command value v _γ ^* is input to the coordinate converter 15. On the other hand, the δ-axis fundamental wave voltage command value v _δf ^* output from the δ-axis current regulator 20b is directly input to the coordinate converter 15 as the δ-axis voltage command value v _δ ^* .

In the coordinate converter 15, the γ-axis voltage command value v _γ ^* and the δ-axis voltage command value v _δ ^* are coordinate-converted using the magnetic pole position calculation value θ ₁ , and the three-phase phase voltage command values v _u ^* and v _v are converted. ^{* And} _vw ^* are calculated.
In the PWM circuit 13 to which the phase voltage command values v _u ^* , v _v ^* , v _w ^* and the DC voltage detection value E _dc are input, the output voltage of the power converter 70 is used as the phase voltage command values v _u ^* , v. _{v ^*,} v _w ^* gate signal for controlling the is generated, the gate signal using by controlling the semiconductor switching elements of the power converter 70, the synchronous motor terminal voltage is the phase voltage command values 80 v _u ^* , V _v ^* , and v _w ^* are controlled.
With the above operation, the rotor speed of the synchronous motor 80 is controlled to the angular speed command value ω _r ^* , and the γ-axis high-frequency current amplitude I _γh for calculating the magnetic pole position and speed of the rotor is set to the amplitude command value I _γh ^*. It is possible to control.

Now, in a synchronous motor such as IPMSM, high-performance torque control and speed control can be realized by controlling based on orthogonal rotation coordinates (dq axis coordinates) synchronized with a rotor. Here, for the dq axis, the direction of the magnetic flux axis of the rotor is defined as the d axis, and the axis in the 90 ° advance direction perpendicular to the d axis is defined as the q axis.
However, in the case of sensorless control in which the synchronous motor is operated without using the magnetic pole position detector, the position of the dq axis cannot be directly detected. Therefore, a method is known in which a γ-δ axis that is an estimated axis of the dq axis is defined, and a control calculation is performed on the γ-δ axis coordinates.

FIG. 2 is an explanatory diagram of the dq axis coordinates and the γ-δ axis coordinates, and shows the angle of the γ axis (magnetic pole position calculation value) θ ₁ and the u phase winding with respect to the u phase winding of the synchronous motor. An angle difference (magnetic pole position calculation error) θ _err with respect to the reference d-axis angle (magnetic pole position) θ _r is defined by Equation 1.
[Equation 1]
θ _err = θ ₁ −θ _r
Further, the angular velocity of the dq axis is defined as ω _r (rotor speed), and the angular velocity (speed calculation value) of the γ-δ axis is defined as ω ₁ .

Next, the principle of position / velocity calculation in this embodiment will be described.
The equation of state for the high-frequency component of the synchronous motor is expressed by Equation 2 when the armature resistance is approximated to zero and the speed is approximated to zero.

Now, the γ-axis and δ-axis high-frequency current amplitudes I _γh and I _δh when the rectangular high-frequency alternating voltage is superimposed on the γ-axis are obtained by integrating the state equation of Equation 2 over ½ period of the high-frequency alternating voltage. , Is derived as in Equation 3.

According to Equation 3, γ and δ-axis high-frequency current amplitudes I _γh and I _δh are functions of the magnetic pole position calculation error θ _err . Here, when the magnetic pole position calculation error θ _err is close to zero, the δ-axis high-frequency current amplitude I _δh can be approximated as Equation 4.

According to Equation 4, the δ-axis high frequency current amplitude I _δh is proportional to the magnetic pole position calculation error θ _err . Therefore, it is possible to obtain the magnetic pole position calculation error θ _err from the δ-axis high-frequency current amplitude I _δh and correct the magnetic pole position calculation value using the magnetic pole position calculation error θ _err .

A position / speed calculation method according to the above principle will be described with reference to FIG.
The angle difference calculator 34 in FIG. 1 uses the δ-axis high-frequency current amplitude I _δh of Equation 4 to obtain the position calculation error calculation value (−θ _errest ) according to Equation 5.

The speed calculator 35 calculates the velocity calculation value omega ₁ from the position calculation error calculation value _(-θ errest), the integrator 36 calculates the magnetic pole position calculation value theta ₁ by integrating the velocity calculation value omega _1.
By these calculations, the speed calculation value ω ₁ and the magnetic pole position calculation value θ ₁ are obtained so that the magnetic pole position calculation error θ _err becomes zero, and these values can be converged to true values.

  Next, an embodiment of the γ-axis high frequency current regulator 31 in FIG. 1 will be described. FIG. 3 shows a γ-axis high-frequency current regulator 311 according to the first embodiment, and FIG. 4 shows a γ-axis high-frequency current regulator 312 according to the second embodiment.

In the γ-axis high-frequency current regulator 311 shown in FIG. 3, a deviation between the γ-axis high-frequency current amplitude command value I _γh ^* and the γ-axis high-frequency current amplitude I _γh is obtained by the adder / subtractor 101 and is integrated so as to make this deviation zero. The regulator 102 operates to calculate the γ-axis high-frequency voltage amplitude command value V _γh ^* . The γ-axis high-frequency voltage amplitude command value V _γh ^* is given to the power converter 70 as the γ-axis voltage command value v _γ ^* via the rectangular wave oscillator 32 and the adder / subtractor 22 shown in FIG. 1, and the synchronous motor 80 is driven. together, the synchronous motor 80 current frequency gamma-axis detected from the current amplitude I _{y H} in is to be fed back to the adder-subtracter 101, the axis RF current amplitude I _{y H} gamma by these series of operations are gamma-axis high frequency voltage amplitude command value V It is controlled in proportion to _γh ^* .
Thus, gamma since feedback control system gamma-axis high frequency current amplitude I _{y H} by axial high frequency current regulator 311 is configured, it is possible to control the gamma-axis high frequency current amplitude I _{y H} to the command value I _{y H} ^*. Since the γ-axis high-frequency current amplitude command value I _γh ^* is a DC amount, the γ-axis high-frequency current amplitude I _γh is the command value I _γh even when the frequency of the superimposed high-frequency voltage is high and the response of the current control system is slow _. Converge to ^* .

Here, in the γ-axis high-frequency current regulator 311 in FIG. 3, in order to increase the responsiveness of the γ-axis high-frequency current amplitude I _γh , the integral regulator 102 needs to be optimally adjusted. The inductance value of the synchronous motor 80 is required. For this reason, when the inductance value is unknown or when the inductance value fluctuates due to magnetic saturation of the motor core, the response of the γ-axis high-frequency current amplitude I _γh becomes slow, or the control system becomes unstable. There are concerns.

Therefore, in the γ-axis high-frequency current regulator 312 of the second embodiment shown in FIG. 4, by applying adaptive control, the response of the γ-axis high-frequency current amplitude I _γh even when the inductance value of the synchronous motor 80 is unknown. It is intended to improve.
In FIG. 4, gamma-axis high frequency current amplitude command value I _{y H} ^* is input to the divider 201, gamma-axis high frequency current amplitude command value I _{y H} ^* parameters division to gamma-axis high-frequency voltage amplitude command by the estimation value theta _1Est The value V _γh ^* is determined. This γ-axis high-frequency voltage amplitude command value V _γh ^* (= ζ ₁ ) is multiplied by the parameter estimated value Θ _1est by the multiplier 202 to _obtain a γ-axis high-frequency current amplitude estimated value I _γhest .

γ-axis high frequency current deviation between the estimated amplitude value I _Ganmahest and γ-axis high frequency current amplitude I _{y H} epsilon is calculated by the adder-subtracter 203, the deviation epsilon, together with γ-axis high frequency voltage amplitude command value V _{γh *} ⁽⁼ ζ ₁₎ Input to the parameter estimator 204.
The parameter estimator 204 amplifies the deviation ε according to Equation 6 and calculates the parameter estimated value Θ _1est .

As a result of the above calculation, the parameter estimated value Θ _1est converges to a true value by calculating so that the deviation ε becomes zero. By using this parameter estimated value Θ _1est , it becomes possible to output the γ-axis high-frequency voltage amplitude command value V _γh ^* for controlling the γ-axis high-frequency current amplitude I _γh to the command value I _γh ^* .

Next, the NS discrimination method in this embodiment will be described.
From Equation 3, since the δ-axis high-frequency current amplitude I _δh is a function that is twice the position calculation error θ _err , the position calculation value θ ₁ in the block diagram of FIG. And the γ-axis may face the S-pole direction instead of the N-pole direction of the rotor. Therefore, the N pole and the S pole are discriminated using the magnetic saturation characteristics of the motor iron core.

FIG. 5 is a diagram showing the principle of NS discrimination. When the d-axis current is controlled to be “positive”, the magnetic flux generated by the permanent magnet and the magnetic flux generated by the current are combined to increase the flux linkage. As a result, the d-axis inductance decreases due to the magnetic saturation characteristics of the electric motor iron core. On the other hand, when the d-axis current is controlled to be “negative”, the magnetic flux generated by the permanent magnet and the magnetic flux generated by the current cancel each other, so the interlinkage magnetic flux decreases. As a result, the d-axis inductance increases.
Using this, after the position calculation value θ ₁ converges, the γ-axis high-frequency voltage when the γ-axis current command value i _γ ^* is controlled to be “positive” using the γ-axis high-frequency current regulator 31 of FIG. Using the amplitude command value V _γhP ^* and the γ-axis high-frequency voltage amplitude command value V _γhN ^* when the γ-axis current command value i _γ ^* is controlled to be “negative”, the N pole and S pole of the rotor magnetic pole are Determine. Specifically, a magnetic pole position calculation value θ _{1 in} which the magnetic pole position calculation error θ _err is corrected is obtained by performing the calculation of Expression 7. (NS discrimination is performed).

[Equation 7]
i) When _VγhP ^* ≦ _VγhN ^*
θ ₁ = θ _{1 (previous value)}
ii) When _VγhP ^* > _VγhN ^*
θ ₁ = θ _{1 (previous value)} + π

数式８より、パラメータ推定値Θ_１ｅｓｔはｄ軸インダクタンスＬ_ｄに反比例する。 Next, another NS determination method in the present embodiment will be described.
In this method, NS discrimination is performed using the parameter estimated value Θ _1est in the γ-axis high-frequency current regulator 312 shown in FIG.
When the magnetic pole position calculation error θ _err is zero or 180 °, the parameter estimated value Θ _1est is expressed by Equation 8.

From Equation 8, the parameter estimated value Θ _1est is inversely proportional to the d-axis inductance L _d .

Therefore, using the magnetic saturation characteristics of the motor iron core, after the position calculation value θ ₁ converges, the γ-axis current command value i _γ ^* is controlled to be “positive” using the γ-axis high-frequency current regulator 312 of FIG. and parameter estimates theta _1EstP upon the gamma-axis current value i _gamma ^* using the parameter estimates theta _1EstN when controlled to "negative", N pole of the rotor poles, to determine the S pole. Specifically, the calculation of Formula 9 is performed to obtain a magnetic pole position calculation value θ _{1 in} which the magnetic pole position calculation error θ _err is corrected (NS discrimination is performed).
[Equation 9]
i) When Θ _1estN ≦ Θ _1estP
θ ₁ = θ _{1 (previous value)}
ii) When Θ _1estN > Θ _1estP
θ ₁ = θ _{1 (previous value)} + π

According to the present embodiment described above, the magnetic pole position and speed of the synchronous motor can be accurately estimated without using the magnetic pole position detector, and current control is performed based on these estimated magnetic pole position and speed. Thus, the speed and torque of the synchronous motor can be controlled with high accuracy.
In the embodiment of the present invention, the case where the γ-axis high-frequency voltage command value v _γh ^* is a rectangular wave has been described, but a sine wave may be used.

50 Three-phase AC power supply 60 Rectifier circuit 70 Power converter 80 IPMSM
11u u phase current detector 11w w phase current detector 12 voltage detector 13 PWM circuit 14, 15 coordinate converters 16, 19a, 19b, 22 adder / subtractor 17 speed regulator 18 current command calculator 20a γ-axis current regulator 20b δ-axis current regulator 21a, 21b Notch filter 31, 311, 312 γ-axis high-frequency current regulator 32 Rectangular wave oscillator 33a, 33b Bandpass filter 34 Angle difference calculator 35 Speed calculator 36 Integrator 101, 203 Adder / subtractor 102 Integral Controller 201 Divider 202 Multiplier 204 Parameter estimator

Claims (8)

A control device for driving the electric motor by a power converter using a magnetic pole position calculation value obtained by using the saliency of the rotor of the permanent magnet type synchronous motor,
The current and voltage are regarded as vectors, and the γ-δ axis as the orthogonal rotational coordinate axis for control is set inside with respect to the dq axis composed of the d axis in the magnetic flux axis direction of the rotor and the q axis orthogonal thereto. In the control device for driving the electric motor by giving each phase voltage command value obtained from the γ-axis voltage command value and the δ-axis voltage command value to the power converter,
means for adding the γ-axis high-frequency voltage command value to the γ-axis fundamental voltage command value to generate the γ-axis voltage command value;
means for detecting the γ-axis high-frequency current amplitude from the γ-axis current detection value;
means for detecting the δ-axis high-frequency current amplitude from the δ-axis current detection value;
means for calculating a γ-axis high-frequency voltage amplitude command value so that the γ-axis high-frequency current amplitude matches the γ-axis high-frequency current amplitude command value;
Means for calculating the γ-axis high-frequency voltage command value from the γ-axis high-frequency voltage amplitude command value;
Magnetic pole position calculating means for calculating the magnetic pole position from the δ-axis high-frequency current amplitude;
A control device for a permanent magnet type synchronous motor. In the control device for the permanent magnet type synchronous motor according to claim 1,
The magnetic pole position calculation means includes
An angle difference calculating means for obtaining an angle difference between the γ-axis and the d-axis from the δ-axis high-frequency current amplitude, and a speed calculating means for obtaining an angular velocity of the γ-δ axis from a magnetic pole position calculation error as the angle difference; And a control unit for the permanent magnet type synchronous motor, comprising: integrating means for obtaining a magnetic pole position from the angular velocity. In the control device for a permanent magnet type synchronous motor according to claim 1 or 2,
The means for calculating the γ-axis high-frequency voltage amplitude command value is:
A control device for a permanent magnet type synchronous motor, characterized by comprising integral adjusting means for adjusting so that a deviation between the γ-axis high-frequency current amplitude command value and the γ-axis high-frequency current amplitude becomes zero. In the control device for a permanent magnet type synchronous motor according to claim 1 or 2,
The means for calculating the γ-axis high-frequency voltage amplitude command value is:
Means for calculating a γ-axis high-frequency current amplitude estimated value from the γ-axis high-frequency voltage amplitude command value and the parameter estimated value;
Means for amplifying a deviation between the γ-axis high-frequency current amplitude estimated value and the γ-axis high-frequency current amplitude to calculate the parameter estimated value;
Means for calculating the γ-axis high-frequency voltage amplitude command value from the γ-axis high-frequency current amplitude command value and the parameter estimation value;
A control device for a permanent magnet type synchronous motor. In the control device for the permanent magnet type synchronous motor according to any one of claims 1 to 4,
means for controlling the value of the γ-axis current to be positive and negative;
means for calculating the magnetic pole position from the γ-axis high-frequency voltage amplitude when the value of the γ-axis current is positive and the γ-axis high-frequency voltage amplitude when the value of the γ-axis current is negative;
A control device for a permanent magnet type synchronous motor. In the control device for the permanent magnet type synchronous motor according to claim 5,
means for positively and negatively controlling the γ-axis current command value;
According to the result of comparing the γ-axis high-frequency voltage amplitude command value when the γ-axis current command value is controlled positively and the γ-axis high-frequency voltage amplitude command value when the γ-axis current command value is controlled negatively, Means for correcting the magnetic pole position calculation value at the current calculation timing by a magnetic pole position calculation value at the previous calculation timing or a value obtained by adding an electrical angle of 180 ° to the magnetic pole position calculation value at the previous calculation timing;
A control device for a permanent magnet type synchronous motor. In the control apparatus for the permanent magnet type synchronous motor according to claim 4,
means for controlling the value of the γ-axis current to be positive and negative;
means for calculating the magnetic pole position from the parameter estimated value when the value of the γ-axis current is positive and the parameter estimated value when the value of the γ-axis current is negative;
A control device for a permanent magnet type synchronous motor. In the control device for the permanent magnet type synchronous motor according to claim 7,
means for positively and negatively controlling the γ-axis current command value;
According to the result of comparing the parameter estimated value when the γ-axis current command value is controlled positively and the parameter estimated value when the γ-axis current command value is controlled negatively, the magnetic pole position calculated value at the previous calculation timing Or means for correcting the magnetic pole position calculation value at the current calculation timing with a value obtained by adding an electrical angle of 180 ° to the magnetic pole position calculation value at the previous calculation timing;
A control device for a permanent magnet type synchronous motor.

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