JP2013251938A - Switching power supply and electronic apparatus including the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a highly efficient and compact switching power supply by an insulated converter of single-stage configuration combining a power factor correction (PFC) function, an output stabilization function and an instantaneous power failure compensation function.SOLUTION: A capacitor 4 for instantaneous power failure compensation is provided on the DC side of a diode bridge 2 rectifying an AC input power, and subjected to step-up charging with power from the tertiary winding N3 of an insulated transformer 9 when correcting the power factor by controlling a switching element 5a during non-power failure. The drive frequency of the switching element 5a when delivering energy stored in the capacitor 4, by detecting power failure, to the insulated secondary via the insulated transformer 9 is set larger than the drive frequency during non-power failure depending on the step-up rate of the capacitor 4.

Description

  The present invention relates to a switching power supply that obtains a DC output insulated from an AC input and improves the power factor of the AC input.

  In order to obtain a DC output by performing rectification and smoothing from a commercial AC power supply (AC power), a configuration using a diode bridge and a smoothing capacitor is the simplest. However, this configuration results in a so-called capacitor input type rectifier circuit in which the input current flows only near the peak of the power supply voltage, resulting in a decrease in power factor and an increase in input harmonics. The problem of input harmonics is regulated by international standards, and measures corresponding to the input power are required. In response to this movement, various circuit type converters called power factor correction (PFC) converters or high power factor converters have been proposed.

  Of these, the most common circuit system is a circuit system called a step-up PFC converter. This is because a series pair of a coil and a switch is connected between the positive side and the negative side of a diode bridge that rectifies alternating current, and the anode side of the boost diode is connected to the connection point between the coil and the switch. In this circuit, the cathode side is connected to the high voltage side of the output smoothing capacitor, and the low voltage side of the output smoothing capacitor is connected to the negative side of the diode bridge. However, since the PFC converter of this circuit system does not have an insulation function and is a step-up type, in order to obtain a voltage such as DC 24V or 12V, an insulation type DC-DC (with an insulation transformer in the subsequent stage of the PFC converter) Direct current to direct current) converter must be connected and converted to the desired DC voltage. In the conventional two-stage configuration of such a converter, since the two conversion circuits are passed before the DC voltage is obtained, the total conversion efficiency is low, and there is a problem in terms of energy saving.

  On the other hand, Patent Document 1 discloses a switching power supply in which a single-stage converter has a power factor improvement function, an insulation function, and an output stabilization function, and also has an instantaneous power failure compensation function. ing. This switching power supply is provided with charge storage means for compensating for instantaneous interruption between the DC output side of the rectifier diode bridge and the primary side of the isolation transformer, and is based on an active clamp type flyback converter. . This achieves power factor improvement by controlling the switching element at high speed and operating the isolation transformer in a discontinuous mode, and there are several types of circuit systems for isolated converters with a power factor improvement function. Proposed.

  Further, Patent Document 1 proposes an operation for suppressing the capacitance of the capacitor by boosting the charge storage means for instantaneous power failure compensation, but details of the driving frequency of the switching element at that time have been clarified. Absent. That is, if the charge storage means is boosted and charged, the current flowing from the charge storage means to the insulation transformer during compensation for instantaneous power failure may become excessive, but no countermeasure is shown.

JP 2010-178521 A

  In the one-stage power factor correction converter, the input power becomes a waveform correlated with the input voltage waveform due to the influence of the power factor correction control, and is sent as it is to the secondary side of the isolation transformer that is the output. On the other hand, the power required by the load is not related to the input waveform. For example, if the maximum output continues on the load side, the difference between the input and output power will be reflected in the output voltage, and output voltage ripple is inevitable. . To reduce this, the capacity of the output capacitor may be increased, but there is a limit to increasing the capacity in consideration of the function and cost. Further, when observing the input power characteristics, although it is a short time, there is a period in which the input power is almost zero before and after the so-called zero cross where the input AC voltage becomes 0 V, which causes the output voltage to decrease. Due to the nonuniformity of the input power characteristics and the independence from the output characteristics, it is difficult in principle to completely suppress the output voltage ripple in the above-described power factor improving converter having a single-stage configuration. There was no choice but to cope with it within the allowable ripple by increasing the capacity.

In addition, in the power factor improvement converter having a single-stage configuration described in Patent Document 1, a circuit configuration for boosting the instantaneous power loss compensation capacity has been proposed as a measure for reducing the capacitance of the capacitor provided on the primary side for instantaneous power failure compensation. . In the proposed active clamp operation, the driving frequency of the switching element Q1 that controls the input current of the isolation transformer is constant, and the input power is controlled by changing the time ratio of the ON period. That is, the current IQ1 of the switching element Q1 is determined by the input voltage Vin, the inductance L1 on the primary side of the isolation transformer, and the on-time t.
IQ1 = Vin / L1 · t
It will be represented by Therefore, if the applied voltage of the capacitor for momentary power loss compensation is significantly increased, the input current during momentary power loss compensation operation will be excessive, causing damage to the element or causing magnetic saturation of the insulation transformer. There is a high possibility of causing a malfunction. As a workaround, it is conceivable to use a part with a higher withstand voltage or to increase the core capacity of the insulation transformer for the compensation operation at the momentary power failure. As a control measure, it is possible to reduce the on-time ratio by reducing the on-time ratio in order to suppress the current level. However, if the remaining period after reducing the on-time is set as the dead time, the floating potential is obtained. Since the potential of the contact vibrates and the low-loss switching condition is lost, there arises a problem that the efficiency is lowered.

  The problem that arises when using the energy of the capacitor for instantaneous power failure compensation as described above is not limited to the flyback converter of the active clamp method, but the power source for the instantaneous power failure compensation is applied to the primary side of the switching power supply using the current resonance method. This is a common problem when a capacitor is provided.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to provide a high-efficiency and small-sized switching power supply having both an output stabilization function and an instantaneous power failure compensation function.

  In order to achieve the above object, a switching power supply according to the present invention includes a rectifier for rectifying input AC power, an insulating transformer having at least a primary winding, a secondary winding, and a tertiary winding. A smoothing means connected to the secondary winding for smoothing and outputting the DC power insulated from the AC power, and connected between the DC side terminal of the rectifying means and the primary winding. A charge storage connected to a DC side terminal of the rectifying means, and a power factor improvement control means for controlling the first switching element to improve the power factor of the input AC power. And a third switching element connected in series to the charge storage means so as to prevent discharge of the charge storage means, and a charging circuit that boosts and charges the charge storage means with power from the tertiary winding; Stop AC power When the power failure detection means detects the AC power failure, the third switching element is controlled so that the charge previously stored in the charge storage means is passed through the insulation transformer. The first switching element is driven at a frequency substantially equal to the normal driving frequency when the AC power failure is not detected (charging of the charge storage means). Charge / discharge control means for raising the voltage at completion / the maximum voltage of the alternating current power) to more than twice.

  According to the present invention, it is possible to provide a highly efficient and small switching power supply having both an output stabilization function and an instantaneous power failure compensation function.

It is a figure which shows the operation | movement waveform of the voltage of the main element of the switching power supply of 1st Embodiment, an electric current, and electric power. It is a circuit diagram which shows the structure of the switching power supply of 1st Embodiment. It is a block diagram which shows schematic structure of the control circuit of the switching power supply of 1st Embodiment. It is a block diagram which shows the input / output signal of the calculator of the switching power supply of 1st Embodiment. It is a figure which shows the electric current path | route when Q1 is an ON state in the normal time operation | movement of the switching power supply of 1st Embodiment. It is a figure which shows the electric current path | route when Q1 is a turn-off state in the normal time operation | movement of the switching power supply of 1st Embodiment. It is a figure which shows the electric current path | route when Q2 and Q4 are in an ON state in the normal time operation | movement of the switching power supply of 1st Embodiment. It is a figure which shows the electric current path | route when Q2 and Q4 are a turn-off state in the normal time operation | movement of the switching power supply of 1st Embodiment. It is a figure which shows the operation | movement waveform of the normal time operation | movement of the switching power supply of 1st Embodiment. It is a figure which shows transition of a gate control signal, a voltage, and an electric current fluctuation | variation when the operation state of the switching power supply of 2nd Embodiment switches from the period A to the period B. It is a figure which shows transition of a gate control signal, a voltage, and an electric current change when the operation state of the switching power supply of 2nd Embodiment switches from the period B to the period A. It is a figure which shows the operation | movement waveform of the voltage of the main element of the switching power supply of 3rd Embodiment, an electric current, and electric power. It is a circuit diagram which shows the structure of the switching power supply of 3rd Embodiment. It is a figure which shows the electric current path | route when Q1 is an ON state in the normal time operation | movement of the switching power supply of 3rd Embodiment. It is a figure which shows the electric current path | route when Q1 is a turn-off state in the normal time operation | movement of the switching power supply of 3rd Embodiment. It is a figure which shows the electric current path | route when Q2 is an ON state in the operation | movement at the time of the steady state of the switching power supply of 3rd Embodiment. It is a figure which shows the electric current path | route when Q2 is a turn-off state in the normal time operation | movement of the switching power supply of 3rd Embodiment. It is a figure which shows the operation | movement waveform of the normal time operation | movement of the switching power supply of 3rd Embodiment. It is a figure which shows transition of a gate control signal, a voltage, and an electric current fluctuation | variation when the operation state of the switching power supply of 4th Embodiment switches from the period A to the period B. It is a figure which shows transition of a gate control signal, a voltage, and an electric current change when the operation state of the switching power supply of 4th Embodiment switches from the period B to the period A. It is the mounting figure of the board | substrate of the switching power supply of 5th Embodiment. It is a figure which shows the example which mounted | wore the electronic device with the board | substrate of the switching power supply of 5th Embodiment. It is a figure which shows the operation | movement waveform of the voltage of the main element of the conventional switching power supply, an electric current, and electric power.

  Hereinafter, modes for carrying out the present invention (hereinafter referred to as the present embodiment) will be described with reference to the drawings as appropriate. However, the embodiment of the present invention is not limited to the present embodiment, and the present invention is not limited thereto. Various modifications can be made without departing from the spirit of the present invention.

  In the following embodiment, an example will be described in which the switching power supply according to the present invention has a thickness of 10 mm or less, thereby contributing to thinning of the set thickness of a liquid crystal television or a plasma television incorporating the switching power supply.

[First Embodiment]
1st Embodiment of this invention is described using FIGS. 1-6 and FIG.

≪Circuit diagram≫
FIG. 2 is a circuit diagram showing a configuration of the switching power supply according to the first embodiment of the present invention. In FIG. 2, the AC power supplied from the AC power source 1 is full-wave rectified by the diode bridge 2, and the output voltage of the diode bridge 2 becomes the full-wave rectified voltage Vac shown in FIG. An input capacitor 16 (Cin) is connected between the positive and negative terminals on the DC side of the diode bridge 2. The input capacitor 16 is for an input filter and has a capacitance of several μF. An input current Isns between the diode bridge 2 and the input capacitor 16 is measured by the current measuring instrument 14.

  A series pair of a power MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) 5c (Q3) and an instantaneous power failure compensation capacitor 4 (Ctk) is connected in parallel to the diode bridge 2 and the input capacitor 16. The power MOSFET 5c is an N-channel power MOSFET. The source is connected to the positive electrode side of the input capacitor 16 and the drain is connected to the instantaneous power failure compensation capacitor 4, so that the instantaneous power failure compensation capacitor 4 is prevented from discharging. The instantaneous power failure compensation capacitor 4 is a capacitor for instantaneous power failure compensation, and its capacity varies depending on the instantaneous power failure compensation time, but is assumed to be about 100 μF to 1000 μF.

  A series pair of the primary winding N1 of the insulating transformer 9 (Tr) and the power MOSFET 5a (Q1) is connected in parallel with the input capacitor 16. At this time, the winding start (the black circle side in FIG. 2) of the primary winding N1 is connected to the positive electrode side of the input capacitor 16. Further, a series pair of a capacitor 8 (Cc) and a power MOSFET 5b (Q2) is connected to both ends of the primary winding N1 of the insulating transformer 9. At this time, the drain of the power MOSFET 5a and the source of the power MOSFET 5b are connected.

  The anode of the diode 10a (D1) is connected to the winding end side of the secondary winding N2 of the insulating transformer 9, and the output smoothing capacitor 11 (Co) is connected between the cathode of the diode 10a and the winding start of the secondary winding N2. Connected. A load 12 is connected in parallel to the output smoothing capacitor 11.

  Furthermore, a charging circuit 15 for charging the instantaneous power failure compensation capacitor 4 is connected to the tertiary winding N3 of the isolation transformer 9, and a current suction port among the negative electrode side of the instantaneous power failure compensation capacitor 4 and the output terminal of the charging circuit 15 is connected. The power MOSFET 5d (Q4) is connected to the corresponding terminal. At this time, the drain of the power MOSFET 5d is connected to the negative electrode side of the instantaneous power failure compensation capacitor 4, and the source is connected to the charging circuit 15 side. The power MOSFET 5 d is an N-channel power MOSFET, and controls the charging current to the instantaneous power failure compensation capacitor 4 by the charging circuit 15.

  In the charging circuit 15, the anode of the diode 10c is connected to the winding end side of the tertiary winding N3, the smoothing capacitor 48 is connected between the cathode and the winding start side of the tertiary winding N3, and the diode 10c. The coil 49 is connected between the cathode of the power supply and the positive electrode side of the instantaneous power failure compensation capacitor 4, and the winding start side of the tertiary winding N3 is connected to the source of the power MOSFET 5d. Thus, the electric charge accumulated in the smoothing capacitor 48 in the charging circuit 15 up to a voltage determined by the winding ratio of the primary winding N1 and the tertiary winding N3 is supplied to the instantaneous power failure compensation capacitor 4 by the control of the power MOSFET 5d. Charged.

  Here, the supply voltage to the load 12 on the secondary side of the insulation transformer 9 is assumed to be 24V, and the load 12 is assumed to be a backlight of a liquid crystal television, a logic circuit, a tuner, etc. It is connected to a load via an inverter and a DC / DC converter. For this reason, the accuracy of the output voltage Vout can be set more loosely than the configuration in which the load is directly connected, and the accuracy in this embodiment is about ± 10%.

  The gates of the power MOSFETs 5a to 5d (Q1 to Q4) are driven by outputs obtained by inputting control signals from the controller 51 to the drivers 29a to 29d, respectively. Among these, the two gate control signals for driving the power MOSFETs 5a and 5b in particular are constituted by a PWM (Pulse Width Modulation) signal having a driving frequency f. In this embodiment, the driving frequency depends on the input voltage and the state of the load 12. f is controlled.

  In this embodiment, a power MOSFET is used as a switching element. However, an IGBT (Insulated Gate Bipolar Transistor) may be used depending on current capacity and voltage conditions. It is also preferable to use a power device made of SiC (Silicon Carbide) including a diode.

≪Control circuit block≫
FIG. 3 is a block diagram illustrating a schematic configuration of a control circuit of the switching power supply according to the first embodiment. As shown in FIG. 3, in the detection system of this embodiment, at least the input voltage Vin (or full-wave rectified voltage Vac) (see FIGS. 1 and 2) and the input current Isns after rectification (see FIGS. 1 and 2). ), The output voltage Vout (see FIGS. 1 and 2), and the voltage of the instantaneous power failure compensation capacitor 4 (momentary power loss compensation capacitance voltage Vtk) (see FIGS. 1 and 2). Yes.

  Among them, the voltage system value has a time margin in the response, so it may be obtained by thinning out once every several control cycles, but the current is obtained in the shortest possible cycle. It is important to obtain at least once every two times.

  Each acquired value is calculated inside the controller 51 (see FIGS. 2 and 4), and a pulse width control signal (PWM signal) for controlling the first to fourth switching elements Q1 to Q4 is output. . These output signals are applied to the gate terminals of the power MOSFETs 5a to 5d (Q1 to Q4) directly or indirectly using a driver IC or the like, and the switching operation of the circuit of FIG. 2 is performed.

≪Control flow≫
The control method of the switching power supply includes digital control and analog control. Hereinafter, the basic control flow will be described with reference to FIG. In FIG. 3, an input voltage Vin is a voltage of the AC power supply 1 that is a commercial AC, and this is input to the power failure detector 21. The output of the power failure detector 21 is connected to the switch 19a and the multiplier 22d.

  The output voltage Vout is connected to the inverting inputs of the amplifier 20a and the amplifier 20b. The output voltage command value is input to the non-inverting inputs of the amplifier 20a and the amplifier 20b. The output of the amplifier 20a is connected to the multiplier 22a. A full-wave rectified voltage Vac is also input to the multiplier 22a. The output of the multiplier 22a is connected to the non-inverting input of the amplifier 20c. A signal obtained by converting the rectified input current Isns into a voltage is input to the inverting input of the amplifier 20c. The output of the amplifier 20c is connected to one input terminal of the switch 19a.

  The output of the amplifier 20b is connected to the other input terminal of the switch 19a. The switch 19a switches the outputs of the amplifier 20b and the amplifier 20c according to the output of the power failure detector 21 as described above, and has a function of switching the operation at the time of normal operation (when energized) and at the time of power failure.

  The output terminal of the switch 19a is connected to the positive input of the PWM comparator 27a. The triangular wave generator 25 is connected to the negative input of the PWM comparator 27a. In FIG. 3, the triangular wave generator 25 is expressed as “triangular wave”, and the PWM comparator 27a is expressed as “PWM CMP”.

  The output of the multiplier 22d is connected to the triangular wave generator 25. When a power failure occurs or the full-wave rectified voltage Vac is lower than the reference voltage and the output of the comparator 27c is present, the slope of the triangular wave is changed to change the driver 29a. And 29b to control the drive frequency f of the switching elements Q1 and Q2.

  The output of the PWM comparator 27a is connected to the switching element Q1, that is, the gate of the power MOSFET 5a via the driver 29a. The output of the PWM comparator 27a is connected to the switching element Q2, that is, the gate of the power MOSFET 5b via the NOT circuit 28 and the driver 29b. Further, the output of the PWM comparator 27a is connected to the multiplier 22b together with the output of the power failure detector 21, and the output of the multiplier 22b is input to the driver 29c. The output of the driver 29c is connected to the switching element Q3, that is, the gate of the power MOSFET 5c. Thereby, the switching element Q1 and the switching element Q2 perform a switching operation exclusively. Further, at the time of power failure or in the vicinity of the zero cross, the switching element Q3 performs a switching operation in synchronization with the switching element Q1.

  The instantaneous blackout compensation capacitor voltage Vtk, which is the voltage of the instantaneous blackout compensation capacitor 4, is connected to the positive input of the comparator 27 b and compared with the upper limit value of the reference voltage (momentary blackout compensation capacitor voltage upper limit value Vtk_lim) connected to the negative input. Is done. If the value is equal to or less than the upper limit value, the output is input to the multiplier 22c, so that the driver 29d outputs a pulse signal having the same phase as the switching element Q2 to the switching element Q4, that is, the gate of the power MOSFET 5d.

  In the case of digital control, the power failure detection signal can be generated by determining the value of the input voltage Vin (or full-wave rectified voltage Vac), which can further simplify the circuit.

<< Calculator >>
FIG. 4 is a block diagram illustrating input / output signals of the arithmetic unit of the switching power supply according to the first embodiment. As shown in FIG. 4, each signal of the input voltage Vin (or full-wave rectified voltage Vac), the rectified input current Isns, the output voltage Vout, and the instantaneous power failure compensation capacitance voltage Vtk is supplied to the arithmetic unit, that is, the controller 51. Based on the information, the controller 51 determines the situation and state and determines and outputs a pulse width control waveform for controlling the switching operation of each of the switching elements Q1 to Q4.

  The computing unit shown in FIG. 4, that is, the controller 51 is a part of the control circuit shown in FIG. As the controller IC constituting the controller 51, a field-programmable gate array (FPGA), an application specific integrated circuit (ASIC), a digital signal processor (DSP), or the like may be used, or an inexpensive microcomputer may be used. is there.

  There are various controller ICs as actual products (parts), and various functions are built in. Depending on what IC component is used as the controller IC (controller 51), the function of the controller 51 in the control circuit of FIG. 3 is different, and the configuration of the control circuit block in FIG. 4 is different.

  For example, an input signal such as the output voltage Vout and the input voltage Vin in FIG. 4 is detected by some detection circuit, but a controller IC that incorporates an ADC (Analog to Digital Converter) in the input unit. If the detected analog signal is directly input, and the controller IC does not include an ADC, a signal obtained as a result of conversion into a digital signal by an external ADC is input.

  Further, as will be described later, in the present invention, the operating conditions are switched between a phase where the input voltage is high and a sufficient input current is obtained, and a phase where the input voltage is low and the input current is hardly obtained (so-called zero cross vicinity). Digital control is preferable. In digital control, an arithmetic processing unit built in the controller 51 executes a control program, and outputs a control signal for each gate as described with reference to FIG. 3 according to each detected value.

<< Operation of switching power supply >>
As described above, the circuit configuration of the switching power supply according to the present embodiment has been described. Next, the relationship between the commercial AC input voltage and the input current (power) and the specific driving method of the switching power supply are described in FIG. For this purpose, description will be made with reference to FIG.

  First, the operation in a steady state will be described. In a steady state, a full-wave rectified voltage Vac that is full-wave rectified by the rectifier of the diode bridge 2 is transmitted to the input capacitor 16 with respect to a commercial alternating current input voltage Vin (100 VAC) of 50 Hz or 60 Hz. Here, when the power factor improvement for satisfying the input harmonic standard is performed according to the process as described in FIG. 4, the rectified input current waveform is approximately sine like the input current Iin in FIG. It can be controlled to have a wavy waveform. Since the commercial alternating current cycle is 50 Hz or 60 Hz, the scale of the horizontal axis in FIG. 1 is on the order of 40 to 50 ms at the maximum.

  Since power is represented by the product of voltage and current, the input power waveform after rectification also shows a substantially sinusoidal waveform like the input power Pin of FIG. That is, in a region where the absolute value of the input voltage Vin is small, there is a phase where the input power Pin becomes 0 or becomes extremely small.

  Hereinafter, in the steady state, the absolute value of the input voltage Vin is relatively large, the period of the input power Pin ≠ 0, the period A, the absolute value of the input voltage Vin is small, and the input power Pin = 0 in the vicinity of the so-called zero cross passing 0V. The description will be continued assuming that the period is referred to as period B, and the period during which the output voltage compensation operation is performed when a power failure is detected is referred to as period C.

  Actually, the boundary between the period A and the period B has a room for fluctuation depending on the operating conditions and cannot be strictly defined. Therefore, the full-wave rectified voltage Vac that is the input voltage waveform after full-wave rectification is used. It is realistic to set an arbitrary threshold and determine whether it is period A or period B by comparison with the threshold. In addition, in consideration of operating conditions and response time at the time of detection, it is possible to provide a hysteresis characteristic so that the threshold value is different between when the full-wave rectified voltage Vac increases and when it decreases.

  The waveform of the transmission power Pt transmitted from the primary side to the secondary side substantially coincides with the input power Pin when the compensation operation from the instantaneous power failure compensation capacitor 4 is not performed in the period B (FIG. 15). For this reason, the transmission power Pt in period B becomes 0, and the output voltage ripple increases.

  In order to avoid this increase in output voltage ripple, in this embodiment, by turning on the power MOSFET 5c serving as the switching element Q3 in the period B, the energy accumulated in the instantaneous power failure compensation capacitor 4 is transmitted from the primary side to the secondary side. To control.

Here, assuming that the on-time of the switching element Q1 when supplying the current in the period A and the period B is both t1, the maximum current ia in the period A and the maximum current ib in the period B are the full-wave rectified voltages, respectively. When the maximum value of Vac is V1 (= 141V), the charging setting voltage of the instantaneous power failure compensation capacitor 4 (Ctk) is V2, and the inductance of the primary winding N1 of the insulating transformer 9 is L1.
ia = V1 / L1 · t1
ib = V2 / L1 · t1
It is expressed by the following formula.

By the way, the voltage of the instantaneous power failure compensation capacitor 4 (Ctk) is held at the charging setting voltage V2 (here, 220V) that is constantly the upper limit by the charging circuit 15 connected to the tertiary winding N3. For,
ib = (V2 / V1) · ia
It becomes. With the above assumptions,
V2 / V1 = 220/141 = 1.56
Therefore, in period B, a current approximately 1.56 times as large as the maximum current in period A is generated. As a result, there is a risk of causing magnetic saturation of the insulating transformer 9, or causing destruction or malfunction of the switching element.

  In order to avoid this problem, the PWM control drive frequency fb is set to the drive frequency in the period A so that the ON time of the switching element Q1 in the period B is shortened and the maximum current level becomes equal to the period A. What is necessary is just to be more than (V2 / V1) times fa. Note that the device tolerance of the circuit components is expected to be several tens of percent, and the on-time ratio of operation dead time includes a period that cannot be incorporated, so the drive frequency magnification is not always strict. Good.

  For example, in a system with a drive frequency of 100 kHz in period A, the time per cycle is 10 μs, but when there is a dead time of 0.5 μs for each ON / OFF, the on-time ratio exceeds 90% at the maximum. There is no. Since the dead time does not depend much on the drive frequency from the element characteristics, increasing the drive frequency corresponds to a reduction in the effective on-time ratio.

  On the other hand, if the drive frequency is further increased in terms of device breakdown voltage, the current becomes smaller and safer, but conversely the energy efficiency decreases due to an increase in loss due to an increase in the number of switching. From the above, it is considered that the drive frequency fb in the period B is desirably controlled within a range in which (V2 / V1) times ± 20% of the drive frequency fa in the reference period A is a guide.

  In the switching power supply according to the present embodiment, the transmission power Pt from the primary side to the secondary side does not have a period of 0 in the period B due to the energy compensated in the period B, as shown in FIG. The energy consumed in the compensation operation is controlled in the middle of the period A, the switching element Q4 is controlled, the instantaneous power failure compensation capacitor Ctk is recharged by the output from the tertiary winding N3, and again charged to the charge setting voltage V2. In order to maintain the state, the waveform is such that the vicinity of the peak is slightly cut off. As a result, the transmission power Pt is leveled and the output voltage ripple is suppressed, so that the capacitance value of the output smoothing capacitor 11 can be reduced.

  Next, an operation at the time of occurrence of a momentary power failure (momentary power failure) during period C will be described. In a state where an instantaneous power failure occurs and the output of the power failure detector 21 (FIG. 3) is high, the input voltage Vin, the input current Iin, and the input power Pin are Therefore, similarly to the period B described above, the switching element Q3 is turned on to supply energy from the instantaneous power failure compensation capacitor Ctk, and control is performed so that the output voltage Vout becomes constant.

  During this time, since the voltage of the instantaneous power failure compensation capacitor Ctk decreases consistently, charging from the tertiary winding N3 is performed during the period A after the power is restored. When the amount of compensation energy during the instantaneous power interruption period is large and the period B is reached before reaching the upper limit voltage, the switching element Q3 is turned off and the charging is stopped. It is assumed that the charging is resumed after the period A is reached again and the period A is reached again. That is, the switching elements Q3 and Q4 are controlled so that the on periods are mutually exclusive. When charging the instantaneous power failure compensation capacitor Ctk, the switching device Q4 is turned on, and when discharging from the power failure compensation capacitor Ctk, the switching device Q4 is turned on. It is necessary to turn on Q3 so that the charge / discharge periods do not overlap.

<Circuit operation at steady state>
Next, details of the circuit operation in a steady state will be described with reference to a circuit diagram and an operation waveform diagram.

(Period A with high input voltage)
First, an operation for each basic switching period in the period A during steady state will be described with reference to FIGS. The switching frequency is approximately in the range of several tens of kHz to 200 kHz. If the driving frequency in period A is 100 kHz, the horizontal axis of the operation waveform diagram of FIG. 6 represents a time of several tens of μs.

  5A to 5D are diagrams schematically showing the current path of the steady state operation of the switching power supply according to the first embodiment. The input capacitor 16 (Cin, FIG. 2) is represented by a variable voltage source named Cin. ing.

  FIG. 5A shows the flow of current when the switching element Q1 is on in the steady state (period A in FIG. 1). In period A (FIG. 1), since full-wave rectified voltage Vac (FIGS. 1 and 2) corresponding to the output of diode bridge 2 (FIG. 2) is large, switching element Q1 is turned on while switching element Q2 is off. As a result, current flows from the variable voltage source Cin to the switching element Q1 via the primary winding (N1, FIG. 2) of the isolation transformer Tr (9, FIG. 2). At this time, the insulating transformer Tr is excited, but since it is in the direction blocked by the diode D1 (10a, FIG. 2), no current flows through the secondary winding (N2, FIG. 2). Power is not transmitted. Instead, the energy stored in the output smoothing capacitor Co (11, FIG. 2) is discharged to the load (12, FIG. 2). Further, since the charging function by the charging circuit (15, FIG. 2) is stopped by turning off the switching element Q4, the instantaneous power failure compensation capacitor Ctk (4, FIG. 2) is not charged and the tertiary winding ( N3, FIG. 2) also does not generate current.

  FIG. 5B shows a current flow when the switching element Q1 is turned off (off) in a steady state (period A in FIG. 1). When the switching element Q1 is turned off in the state of FIG. 5A, the current flowing through the switching element Q1 can no longer flow through the switching element Q1, as shown in FIG. 5B, and is commutated to the parasitic diode of the switching element Q2. As shown in the waveform of IQ2 shown in FIG. 6 to be described later, the current of the switching element Q2 greatly fluctuates on the negative side.

  FIG. 5C shows a current flow when the switching element Q1 is off and the switching elements Q2 and Q4 are on in the steady state (period A in FIG. 1). When switching elements Q2 and Q4 are turned on in the state of FIG. 5B, as shown in FIG. 5C, current is supplied to the load on the secondary side and from the tertiary winding (N3, FIG. 2) on the primary side. The instantaneous charging compensation capacitor Ctk (4, FIG. 2) is charged by the current. Further, in the state of FIG. 5B, current is commutated to the parasitic diode of the switching element Q2, which suggests that no current flows through the switching element Q2. Therefore, turning on the switching element Q2 in the state of FIG. 5B means turning on the switching element Q2 when the current flowing through the switching element Q2 is 0, resulting in ZCS (Zero Current Switching), and switching loss is reduced. It can be suppressed and the efficiency can be improved.

  FIG. 5D shows a current flow when the switching element Q1 is turned off and the switching elements Q2 and Q4 are turned off (off) in a steady state (period A in FIG. 1). In the state shown in FIG. 5C, when the voltage of the instantaneous power failure compensation capacitor Ctk (4, FIG. 2) reaches a separately set value, the switch of the switching element Q4 is turned off to stop charging the instantaneous power failure compensation capacitor Ctk. To do. After that, when the switching element Q2 is turned off, as shown in FIG. 5D, the current flowing through the switching element Q2 is commutated to the parasitic diode of the switching element Q1, and the switching is performed as shown in the waveform of IQ1 shown in FIG. The current of the element Q1 greatly fluctuates on the negative side. If the switching element Q1 is turned on again at this timing, the switching element Q1 is turned on when the current flowing through the switching element Q1 is 0. Therefore, the switching loss is suppressed and the efficiency is improved as in the case of the above. can do.

(Period B / low input voltage)
On the other hand, in the period B in which the absolute value of the commercial AC input voltage Vin is small, the input current is hardly generated as it is or the current value is small as described above. The transformer 9 is not excited. However, in the present invention, the switching element Q3 is turned on before the switching element Q1 is turned on, so that the instantaneous power failure compensation capacitor Ctk connected in series with the switching element Q3 functions as an input power supply in the primary side circuit. To do. Therefore, in the period B, instead of the variable voltage source Cin corresponding to the rectified output of the input power source in FIG. 5A, current is supplied from the instantaneous power failure compensation capacitor Ctk, and the operation described in FIGS. 5A to 5D is realized. Then, energy is transmitted to the secondary side.

  The circuit operation at the time of the low input voltage (period B) is the same as the circuit operation at the time of instantaneous power failure detection (period C) described later, but the energy supply time from the instantaneous power failure compensation capacitor Ctk is short. As a result, the output voltage Vout is less decreased and the charging time of the instantaneous power failure compensation capacitor Ctk is also shortened.

<Operation waveform in period A>
FIG. 6 is a diagram illustrating operation waveforms in the period A when the switching power supply according to the first embodiment is stationary. In FIG. 6, the period during which the Q1 gate is high and the switching element Q1 is on corresponds to the state of FIG. 5A. The Q1 gate, Q2 gate, and Q4 gate are low and the switching elements Q1, Q2 are low. , Q4 is OFF corresponding to the state of FIG. 5B, the switching element Q1 is OFF when the Q1 gate is LOW, and the switching elements Q2 and Q4 are ON when the Q2 gate and Q4 gate are HIGH. Corresponds to the state of FIG. 5C, the switching element Q1 is off when the Q1 gate is low, the switching elements Q2 and Q4 are off when the Q2 gate and Q4 gate are low. The period corresponds to the state of FIG. 5D.

  VQ1 represents the voltage applied between the source and drain of the switching element Q1. When the Q1 gate, which is the gate potential of the switching element Q1, is high, VQ1 is 0 because the switching element Q1 is on, and the current IQ1 flows between the source and drain of the switching element Q1. It is shown.

  Further, when the gate Q1 which is the gate potential of the switching element Q1 is low, the switching element Q1 is off, so that VQ1 shows a high voltage, and the current IQ1 between the source and drain of the switching element Q1 is 0.

  The relationship between the Q2 gate, VQ2, and IQ2 in the switching element Q2 is substantially the same as the relationship in the switching element Q1. However, at the moment when the Q1 gate is turned off while the Q2 gate is in a low state, a high voltage is applied between the source and drain of the switching element Q2, so IQ2 indicates a negative current value. This indicates that a current flows through the parasitic diode of the switching element Q2.

  Note that in the period A, the Q3 gate is low, and the switching element Q3 is kept off.

<Power factor correction operation>
Next, the power factor improving operation will be described with reference to FIGS.
The period A in FIG. 1 is approximately 10 ms (in the case of AC input 50 Hz), and the period in which the Q1 and Q2 gates in FIG. 6 repeat high and low is approximately 10 μs (in the case of the drive frequency f = 100 kHz). In other words, by repeating the operation in which the Q1 and Q2 gates in FIG. 6 change to high and low within a period of about 10 ms in the period A in FIG. 1, the on / off operation of the switching elements Q1 and Q2 (FIG. 2) is about 1000 times. It will be repeated about once.

  At this time, if the period in which the Q1 and Q2 gates in FIG. 6 are changed from high to low is changed, or the ratio between the high period and the low period is changed, the current waveform of the harmonic component in the input current Iin changes. . Using this principle, the controller 51 (FIG. 4) optimally controls on / off including the pulse widths of the switching elements Q1, Q2, Q3, and Q4 (FIG. 2) according to the situation. The harmonic component of Iin can be removed and the power factor can be improved.

<Circuit operation when instantaneous power failure is detected (period C)>
Next, details of the circuit operation at the time of instantaneous power failure detection (period C) will be described. When the power failure detector 21 (FIG. 3) detects a sudden drop in the input voltage Vin of commercial AC (AC power supply 1) (period C in FIG. 1), the input current Iin also decreases as shown in FIG. Since the input power Pin is interrupted, power cannot be transmitted to the secondary side.

  If the output voltage Vout (FIG. 1) remains as it is, the switching element Q3 is turned on and energy is supplied from the instantaneous power failure compensation capacitor 4 (Ctk, FIG. 2). At this time, if the instantaneous power failure compensation capacitor 4 (Ctk, FIG. 2) is charging from the tertiary winding N3 (FIG. 2), although not shown in the figure, the switching element Q4 is first turned off to stop the charging. After that, the switching element Q3 is turned on.

  Thereafter, the switching element Q1 is turned on to generate a current in the insulating transformer 9 (Tr, FIG. 2). Subsequently, the switching element Q1 is turned off and the switching element Q2 is turned on to obtain the secondary. On the side, current is supplied to the output smoothing capacitor 11 (Co, FIG. 2) and the load 12 (FIG. 2) via the diode D1, and the output voltage Vout can be maintained within the control range.

  Accordingly, the basic operation after turning off the switching element Q4 and turning on the switching element Q3 is common to the period A except that the drive frequency of the switching elements Q1 and Q2 is increased. That is, in FIG. 1, the full-wave rectified voltage Vac and the input current Iin become 0 with the occurrence of a power failure, the switching element Q3 is turned on, and energy is supplied from the instantaneous power failure compensation capacitor Ctk. The output voltage Vout gradually decreases, but the supply of power to the load 12 (FIG. 2) is continued. Moreover, it returns to the operation | movement at the time of a steady state (period A and period B of FIG. 1) with the power recovery of AC power supply 1. FIG. In this period C, the switching element Q4 may be turned off and the switching element Q3 may be kept on throughout the period during which the compensation operation is performed.

[Second Embodiment]
Next, a second embodiment of the present invention will be described with reference to FIGS. 2, 7A, and 7B. In the second embodiment, the circuit configuration and basic driving procedure are substantially the same as those in the first embodiment, but the process at the time of transition between the period A and the period B is defined in more detail.

  As described in the first embodiment, the frequency of the AC power supply 1 is approximately 50 to 60 Hz, whereas the drive frequency f in the period A during the PFC (power factor correction) operation is about 100 kHz. It is very different. Therefore, in FIG. 1, 1000 switching operations are performed in a period of 10 μs within the period A of about 10 ms. Here, if the drive frequency fa in the period A is 100 kHz, the frequency of the AC power supply 1 is 50 Hz and the period A is 9 ms in length, although it actually depends on the operation status of the PFC and the reference voltage in FIG. If there is, 900 control pulses are output to the Q1 gate during this period.

  On the other hand, in the remaining 1 ms period B, when the charging set voltage V2 of the instantaneous power failure compensation capacitor Ctk is 220V, the driving frequency fb is about 150 kHz from V2 / V1 = 1.56, so the Q1 gate has a period. During B, about 150 control pulses are output. As described above, since the horizontal axis of FIG. 1 is greatly different from the scale of the control pulse signal of the Q1 gate, the operation at the time of switching will be described in more detail with reference to FIG.

  In the circuit diagram of FIG. 2, once again confirming the operation when shifting from the period A to the period B, the input power source viewed from the switching elements Q1 and Q2 at the timing when the switching element Q3 is turned on is The output is switched to the instantaneous power failure compensation capacitor Ctk, the drive frequency f is switched from fa to fb, and then the current IQ1 of the switching element Q1, which has approached 0 as the input voltage decreases, is rapidly raised to the peak current level.

  This operation will be described with reference to the waveform diagram of FIG. 7A. When the voltage of the input capacitor 16 (Cin) (= the full-wave rectified voltage Vac) decreases as the input voltage Vin decreases, the current IQ1 of the switching element Q1 also decreases. The switching period at this time is ta (= 1 / fa).

  Here, when the switching element Q3 is turned on and the instantaneous power failure compensation capacitor Ctk charged to the charge setting voltage V2 is connected, the input capacitor Cin is charged so that the potentials of both are equal, and the Cin voltage rises. In FIG. 7A, for the sake of safety, the operation of the switching elements Q1 and Q2 (not shown) is stopped for two periods until the Cin voltage becomes equal to the voltage of the instantaneous power failure compensation capacitor Ctk.

  In addition, since the rapid change of the current IQ1 is suppressed by the inductance of the insulating transformer 9 (Tr), the two periods of the stop period are not necessarily set. Further, a longer stop period can be provided depending on circuit conditions.

  Next, the drive frequency f of the Q1 gate and the Q2 gate is increased to fb, but the purpose is to prevent the peak current level from becoming excessive. Since the current IQ1 of the switching element Q1 is also affected by the state immediately before (= initial condition), if the current value does not immediately become excessive, the cycle and the on-time ratio are stepped over several cycles as shown in FIG. 7A. The drive frequency may be continuously increased by narrowing down to.

  Next, an operation when shifting from the period B to the period A will be described with reference to FIG. 7B. The Cin voltage that is the voltage of the input capacitor 16 (Cin) and the instantaneous power failure compensation capacitor 4 (Ctk) during the period B is slightly lower than the charge setting voltage V2 (= Vtk_lim) due to the discharging process in the period B. Since energy for compensating for the instantaneous power interruption period of about 20 ms is accumulated, the amount of change in the period B which is only about 1 ms is small.

  In FIG. 7B, the switching element Q3 is turned off several cycles earlier than the period A is reached. As a result, the energy source that can be discharged to the secondary side is limited to the input capacitor Cin only, and the Cin voltage and the current IQ1 of the switching element Q1 depending on the Cin voltage are also greatly reduced in the following several cycles. As the current decreases, the period tb (= 1 / fb) and the on-time ratio are gradually increased to the period ta (1 / fa) and the on-time ratio of the period A, so that the voltage of the input capacitor Cin becomes the diode bridge 2. When the output coincides with the output, the operation mode of period A is switched.

  In this way, the switching element Q3 is turned off early so that the voltage of the input capacitor Cin smoothly matches the output voltage of the diode bridge 2 so that the current IQ1 also changes continuously. Can do.

  In the transition from the period B to the period A, a period for turning off the switching element Q1 (and the switching element Q2 (not shown)) is not provided as in FIG. 7A, but a several period stop period is used for timing adjustment. It may be provided. Further, except for the suspension period, the switching element Q3 may be controlled by a control pulse signal synchronized with the switching element Q1 as in the first embodiment.

  The detailed operation at the time of transition between the period A and the period B has been described above. In the scale of FIG. 1, the digit of the period is different, so the drive frequency f appears to change discontinuously, and such control is possible. However, as in the second embodiment, it is continuous in a short period. It goes without saying that even if such control is performed, it is possible to achieve overcurrent prevention and suppression of output voltage ripple, which are the objects of the present invention.

  Further, here, the transition between the period A and the period B has been described. However, when a momentary power failure is detected and the period A is shifted to the period C, or when the power supply is restored (power recovery), the period C to the period C It goes without saying that the control as described above can also be performed when the process shifts to A.

[Third Embodiment]
Next, 3rd Embodiment is described using FIG. 8, FIG. 9, FIG. 10A-10D, and FIG. The third embodiment employs a power circuit using a current resonance type converter, whereas the first embodiment is a power circuit using an active clamp type flyback converter.

  FIG. 9 is a circuit diagram showing a configuration of the switching power supply according to the third embodiment. In FIG. 9, the input of the AC power source 1 is converted into a full-wave rectified waveform via the diode bridge 2. The input capacitor 16 (Cin) is connected to the DC side of the diode bridge 2 as in the first embodiment. The input capacitor 16 is for an input filter and has a capacitance of several μF.

  Further, a series pair of a power MOSFET 5 c (Q 3) and an instantaneous power failure compensation capacitor 4 (Ctk) is connected in parallel to the diode bridge 2 and the input capacitor 16. The power MOSFET 5c is an N-channel power MOSFET. The source is connected to the positive electrode side of the input capacitor 16 and the drain is connected to the instantaneous power failure compensation capacitor 4, so that the instantaneous power failure compensation capacitor 4 is prevented from discharging. The instantaneous power failure compensation capacitor 4 is a capacitor for instantaneous power failure compensation, and its capacity varies depending on the instantaneous power failure compensation time, but is assumed to be about 100 μF to 1000 μF.

  A series pair of a power MOSFET 5a (Q1) and a power MOSFET 5b (Q2) is further connected in parallel to the diode bridge 2 and the input capacitor 16, and the power MOSFET 5b has a primary winding N1 of an insulating transformer 9 (Tr). And a resonance capacitor 17 (Cr) are connected in parallel.

  Each terminal of the secondary winding N2 of the insulating transformer 9 is connected to the anodes of the diodes 10a (D1) and 10b (D2), and the cathodes of the diodes 10a and 10b are both connected to the output smoothing capacitor 11 (Co ) To the positive electrode side. The middle point of the secondary winding N2 is grounded together with the negative electrode side of the output smoothing capacitor 11. Further, a load 12 is connected to the output smoothing capacitor 11.

  A charging circuit 15 for charging the instantaneous power failure compensation capacitor 4 is connected to the tertiary winding N3 of the isolation transformer 9, and a current suction port among the negative electrode side of the instantaneous power failure compensation capacitor 4 and the output terminal of the charging circuit 15 is connected. The power MOSFET 5d (Q4) is connected to the corresponding terminal. At this time, the drain of the power MOSFET 5d is connected to the negative electrode side of the instantaneous power failure compensation capacitor 4, and the source is connected to the charging circuit 15 side. The power MOSFET 5 d is an N-channel power MOSFET, and controls the charging current to the instantaneous power failure compensation capacitor 4 by the charging circuit 15.

  In the charging circuit 15, both ends of the tertiary winding N3 are connected to the anodes of the diode 10c (D3) and the diode 10d (D4), and each cathode is commonly connected to the positive side of the instantaneous power failure compensation capacitor 4 via the coil. Connected. Further, the cathodes of the diode 10c and the diode 10d are connected to the midpoint of the tertiary winding N3 and the source of the power MOSFET 5d through the smoothing capacitor 48.

  The charging circuit 15 is different from the first embodiment (FIG. 2) in that a charging path via the diode 10d is added in accordance with the current resonance type, but the instantaneous power failure compensation capacitor 4 is charged. Is a characteristic that should be performed more slowly in the background than power transmission to the secondary side, and since the output current and operation period are short, the instantaneous power failure compensation capacitor 4 can be charged even with one system. In this case, the diode 10d can be omitted. In this case, the connection between the smoothing capacitor 48 connected via the diode 10c and the tertiary winding N3 is not the midpoint of the tertiary winding N3 but the terminal opposite to the terminal connected to the anode of the diode 10c. Will be done.

  The positive electrode side of the smoothing capacitor 48 and the positive electrode side of the instantaneous power failure compensation capacitor 4 of the primary circuit are connected via a coil 49 so that the charging current does not become excessive. The negative side of the smoothing capacitor 48 is connected to the source of the power MOSFET 5d, and the operation of the charging circuit 15 can be controlled by the control of the power MOSFET 5d.

  The gates of the power MOSFETs 5a to 5d (Q1 to Q4) are driven by outputs obtained by inputting control signals from the controller 51 to the drivers 29a to 29d, respectively. Among these, especially the two signals for driving the power MOSFETs 5a and 5b are constituted by PWM signals of the drive frequency f, and in this embodiment, the drive frequency f is controlled by the input voltage and the state of the load 12.

  In this embodiment, the power MOSFET is used as the switching element, but an IGBT may be used depending on the current capacity and voltage conditions. It is also suitable to use a power device made of SiC including a diode.

  The input / output of the controller 51 is the same as in FIG. 4 of the first embodiment. The control circuit block is substantially the same as that of the first embodiment shown in FIG. 3, but in the current resonance method, the input current is controlled not by the on-time ratio of the switching elements Q1 and Q2, but by changing the drive frequency. Therefore, the difference is that the output of the amplifier 20c in FIG. 3 is connected to the triangular wave generator 25, and the drive frequency is changed by changing the inclination of the output triangular wave.

  Further details of the operation will be described with reference to FIG. In the current resonance type power supply circuit, the ON periods of the switching elements Q1 and Q2 are in a complementary relationship. If the dead time is ignored, the ON time ratio of each element is normally driven at 50%, respectively. The flow control of the input current Iin is different from the first embodiment in that this is controlled by increasing / decreasing the period, that is, by changing the drive frequency.

Similarly to the first embodiment, if each stage when viewed in an AC cycle according to the state of input power is a period A, a period B, and a period C, in the period A, the current is obtained by the effect of the power factor improving operation together with the input voltage. In order to increase or decrease, the driving frequency f in the period A changes in conjunction with the input voltage. Among these, the drive frequency becomes maximum (= f ₀ ) almost coincides with the phase where the input voltage takes the peak value V1.

Next, in the period B, the switching element Q3 is turned on, and the compensation operation from the instantaneous power failure compensation capacitor 4 boosted to the charge setting voltage V2 is performed. In the case of a current resonance type power supply, the immediately preceding drive frequency f is rather lower than f ₀ , and therefore, if the same period is applied to the on-time, the synergistic effect of the increment of the applied voltage and the increment of the on-time is even greater. Electric current is generated.

Therefore, in order to suppress the peak current level to be equal to the maximum level of the period A, the drive frequency fb of the switching elements Q1, Q2 is set to fb = (V2 / V1) · f ₀
Need to be raised. Incidentally, as described in the first embodiment, the magnification of the driving frequency fb when change is not necessarily exact, as a guide with respect to f ₀ as the base of (V2 / V1) times ± 20% It may be desirable to control the range.

The energy output from the instantaneous power failure compensation capacitor 4 in the compensation operation in the period B is charged in the subsequent period A. Similarly, in the period C after the occurrence of the instantaneous power failure is detected, the drive frequency fc of the switching elements Q1 and Q2 is fc = f ₀ × V2 / V1.
It is necessary to raise so that it becomes. Note that the magnification of the driving frequency is not strict, and in FIG. 8, it is shown as being almost constant at fc throughout the period C. However, depending on the length of the instantaneous power failure compensation elapsed time, Since the voltage of Ctk decreases from the charge setting voltage V2, it is also possible to perform an operation of decreasing the drive frequency fc in accordance with the voltage applied to the instantaneous power failure compensation capacitor Ctk within a range satisfying the above equation.

  FIGS. 10A to 10D are diagrams schematically showing a current path of steady-state operation of the switching power supply according to the second embodiment. The input capacitor 16 (Cin, FIG. 9) is represented by a variable voltage source named Cin. ing.

  FIG. 10A shows the flow of current when the switching element Q1 is on in the steady state (period A in FIG. 8). In period A (FIG. 8), since full-wave rectified voltage Vac (FIGS. 8 and 9) corresponding to the output of diode bridge 2 (FIG. 9) is large, switching element Q1 is turned on while switching element Q2 is off. As a result, current flows from the variable voltage source Cin to the primary winding (N1, FIG. 9) of the insulating transformer Tr (9, FIG. 9) via the switching element Q1. At this time, the resonant capacitor Cr (17, FIG. 9) is charged, the insulating transformer Tr is excited, and the secondary winding (N2, FIG. 9) is blocked by the diode D2 (10b, FIG. 9). A current flows through the unblocked diode D1 (10a, FIG. 9), and power is supplied to the output smoothing capacitor Co (11, FIG. 9) and the load (12, FIG. 9).

  At this time, in the tertiary winding (N3, FIG. 9), if the switching element Q4 is turned on, a current flows through the diode D3 (10c, FIG. 9), and the instantaneous power failure compensation capacitor Ctk (4, FIG. 9). Is charged.

  FIG. 10B shows the flow of current when the switching element Q1 is turned off (off) in a steady state (period A in FIG. 8). When the switching element Q1 is turned off in the state of FIG. 10A, the current flowing through the switching element Q1 cannot flow through the switching element Q1 as shown in FIG. 10B, and therefore is commutated to the parasitic diode of the switching element Q2. As will be described later with reference to the waveform of IQ2 shown in FIG. 11, the current of the switching element Q2 once swings to the minus side.

  FIG. 10C shows a current flow when the switching element Q1 is off and Q2 is on in the steady state (period A in FIG. 8). When the switching element Q2 is turned on in the state of FIG. 10B, as shown in FIG. 10C, the charge charged in the resonance capacitor Cr (17, FIG. 9) is discharged, and current is supplied through the diode D2 on the secondary side. In addition, if the switching element Q4 is in the ON state, the current from the tertiary winding (N3, FIG. 9) flows through D4 on the primary side, so that the instantaneous power failure compensation capacitor Ctk (4, FIG. 9) is charged. Is done. Further, in the state of FIG. 10B, the current is commutated to the parasitic diode of the switching element Q2, which suggests that no current flows through the switching element Q2. Therefore, turning on the switching element Q2 in the state of FIG. 10B means turning on the switching element Q2 when the current flowing through the switching element Q2 is 0, which becomes ZCS, suppressing switching loss and improving efficiency. Can be improved.

  FIG. 10D shows a current flow when the switching element Q1 is turned off and the switching element Q2 is turned off (off) in a steady state (period A in FIG. 8). When the switching element Q2 is turned off in the state of FIG. 10C, as shown in FIG. 10D, the current flowing through the switching element Q2 is commutated to the parasitic diode of the switching element Q1, and the waveform of IQ1 shown in FIG. The current of the switching element Q1 once swings to the minus side. If the switching element Q1 is turned on again at this timing, the switching element Q1 is turned on when the current flowing through the switching element Q1 is 0. Therefore, the switching loss is suppressed and the efficiency is improved as in the case of the above. can do.

  As described above, when the voltage of the instantaneous power failure compensation capacitor Ctk (4, FIG. 9) charged by turning on the switching element Q4 at the start of the period A reaches a separately set value. Q4 is turned off to stop charging the instantaneous power failure compensation capacitor Ctk. After that, since the output from the charging circuit 15 is stopped even when the processes of FIGS. 10A to 10D are repeated, when the voltage of the smoothing capacitor (48, FIG. 9) of the charging circuit 15 is saturated, the diodes D3 and D4 The current becomes 0, and the standby state is maintained until the next charging process is executed.

  Next, in the period B (FIG. 8), the switching element Q3 is turned on, so that the role of the input capacitor Cin (16, FIG. 9) is replaced by the instantaneous power failure compensation capacitor Ctk (4, FIG. 9). The operation of 10A to 10D is repeated. The same applies to the scene in which the period C (FIG. 8) is reached due to the power failure detection. However, in the period B and the period C, the switching element Q4 is maintained in the off state, and the charging circuit 15 stops the charging function.

  FIG. 11 is a diagram illustrating operation waveforms in the period A when the switching power supply according to the third embodiment is in a steady state. In FIG. 11, the period during which the Q1 gate is high, the switching element Q1 is on, the Q2 gate is low, and the switching element Q2 is off corresponds to the state of FIG. 10A, and the Q1 gate and Q2 gate The period during which switching elements Q1 and Q2 are off corresponding to the low level corresponds to the state of FIG. 10B. Switching is performed when the Q1 gate is low, the switching element Q1 is off, and the Q2 gate is high. The period during which the element Q2 is on corresponds to the state of FIG. 10C. The switching element Q1 is off when the Q1 gate is low, and the switching elements Q2 and Q4 are off when the Q2 gate is low. The period of time corresponds to the state of FIG. 10D.

  FIG. 11 shows an operation waveform in a state where the switching element Q4 is on and the charging circuit 15 is performing the charging operation before the voltage of the instantaneous power failure compensation capacitor Ctk reaches the charge setting voltage V2 (= Vtk_lim). When the switching element Q1 is on, a current ID1 of the secondary diode D1 and a current ID3 of the diode D3 of the charging circuit 15 are generated, and when the switching element Q2 is on, the current ID2 of the secondary diode D2 and A current ID4 of the diode D4 of the charging circuit 15 is generated.

  Although illustration is omitted, when the voltage of the instantaneous power failure compensation capacitor Ctk reaches the charge setting voltage V2 and the charging is completed, the switching element Q4 is turned off and the charging operation of the charging circuit 15 is stopped. As a result, both the current ID3 and the current ID4 in FIG.

  The operation waveform in the period B (FIG. 8) following the period A or the period C (FIG. 8) when a power failure occurs is that the driving frequency of the switching elements Q1 and Q2 is higher than the period A and the switching element Q4 is turned off. On the other hand, except that the switching element Q3 is turned on, the current ID3 and the current ID4 are both 0 in FIG.

[Fourth Embodiment]
Next, a fourth embodiment of the present invention will be described with reference to FIGS. 9, 12A and 12B. In the fourth embodiment, the circuit configuration and the basic driving procedure are almost the same as those in the third embodiment, but the process at the time of transition between the period A and the period B is the same as in the second embodiment. It is specified in detail.

  As described in the third embodiment, while the frequency of the AC power supply 1 is approximately 50 to 60 Hz, the drive frequency f in the period A during the PFC (power factor correction) operation is 60 to 100 kHz. It changes greatly in the range before and after. That is, since the horizontal axis of FIG. 8 is greatly different from the scale of the control pulse signal of the Q1 gate, the operation at the time of switching will be described in more detail using FIG. 12A.

In the circuit diagram of FIG. 9, once again confirming the operation when shifting from the period A to the period B, at the timing when the switching element Q3 is turned on, the input power source viewed from the switching elements Q1 and Q2 is connected to the diode bridge 2. The output is switched to the instantaneous power failure compensation capacitor Ctk, the drive frequency f is switched from f ₀ or less to fb, and the current IQ1 of the switching element Q1, which has approached 0 as the input voltage Vin decreases, rapidly reaches the peak current level. Launched.

This operation will be described with reference to the waveform diagram of FIG. 12A. When the voltage of the input capacitor 16 (Cin) (= the full-wave rectified voltage Vac) decreases as the input voltage Vin decreases, the current IQ1 of the switching elements Q1, Q2 operating complementarily at an on-time ratio of 50%, respectively. , IQ2 (not shown) also decreases. The switching period at this time is ta (> 1 / f ₀ ).

  Here, when the switching element Q3 is turned on and the instantaneous power failure compensation capacitor Ctk charged to the charge setting voltage V2 is connected, the input capacitor Cin is charged so that the potentials of both are equal, and the Cin voltage rises. In FIG. 12A, for safety, the operations of the switching elements Q1 and Q2 (not shown) are stopped for two periods until the Cin voltage becomes equal to the voltage of the instantaneous power failure compensation capacitor Ctk.

  In addition, since the rapid change of the current IQ1 is suppressed by the inductance of the insulating transformer 9 (Tr), the two periods of the stop period are not necessarily set. Further, a longer period can be provided depending on circuit conditions.

  Next, the drive frequency f of the Q1 gate and the Q2 gate is increased to fb, but the purpose is to prevent the peak current level from becoming excessive. Since the current IQ1 of the switching element Q1 is also affected by the state immediately before (= initial condition), if the current value does not immediately become excessive, the cycle is gradually reduced over several cycles as shown in FIG. The drive frequency may be increased continuously.

  Next, an operation when shifting from the period B to the period A will be described with reference to FIG. The Cin voltage, which is the voltage of the input capacitor Cin and the instantaneous blackout compensation capacitor Ctk in the period B, is slightly lower than the charge setting voltage V2 due to the discharging process in the period B, but originally compensates the instantaneous blackout period of about 20 ms. Therefore, the amount of change in the period B which is only about 1 ms is small.

In FIG. 12B, the switching element Q3 is turned off several cycles earlier than the period A is reached. As a result, the energy source that can be discharged to the secondary side is limited to the input capacitor Cin only, and the Cin voltage and the current IQ1 of the switching element Q1 depending on the Cin voltage are also greatly reduced in the following several cycles. As the current decreases, the period tb (= 1 / fb) is gradually increased to the period ta (> 1 / f ₀ ) of the period A, and the voltage of the input capacitor Cin (Cin voltage) is output from the diode bridge 2. When it coincides with the voltage (Vac), the mode is switched to the operation mode of period A.

  In this way, the switching element Q3 is turned off early so that the voltage of the input capacitor Cin smoothly matches the output voltage of the diode bridge 2 so that the current IQ1 also changes continuously. Can do.

  In the transition from the period B to the period A, a period for turning off the switching element Q1 (and the switching element Q2 (not shown)) is not provided as in FIG. 12A, but several periods of stop periods are provided for timing adjustment. It may be provided.

  The detailed operation at the time of transition between the period A and the period B has been described above. In the scale of FIG. 8, since the digit of the period is different, the drive frequency f seems to change discontinuously, and such control is possible. However, as in the fourth embodiment, it is continuous in a short period. It goes without saying that even if such control is performed, it is possible to achieve overcurrent prevention and suppression of output voltage ripple, which are the objects of the present invention.

  Further, here, the transition between the period A and the period B has been described. However, when a momentary power failure is detected and the period A is shifted to the period C, or when the power is restored (recovered) and the period C is exceeded. It goes without saying that the control as described above can also be performed when the period A is shifted.

[Fifth Embodiment]
Next, a fifth embodiment of the present invention will be described with reference to FIGS.
FIG. 13 shows a top view of the switching power supply substrate of the first embodiment. The circuit configuration of the power supply substrate 32 is basically the same as the circuit diagram of FIG. 2, and in FIG. 13, the same components as those in the circuit diagram of FIG. Also, new components not shown in FIG. 2 are shown.

  FIG. 14 shows a form in which the power supply substrate 32 shown in FIG. 13 is mounted on a thin television. In FIG. 14, the power supply substrate 32 is a surface mounting substrate, and the power supply substrate 32 is mounted so that the upper side of the drawing faces the liquid crystal panel 33.

  In FIG. 13, there is an input connector 30 below the power supply board 32, and in the vicinity thereof, the diode bridge 2, the input capacitor 16, the charging circuit 15, and the power MOSFETs 5a (switching element Q1), 5b (switching element Q2), 5c ( Switching elements Q3) and 5d (switching element Q4) are arranged. Among these, the diode bridge 2 and the power MOSFETs 5a and 5b, which are heat generating components, are mounted on an aluminum plate 31 having a thickness of 1 to 2 mm for heat dissipation. The power MOSFETs 5c (switching element Q3), 5d (switching element Q4) and the diode 10c are mounted directly on the substrate because they hardly generate heat.

  Near the center of the power supply board 32, a controller 51 is arranged, and detection signals from each part are input, and an output control signal of a driving frequency f is connected to a driver IC 29e having two outputs, a high side and a low side. Switching elements Q1-Q4 are driven by the output of driver IC 29e.

  Further, near the center of the power supply substrate 32, the instantaneous power failure compensation capacitor 4 is divided into several pieces and mounted. A capacitor 8 is mounted. An insulating transformer 9 is mounted on the power supply board 32 above the capacitor mounting position. In the power supply substrate 32, the upper part of the insulating transformer 9 is the secondary side, and the diode 10 a is attached to and mounted on an aluminum plate 31 a different from the aluminum plate 31 in this area. In the vicinity of the diode 10a, the output smoothing capacitor 11 is divided into several pieces and mounted. Output connectors 40a, 40b, and 40c are mounted on the top of the power supply board 32. In addition, the diode 10 c, the smoothing capacitor 48, and the coil 49 that constitute the charging circuit 15 are also mounted on the power supply substrate 32.

  As described above, according to the switching power supply device according to the present invention, it is possible to reduce the required capacitor capacity by adopting a high-efficiency circuit system, and by mounting thin type capacitors in parallel, etc. The entire thickness of 32 can be suppressed to less than 10 mm.

  Next, FIG. 14 will be described. In the center of FIG. 14, a view of the thin liquid crystal television set as viewed from the back, a view as viewed from above at the lower side, and a view as viewed from the side at the right side are shown.

  In the figure seen from the back side, the television set is shown with the back cover removed, and the power supply board 32 is mounted between the central column 34b and the column 34c on the right side. A power supply cable 38 is input to the television set and is connected to a filter substrate 39 mounted below the power supply substrate 32. An output cable from the filter substrate 39 is connected to the input connector 30 (FIG. 13) of the power supply substrate 32.

  The output connectors 40a, 40b, 40c (FIG. 13) of the power supply board 32 are connected to the LED driver board 35b, the LED driver board 35a, and the circuit board 36, respectively. The LED driver boards 35b and 35a are boards on which a DC voltage of 24V output from the power supply board 32 is input, and a converter (not shown) for stepping up or down to a voltage required for lighting an LED backlight (not shown) is mounted. Yes, it is possible to adjust the brightness of the LED by controlling the current flowing through the LED. The LED backlight (not shown) itself is between the power supply board 32 and the circuit board 36 and the liquid crystal panel 33, and has a thickness of about 10 mm.

  The T-con (timing controller) board 37 is supplied with electric power from the circuit board 36 after converting a DC voltage of 24V input from the power board 32 to the circuit board 36 into a necessary voltage.

  Further, as shown in FIG. 14, a power supply substrate 32 having a thickness of less than 10 mm is mounted on the back of a display panel unit of a television set using a liquid crystal panel 33 or other thin image monitor device (not shown). In addition, by using an LED backlight (not shown) having a thickness of about 10 mm driven by the LED driver, the set thickness of the display panel portion can be reduced to 20 mm or more and 30 mm or less.

[Other Embodiments]
In the first to fifth embodiments described above, the N-channel power MOSFET is used as the switching element, but a P-channel switching element may be used depending on the application. In the first and third embodiments, the control of the switching power supply has been described by taking the configuration of the analog circuit as an example, but digital control may be used. Even in the case of digital control, various forms of control algorithms can be used.

  In addition, the AC power supply source has been described on the premise of commercial AC. However, it is not limited to commercial AC, and private power generation or the like may be used, which is widely applied to general cases where AC power is used as a power source. it can. In addition, examples of the power supply voltage and the voltage applied to each element, and numerical values of specific forms in the display device and the mounting substrate are given as examples, but these are design matters, and other voltage values and shape values However, by applying the present invention, the switching power supply can be reduced in size, and the apparatus equipped with the switching power supply can be reduced in size, weight, and thickness.

  In the fifth embodiment, the thin liquid crystal television set mounted with the switching power supply substrate according to the present invention has been described. However, this is merely an example, and the use of the switching power supply according to the present invention reduces the size of various electronic devices. It is possible to reduce the weight and thickness.

  As described above, according to these embodiments, in an AC input isolated switching power supply having a harmonic suppression function, output voltage fluctuations are suppressed to around ± 10%, and in the conventional case, the input power is almost equal. Since power can be transmitted to the output side insulated by the charge storage means even in the phase of the low input voltage that was interrupted, the peak value of the general input current is suppressed, switching loss is suppressed, and one-stage conversion is performed. It is possible to improve the power supply efficiency together with the effect of the above.

  In addition, since the charge storage means is boosted and charged with energy from the tertiary winding, the capacity of the charge storage means can be reduced, the mounting volume of the switching power supply can be reduced, and the output density of the power supply can be improved. . Further, when using the energy of the charge storage means that has been boosted and densified, the drive frequency of the switching element is set high, so that malfunctions such as element destruction and magnetic saturation can be avoided.

  In recent years, digital control has been introduced in many devices. For example, when a control IC system that operates at a low voltage of 5 V or less and a main system that requires power of 20 V are controlled by an output signal of the control IC, If the power supply of the control system is cut off due to the above, it is easy to cause a big trouble as compared with the case where it is configured with an analog control system. By using the switching power supply having the instantaneous power failure compensation function as in the embodiment of the present invention, it is possible to avoid the sudden stop of the operation of the control system, and it is possible to construct a highly efficient system as a whole.

  The present invention can be widely applied to various electric devices, air conditioners, home appliances, information devices such as personal computers and servers that operate by inputting AC power.

1 AC power supply 2 Diode bridge (rectifying means)
4, Ctk Instantaneous power failure compensation capacitor (charge storage means)
5a, Q1 power MOSFET (first switching element)
5b, Q2 Power MOSFET (second switching element)
5c, Q3 Power MOSFET (third switching element)
5d, Q4 power MOSFET (fourth switching element)
8, Cc capacitor 9, Tr isolation transformer 10a, 10b, D1, D2 diode (smoothing means)
10c, 10d, D3, D4 Diode 11, Co Output smoothing capacitor (smoothing means)
DESCRIPTION OF SYMBOLS 12 Load 14 Current detector 15 Charging circuit 16, Cin Input capacitor 17 Resonant capacitor 19a Switch 20a, 20b, 20c Amplifier 21 Power failure detector (power failure detection means)
22a, 22b, 22c, 22d multiplier 25 triangular wave generator 27a PWM comparator 27b, 27c comparator,
28 NOT circuit 29a, 29b, 29c, 29d, 29e Driver 30 Input connector 31, 31a Aluminum plate 32 Power supply board 33 Liquid crystal panel 34a, 34b, 34c Post 35a, 35b LED driver board 36 Circuit board 37 T-con (timing controller) Substrate 38 Power cable 39 Filter substrate 40a, 40b, 40c Output connector 48 Smoothing capacitor 49 Coil 51 Controller, calculator (power factor improvement means, charge / discharge control means)
N1 Primary winding N2 Secondary winding N3 Tertiary winding

Claims (7)

Rectifying means for rectifying input AC power;
An insulating transformer having at least a primary winding, a secondary winding, and a tertiary winding;
Smoothing means connected to the secondary winding and smoothing and outputting DC power insulated from the AC power;
A first switching element connected between a DC side terminal of the rectifying means and the primary winding;
Power factor improvement control means for controlling the first switching element to improve the power factor of the input AC power;
A charge storage means connected to the DC side terminal of the rectifier means and a third switching element connected in series to the charge storage means so as to prevent discharge of the charge storage means;
A charging circuit that boosts and charges the charge storage means with electric power from the tertiary winding;
A power failure detection means for detecting a power failure of the AC power;
When the power failure detection means detects a power failure of the AC power, the third switching element is controlled so that the charge stored in the charge storage means in advance through the insulation transformer At this time, the drive frequency of the first switching element is approximately the normal drive frequency when the AC power outage is not detected (the voltage at the completion of charging of the charge storage means / the AC power Charge / discharge control means to raise the maximum voltage) times,
A switching power supply comprising: The switching power supply according to claim 1,
The charging circuit is
A diode connected to the input terminal of the tertiary winding;
A capacitor connected in parallel in a series pair of the diode and the tertiary winding;
A coil connected between the capacitor and the charge storage means;
A switching power supply comprising: In the switching power supply according to claim 1 or 2,
A fourth switching element connected between the charging circuit and the charge storage means;
The charge / discharge control means includes
When the input voltage of the AC power is higher than a predetermined voltage, the third switching element is stopped, and the fourth switching element is controlled and charged until the voltage of the charge storage means reaches a set value. ,
When the input voltage of the AC power is lower than a predetermined voltage, the fourth switching element is stopped, the third switching element is controlled, and the charge previously stored in the charge storage means is Discharge to the insulated DC output side through a transformer, and at this time, the drive frequency of the first switching element is substantially equal to the normal drive frequency when the AC power failure is not detected (the charge storage means) The switching power supply is characterized in that it is raised to a voltage at the time of completion of charging / the maximum voltage of the AC power) times or more. Rectifying means for rectifying input AC power;
An insulating transformer having at least a primary winding, a secondary winding, and a tertiary winding;
Smoothing means connected to the secondary winding and smoothing and outputting DC power insulated from the AC power;
A first switching element and a second switching element connected in series to a DC side terminal of the rectifying means;
A series resonant circuit comprising the primary winding and a resonant capacitor connected in parallel to the first switching element or the second switching element;
Power factor improvement control means for controlling the first switching element and the second switching element to improve the power factor of the input AC power;
A charge storage means connected to the DC side terminal of the rectifier means and a third switching element connected in series to the charge storage means so as to prevent discharge of the charge storage means;
A charging circuit that boosts and charges the charge storage means with electric power from the tertiary winding;
A power failure detection means for detecting a power failure of the AC power;
When the power failure detection means detects a power failure of the AC power, the third switching element is controlled so that the charge stored in the charge storage means in advance through the insulation transformer At this time, the driving frequency of the first switching element and the second switching element is substantially equal to the normal driving frequency when the AC power failure is not detected (the charge storage means is completely charged). Charge / discharge control means for raising the voltage to the time voltage / maximum voltage of the AC power) times or more,
A switching power supply comprising: The switching power supply according to claim 4,
The charging circuit is
At least one diode connected to an input terminal of the tertiary winding;
A capacitor having a positive side connected to a cathode of the at least one diode and a negative side connected to a grounding point of the tertiary winding;
A coil connected between the capacitor and the charge storage means;
A switching power supply comprising: In the switching power supply according to claim 4 or 5,
A fourth switching element connected between the charging circuit and the charge storage means;
The charge / discharge control means includes
When the input voltage of the AC power is higher than a predetermined voltage, the third switching element is stopped, and the fourth switching element is controlled and charged until the voltage of the charge storage means reaches a set value. ,
When the input voltage of the AC power is lower than a predetermined voltage, the fourth switching element is stopped, the third switching element is controlled, and the charge previously stored in the charge storage means is Discharge to the insulated DC output side through a transformer, and at this time, the driving frequency of the first switching element and the second switching element is set to the normal driving frequency when the AC power failure is not detected. A switching power supply characterized in that the switching power supply is increased to a value approximately equal to or higher than (the voltage at the completion of charging of the charge storage means / the maximum voltage of the AC power).   The electronic device provided with the switching power supply as described in any one of Claims 1-6.

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