JP2012217121A - Digital phase-locked loop and physical quantity detection sensor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a digital phase-locked loop and a physical quantity detection sensor that implement phase locking in a simple configuration.SOLUTION: A phase comparison section calculates a phase difference between a code clock CLK1 indicating code information about a sample value and a divided clock CLK2 from a frequency divider. Specifically, the phase comparison section counts down from a rise timing of the code clock CLK1, then counts up at a fall timing of the divided clock CLK2 from the frequency divider, and outputs a count value at a rise timing of the code clock CLK1 as a phase comparison count value. An adder adds a phase correction value from a phase correction section to the phase comparison count value before outputting it to a loop filter.

Description

  The present invention relates to a digital phase synchronization circuit and a physical quantity detection sensor that output a predetermined frequency signal in phase synchronization.

  In general, when an analog circuit is replaced with a digital circuit, a processing circuit of a physical quantity detection sensor can be manufactured using a fine semiconductor manufacturing process, so that an IC chip can be reduced in size and cost can be reduced. In addition, if the input analog signal can be converted into a digital signal processing system at an early stage, the effect is greater.

  Patent Document 1 shows an example of a technology that employs an analog PLL, and FIG. 18 shows a block configuration of a PLL circuit to which the technology of Patent Document 1 is applied. The PLL circuit 1 outputs an output signal based on an input signal input through the waveform shaping circuit 2, and includes a voltage controlled oscillator (VCO) 3 and an oscillation control circuit 4.

  Here, the oscillation control circuit 4 includes a low-pass filter (LPF) 5 that generates a control voltage, a frequency divider (DIV) 6 that divides the oscillation signal of the voltage-controlled oscillator 3 to generate a divided signal, and an input A phase detection circuit (PFD) 7 that outputs a charge signal or a discharge signal based on the phase difference between the signal and the frequency-divided signal, and a charge pump that increases or decreases the control voltage of the low-pass filter 5 according to the charge signal or the discharge signal ( CP) 8. When the input signal of an analog circuit having such a block configuration is AD-converted and the analog circuit is digitized, a phase error due to digital discretization occurs in the phase comparison result of the phase detection circuit, so that accurate phase synchronization cannot be performed. Produce. In order to eliminate the phase error with this configuration, it is necessary to increase the number of samplings and perform high-speed sampling in order to increase the accuracy.

  Moreover, the digital phase locked loop circuit relevant to this application is also disclosed by patent documents 2-4. In the technical idea described in Patent Document 2, the input signal is an aperiodic signal, and a phase comparison and a frequency comparison must be performed every time a zero cross occurs aperiodically. In addition to the frequency comparator and the low-pass filter, Since it is necessary to provide a circuit for performing on / off control of the frequency comparator with a switch, the processing is complicated and the circuit size is large.

  Further, according to the technical idea described in Patent Document 3, in the digital phase synchronization circuit, the amplitude ratio of the input signal at the two time points sandwiching the zero cross point in the input signal having the amplitude value only at the time axis discrete time points is obtained to obtain the zero cross. And a phase amount signal corresponding to the time difference is output.

  However, even in this technical idea, although the phase of the input signal is calculated by zero-cross estimation, the phase of the comparison signal is calculated only by counting, so that it is necessary to speed up the counting clock for high accuracy. The timing design is difficult because it is necessary to estimate the zero-cross point and reflect it in the dense timing with the reset signal of the counter. In the technical idea described in Patent Document 4, the phase of the zero-cross point with respect to the sample point is calculated and the phase of the clock is controlled by numerical calculation processing using discrete input signals before and after the zero-cross point in the phase synchronization circuit. . However, a configuration capable of performing accurate phase synchronization with a simple configuration is desired.

JP 2009-281888 A JP-A-10-107623 JP 55-115737 A Japanese Patent Laid-Open No. 06-119720

  The present invention has been made in view of the above circumstances, and an object thereof is to provide a digital phase synchronization circuit and a physical quantity detection sensor capable of performing accurate phase synchronization with a simple configuration.

  According to invention of Claim 1, it acts as follows. The phase correction unit linearly approximates the values before and after the zero crossing of the sampling value of the AD converter and calculates a phase correction value at the zero crossing timing. On the other hand, the clock generation unit generates a code clock according to the code information of the sampling value of the AD converter, and the phase comparison unit calculates the phase difference between the code clock of the clock generation unit and the frequency-divided clock of the frequency-dividing unit. Then, it is possible to detect the phase difference between the timing at which the sign of the sampling value of the AD converter is switched and the timing of the divided clock of the frequency dividing unit.

  Since the adjustment output unit is given a value obtained by correcting the phase difference of the phase comparison unit with the phase correction value of the phase correction unit, the adjustment output unit uses the adjustment amount corresponding to this value as the input adjustment value of the control oscillation unit. The control oscillation unit can oscillate and output an N-multiplied frequency signal according to the adjustment amount.

  Therefore, the feedback loop by the phase comparison unit, addition / subtraction unit, adjustment output unit, control oscillation unit, and frequency division unit corrects the phase of the code clock generated by the clock generation unit with the phase correction value of the phase correction unit, The phase can be accurately synchronized, and the control oscillation unit can oscillate and output a phase-synchronized stable N-multiplied frequency signal. Since phase synchronization is mainly achieved by the phase correction unit and the phase comparison unit, accurate phase synchronization processing can be realized.

  In addition, the up / down counter starts counting in one direction from the count start timing by the code clock of the clock generation unit, reverses the count in the reverse direction at the count conversion timing by the frequency division clock of the frequency division unit, The count value obtained at the count end timing by the code clock is output, and this count value is used as the phase calculation value. Thereby, the phase difference between the code clock and the divided clock can be calculated with a simple configuration.

  According to the second aspect of the invention, the up / down counter counts down according to a predetermined reference clock from the rising timing of the code clock of the clock generation unit, and after the down-counting, the up / down counter uses the divided clock of the frequency division unit. When the falling timing is received, the count-up is performed according to the reference clock from the falling timing. Then, the phase comparison unit ends the up-counting by the up-down counter at the rising timing of the code clock of the clock generation unit, and outputs the obtained count value as a phase calculation value. Thereby, the phase difference between the code clock and the divided clock can be calculated with a simple configuration.

As shown in the third aspect of the present invention, the up / down counter may be configured to down-count and up-count by combining two identical counters.
According to the fourth aspect of the present invention, the first subtractor of the phase correction unit uses the delay sampling value x (n−1) (a negative number at the rising zero crossing) and the current sampling value x (n) (the rising edge of the AD converter). Subtraction value x (n) -x (n-1) can be calculated by subtracting from a positive number at the time of zero crossing. Then, when the first divider divides the current sampling value x (n) of the AD conversion unit by the subtraction value (x (n) −x (n−1)) of the first subtractor, x (n) / (x (N) -x (n-1)) can be calculated. This value corresponds to the amplitude ratio of the sampling value before and after the zero crossing of the AD converter, and corresponds to the ratio of the time from the current sampling timing to the zero crossing timing with respect to the time between the current sampling timing and the delayed sampling timing.

  The increment of the downcount of the up / down counter of the phase comparator becomes the increment of the upcount, and conversely, the increment of the downcount becomes the increment of the upcount. Therefore, the count value corresponding to the phase difference obtained by the phase comparator is counted. The count value is twice that of the clock.

  Therefore, the first multiplier obtains a value obtained by multiplying the count value of the count clock corresponding to the sampling period by -2. The divided value x (n) / (x (n) -x (n-1)) of the first divider. And the phase value obtained by matching the count value increase / decrease degrees of the phase comparison unit and the phase correction unit can be calculated. Therefore, the phase value corresponding to the zero cross timing can be accurately output by adding and subtracting the phase correction value calculated by the phase correction unit to the phase calculation value calculated by the phase comparison unit.

  According to the fifth aspect of the present invention, at least one of the first subtractor, the first divider, and the first multiplier of the phase correction unit has a calculation permission / stop switching function. At least one of the first subtractor, the first divider, and the first multiplier can be stopped, and the power consumption during the stop of the calculation can be reduced.

  According to the sixth aspect of the present invention, since the fixed value setter sets the input of the arithmetic unit to be stopped to a fixed value, for example, the switching loss caused when the transistors constituting the arithmetic unit are turned on and off is reduced. It is possible to reduce the power consumption when the computation is stopped.

  According to the seventh aspect of the present invention, when the first subtracter, the first divider, and the first multiplier hold data in synchronization with the clock, the first subtracter, the first divider, and the first divider While at least one of the operations of one multiplier is stopped, the operation clock of the operation unit to be stopped is preferably stopped. In this case, for example, switching loss at the time of turning on and off a transistor that operates according to a clock can be reduced, and power consumption when computation is stopped can be reduced.

  According to the eighth aspect of the present invention, the zero-cross detection unit detects the timing at which the sampling value of the AD converter has zero-crossed, and the first cross-phase signal of the phase correction unit is detected using the zero-cross timing signal detected by the zero-cross detection unit. A period for stopping at least one of the subtractor, the first divider, and the first multiplier is set.

According to the ninth aspect of the present invention, the period during which at least one of the first subtractor, the first divider, and the first multiplier of the phase correction unit is stopped using the code clock of the clock generation unit. Set.

  According to the tenth aspect of the present invention, the second adder of the phase correction unit is configured such that the delay sampling value | x (n−1) | (where x (n−1) is a negative number at the rising zero cross) of the second delay unit. The calculated value | x (n) | (x (n) is a positive number at the rising zero crossing) is added to the absolute value calculator, and the second divider calculates the calculated value | x (n) | Dividing by the addition value (| x (n) | + | x (n-1) |) of the two adders gives | x (n) | / (| x (n) | + | x (n-1) |) Can be calculated. Therefore, similarly to the invention according to claim 4, the second multiplier obtains a value obtained by multiplying the count value of the count clock corresponding to the sampling period by -2 by the divided value | x (n) | / ( | X (n) | + | x (n−1) |) can be used to calculate a phase value that matches the count value increase / decrease degree of the phase comparison unit and the phase correction unit, and similarly according to the zero cross timing. The phase value can be output accurately.

  According to the eleventh aspect of the present invention, at least one of the second subtractor, the second divider, and the second multiplier of the phase correction unit has a function of switching between operation permission / operation stop. , The second subtractor, the second divider, and the second multiplier can stop the operation, and the power consumption during the operation stop can be reduced.

  According to the twelfth aspect of the present invention, the fixed value setter sets the input of the computing unit to be stopped to a fixed value, so that, for example, the switching loss caused when the transistors constituting the computing unit are turned on and off is reduced. It is possible to reduce power consumption while the computation is stopped.

  According to a thirteenth aspect of the present invention, when the second subtractor, the second divider, and the second multiplier are configured to hold data in synchronization with the clock, the second subtracter, the second divider, and the second divider While at least one of the operations of the two multipliers is stopped, the operation clock of the operation unit to be stopped is preferably stopped. In this case, for example, switching loss at the time of turning on and off a transistor that operates according to a clock can be reduced, and power consumption during computation stop can be reduced.

  According to the fourteenth aspect of the present invention, the zero-cross detection unit detects the timing at which the sampling value of the AD converter has zero-crossed. The second cross-phase signal of the phase correction unit is detected using the zero-cross timing signal detected by the zero-cross detection unit. A period during which at least one of the subtracter, the second divider, and the second multiplier is stopped is set.

  According to the fifteenth aspect of the present invention, the period in which at least one of the second subtracter, the second divider, and the second multiplier of the phase correction unit is stopped using the code clock of the clock generation unit. Set.

  The invention may be applied to a physical quantity detection sensor including an AGC, a signal detection unit, and the digital phase synchronization circuit according to any one of claims 1 to 15 as in the invention described in claim 16.

The block diagram which shows the electrical constitution of the vibration type angular velocity sensor in 1st Embodiment of this invention. Hardware diagram of phase synchronization circuit Hardware diagram of phase correction unit Illustration of phase correction value The figure which shows the correspondence of code information and code clock Hardware diagram of phase comparator Explanatory drawing of the phase comparison count value of a phase comparison part ((a) is a reference output, (b) is a phase delay, (c) is a phase advance) Explanatory drawing of the phase comparison count value of a phase comparison part ((a) is a reference output, (b) is high frequency, (c) is low frequency) FIG. 3 equivalent view showing a second embodiment of the present invention. FIG. 6 equivalent view showing a third embodiment of the present invention. FIG. 6 equivalent view showing a fourth embodiment of the present invention. FIG. 6 equivalent view showing a fifth embodiment of the present invention. Explanatory drawing which shows 6th Embodiment of this invention ((a) is an equivalent figure of FIG. 3, (b) (c) is a structural example of a data mask circuit.) Timing chart schematically showing the operation FIG. 3 equivalent view showing a seventh embodiment of the present invention Timing chart schematically showing the operation FIG. 9 equivalent view showing an eighth embodiment of the present invention. FIG. 2 equivalent diagram showing a conventional example

(First embodiment)
Hereinafter, a first embodiment in which a physical quantity detection sensor of the present invention is applied to a vibration type angular velocity sensor will be described with reference to the drawings. In the vehicle, the spin and drift of the vehicle are suppressed by appropriately controlling the drive torque and brake force of each wheel from the information of the sensors of the steering angle, wheel speed, and acceleration. Therefore, the vibration type angular velocity sensor 10 is an important sensor for improving the vehicle performance.

FIG. 1 shows an electrical block configuration of the vibration type angular velocity sensor. The vibration type angular velocity sensor 10 is used especially as a yaw rate sensor for a vehicle, and is configured by connecting a vibrator 11, a signal detection circuit 12, a phase synchronization circuit 13, a timing generation unit 13a, an AGC 14, and an angular velocity detection unit 15. The The vibrator 11 vibrates in the reference direction when a drive signal is given from the AGC 14. When an angular velocity Ω is applied to the vibrator 11 that vibrates, Coriolis force acts in the detection direction orthogonal to the reference direction. For example, when an object of mass m moves with a velocity vector v on a plane rotating with an angular velocity vector Ω, the Coriolis force Fc generated on the object is
Fc = 2mv × Ω (× indicates an outer product) (1)
It is represented by The signal detection circuit 12 is a circuit that shapes the waveform of the vibration signal in the reference direction. The vibrator 11 includes electrodes, and the signal detection circuit 12 detects a change in the capacitance of the electrode of the vibrator 11 that vibrates in the reference direction as a voltage value by the CV conversion circuit, thereby causing a vibration signal in the reference direction (hereinafter referred to as a reference value). Monitor signal) and output through a filter, amplifier, etc.

  A digital phase synchronization circuit (hereinafter referred to as a phase synchronization circuit) 13 is a so-called digital PLL (Phase Locked Loop) circuit that detects a phase difference between an input signal and an output signal and oscillates and outputs a synchronized frequency signal. The detailed configuration of the phase synchronization circuit 13 will be described later. The timing generator 13a generates timing signals required for the operation of the sensor circuit, such as a drive timing signal, a sampling clock, and a count clock. Based on the output signal amplitude of the signal detection circuit 12, an AGC (Auto Gain Control) 14 is used for the drive timing signal of the timing generator 13a so that the vibration in the reference direction of the vibrator 11 becomes constant in a desired vibration state. A drive signal whose amplitude is controlled is applied to the vibrator 11.

  The angular velocity detection unit 15 detects an angular velocity vibration signal corresponding to the Coriolis force generated in the vibrator 11 as a voltage value by a process equivalent to the signal detection circuit 12, and uses a synchronous detection circuit and an LPF (Low Pass Filter). Only the angular velocity vibration component is extracted and output as angular velocity information (sensor signal).

  The vibrator 11 has a natural frequency, and in order to efficiently detect the angular velocity vibration signal, a drive signal having an appropriate frequency may be applied to the vibrator 11, and the drive signal is appropriately set using the phase synchronization circuit 13. It is good to control to a proper frequency and phase.

FIG. 2 shows a schematic block diagram of the electrical configuration of the phase synchronization circuit. In the present embodiment, an embodiment in which all components are configured by digital blocks is shown.
The phase synchronization circuit 13 includes an AD converter 16, a phase correction unit 17, a clock generation unit 18, a phase comparison unit 19, an adder 20 as an addition / subtraction unit, a loop filter 21 as an adjustment output unit, and a DCO (control oscillator). A digitally controlled oscillator) 22 and a frequency divider 23 as a frequency divider are connected. The phase synchronization circuit 13 outputs a PLL output clock to the AGC 14 according to the sampling value obtained by sampling the monitor signal by the AD converter 16.

  The AD converter 16 samples and holds the monitor signal in accordance with the given sampling clock, and uses it as the input digital value of the phase correction unit 17 and the clock generation unit 18. The phase correction unit 17 outputs a phase correction value for correcting the phase of the zero cross timing according to the sampling value before and after the sampling value is zero crossed. A specific configuration example of the phase correction unit 17 is shown in FIG.

  As shown in FIG. 3, the phase correction unit 17 includes a delay unit (corresponding to the first delay unit) 24, a subtracter (corresponding to the first subtractor) 25, a divider (corresponding to the first divider) 26, and multiplication. A hardware (a first multiplier) 27, a sample hold circuit 28, and a rising zero-cross detection circuit 29 are combined. The phase correcting unit 17 calculates the phase value of the zero cross timing by linearly approximating the sampling value of the AD converter 16 before and after the zero cross.

  The delay unit 24 delays the sampling value x (n) of the AD converter 16 by one sampling clock and outputs it to the subtracter 25. The subtracter 25 subtracts the delayed sampling value x (n−1) delayed by one sampling clock from the current sampling value x (n) of the AD converter 16 (that is, x (n) − (− | x (n− 1) |)) and output to the divider 26.

  The divider 26 divides the current sampling value x (n) by the result x (n) −x (n−1) obtained by subtracting by the subtractor 25 and outputs the result to the multiplier 27. The multiplier 27 sets the constant (a value obtained by multiplying the count value of the count clock corresponding to the sampling period by -2) to the division value x (n) / (x (n) −x (n−1)) of the divider 26. Multiply and give to the sample and hold circuit 28. The sample hold circuit 28 samples and holds the multiplication value of the multiplier 27 at the timing detected by the rising zero cross detection circuit 29.

  FIG. 4 shows an operation explanatory diagram of the phase correction unit 17. FIG. 4 shows an example in which, for example, the maximum frequency (minimum cycle) of the analog value of the monitor signal is sampled eight times (one digit number sampling). The sampling clock is synchronized with the count clock, and the sampling clock cycle is set to a time several tens to several hundred times the count clock cycle.

  If the sampling values before and after the zero cross timing are x (n-1) and x (n), the amplitude ratio of these x (n-1) and x (n) is the AD conversion of x (n-1). The ratio of the time T1 from the timing to the zero crossing and the time T2 from the zero crossing timing to the x (n) AD conversion timing is substantially the same. Therefore, the time corresponding to the time T2 can be calculated by multiplying the sampling interval of the AD converter 16 by x (n) / (x (n) −x (n−1)).

  The rising zero-cross detection circuit 29 outputs a sampling pulse to the sample hold circuit 28 at a timing when the current sampling value x (n) is a positive number and the delay sampling value x (n−1) is a negative number. At this timing, x (n) / (x (n) −x (n−1)) × constant (count number of count clock corresponding to sampling period × (−2)) is acquired.

  Here, “−1” in (−2) indicates that the time goes back from the sampling point A in FIG. 4, and “2” indicates a coefficient value depending on the system. This constant is set in accordance with the counting method (see later) of the phase comparison count value of the count clock of the clock generator 18 and the phase comparator 19.

  Now, when the correction method by the phase correction unit 17 is applied, the zero cross timing when the sampling points are linearly approximated can be calculated. For example, the vicinity of the zero cross of the sine wave can be approximated and obtained when approximated. In the present embodiment, by providing the clock generation unit 18, the phase comparison unit 19, and the adder 20, accurate zero cross timing can be easily derived even if the number of samplings of the AD converter 16 is small. Hereinafter, description will be given with reference to FIG.

  The clock generation unit 18 binarizes the sign information of the sampling value of the AD converter 16 of the monitor signal and outputs it to the phase comparison unit 19. For example, “+” where the sampling value of the AD converter 16 is 0 or more. "H" level is output when the signal is "L", and "L" level is output when "-" is less than zero. Thereby, the code clock CLK1 is generated. FIG. 5 shows the relationship between the sign information of the sampling value of the AD converter 16 and the signal waveform of the sign clock CLK1. The rise timing of the code clock CLK1 has an error corresponding to the discretization.

  In order to perform accurate phase comparison, it is necessary to make the sampling frequency of the AD converter 16 faster. In order to perform accurate phase comparison while preventing high-speed sampling, the above-described phase correction unit 17 is provided, and a phase comparison unit 19 and an adder 20 described later are further configured.

  As shown in FIG. 2, the phase comparison unit 19 is provided with a sign clock CLK <b> 1 and a frequency-divided clock CLK <b> 2 obtained by frequency-dividing the PLL output clock of the DCO 22 by the frequency divider 23. The DCO 22 oscillates and outputs a PLL output clock obtained by multiplying the code clock CLK1 by N as an N-multiplied frequency signal, and the frequency divider 23 divides the PLL output clock by N and outputs the divided clock CLK2 to the phase comparator 19.

  FIG. 6 shows a hardware configuration of the phase comparison unit 19. The phase comparison unit 19 includes an up / down selection circuit 30, a selector 31, an adder 32, a sample hold circuit 33, a rising edge detection circuit 34, and a sample hold circuit 35. The count clock is supplied to the up / down selection circuit 30, the selector 31, the sample hold circuit 33, and the rising edge detection circuit.

  The up / down selection circuit 30 receives the sign clock CLK1 and the divided clock CLK2, and switches and outputs a selection signal at the input timing of the count clock. This up / down selection circuit 30 determines whether or not the sign clock CLK1 has risen at the rise timing of the count clock, and when the rising edge of the sign clock CLK1 is detected, the selector 31 selects “−1” for selection. Output a signal.

  Further, the up / down selection circuit 30 determines whether or not the divided clock CLK2 has fallen at the rising timing of the count clock, and selects “+1” for the selector 31 when detecting the falling of the divided clock CLK2. A selection signal for output.

  The selector 31 outputs “−1” or “+1” according to the selection signal, and the adder 32 adds the output value (“−1” or “+1” of the selector 31 to the output value held by the sample hold circuit 33. The sample hold circuit 33 samples and holds the output value at the next rise timing of the count clock.

  Therefore, the up / down selection circuit 30 changes the counting direction in a predetermined direction (decrease direction in the present embodiment) in accordance with the change of the sign clock CLK1, and changes the divided clock CLK2 (count conversion timing). ), The counting direction is changed in the direction opposite to the one direction (in the case of the present embodiment, the increasing direction).

  Then, the adder 32 and the sample hold circuit 33 add the increment value (“1”) or the decrease value (“−1”) of the selector 31 at the rising timing of the count clock. That is, the selector 31, the adder 32, and the sample hold circuit 33 operate as an up / down counter 36, and output the count value to the sample hold circuit 35 as a phase comparison count value.

  The rising edge detection circuit 34 detects the rising edge of the code clock CLK 1 at the rising timing of the count clock and outputs it as the sample timing of the sample hold circuit 35. The sample hold circuit 35 samples and holds the phase comparison count value at the rising edge timing of the code clock CLK1 detected by the rising edge detection circuit 34, and outputs it as a phase value.

  FIG. 7A to FIG. 7C show the relationship between each clock and the phase comparison count value when the frequency and phase are synchronized, when phase delay occurs, or when phase advance occurs. . FIGS. 8A to 8C show the clocks when the frequency and phase are synchronized, when the frequency of the divided clock CLK2 is high, and when the frequency of the divided clock CLK2 is low. The relationship of the phase comparison count value is shown.

  As shown in FIGS. 7 (a) to 7 (c) and FIGS. 8 (a) to 8 (c), when the sign clock CLK1 rises, the up / down counter 36 starts down counting and counts down. Every time a clock is received, it counts down. When the falling edge of the frequency-divided clock CLK2 is detected, the count is stopped. Thereafter, the count-up is performed every time the count clock is received, and when the code clock CLK1 rises again, the sample hold circuit 35 converts the phase comparison count value to the phase value. As a sample hold output, the output is cleared.

  Therefore, as shown in FIG. 7A or FIG. 8A, when the sign clock CLK1 and the divided clock CLK2 have the same frequency and the same phase, for example, when the phase comparison count value is, for example, 0 as a reference output. The sample hold circuit 35 samples and holds and outputs it as a phase value.

  Further, as shown in FIG. 7B, when the phase of the divided clock CLK2 is delayed compared to the sign clock CLK1, the downcount number is larger than the reference output, and conversely, the upcount number is the reference. The sample hold circuit 35 samples and holds and outputs as a phase value when a value less than the output and a value less than the reference output (for example, a negative value) is output as the phase comparison count value.

  At this time, since the finally obtained phase value is a count number obtained by adding the increase count of the down count to the decrease count of the up count, it is counted twice as many as the count clock number corresponding to the actual phase delay. Become. In the schematic example shown in FIG. 7B, the divided clock CLK2 is delayed by “3” counts from the sign clock CLK1, and the phase value becomes “−6”.

  On the contrary, as shown in FIG. 7C, when the phase of the divided clock CLK2 advances compared to the sign clock CLK1, the down count number becomes smaller than the reference output, and conversely, the up count number is the reference. Since it exceeds the output, the sample hold circuit 35 samples and holds the phase comparison count value when it outputs a value (for example, a positive value) greater than the reference output as the phase comparison count value.

  As described above, the finally obtained phase value is a count number obtained by adding the increase number of the up count to the decrease number of the down count, and thus is counted twice the number of count clocks corresponding to the actual phase advance. It will be. In the schematic example shown in FIG. 7C, the frequency-divided clock CLK2 advances by “3” counts to the sign clock CLK1, and the phase value becomes “+6”.

  Similarly, as shown in FIG. 8B, when the frequency of the divided clock CLK2 is higher than that of the sign clock CLK1, the downcount number is smaller than the reference output, and conversely, the upcount number is the reference output. Therefore, a larger value (for example, a positive value) than the reference output is output as the phase value.

  At this time, the output phase value is a count number obtained by adding the increase number of the up count to the decrease number of the down count, and therefore is counted twice the number of count clocks corresponding to the actual frequency deviation. In the schematic example shown in FIG. 8B, the frequency-divided clock CLK2 has a frequency twice that of the sign clock CLK1, the count clock number has advanced by “3”, and the phase value becomes “+6”. .

  Further, as shown in FIG. 8C, when the frequency of the divided clock CLK2 is lower than that of the sign clock CLK1, the down count number is larger than the reference output, and conversely, the up count number is larger than the reference output. Since it decreases, a negative value is output as the phase value.

  The phase value output at this time is a count number obtained by adding the decrease count of the up count to the increase count of the down count, and is therefore counted twice the count clock number corresponding to the actual frequency deviation. In the example of FIG. 8C, the frequency-divided clock CLK2 has a frequency half that of the sign clock CLK1, the count clock number is delayed by “3” counts, and the phase value is “−6”.

  The phase comparison count value of the phase comparison unit 19 counts twice the count clock, and the phase correction value by the phase correction unit 17 corresponding to the time T2 described above is associated with the count clock and counted twice. ing. Therefore, in the above description, the coefficient value dependent on the up / down counting system of “2” is adopted.

<Summary of First Embodiment>
The phase correction unit 17 linearly approximates the values before and after the zero crossing of the sampling value of the AD converter 16 to calculate a phase correction value at the zero crossing timing. On the other hand, the clock generation unit 18 encodes the sign information of the sampling value of the AD converter 16. The phase comparator 19 calculates the phase difference between the code clock CLK1 of the clock generator 18 and the frequency-divided clock CLK2 of the frequency divider 23. Thereby, the phase difference between the timing at which the sign of the input sampling value of the AD converter 16 is switched and the timing of the divided clock of the frequency divider 23 can be detected.

  Since the loop filter 21 is given a value obtained by correcting the phase difference of the phase comparison unit 19 with the phase correction value of the phase correction unit 17, the loop filter 21 outputs an adjustment amount corresponding to the output value to the DCO 22, The DCO 22 can oscillate and output an N-multiplied frequency signal in accordance with the adjustment amount.

  Therefore, the feedback loop including the phase comparison unit 19, the adder 20, the loop filter 21, the DCO 22, and the frequency divider 23 corrects the phase of the code clock CLK 1 generated by the clock generation unit 18 with the phase correction value of the phase correction unit 17. Therefore, phase synchronization can be achieved, and the DCO 22 can oscillate and output a phase-stable and stable N-multiplied frequency signal. Therefore, since the phase correction unit 17 and the phase comparison unit 19 are mainly configured to achieve phase synchronization, accurate phase synchronization can be realized.

  The up / down counter 36 starts counting in one direction from the count start timing by the sign clock CLK1 of the clock generator 18, reverses the count in the reverse direction at the count conversion timing by the divided clock CLK2 of the frequency divider 23, The count value obtained at the count end timing by the code clock CLK1 of the clock generator 18 is output as a phase calculation value. Thereby, the phase difference between the sign clock CLK1 and the divided clock CLK2 can be calculated with a simple configuration.

  Specifically, the up / down counter 36 counts down according to the count clock from the rising timing of the sign clock CLK1 of the clock generator 18, and the up / down counter 36 divides the frequency of the frequency divider 23 after being counted down. When the falling timing by the clock CLK2 is received, the count-up is performed according to the count clock from the falling timing. Then, the phase comparison unit 19 outputs a count value obtained by completing the counting by the up counter at the rising timing of the code clock CLK1 of the clock generation unit 18 as a phase calculation value. Thereby, the phase difference between the sign clock CLK1 and the divided clock CLK2 can be calculated with a simple configuration.

  When the subtracter (adder / subtractor) 25 of the phase correction unit 17 subtracts the delay sampling value and the current sampling value of the AD converter 16, x (n) -x (n-1) can be calculated. Then, when the divider 26 divides the current sampling value of the AD converter 16 by the addition / subtraction value of the subtracter (adder / subtractor) 25, x (n) / (x (n) −x (n−1)) can be calculated. .

  The increment of the down count of the up / down counter 36 of the phase comparator 19 becomes the increment of the up count, and conversely, the increment of the down count becomes the increment of the up count. Therefore, the count corresponding to the phase difference obtained by the phase comparator 19 The value is a count value that is twice the count clock.

  Therefore, if the multiplier 27 multiplies the value obtained by multiplying the count value of the counter clock corresponding to the sampling period by −2 with the division value of the divider 26, the count value of the count value of the phase comparison unit 19 and the phase correction unit 17 It can be calculated by matching the degree of increase / decrease, and by adding the phase correction value calculated by the phase correction unit 17 to the phase calculation value calculated by the phase comparison unit 19, the phase value corresponding to the zero cross timing can be output accurately. .

  With the above operation, accurate phase synchronization is possible even when a phase error due to discretization occurs in low-speed sampling. Further, since the phase correction accuracy is determined by the resolution of the AD converter 16, the count clock speed can be minimized by setting an appropriate AD resolution.

  The technical idea described in Patent Document 2 includes a circuit that performs on / off control of the frequency comparator with a switch in addition to the frequency comparator and the low-pass filter, so that the processing becomes complicated and the circuit size increases. In the present embodiment, there is no need to provide a circuit for turning on and off the frequency comparator with a switch, and it is simpler and more compact. Further, since a high-speed clock is not required, the phase correction unit 17 and the phase comparison unit 19 can be reduced in size and power consumption.

  In addition, since the phase correction value calculated by the phase correction unit 17 can be performed only by adding digital values instead of a precise timing operation (for example, circuit setting, resetting, etc.), circuit design is simplified.

(Second Embodiment)
FIG. 9 shows a second embodiment of the present invention. The difference from the previous embodiment is that the configuration of the phase correction unit is changed. The same parts as those of the above-described embodiment are denoted by the same reference numerals and description thereof is omitted, and different parts will be described below.

  FIG. 9 shows a modification of the phase correction unit. The phase correction unit 37 includes an absolute value calculator 38, a delay unit (corresponding to the second delay unit) 39, an adder (corresponding to the second adder) 40, a divider (corresponding to the second divider) 41, and a multiplier. (Corresponding to a second multiplier) 42 and the like are combined, and a phase correction value is output via a sampling hold circuit (not shown). The sampling value of the monitor signal by the AD converter 16 is input to the absolute value calculator 38.

  The absolute value calculator 38 calculates the absolute value of the sampling value and outputs the calculated value to the delay unit 39, the adder 40, and the divider 41. The delay unit 39 delays the calculation value of the absolute value calculation unit 38 by one sampling clock and outputs it to the adder 40. The adder 40 adds the absolute value | x (n−1) | of the delayed sampling value and the absolute value | x (n) | of the current sampling value and outputs the result to the divider 41. The divider 41 divides the absolute value | x (n) | of the current sampling value by the addition value | x (n) | + | x (n−1) | of the adder 40 and outputs the result to the multiplier 42. The multiplier 42 divides the division value | x (n) | / (| x (n) | + | x (n−1) |) by the divider 41 and a constant (the count value of the count clock corresponding to the sampling period − A value obtained by multiplying (doubled value) is output as a phase correction value. Then, | x (n) | / (| x (n) | + | x (n−1) |) × constant (count number of count clock corresponding to sampling period × (−2)) can be calculated.

  In this case, if sampling is performed at the rising timing (“L” → “H”) of the code clock CLK1 immediately after the zero crossing timing output by the clock generation unit 18 and output as a phase correction value, the phase correction unit as in the above-described embodiment. 17 phase correction values can be acquired appropriately.

(Third embodiment)
FIG. 10 shows a third embodiment of the present invention. The difference from the previous embodiment is that the configuration of the up / down counter of the phase comparison unit is changed.

  FIG. 10 shows an electrical configuration of a phase comparison unit 43 that replaces the phase comparison unit 19 of the first embodiment. An up / down counter 44 shown in FIG. 10 is configured in place of the up / down counter 36 of the first embodiment. When an up / down selection circuit 30 receives an up / down count selection signal, the up / down counter 44 switches the function of the up / down counter, and counts up or down according to the input count clock pulse. The phase comparison count value is output to the sample hold circuit 35.

  The up / down counter 44 is configured by connecting flip-flops such as JK flip-flops, for example. Even if the phase comparison unit 43 is configured in this way, the same effects as those of the above-described embodiment can be obtained.

(Fourth embodiment)
FIG. 11 shows a fourth embodiment of the present invention. The difference from the previous embodiment is that the up / down counter of the phase comparison unit is configured by combining two up counters. The same parts as those of the above-described embodiment are denoted by the same reference numerals and description thereof is omitted, and different parts will be described below.

  FIG. 11 shows the electrical configuration of the phase comparison unit 45. The phase comparison unit 45 includes a counter selection circuit 46 that replaces the up / down selection circuit 30 of the above-described embodiment. Further, an up / down counter 47 in place of the up / down counter 36 or 44 is provided.

  The counter selection circuit 46 outputs an “H” level as a valid / invalid switching signal (enable signal) when receiving the rising edge of the sign clock CLK1, and receives a valid / invalid switching signal (enable signal) when receiving the falling edge of the divided clock CLK2. The “L” level is output to the up / down counter 47 as an enable signal).

  The up / down counter 47 includes two up counters 48a and 48b. The up counter 48a adds “+1” to the output of the sample hold circuit 51a by the adder 50a, and inputs the result to the sample hold circuit 51a again. The up counter 48a inputs a count clock to the clock terminal, and counts up according to the pulse of the count clock.

  The sample hold circuit 51a of one up counter 48a is provided with an enable terminal EN, and an enable / disable switching signal (enable signal) output from the counter selection circuit 46 is input to the enable terminal EN.

  On the other hand, the up counter 48b is configured to add “+1” to the output of the sample and hold circuit 51b by the adder 50b and input it again to the sample and hold circuit 51b. The up counter 48b inputs a count clock to the clock terminal, and counts up according to the pulse of the count clock.

  The sample hold circuit 51b of the other up counter 48b is also provided with an enable terminal EN, and an enable / disable switching signal (enable signal) output from the counter selection circuit 46 is passed through the inverting gate 49 to the enable terminal EN. Have been entered.

  That is, one of the two up counters 48a and 48b operates in accordance with the valid / invalid switching signal (enable signal) output from the counter selection circuit 46. The outputs of these up counters 48a and 48b are given to a subtractor 52. The subtracter 52 subtracts the output value of the up counter 48a from the output value of the up counter 48b, and outputs it to the sample hold circuit 35 as a phase comparison count value. The sample hold circuit 35 samples and holds in accordance with the output pulse of the rising edge detection circuit 34 and outputs it as a phase value.

  Therefore, by switching the two up counters 48a and 48b and causing them to function as an up counter and a down counter, respectively, the same operation as in the previous embodiment can be realized. Even if the up / down counter 47 of the phase comparison unit 45 includes the two up counters 48a and 48b as in the present embodiment, the same operational effects as those of the previous embodiment can be obtained.

(Fifth embodiment)
FIG. 12 shows a fifth embodiment of the present invention. The difference from the fourth embodiment is that two up counters are separately configured.
FIG. 12 shows an electrical configuration of a phase comparator 53 that replaces the phase comparator 45 of the fourth embodiment. The up / down counter 54 shown in FIG. 12 is configured in place of the up / down counter 47 of the fourth embodiment. The up / down counter 54 includes two up counters 55 a and 55 b together with the inverting gate 49 and the subtractor 52. That is, in this embodiment, up counters 55a and 55b are provided instead of the up counter 48a and the up counter 48b of the fourth embodiment.

  Each of the up counters 55a and 55b is configured by connecting flip-flops such as JK flip-flops, for example. Even if the phase comparator 53 is configured in this way, the same effects as those of the above-described embodiment can be obtained.

(Sixth embodiment)
FIGS. 13 (a) to 13 (c) and FIG. 14 show a sixth embodiment of the present invention. The difference from the first embodiment is that a subtracter 25 (in the first subtractor) of the phase correction unit. Equivalent) and the divider 26 (corresponding to the first divider) are configured to be capable of switching between calculation enable / stop calculation. In particular, the subtractor 25 and the divider 26 are characterized in that the calculation is stopped in the invalid period (low period) of the rising zero cross detection signal detected by the rising zero cross detection circuit 29. Parts that are the same as or similar to those in the previous embodiment are assigned the same or similar reference numerals, and descriptions thereof are omitted. Hereinafter, different parts will be described.

  FIG. 13A schematically shows a block configuration of a phase correction unit instead of FIG. As shown in FIG. 13A, a phase correction unit 17a in place of the phase correction unit 17 includes a delay unit 24, a subtracter 25, a divider 26, a multiplier 27, a sample hold circuit 28, and a rising zero cross detection circuit 29. The data mask circuits 56 to 58 are added and combined as a fixed value setter.

  As described in the above embodiment, the phase correction unit 17a calculates the phase value of the zero cross timing by linearly approximating the sampling value of the AD converter 16 before and after the zero crossing. Since the calculation function is not required during a period other than the period when the sampling value is acquired and the phase value is calculated, the calculation functions of the subtracter 25 and the divider 26 are stopped by providing the data mask circuits 56 to 58.

  A configuration example of each of the data mask circuits 56 to 58 is shown in FIG. 13 (b) or FIG. 13 (c). The data mask circuits 56 to 58 may be configured by an AND gate 59 as shown in FIG. 13B, or a combination of a NOT gate 60 and an OR gate 61 as shown in FIG. 13C. Also good.

  When the circuit example shown in FIG. 13B is applied, the input data is output as it is when the output of the rising zero-cross detection circuit 29 is high, but the data mask circuit 56 when the output of the rising zero-cross detection circuit 29 is low. The output of ˜58 can be fixed (masked) to “low”.

  When the circuit example shown in FIG. 13C is applied, the input data is output as it is when the output of the rising zero cross detecting circuit 29 is high, but when the output of the rising zero cross detecting circuit 29 is low, the data mask circuit 56 is output. The output of ˜58 can be fixed (masked) to “high”.

  Therefore, as shown in the timing chart of FIG. 14, when the sampling value of the monitor signal changes from negative to positive, the sign information is switched from negative to positive, and the output of the rising zero-cross detection circuit 29 becomes “high”. The data mask circuits 56 to 58 provide input data to the subtracter 25 and the divider 26, respectively, without masking the input data.

  However, in other periods (low period of the output of the rising zero cross detection circuit 29), the data mask circuits 56 to 58 mask the input data, so that the data mask circuits 56 to 58 include the subtractor 25 and the divider. A fixed value is given to each of 26.

  The subtractor 25 and the divider 26 are each configured by combining a plurality of switching elements, and these arithmetic units perform arithmetic processing by switching on and off a large number of switching elements, but switching loss increases when arithmetic processing is performed. .

  Therefore, while a fixed value is given to the subtractor 25 and the divider 26, switching elements are not switched on and off, so that no switching loss occurs (see the low power consumption period in FIG. 14). Thereby, in the low power consumption period, the subtracter 25 and the divider 26 do not perform arithmetic processing, and the power consumption can be reduced.

As shown in FIG. 14, since the subtractor 25 and the divider 26 only perform the arithmetic processing once in one cycle of the monitor signal, the power consumption can be reduced.
(Seventh embodiment)
15 and 16 show a seventh embodiment of the present invention. The difference from the first embodiment is that a subtractor 25 (corresponding to the first subtractor) and a divider 26 (first subtractor) of the phase correction unit. (Corresponding to a divider) is set as the operation stop target, and is configured to be switchable between operation permission and operation stop. In particular, the subtractor 25 and the divider 26 are characterized in that the calculation is stopped in the low period (negative period of the code information) of the code clock CLK1 generated by the clock generator 18. Parts that are the same as or similar to those in the previous embodiment are assigned the same or similar reference numerals, and descriptions thereof are omitted. Hereinafter, different parts will be described.

  FIG. 15 schematically shows a block configuration of a phase correction unit instead of FIG. As in the previous embodiment, the phase correction unit 17a is provided with data mask circuits 56 to 58, and the code mask CLK1 generated by the clock generation unit 18 is used as a mask signal in these data mask circuits 56 to 58. Is given.

  As shown in the timing chart of FIG. 16, while the clock generator 18 outputs the sign clock CLK <b> 1 high, each data mask circuit 56 to 58 supplies the input data as it is to the subtracter 25 and the divider 26. However, while the clock generation unit 18 outputs the code clock CLK1 low, each of the data mask circuits 56 to 58 masks the input data and supplies a fixed value to the subtracter 25 and the divider 26. Therefore, while fixed values are given to the subtractor 25 and the divider 26, the power consumption can be reduced as in the above-described embodiment.

(Eighth embodiment)
FIG. 17 shows an eighth embodiment of the present invention. The difference from the previous embodiment is that the adder 40 (corresponding to the second adder) and the divider 41 (second division) described in the second embodiment. The feature of the sixth embodiment or the seventh embodiment is applied. The same or similar parts as those in the above-described embodiment are denoted by the same or similar reference numerals, and the description thereof is omitted. Hereinafter, different parts will be described.

  FIG. 17 schematically shows a block configuration of a phase correction unit instead of FIG. As shown in FIG. 17, a phase correction unit 37a instead of the phase correction unit 37 includes a data mask as a fixed value setter together with an absolute value calculator 38, a delay unit 39, an adder 40, a divider 41, and a multiplier 42. It is configured by hardware in which circuits 62 to 64 are added and combined, and outputs a phase correction value via a sampling hold circuit (not shown).

  Each of the data mask circuits 62 to 64 is configured by a circuit similar to the data mask circuits 56 to 58 described in the above embodiment, and these data mask circuits 62 to 64 correspond to the low power consumption control signal. The input data is masked and output to the adder 40 and the divider 41.

  This “low power consumption control signal” may be the output of the rising zero cross detection circuit 29 as shown in the sixth embodiment, or the code of the clock generator 18 as shown in the seventh embodiment. It may be the output of the clock CLK1.

  That is, as described in the sixth embodiment, when the rising zero cross detection signal of the rising zero cross detection circuit 29 is in the high period, the data mask circuits 62 to 64 output the input data as they are to the adder 40 and the divider 41, When the zero-cross detection signal is in the low period, the data mask circuits 62 to 64 mask the input data and output a fixed value to the adder 40 and the divider 41. In this case, low power consumption can be achieved as in the above description.

  Further, as described in the seventh embodiment, when the output of the sign clock CLK1 of the clock generation unit 18 is in the high period, the data mask circuits 62 to 64 output the input data as they are to the adder 40 and the divider 41, When the output of the sign clock CLK1 is in the low period, the data mask circuits 62 to 64 mask the input data and output a fixed value to the adder 40 and the divider 41. In this case as well, low power consumption can be achieved.

(Other embodiments)
The present invention is not limited to the above-described embodiments, and for example, the following modifications or expansions are possible.
The vibration type angular velocity sensor 10 may be applied to any of ceramic tuning fork type, silicon tuning fork type, capacitive type, and the like.
The up / down counters 36, 44, 47, and 54 are configured to start counting up at the count conversion timing after counting down, respectively, but are configured to start counting down at the count conversion timing after counting up. You may apply to.

  In the above-described embodiment, the phase synchronization circuit 13 is an embodiment in which all the components are configured by digital blocks. However, a part of them may be configured by analog blocks. For example, the loop filter 21 may be an analog low-pass filter, and the DCO 22 may be a VCO (Voltage Controlled Oscillator). In such a case, a D / A converter, an A / D converter, or the like may be inserted separately.

  In the sixth embodiment and the seventh embodiment, the subtracter 25 and the divider 26 are configured as operation stop targets and the operation is allowed to be enabled / stopped. However, only one of them is switched. It may be configured so that other arithmetic units (for example, the multiplier 27 and the like) may have a calculation stop function. Similarly, in the eighth embodiment, the adder 40 and the divider 41 are configured so that the calculation can be permitted / stopped. However, the adder 40 and the divider 41 can be switched by only one of them. In addition, other arithmetic units (for example, the multiplier 42) may have a calculation stop function.

  In particular, in the first embodiment, when the subtractor 25, the divider 26, and the multiplier 27 are configured to hold data in synchronization with the clock, at least one of the subtractor 25, the divider 26, and the multiplier 27 is used. While one operation is stopped, the operation clock of the operation unit to be stopped may be stopped.

  Similarly in the second embodiment, the adder 40, the divider 41, and the multiplier 42 are configured to hold data in synchronization with the clock. While at least one of the operations is stopped, the operation clock of the operation unit to be stopped may be stopped. In such a case, the switching loss at the on / off time of the transistor that operates according to the operation clock of the arithmetic unit can be reduced, and the power consumption during the operation stop can be reduced.

  In the drawings, 10 is a vibration type angular velocity sensor (physical quantity detection sensor), 11 is a vibrator, 12 is a signal detection circuit (signal detection unit), 13 is a phase synchronization circuit, 13a is a timing generation unit, 14 is AGC, and 15 is an angular velocity. Detection unit, 16 AD converter, 17, 37 phase correction unit, 18 clock generation unit, 19, 43, 45, 53 phase comparison unit, 20 adder (addition / subtraction unit), 21 loop filter (adjustment) (Output unit), 22 is a DCO (control oscillator), 23 is a frequency divider (frequency divider), 24 is a delay device (first delay device), 25 is a subtractor (first subtractor), and 26 is a divider. (First divider), 27 is a multiplier (first multiplier), 36, 44, 47 and 54 are up / down counters, 38 is an absolute value calculator, 39 is a delay unit (second delay unit), 40 is Adder (second adder), 41 is a divider (second divider), 42 is a multiplier Vessel (second multiplier), 48a, 48b, 55a, 55b are up-counter, 56～58,62～64 shows the data masking circuit (fixed value setting device).

Claims (16)

A control oscillation unit for oscillating and outputting an N-multiplied frequency signal adjusted according to an input adjustment value;
A frequency divider that outputs a frequency-divided clock obtained by dividing the N-multiplied frequency signal of the control oscillator by N;
A phase correction unit that linearly approximates a digital value before and after the zero crossing of the sampling value of the AD converter and calculates a phase correction value of the zero crossing timing;
A clock generation unit that generates a code clock according to code information of a sampling value of the AD converter;
A phase comparator that calculates a phase difference between the code clock of the clock generator and the divided clock of the divider;
An addition / subtraction unit for adding / subtracting the phase correction value of the phase correction unit to / from the phase calculation value of the phase comparison unit;
An adjustment output unit that outputs an adjustment amount according to an output value of the addition / subtraction unit as an input adjustment value of the control oscillation unit, and
The phase comparison unit includes:
Counting by the count clock is started in one direction from the count start timing by the code clock of the clock generator, and counting by the count clock is performed in the direction opposite to the one direction at the count conversion timing by the divided clock of the frequency divider. An up / down counter that reverses and counts up to a count end timing by the code clock of the clock generation unit,
When calculating the phase difference between the code clock of the clock generator and the divided clock of the divider, the count value obtained at the count end timing by the up / down counter is output as the phase calculated value. Digital phase synchronization circuit. The phase comparison unit includes:
The configuration is such that the count value when the frequency and phase of the code clock of the clock generation unit and the frequency division clock of the frequency division unit are the same as each other is output as a reference
The up / down counter counts down according to a predetermined count clock from the rising timing of the code clock of the clock generation unit, and receives the falling timing by the divided clock of the frequency dividing unit after the down counting. 2. The digital phase locked loop circuit according to claim 1, wherein up-counting is performed in accordance with the count clock from a falling timing.   3. The digital phase synchronization circuit according to claim 1, wherein the up / down counter performs counting by combining two same counters. The phase correction unit is
A first delay unit that delays the sampling value of the AD converter that has sampled the analog value by one sampling clock;
A first subtractor for subtracting a delay sampling value of the first delay unit from a current sampling value of the AD converter;
A first divider for dividing the current sampling value of the AD converter by the subtracted value of the first subtractor;
2. A first multiplier for multiplying a constant value obtained by multiplying a count value of the count clock corresponding to a sampling period of the AD converter by -2 with a division value of the first division unit. 4. The digital phase synchronization circuit according to any one of 1 to 3.   5. The digital phase synchronization according to claim 4, wherein at least one of the first subtractor, the first divider, and the first multiplier of the phase correction unit has a function of switching operation permission / operation stopped. circuit.   Fixed value setting for setting the input of the arithmetic unit to be stopped to a fixed value when stopping at least one of the first subtractor, the first divider, and the first multiplier of the phase correction unit 6. The digital phase locked loop circuit according to claim 5, further comprising a counter. When the first subtractor, the first divider and the first multiplier of the phase correction unit are configured to hold data in synchronization with the clock,
7. The operation clock of an arithmetic unit to be stopped is stopped while at least one of the first subtractor, the first divider, and the first multiplier is stopped. The digital phase synchronization circuit described. A zero-cross detector that detects the timing at which the sampling value of the AD converter has zero-crossed,
Using the zero-cross timing signal detected by the zero-cross detection unit, setting a period for stopping the calculation of at least one of the first subtractor, the first divider, and the first multiplier of the phase correction unit. 8. A digital phase locked loop according to claim 5, wherein   The period for stopping the calculation of at least one of the first subtractor, the first divider, and the first multiplier of the phase correction unit is set using the code clock of the clock generation unit. 8. The digital phase locked loop circuit according to any one of 5 to 7. The phase correction unit is
An absolute value calculator that calculates the absolute value of the sampling value of the AD converter that samples the analog value;
A second delay unit for delaying the operation value of the absolute value operation unit by one sampling clock;
A second adder for adding the delay sampling value of the second delay unit and the operation value of the absolute value calculator;
A second divider for dividing the calculated value of the absolute value calculator by the added value of the second adder;
2. A second multiplier that multiplies a constant value obtained by multiplying a count value of the count clock corresponding to a sampling period of the AD converter by -2 with a division value of the second division unit. 4. The digital phase synchronization circuit according to any one of 1 to 3.   11. The digital phase synchronization according to claim 10, wherein at least one of the second subtracter, the second divider, and the second multiplier of the phase correction unit has a function of switching operation permission / operation stopped. circuit.   Fixed value setting that sets the input of the operation unit to be stopped to a fixed value when stopping the operation of at least one of the second subtractor, the second divider, and the second multiplier of the phase correction unit 12. The digital phase locked loop circuit according to claim 11, further comprising a counter. When the second subtractor, the second divider, and the second multiplier of the phase correction unit are data holding circuits that hold data in synchronization with a clock,
13. The input clock of an arithmetic unit to be stopped is stopped while at least one of the second subtracter, the second divider, and the second multiplier is stopped. The digital phase synchronization circuit described. A zero-cross detector that detects the timing at which the sampling value of the AD converter has zero-crossed,
Using the zero-cross timing signal detected by the zero-cross detection unit, setting a period for stopping the calculation of at least one of the second subtractor, the second divider, and the second multiplier of the phase correction unit. 14. The digital phase locked loop circuit according to claim 11, wherein the digital phase locked loop circuit is a digital phase locked loop.   The period during which at least one of the second subtracter, the second divider, and the second multiplier of the phase correction unit is stopped is set using the code clock of the clock generation unit. The digital phase locked loop circuit according to any one of 11 to 13. A physical quantity detection sensor comprising an vibrator that vibrates in a reference direction when given with a drive signal, and an angular velocity detection unit that outputs a sensor signal according to the physical quantity given to the vibrator,
AGC for controlling the amplitude of the input signal to be constant and applying a drive signal to the vibrator;
A signal detector for detecting a vibration signal in the reference direction;
16. The digital phase synchronization circuit according to claim 1, wherein the detection signal of the signal detection unit is input, the input signal is phase-synchronized and output as an input signal of the drive signal output unit. A physical quantity detection sensor characterized by.

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