JP2012042214A - Radar device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To solve the problem that, in a radar for calculating a target distance by combining beat frequencies of reception beat signals modulated at different frequency modulation rates, targets can not be rightly combined when amplitude values and angle measurements of different targets are within a threshold value, and signals can not be separated when a beat frequency difference of a plurality of targets is less than or equal to a beat frequency resolution.SOLUTION: A carrier signal is radiated which is frequency-modulated in a chirp shape over a plurality of PRIs at a frequency modulation rate selected by parameters from a distance resolution with no ambiguity and a beat frequency resolution, a reflection signal in a target is received, and target candidates are combined with each other from an in-PRI distance-beat frequency map created by performing inter-PRI FFT on a reception beat signal converted by the carrier signal. A distance with no ambiguity and a speed are calculated from a combination of beat frequencies of target candidates, and the distance ambiguity of the in-PRI distance is canceled by the distance with no ambiguity, thereby calculating a distance to the target.

Description

  The present invention relates to a radar apparatus that determines a combination of beat frequencies of received beat signals that are frequency-modulated at different frequency modulation rates and calculates a distance to a target.

  Here, a radar shown in Japanese Patent Laid-Open No. 08-262131 (Patent Document 1) that determines a combination of beat frequencies according to the present invention using an amplitude value or an angle measurement value and measures a distance to a plurality of targets. The apparatus will be described.

  The radar apparatus disclosed in Patent Literature 1 transmits a signal subjected to up-chirp frequency modulation, receives the reflected wave from the target, and then performs down-chirp frequency modulation with the same absolute value of the frequency modulation rate. The signal is transmitted, the reflected wave is received from the target, and the received signal is subjected to fast Fourier transform (FFT) to obtain the beat frequency. In the case of a plurality of targets, a peak of a plurality of beat frequencies can be obtained by transmitting and receiving a signal subjected to up-chirp and down-chirp frequency modulation, so it is necessary to obtain a combination of the beat frequencies.

  Here, with reference to FIG. 26 and FIG. 27, a method for determining a combination of beat frequencies of the received beat signal using the amplitude value or the angle measurement value of the received beat signal in the radar apparatus disclosed in Patent Document 1. explain. 26 shows a method for determining the beat frequency combination of the received beat signal using the amplitude value of the received beat signal, and FIG. 27 shows a method for determining the beat frequency combination of the received beat signal using the angle measurement value. Show.

In FIG. 26, the horizontal axis represents the beat frequency of the received beat signal, and the vertical axis represents the amplitude. The horizontal axis in FIG. 27 is the beat frequency of the received beat signal, and the vertical axis is the angle measurement value. In FIG. 26, fd _u1 is the beat frequency at the time of up chirping of target 1, fd _u2 is the beat frequency at the time of up chirping of target 2, fd _d1 is the beat frequency at the time of down chirping of target 1, and fd _d2 is the down frequency of target 2 Beat frequency at chirp, A _u1 is the amplitude of beat frequency fd _u1 at the time of up chirping of target 1, A _u2 is the amplitude of beat frequency fd _u2 at the time of up chirping of target 2, and A _d1 is at the time of down chirping of target 1 The amplitude of the beat frequency fd _d1 and A _d2 are the amplitude of the beat frequency fd _d2 when the target 2 is down-chirped.

In FIG. 27, fd _u1 is the beat frequency at the time of up chirping of target 1, fd _u2 is the beat frequency at the time of up chirping of target 2, fd _d1 is the beat frequency at the time of down chirping of target 1, and fd _d2 is the target 2 Beat frequency during down-chirping, θ _u1 is the angle value of beat frequency fd _u1 during target 1 up-chirp, θ _u2 is the angle value of beat frequency fd _u2 during target 2 up-chirp, and θ _d1 is the target value The angle value of beat frequency fd _d1 at the time of down chirp of 1 and θ _d2 are the angle value of beat frequency fd _d2 at the time of down chirp of target 2.

In Figure 26, the beat frequency fd _u1 at the target 1 up-chirp amplitude A _u1 and target 1 when the down-chirp beat frequency fd _d1 amplitude A _d1 and the amplitude A of the target 2 in the up-chirp when the beat frequency fd _u2 difference between _u2 and amplitude a _d2 of the beat frequency fd _d2 when the down-chirp target 2 and becomes less the threshold, otherwise, for example, the amplitude a _u1 and objectives of the up chirp when the beat frequency fd _u1 target 1 The difference between the beat frequency fd _d2 amplitude A _d2 when down chirping 2 and the amplitude A _d1 of the beat frequency fd _d1 when down chirping target 1 and the beat frequency fd _{u2 and} amplitude A _u2 when up chirping target 2 Between the beat frequency fd _u1 for target 1 up-chirp and the beat frequency fd _{d1 for} target 1 down-chirp and the beat frequency fd _u2 for target 2 up-chirp and target 2 Beat during down chirp Performing a combination wave number fd _d2.

In FIG. 27, as in FIG. 26, the angle measurement value θ _u1 of the beat frequency fd _u1 at the time of up chirping of the target 1 and the angle measurement value θ _d1 of the beat frequency fd _d1 at the time of down chirping of the target 1 and the up chirp of the target 2 has a less difference threshold between the measured angle value theta _d2 beat frequency fd _d2 when the down-chirp beat frequency fd measured angle value of _u2 theta _u2 and the target 2 when otherwise, for example, up target 1 angle measurement of angle measurement value theta _u1 and the difference between the measured angle value theta _d2 and the target 2 of the down-chirp when the beat frequency fd _d2, beat frequency fd _d1 at the down-chirp target 1 beat frequency fd _u1 during chirp since the difference between the measured angle value theta _u2 of _the beat frequency fd _u2 during the up-chirp value theta _d1 and the target 2 is equal to or greater than the threshold, the beat frequency fd _u1 and the target 1 of the down-chirp when the target 1 up-chirp beat frequency at the time of the beat frequency fd _d1 and goal 2 up-chirp when carry out the combination for fd _u2 and the beat frequency at the time of down-chirp of the target 2 fd _d2.

  In this radar apparatus, distance measurement is performed by combining the beat frequencies of signals whose difference between the amplitude values or the angle measurement values of the beat signals is within the threshold and eliminating the velocity term of the beat frequency by the above method.

Japanese Patent Laid-Open No. 08-262131

  However, as shown in FIGS. 28 and 29, when the amplitude value and the angle measurement value of different targets are within the threshold, there is a problem that the targets cannot be combined correctly.

  In addition, when the beat frequency difference between a plurality of targets is equal to or less than the resolution of the beat frequency, this radar apparatus has a problem that signals cannot be separated as shown in FIG.

  The object of the present invention is to determine the combination of beat frequencies, calculate the distance to the target with high resolution, even when the difference between the amplitude value and the angle measurement value of different targets is within the threshold, and to beat multiple targets. To provide a radar apparatus capable of separating signals and calculating respective distances to a target even when the frequency difference is equal to or less than the resolution of the beat frequency.

The radar apparatus according to the present invention is
PRI (Pulse Repetition Interval) Different frequency modulation rates of transmitted signals based on the resolution of the distance without ambiguity that can eliminate the distance ambiguity of the inner distance and the parameter that makes the PRI inner distance resolution more accurate than the beat frequency resolution. Transmitting means for radiating a carrier signal frequency-modulated on the chirp across a plurality of PRIs with a plurality of selected and frequency modulation rates obtained;
A transmission signal obtained from the transmission means reflected back from the target is received as a reception signal, and the received signal is down-converted using the carrier signal obtained from the transmission means and converted into a reception beat signal. Receiving means;
Create PRI intra-distance-beat frequency map to calculate the relative distance to the target by performing inter-PRI FFT (Fast Fourier Transform) on the received beat signal obtained from the receiving means. Means,
Target candidate detection means for detecting a target candidate based on the signal strength for the PRI internal distance-beat frequency map obtained from the PRI internal distance-beat frequency map creating means;
For the target candidate obtained from the target candidate detecting means, a combination means for combining the target candidates using the PRI internal distance;
Using the combination obtained by the combination means, combining the beat frequencies of the target candidates, distance speed calculation means for calculating the distance and speed without ambiguity,
Distance high-resolution means for eliminating the distance ambiguity of the PRI internal distance and calculating the relative distance to the target using the distance without ambiguity obtained by the distance speed calculation means.

According to the radar apparatus according to the present invention,
By using the PRI intra-distance information that allows multiple targets to have high resolution, it is possible to obtain a radar apparatus that can handle multiple targets and that can calculate a high-resolution distance without distance ambiguity. Further, even when the difference between the amplitude value and the angle measurement value is small, it is possible to obtain a plurality of target distances by using the high-resolution PRI internal distance. Further, in the conventional radar apparatus, the signal separation performance depends on the resolution of the beat frequency having a low resolution, and when the difference between the beat frequencies is smaller than the frequency resolution, the signal cannot be separated. In such a radar apparatus, since signals can be separated with high resolution within the PRI distance, target separation performance is highly effective.

1 is a configuration diagram of a radar apparatus according to Embodiment 1 of the present invention. It is explanatory drawing of the relationship between the transmission signal and reception signal which carried out the frequency modulation | alteration and the intra-pulse code modulation in the case of up-chirp. It is explanatory drawing of PRI (Pulse Repetition Interval) distance-beat frequency map in the case of up chirp. 6 is an explanatory diagram of a frequency time change of a local oscillation signal (carrier signal) in the case of up-chirp and down-chirp having the same transmission signal bandwidth in Embodiment 1. FIG. It is explanatory drawing of the frequency time change (in the case of up chirp and down chirp in which transmission signal bandwidths differ) of a local oscillation signal. It is explanatory drawing of the frequency time change (in the case of up-chirps from which a transmission signal bandwidth differs) of a local oscillation signal. It is explanatory drawing of the frequency time change (in the case of down chirp from which a transmission signal bandwidth differs) of a local oscillation signal. It is explanatory drawing of the 4-bit Barker code which an intra-pulse modulation signal generator produces | generates. It is a processing block diagram with respect to the reception beat signal of a signal processor. It is explanatory drawing of the attention cell, guard cell, and sample cell regarding a CFAR process. CFAR processing according to the results CFAR threshold CFAR_th (k _h, l) is a diagram that cell exceeds explains a procedure in the case where the aggregate. It is explanatory drawing of the distance determination method in PRI by a combination means. It is explanatory drawing of the beat frequency combination method in the case of the multiple target by a combination means. It is a block diagram of the radar apparatus which concerns on Embodiment 2 of this invention. FIG. 10 is a processing block diagram for a received beat signal of a signal processor in a second embodiment. It is explanatory drawing which shows the influence of a ground clutter side lobe. It is a figure which shows the case where a target is buried in a ground clutter side lobe. It is explanatory drawing which shows the filter characteristic of a filter process means. It is explanatory drawing which shows the effect of the filter process by a filter process means. It is a block diagram of the radar apparatus which concerns on Embodiment 3 of this invention. FIG. 10 is a processing block diagram for a received beat signal of a signal processor according to Embodiment 3. It is explanatory drawing which shows the effect of performing a window function process in the case of a correlation process. It is explanatory drawing which shows the effect of performing a window function process in the case of FFT between PRI. It is a block diagram of the radar apparatus which concerns on Embodiment 4 of this invention. It is explanatory drawing of the relationship between the transmission signal and reception signal which carried out the stepwise frequency modulation and the intra-pulse code modulation. It is explanatory drawing of the beat frequency combination method using the amplitude value of the conventional radar apparatus shown by patent document 1. FIG. It is explanatory drawing of the beat frequency combination method using the angle value of the conventional radar apparatus shown by patent document 1. FIG. It is explanatory drawing when a combination cannot be solved with the amplitude value of the conventional radar apparatus shown by patent document 1. FIG. It is explanatory drawing when a combination cannot be solved with the angle measurement value of the conventional radar apparatus shown by patent document 1. FIG. It is explanatory drawing of the target isolation | separation performance of the conventional radar apparatus at the time of the modulation frequency shown by patent document 1 up-chirping.

Embodiment 1 FIG.
A radar apparatus according to Embodiment 1 of the present invention will be described with reference to FIGS. 1 is a processing block diagram showing a configuration of a radar apparatus according to Embodiment 1 of the present invention. In addition, in each figure, the same code | symbol shows the same or equivalent part.

  As shown in FIG. 1, the radar apparatus according to Embodiment 1 of the present invention includes an antenna 1, a transmission means 2, a transmission / reception switch 7, a reception means 100, a signal processor 200, and a display 9. It is composed of

  The transmission means 2 is provided with a transmitter 3, an intra-pulse modulation signal generator 4, a pulse modulator 5, and a local oscillator 6.

  The signal processing means 200 includes a computer having a CPU (Central Processing Unit), a RAM (Random Access Memory), a ROM (Read Only Memory), and an interface circuit. The CPU performs arithmetic processing according to a program stored in the ROM. Is called. The processing result of the signal processing means 200 is recorded in a memory such as a RAM.

  The signal processing means 200 is provided with PRI internal distance-beat frequency map creation means 201, target candidate detection means 220, combination means 221, distance speed calculation means 222, and distance high resolution means 223.

  The PRI internal distance-beat frequency map creating means 201 is provided with a correlating means 202 and a beat frequency calculating means 203.

  Next, the operation of the radar apparatus according to the first embodiment will be described with reference to the drawings.

  First, an operation until a reception beat signal is generated will be described with reference to FIG. FIG. 2 is a diagram for explaining the relationship between a frequency-modulated and intra-pulse code modulated transmission signal and reception signal. However, FIG. 2 shows the case of up-chirp.

In FIG. 2, f ₀ is the transmission start frequency, T _pri is the pulse repetition period (PRI), T _L is the transmission signal frequency sweep time, B _L is the transmission signal bandwidth, T ₀ is the beat signal observation time, and B ₀ is the beat Transmission signal bandwidth at time interval of signal observation time T ₀ , T _p is transmission pulse width, φ ₀ is initial phase of transmission signal and local oscillation signal (carrier signal), R ₀ is target of time t = 0 The initial relative distance, R _max is the maximum detection distance, T _offset is the first transmission pulse transmission start time at time 2R _max / c or more, and c indicates the speed of light.

The time 2R _max / c or more first transmission pulse transmission start time T _offset, the maximum detectable range R _max, the transmission signal frequency sweep time T _L, the transmission of a time interval of T _L signal bandwidth B _L, the beat signal observed The transmission signal bandwidth B ₀ at the time intervals T ₀ and T ₀ has a relationship represented by the following equations (1), (2), and (3).

  In FIG. 2, for simplicity, the local oscillation signal is up-chirp and only the received beat signal from a single target is shown. However, in the first embodiment, as the local oscillation signal, an up-chirp and a down-frequency having the same absolute value of the frequency modulation factor are shown. In the following description, the chirp signal is used and the target number is 2.

Next, a method for setting parameters related to the transmission signal will be described with reference to FIG. FIG. 3 shows a PRI internal distance-beat frequency map in the case of up-chirp. Here, Ru1 'is the target 1 PRI distance (when the frequency modulation rate is up-chirped), _Ru2 ' is the target 2 PRI distance (when the frequency modulation rate is up-chirped), and Δf _FM is the beat frequency sampling frequency It is an interval.

Equation (4), Equation (5), and Equation (6) are satisfied so that the distance resolution after correlation processing is higher than the resolution of the distance within the PRI distance and the distance resolution without the ambiguity. Parameters for calculating beat frequency frequency resolution Δf _FM ′, beat frequency sampling frequency interval Δf _FM , distance resolution after correlation processing ΔR _PC ′, and 1PRI folding distance R _pri are set. The parameters for calculating the aliasing distance R _pri of 1PRI, the beat signal observation time T _0, the transmission signal bandwidth B ₀ in a time interval of the beat signal observation time T ₀ is set.

Further, the beat frequency distance resolution Δf _FM ′ is the expression (7), the PRI folding distance R _pri is the expression (8), the sampling interval Δf _FM is the expression (9), and the distance resolution ΔR _PC ′ after the correlation processing is the expression ( 10). However, it is assumed that code modulation is performed for each sample. Here, H represents the number of FFT points between PRIs. The transmission means 2 operates based on the parameters set as described above.

  Next, a method for setting the frequency modulation rate change of the local oscillation signal will be described with reference to FIGS.

The local oscillator 6 generates a local oscillation signal L ₀ (t) expressed by the following equation (11) that is frequency-modulated with the transmission signal frequency sweep time T _L and the transmission signal bandwidth B _L , and the pulse modulator 5 To the receiving means 100.

However, _AL is the amplitude of the local oscillation signal, φ ₀ is the initial phase of the transmission RG signal, and the sign of ± is + for up-chirp and-for down-chirp.

FIG. 4 shows the frequency time change of the local oscillation signal. As shown in FIG. 4, the local oscillator 6 generates an up-chirp local oscillation signal in the first transmission signal frequency sweep time _TL , and a down-chirp local oscillation signal in which the absolute value of the frequency modulation rate is equal in the next _TL. To do. In the first embodiment, the local oscillator 6 operates based on the frequency modulation rate of the local oscillation signal set as described above.

As shown in FIG. 5, the frequency modulation rate set in the local oscillator 6 is different from the transmission signal bandwidth B _L in the time interval of the transmission signal frequency sweep time T _L of up-chirp or down-chirp. A combination of up-chirp and down-chirp local oscillation signals changed to B _L 'may be used. Further, as shown in FIG. 6, the local oscillation signal of the down-chirp of a transmission signal bandwidth B _L, may be a combination of up-chirp between changing the local oscillation signal of the up-chirp of the different transmission signal bandwidths B _L '. Furthermore, as shown in FIG. 7, the transmission signal bandwidth B _L local oscillation signal up chirp may be a combination of the down-chirp between changing the local oscillation signal of the down-chirp of the different transmission signal bandwidths B _L '. However, it is assumed that the transmission signal bandwidth B _L ′ used satisfies the conditions of the equations (4), (5), and (6).

The pulse modulator 5 performs pulse modulation on the local oscillation signal L ₀ (t) input from the local oscillator 6, and the pulse-modulated local oscillation signal L ′ ₀ (t) expressed by the following equation (12). ) And output to the transmitter 3.

Here, n _L represents the pulse number within the transmission signal frequency sweep time _TL , and N _L represents the number of pulses within the transmission signal frequency sweep time _TL .

  The intra-pulse modulation signal generator 4 generates the 4-bit Barker code shown in FIG. 8 as the intra-pulse modulation signal φ (t) and outputs it to the transmitter 3. Another bit Barker code may be used as the intra-pulse modulation signal. Further, other code modulation or frequency modulation may be used as the intra-pulse modulation signal.

The transmitter 3 uses the intra-pulse modulation signal φ (t) input from the intra-pulse modulation signal generator 4 to the pulse-modulated local oscillation signal L ′ ₀ (t) input from the pulse modulator 5. Then, intra-pulse modulation is performed, and a transmission RF signal Tx (t) expressed by the following equation (13) is output to the transmission / reception switch 7.

Here, _AT represents the amplitude of the transmission RF signal, and rem (X, Y) represents the remainder when variable X is divided by variable Y.

  The transmission / reception switch 7 outputs the transmission RF signal input from the transmitter 3 to the antenna 1. Then, the transmission RF signal is radiated from the antenna 1 into the air.

  The transmission RF signal radiated into the air is reflected by the target and enters the antenna 1 as a reflected RF signal. Therefore, the antenna 1 receives the incident reflected RF signal and outputs it to the transmission / reception switch 7 as a received RF signal.

The transmission / reception switch 7 outputs the reception RF signal input from the antenna 1 to the reception means 100. When the transmission RF signal Tx (t) represented by the equation (13) receives the reflected RF signal from the target at the relative distance R (t), the reception RF signal Rx _tgt (n _L , t) is It is represented by Formula (14).

Here, _Atgt ( _ntgt ) represents the amplitude of the received RF signal, _ntgt represents the target number, target 1 corresponds to _ntgt = 1, and target 2 corresponds to _ntgt = 2. Further, the target relative distance R (t, n _tgt ) at time t is expressed by the following equation (15).

Here, R ₀ (1) represents the distance of target 1, v (1) represents the speed of target 1, R ₀ (2) represents the distance of target 2, and v (2) represents the speed of target 2.

The received RF signal Rx _tgt (n, t) observed within the beat signal observation time T ₀ is expressed by the following equation (16).

Here, n represents the pulse number within the beat signal observation time T ₀ , and N represents the number of pulses within the beat signal observation time T ₀ .

The receiving means 100 is constituted by a receiver 8. The receiver 8 down-converts the received RF signal Rx _tgt (n, t) input from the transmission / reception switch 7 using the local oscillation signal L ₀ (t) input from the local oscillator 6 and then amplifies it. Then, phase detection is performed, and the signal is output to the signal processing means 200 as a received beat signal S (n, t) represented by the following equation (17).

  Further expanding the equation (17), the received beat signal S (n, t) is expressed as the following equation (18).

Hereinafter, the sampled received beat signal S (n, m) represented by the following equation (19) is expressed by using m as the sampling number within 1PRI, M as the number of sampling points within 1PRI, and Δt as the sampling time within 1PRI. The explanation will be given. However, the sampling time t _samp (n, m) within T ₀ is expressed by the following equation (20).

  Hereinafter, the processing content of each means for the received beat signal shown in the processing block diagram of FIG. 9 will be described.

  The PRI internal distance-beat frequency map creation means 201 is composed of a correlation means 202 and a beat frequency calculation means 203, which performs a correlation process and an inter-PRI FFT on the received beat signal and calculates a relative distance from the target. Create an internal distance-beat frequency map.

Correlation means 202 performs correlation processing of received beat signal S (n, m) and reference signal Ex (m _τ ) expressed by equation (18), that is, pulse compression. Here, correlation processing in the frequency domain will be described.

The reference signal Ex (m _τ ) after A / D conversion which is the same as the intra-pulse modulation component of the transmission RF signal is expressed by Expression (21). Here, m _τ represents a sampling number, M _τ represents the number of sampling points of 1 PRI (formula (22)), and floor (X) represents the maximum integer that does not exceed the variable X.

The correlator 202 performs multiplication after multiplying the received beat signal S (n, m) and the reference signal Ex (m _τ ) by Expression (23) and Expression (24), respectively (Expression (25)). Here, * represents a complex conjugate, l represents a sampling number in PRI, and L ′ represents an FFT score for correlation processing. However, when L ′> M, 0 is substituted into S (n, m), and when L ′> M _τ , 0 is substituted into Ex (m _τ ).

  Correlation means 202 sets 0 by equation (26) when sampling the signal after correlation processing with higher accuracy than the sampling interval of the received beat signal. Here, L is the inverse fast Fourier transform (IFFT) point of correlation processing, and is expressed by equation (27). However, q is an integer of 0 or more. When q = 0, the sampling interval is the same as the sampling interval of the received beat signal.

Finally, the correlator 202 performs an IFFT on the multiplication result F _{V · Ex} (n, k _r ) according to the equation (28), and obtains the result of correlation processing, that is, the signal R _{V · Ex} (n after pulse compression). , l). Further, the relative distance R _PC (l) in PRI corresponding to the sampling number l of the signal R _{V · Ex} (n, l) after the pulse compression is expressed by Expression (29).

The beat frequency calculation means 203 performs PRIFT distance-beat frequency map R _FM R _PC (k) by performing IFFT on the signal R _{V · Ex} (n, l) after pulse compression in the PRI direction at the H point according to the equation (30). Output _h , l). However, when H> N, R _{V · Ex} (n, l) is zero-padded. Here, k _h represents the sampling number after the inter-PRI FFT. Also, the beat frequency f (k _h ) corresponding to the sampling number after the inter-PRI FFT is expressed by equation (31).

  As described above, the PRI intra-distance-beat frequency map creating means 201 performs the intra-PRI pulse compression and the inter-PRI FFT to calculate the relative distance to the target shown in FIG. The map is output to the target candidate detection means 220.

  Next, the target candidate detection means 220 performs a CFAR (Constant False Alarm Rate) process for detecting a target candidate based on the signal strength.

The contents of target candidate detection processing by CFAR processing will be described with reference to FIGS. 10 and 11. FIG. 10 is a diagram for explaining a cell of interest, a guard cell, and a sample cell related to the CFAR process. FIG. 11 is a diagram for explaining the processing contents when cells exceeding the CFAR threshold value CFAR_th (k _h , l) are gathered as a result of the CFAR processing.

The target candidate detection means 220 performs the CFAR process on the PRI internal distance-beat frequency map R _FM R _PC (k _h , l) obtained from the PRI internal distance-beat frequency map creating means 201 by the equation (32), Detect target candidates. Here, R _FM R _{PC, CFAR} (k _h , l) represents a target candidate detection result by CFAR processing, and 0 is set as the target candidate. Here, abs (X) represents the absolute value of the constant X.

The CFAR threshold value CFAR_th (k _h , l) is calculated by the following equation (33). Here, CFAR_cor the CFAR factor, Samp_cell (k _h, l) is the sample cell, ave (Z (p)) represents the average value of the sequence Z (p). However, when cells exceeding the CFAR threshold value CFAR_th (k _h , l) are gathered as shown in FIG. 11, a cell having the maximum amplitude value is detected as a target candidate. Even when there are a plurality of sets, a cell indicating the maximum value of the amplitude is detected as a target candidate.

  Here, in this method, as shown in FIG. 3, the correlator 202 calculates a high-resolution PRI distance. Therefore, unlike the radar apparatus disclosed in Patent Document 1 in this method, the target candidate detection means 220 separates signals in the PRI inner distance direction by the correlation means 202 even if the difference in beat frequency is less than the frequency resolution, and the CFAR Two targets can be detected by processing.

  The target candidates obtained by the above processing are the peak u1 and the peak u2 in the cell where the amplitude value is peak when the frequency modulation is up-chirped, and the peak d1 is the cell where the amplitude value is peaked when the frequency modulation is down-chirped. Assuming peak d2, peak u1 is represented by the following equation (34), peak u2 is represented by the following equation (35), peak d1 is represented by the following equation (36), and peak d2 is represented by the following equation (37).

However, the beat frequency corresponding to the cell of peak u1 is fd _u1 , the PRI internal distance is R _u1 ', the beat frequency corresponding to the peak u2 cell is fd _u2 , the PRI internal distance is R _u2 ', and the peak d1 cell is compatible Beat frequency fd _d1 , PRI internal distance R _d1 ′, peak 2 cell and corresponding beat frequency fd _d2 , PRI internal distance R _d2 ′, target 1 PRI internal distance sampling number l _tgt1 , target The sampling number of the PRI internal distance of 2 was set to l _tgt2 . K _r and K _v are expressed by the following equations (38) and (39).

  The processing contents of the combination means 221 will be described with reference to FIGS. FIG. 12 is a diagram for explaining a PRI distance determination method, and FIG. 13 is a diagram for explaining a beat frequency combination method in the case of multiple targets.

The combination means 221 obtains a combination for the target candidates input from the target candidate detection means 220 using the relative distance R _PC (l) within PRI.

FIG. 12 shows a method for determining the same target. Here, R _u1 ′ is the target 1 PRI distance (when the frequency modulation rate is up-chirped), R _d1 ′ is the target 1 PRI distance (when the frequency modulation rate is down-chirped), and R _d2 ′ is the target 2 The distance within the PRI (when the frequency modulation rate is down-chirped).

The combination means 221 determines that the target is the same when the difference in the PRI distance is within the threshold γ as in R _u1 ′ and R _d1 ′ in FIG. Further, the combination means 221 determines that the target is different when the difference in the PRI distance is equal to or larger than the threshold γ as in R _u1 ′ and R _d2 ′ in FIG. 12.

  FIG. 13 shows a combination method in the case of multiple targets. The combination means 221 uses the determination method shown in FIG. 12 to combine the beat frequencies of the target candidates whose PRI distance difference is within the threshold. Hereinafter, target 1 will be described, but a similar method can be used for target 2 as well.

  The distance / velocity calculation means 222 uses the combination of beat frequencies input from the combination means 221 to calculate a distance and speed without ambiguity. The speed calculation formula is shown in Formula (40), and the distance calculation formula without ambiguity is shown in Formula (41).

  As described above, in this method, even if the speed is not obtained in advance, the speed term can be eliminated by the combination of beat frequencies and the distance without ambiguity can be calculated. The distance can be calculated without any problem.

  The distance high resolution means 223 performs a distance high resolution process.

First, the distance high resolution means 223 calculates the number of pulse wraps. The target distance is expressed by the following equation (42) using the number of pulse wraps and the PRI internal distance. Therefore, the number of pulse wrapping is calculated by the equation (43) using the distance without ambiguity input from the distance / velocity calculation means 222 and the relative distance R _FM (k _h ) between PRIs input from the correlation means 202. expressed. Here, N _rtn (1) is the target 1 pulse folding number. Further, R _u1 ′ may be used instead of R _d1 ′ in the equation (42).

However, the target 1 pulse folding number N _rtn (1) calculated by the distance high-resolution means 223 includes a sampling error of the beat frequency.

Therefore, the distance high resolution means 223 _obtains the pulse folding number N _rtn ′ (1) of the target 1 excluding the sampling error by the equation (44). Here, round (X) is an integer obtained by rounding off the variable X.

Further, the distance resolution _increasing means 223 _{obtains the} high-resolution target relative distance R ₀ ′ (1) from the equation (45) using the pulse wrap count N _rtn ′ (1) of the target 1 excluding the error. Again, R _u1 ′ may be used instead of R _d1 ′ in Equation (45).

  If there is a PRI distance-beat frequency map and target candidates, the display 9 displays the target relative distance as target information on the screen.

  Although the case where the target number is 2 has been described in the first embodiment, the same effect can be obtained when the target number is 1 or 3 or more.

  As described above, according to the first embodiment, up-chirp frequency modulation is performed in order to obtain a beat frequency, and transmission RF signals subjected to intra-pulse modulation in order to obtain PRI internal distance are transmitted and received. For the received beat signal, pulse compression is performed within PRI and FFT is performed between PRIs to create a PRI distance-beat frequency map, and target candidates are calculated for the PRI distance-beat frequency map by CFAR processing. The same processing is performed on the transmission signal that has undergone down-chirp frequency modulation, and the combination of up-chirp and down-chirp beat frequencies is determined by combination processing using the PRI inner distance, and obtained by combination processing. From the beat frequency combination, the distance and speed without the target ambiguity are calculated, and the target PRI internal distance ambiguity is the distance without ambiguity. Solved by using, to calculate the target relative distance.

  As a result, it is possible to obtain a radar apparatus that can deal with multiple targets and can calculate a high-resolution distance without distance ambiguity by using high-resolution PRI distance information for a plurality of targets. Further, in the above-mentioned Patent Document 1, a plurality of targets having a small difference between an amplitude value and an angle measurement value could not distinguish a combination of beat frequencies, but in Embodiment 1, an amplitude value and an angle measurement value are not distinguished. Even when the difference is small, it is possible to obtain a plurality of target distances by using a high-resolution PRI internal distance. Furthermore, in the previous Patent Document 1, the signal separation performance depends on the resolution of the beat frequency with a low resolution, and when the difference between the beat frequencies is smaller than the frequency resolution, the signal cannot be separated. In the first mode, since the signal can be separated with a high resolution within the PRI distance, the target separation performance is high.

Embodiment 2. FIG.
A radar apparatus according to Embodiment 2 of the present invention will be described with reference to FIGS. FIG. 14 is a processing block diagram showing the configuration of the radar apparatus according to Embodiment 2 of the present invention.

  In FIG. 14, the radar apparatus according to Embodiment 2 of the present invention is provided with an antenna 1, a transmission means 2, a transmission / reception switch 7, a reception means 100, a signal processor 200B, and a display 9. ing.

  The radar device according to the second embodiment is different from the radar device according to the first embodiment except for the signal processor 200B, and the other components are the same.

  That is, as shown in FIG. 14, the signal processor 200B replaces the PRI internal distance-beat frequency map creating means 201 of the signal processor 200 according to Embodiment 1 with the PRI internal distance added with the filter processing means 204. -The difference is that it has the beat frequency map creating means 201B, and the others are the same, so the same reference numerals are given to the same parts, and the description will be omitted.

  The signal processor 200B performs the process shown in FIG. 15 on the received beat signal.

  In the second embodiment, a ground clutter side lobe is further assumed as a clutter, and a filter process is performed for the purpose of reducing the ground clutter side lobe. When there is a ground clutter having a high strength, the target may be buried in the ground clutter side lobe as shown in FIG. 17 even when the target is different from the ground clutter main lobe as shown in FIG.

The filter processing means 204 calculates the parameters of the filter that attenuates the ground clutter main lobe and attenuates the ground clutter side lobe according to the following equations (46), (47), (48), and (49). To do. Here, f _c1 and f _c2 are stop band edge frequencies, f _a1 and f _a2 are cut-off frequencies, α is a bandwidth giving a margin to a preset stop band bandwidth, β is a preset transition bandwidth, A _c is a preset passband ripple amount, and A _a is a preset stopband attenuation amount. Min (X, Y) represents the minimum value of variables X and Y, and max (X, Y) represents the maximum value of variables X and Y.

The filter processing means 204 creates an N _flt- order filter flt (p) having parameters calculated by the equations (46) to (49) and preset filter characteristics.
The second embodiment will be described using a FIR (Finite Impulse Response) filter as a filter, but the same effect can be obtained when an IIR (Infinite Impulse Response) filter is used.

  As shown in FIG. 18, since the filter processing means 204 has a filter characteristic that attenuates the ground clutter main lobe, it is possible to attenuate the ground clutter main lobe and the ground clutter side lobe after the filter processing. become.

For the purpose of reducing ground clutter side lobes, the filter processing means 204 performs a filter process in the PRI direction on the signal R _{V · Ex} (n, l) after pulse compression in the PRI direction by the following equation (50). The processed signal R _flt (n, l) is output.

The beat frequency calculation means 203 performs the IFFT at the H point in the PRI direction on the input filtered signal R _flt (n, l), so that the PRI internal distance-beat frequency map R _FM R _PC (k _h , l) is output.

  As shown in FIG. 19, by performing the filter processing, the ground clutter main lobe and side lobes in the beat frequency direction of the PRI intra-distance-beat frequency map are reduced, and the target detection performance is improved.

  As described above, according to the second embodiment, since the filter processing unit 204 is provided to perform the filter process so that the target is not buried in the ground clutter side lobe, the clutter separation performance is improved, that is, the target detection performance is improved. Can be obtained.

Embodiment 3 FIG.
A radar apparatus according to Embodiment 3 of the present invention will be described with reference to FIGS. FIG. 20 is a processing block diagram showing the configuration of the radar apparatus according to Embodiment 3 of the present invention.

  20, the radar apparatus according to Embodiment 3 of the present invention includes an antenna 1, a transmission means 2, a transmission / reception switch 7, a reception means 100, a signal processor 200C, and a display 9. It has been.

  Since the radar apparatus according to the third embodiment is different from the radar apparatus according to the first embodiment except for the signal processor 200C, the same components are denoted by the same reference numerals, and description thereof is omitted. To do.

  That is, as shown in FIG. 20, the signal processor 200C replaces the beat frequency calculation means 203 of the PRI internal distance-beat frequency map creation means 201 of the signal processor 200 according to Embodiment 1 with a distance-beat frequency. The difference is that the map creation means 201C has the beat frequency calculation means 203B, and the others are the same, so the same parts are denoted by the same reference numerals, and description thereof will be omitted.

  The signal processor 200C performs the process shown in FIG. 21 on the received beat signal.

The beat frequency calculation means 203B calculates the window function wind _FM (n) for the inter-PRI FFT by the following equation (51). Here, the window function for the inter-PRI FFT is described as a Hamming window, but the same effect can be obtained when other window functions are used. Further, in Embodiment 3, code modulation is performed in PRI, but when modulation other than code modulation is performed in PRI, the same effect can be obtained by using a window function in PRI as shown in FIG. It is possible to obtain

The beat frequency calculation means 203B multiplies the window function wind _FM (n) for the FFT between the PRIs subjected to the window function processing by the following equation (52) and performs the window function processing in the PRI direction after the pulse compression. The signal R ″ _{V · Ex} (n, l) is calculated.

Beat frequency calculation unit 203B, the signal after the pulse compression _{R V · Ex (n, l} ) signal after pulse compression performing window function processing in PRI direction instead of the R "V _{· Ex} the (n, l) the formula PRI within a distance by IFFT at point H in the PRI direction according (30) - the beat frequency map _{R FM R PC (k h,} l) to output a.

  As shown in FIG. 23, by performing window function processing in the case of FFT between PRIs, ground clutter side lobes are reduced and target detection performance is improved.

  As described above, according to the third embodiment, the beat frequency calculation means 203B is provided to perform the window function processing so that the target is not buried in the ground clutter side lobe, so that the clutter separation performance is improved, that is, the target A radar device with improved detection performance can be obtained. Further, when the beat frequency calculating means 203 is used instead of the beat frequency calculating means 203B, the same effect can be obtained for each.

Embodiment 4 FIG.
A radar apparatus according to Embodiment 4 of the present invention will be described with reference to FIGS. FIG. 24 is a processing block diagram showing the configuration of the radar apparatus according to Embodiment 4 of the present invention.

  In FIG. 24, the radar apparatus according to Embodiment 4 of the present invention is provided with an antenna 1, a transmission means 2B, a transmission / reception switch 7, a reception means 100, a signal processor 200, and a display 9. ing.

  The radar apparatus according to the fourth embodiment of the present invention is different from the radar apparatus according to the first embodiment in the transmission means 2B, and is otherwise the same. .

  That is, as shown in FIG. 24, the transmission means 2B has a local oscillator 6B instead of the local oscillator 6 of the transmission means 2 according to Embodiment 1, and the other parts are the same. The same reference numerals are added and description thereof is omitted.

In the transmission means 2B, parameters of beat frequency resolution Δf _FM ′, sampling interval Δf _FM , distance resolution ΔR ′ _PC after correlation processing, and 1PRI folding distance R _pri satisfying equations (4) to (6) are set. .

Hereinafter, the operation of the local oscillator 6B will be described with reference to FIG. The local oscillator 6B generates a local oscillation signal L ₀ (t) expressed by the following equation (53) that is stepwise frequency modulated at the PRI interval with the transmission signal frequency sweep time T _L and the transmission signal bandwidth B _L. And output to the pulse modulator 5 and the receiving means 100. Here, the frequency modulation width Δf of the PRI interval is calculated by the equation (54).

The local oscillator 6B performs frequency modulation step by step, but the processing for the received beat signal performs the same processing, so the signal processor 200 performs the processing shown in FIG. 9 on the received beat signal. However, since frequency modulation is performed in stages, the sampling interval of the beat frequency is the same as the beat frequency resolution Δf _FM '. The need for frequency modulation within the pulse is eliminated, and the hardware scale can be reduced.

  As described above, according to the fourth embodiment, it is not necessary to perform frequency modulation within the pulse, the hardware scale can be reduced, and the target relative distance can be calculated with high accuracy. Can be obtained.

  The radar apparatus of the present invention can measure distances to a plurality of targets, and may be used for target detection of, for example, a plurality of ships and a plurality of aircraft.

  1 Antenna, 2, 2B transmission means, 3 Transmitter, 4 In-pulse modulation signal generator, 5 Pulse modulator, 6, 6B Local oscillator, 7 Transmission / reception switcher, 8 Receiver, 9 Display, 100 Receiving means, 200 , 200B, 200C signal processing means, 201, 201B, 201C PRI internal distance-beat frequency map creation means, 202 correlation means, 203, 203B beat frequency calculation means, 204 filter processing means, 220 target candidate detection means, 221 combination means, 222 Distance speed calculation means, 223 distance high resolution means.

Claims (5)

PRI (Pulse Repetition Interval) Different frequency modulation rates of transmitted signals based on the resolution of the distance without ambiguity that can eliminate the distance ambiguity of the inner distance and the parameter that makes the PRI inner distance resolution more accurate than the beat frequency resolution. Transmitting means for radiating a carrier signal frequency-modulated on the chirp over a plurality of PRIs with a plurality of selected frequency modulation rates;
A transmission signal obtained from the transmission means reflected back from the target is received as a reception signal, and the received signal is down-converted using the carrier signal obtained from the transmission means and converted into a reception beat signal. Receiving means;
Create PRI intra-distance-beat frequency map to calculate the relative distance to the target by performing inter-PRI FFT (Fast Fourier Transform) on the received beat signal obtained from the receiving means. Means,
Target candidate detection means for detecting a target candidate based on the signal strength for the PRI internal distance-beat frequency map obtained from the PRI internal distance-beat frequency map creating means;
For the target candidate obtained from the target candidate detecting means, a combination means for combining the target candidates using the PRI internal distance;
Using the combination obtained by the combination means, combining the beat frequencies of the target candidates, distance speed calculation means for calculating the distance and speed without ambiguity,
A radar having a high-resolution distance means for canceling the distance ambiguity of the intra-PRI distance and calculating a relative distance to the target using a distance without ambiguity obtained by the distance speed calculating means apparatus.   The radar apparatus according to claim 1, wherein the PRI internal distance-beat frequency map creating unit includes a filter processing unit that performs a filter process for reducing an influence of a clutter based on a relationship between the radar apparatus and the clutter. The PRI internal distance-beat frequency map creating means is:
Correlation means for performing correlation processing between the reference signal and the received beat signal from the receiver;
The beat frequency calculating means for performing window function processing for reducing the influence of clutter on the signal subjected to correlation processing by the correlation means, performing IFFT processing in the PRI direction, and outputting a PRI internal distance-beat frequency map. The radar apparatus according to 1. .   In place of the transmission means, the distance resolution of the distance without PRI that can eliminate the distance ambiguity of the PRI distance within PRI, and the distance resolution of PRI within the distance resolution of the distance without ambiguity is more accurate. 2. The radar apparatus according to claim 1, further comprising: a transmission unit that radiates a transmission signal that is pulse-modulated with PRI with respect to a carrier signal that is frequency-modulated stepwise over a plurality of PRIs based on parameters. . The transmission means applies intra-pulse modulation with PRI to the carrier signal,
5. The radar apparatus according to claim 1, wherein the PRI intra-distance-beat frequency map creation unit adds a correlation process with intra-pulse modulation to the received beat signal. 6. .

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