JP2011223429A - Filter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a filter capable of substantially reducing variation of characteristics in comparison with a conventional filter.SOLUTION: The filter includes: a filter part 22 that compensates the frequency characteristics of a input Vin and outputs the input signal; and a controller 23 that controls characteristics of the filter part 22. The filter part 22 is a filter circuit having a gyrator system with a transconductance circuit 24 that varies a mutual conductance with a control current Iand outputs a current depending on an input voltage. The controller 23 generates the control current Iby frequency-current conversion for a control clock signal CLK.

Description

  The present invention relates to a gyrator bandpass filter, a band elimination filter, and the like.

  Conventionally, various band-pass filters are used in electronic devices. As the band pass filter, an active filter composed of an amplifying element, a resistor and a capacitor, a tuned band pass filter using a resonance phenomenon of a coil and a capacitor, a band pass filter using a piezoelectric element, and the like are known.

  FIG. 9 is a connection diagram showing a band-pass filter using an active filter. This band pass filter 1 is a so-called voltage feedback type band pass filter. In the band pass filter 1, the inverting input terminal of the operational amplifier 2 is grounded by the ground resistor R 4, and the inverting input terminal of the operational amplifier 2 is connected to the output terminal of the operational amplifier 2 through the feedback resistor R 5. Further, the non-inverting input terminal of the operational amplifier 2 is grounded by a parallel circuit of the capacitor C2 and the resistor R3, and this non-inverting input terminal is connected to the input terminal Vin of the band pass filter 1 through the capacitor C1 and the resistor R1 in order. Is done. Further, the midpoint of connection between the resistor R1 and the capacitor C1 is connected to the output terminal of the operational amplifier 2 via the feedback resistor R2.

In the band pass filter 2, ω ₀ and Q can be expressed by the following equations, respectively. Note that ω ₀ is the central angular frequency of the pass band, and Q is the ratio of the bandwidth where the gain is −3 [dB] to the gain of the pass band and the central angular frequency ω ₀ of the pass band.

  FIG. 10 is a connection diagram illustrating another example of a bandpass filter using an active filter. This band pass filter 6 is a so-called multiple feedback type band pass filter. In the band pass filter 6, the non-inverting input terminal of the operational amplifier 7 is grounded by a resistor R3. Further, the inverting input terminal of the operational amplifier 7 is connected to the input terminal Vin of the band-pass filter 6 through the capacitor C2 and the resistor R1 in order, and the connection middle point of the resistor R1 and the capacitor C2 and the inverting input terminal of the operational amplifier 7 are respectively connected to the capacitor. The output terminal of the operational amplifier 7 is connected by C1 and a resistor R2. The resistors R2 and R3 are set to the same resistance value.

In this band pass filter 6, ω ₀ and Q can be expressed by the following equations, respectively.

FIG. 11 is a diagram showing a tunable bandpass filter. The band pass filter 11 is provided with a parallel resonance circuit (also called a tank circuit) including a resistor R, a capacitor C, and an inductor L, and this parallel resonance circuit is connected in parallel to a signal source Iin and the like. In this band pass filter 11, ω ₀ and Q can be expressed by the following equations, respectively.

FIG. 12 is a diagram showing a bandpass filter using a piezoelectric element. In the band-pass filter 15, a ceramic filter CF is applied as a piezoelectric element. In the band pass filter 15, the input end of the ceramic filter CF is connected to the signal source Vin via the matching resistor R, and the output end of the ceramic filter CF is terminated by the matching resistor R and connected to the buffer amplifier 16. In the band pass filter 15, ω ₀ and Q are determined by the characteristics of the ceramic filter CF. FIG. 13 shows an example of the frequency characteristics of the ceramic filter.

  Regarding such a filter, Japanese Patent Laid-Open No. 05-259808 proposes a device for adjusting the characteristics of a filter by a gyrator.

Japanese Patent Laid-Open No. 05-259808

  However, the characteristics of the filters described above with reference to FIGS. 9 to 12 cannot be set accurately unless the accuracy of the components is increased.

In other words, in the bandpass filters 1 and 6 (FIGS. 9 and 10), in order to reduce the variation in ω ₀ and Q, the resistances R1 and R2 and the capacitors C1 and C2 are increased according to the equations (1) to (4). It is necessary to use precision elements. Further, in the tunable bandpass filter 11 (FIG. 11), in order to reduce variations in ω ₀ and Q, it is necessary to increase the accuracy of the inductor L, the capacitor C, and the resistor R that constitute the resonance circuit. The inductor L uses a magnetic core material to ensure a high inductance value due to its small size, and the core material has a large variation in magnetic permeability. Therefore, in the tunable bandpass filter 11, the variation of the inductor L is large, and consequently, adjustment of the inductance value is unavoidable. Further, in the band-pass filter 15 (FIG. 12) using a piezoelectric element, it is necessary to reduce variations in characteristics of the piezoelectric element, and for this purpose, it is necessary to increase the processing accuracy of the piezoelectric element.

  As a result, these conventional filters have a problem in which variations in characteristics cannot be avoided. In the case where many products have the same characteristics, it is necessary to select parts after all.

  As one method for solving this problem, it is conceivable to control the characteristics by a microcomputer or the like to correct the variation, but in practice, this method cannot be applied. That is, in the bandpass filters 1 and 6 and the tunable bandpass filter 11 using the active filter, as is apparent from the equations (1) to (6), it is necessary to adjust the circuit constants such as capacitors after all. In practice, the characteristics cannot be controlled by a microcomputer or the like. In the band-pass filter 15 using a piezoelectric element, the characteristics of the filter are determined by the characteristics of the piezoelectric element itself. Therefore, the characteristics cannot be controlled by a microcomputer or the like.

Thus, when considering the case where the filter is integrated, the resistance value and capacitance inside the integrated circuit vary by about ± 20 [%] in absolute value, and relative variation within the same wafer by ± 5 [%]. It will be about. Considering this relationship by applying the equations (3) to (6), ω ₀ varies from 0.694 to 1.56 (−30 [%] to +56 [%]) due to the integration of the integrated circuit, and Q is 0. 952 to 1.052 (-4.3 [%] to +5.2 [%]). On the other hand, the variation of ω ₀ of the ceramic filter is ± 0.5 [%]. As a result, with the conventional active filter technology using semiconductors, frequency characteristics similar to those of ceramic filters could not be realized when integrated circuits were realized.

Note that each of the above-described filters has respective disadvantages and advantages. That is, with respect to Q, in the bandpass filters 1 and 6 and the tunable bandpass filter 11 using the active filter, the upper limit value of the Q value is limited due to variations in circuit constants. On the other hand, in the band pass filter 15 using a piezoelectric element, the Q value is about several tens from the balance with the variation of ω ₀ .

Further, in the tunable bandpass filter 11 that needs to be provided with an inductor, the inductor becomes large when ω ₀ is low, and as a result, the board mounting area becomes large. In addition, the band-pass filter 15 using a piezoelectric element has a remarkable deterioration in characteristics when used in a place where vibration occurs, and needs to cope with secondary and tertiary spurious. The band-pass filter 15 using a piezoelectric element requires a matching circuit. Accordingly, the degree of freedom in circuit design related to the input / output circuit is reduced, and the board mounting area is increased by the matching circuit.

  The present invention has been made in consideration of the above points, and an object of the present invention is to provide a filter capable of dramatically reducing the variation in characteristics as compared with the prior art. That is, this invention is comprised from the following technical matters.

(1) a filter unit for correcting and outputting the frequency characteristics of the input signal;
A control unit for controlling the characteristics of the filter unit,
The filter unit is
The frequency characteristic of the input signal is corrected by a gyrator type filter circuit using a transconductance circuit that outputs a current according to an input voltage by changing a mutual conductance by a control current,
The controller is
A filter comprising: a frequency-current conversion circuit that generates a control current by performing frequency-current conversion on a control clock signal.

  According to (1), even when the capacitance of the capacitor varies in the filter unit, this variation can be corrected and the frequency characteristic of the input signal can be corrected by a desired characteristic. Variations in characteristics can be reduced.

(2) In (1),
The filter unit is
A plurality of the gyrator type filter circuits are provided.

  According to (2), it can be applied to the case where the frequency characteristic of the input signal is corrected by the total characteristic of the plurality of filter circuits, and the characteristic variation can be significantly reduced as compared with the conventional case.

(3) In (2),
The controller is
A frequency adjustment control unit that outputs a control current for frequency adjustment that varies the center frequency or cut-off frequency of the plurality of filter circuits in the same direction;
A band adjustment control unit that outputs to each filter circuit a control current for band adjustment that varies the band according to the overall characteristics of the plurality of filter circuits;
A variation correction control unit that outputs a control current for variation correction of a center frequency or a cutoff frequency in the plurality of filter circuits to each filter circuit;
The frequency adjustment control current, band adjustment control current, and variation correction control current are added to each filter circuit and supplied to the corresponding filter circuit.

According to (3)
The center frequency or cut-off frequency, band, and variation can be individually adjusted and corrected.

(4) In (2) or (3),
The plurality of filter circuits are
A bandpass filter circuit;
A high-frequency band elimination filter that emphasizes attenuation characteristics on the high-frequency side of the band-pass filter circuit;
A low-frequency band elimination filter that emphasizes a low-frequency attenuation characteristic of the band-pass filter circuit.

  According to (4), when the band-pass filter is configured with emphasis on the attenuation characteristic, the characteristic variation can be significantly reduced as compared with the conventional case.

  According to the present invention, the variation in characteristics can be remarkably reduced as compared with the prior art. Further, it is possible to improve the frequency characteristics of the filter without generating spurious.

It is a connection diagram which shows the filter of the 1st Embodiment of this invention. It is a characteristic curve figure which shows the characteristic which concerns on the center frequency of the filter of FIG. It is a characteristic curve figure which shows the characteristic concerning Q of the filter of FIG. It is a characteristic curve figure which shows the characteristic which concerns on the center frequency of the filter of FIG. 1 at the time of changing a clock frequency. It is a characteristic curve figure with which it uses for description of the relationship between a pass band and an interference wave. It is a connection diagram which shows the filter of the 2nd Embodiment of this invention. It is a characteristic curve figure which shows the characteristic which concerns on the center frequency of the filter of FIG. It is a figure which shows the frequency characteristic of the filter of FIG. It is a connection diagram which shows the bandpass filter by the conventional active filter. It is a connection diagram which shows the other example of the band pass filter by the conventional active filter. It is a figure which shows the conventional tuning type | mold band pass filter. It is a figure which shows the bandpass filter using the conventional piezoelectric element. It is a figure which shows one example of the frequency characteristic of a ceramic filter.

(1) First Embodiment FIG. 1 is a diagram showing a filter according to a first embodiment of the present invention. The entire filter 21 is formed on the same wafer by an integrated circuit. In the filter 21, the frequency characteristic of the input signal is corrected by the filter unit 22, and the characteristic of the filter unit 22 is controlled by the control unit 23.

  Here, the filter unit 22 is a secondary filter using a gyrator, and in this embodiment, is configured by a band pass filter. Therefore, the filter unit 22 is configured by connecting transconductance circuits 24 and 26 and buffer circuits 25 and 27 in a ring shape.

  Here, the transconductance circuits 24 and 26 are differential amplifier circuits that change the mutual conductance by a control current and convert a differential input voltage into an output current. In the filter unit 22, the non-inverting input terminal of the first transconductance circuit 24 is connected to a predetermined reference power supply, and the inverting input terminal is connected to the output terminal of the buffer circuit 27. In the filter unit 22, the output end of the first transconductance circuit 24 is connected to the input end of the buffer circuit 25, and the input end of the buffer circuit 25 is connected to the signal source 28 to be processed via the capacitor C1. Connected. Further, in the filter unit 22, the output terminal of the buffer circuit 25 is connected to the non-inverting input terminal of the second transconductance circuit 26, and the output terminal of the buffer circuit 27 is connected to the inverting input terminal of the second transconductance circuit 26. Is done. Further, the output terminal of the second transconductance circuit 26 is input to the buffer circuit 27, and the input terminal of the buffer circuit 27 is grounded by the capacitor C2.

  The transfer function H (ω) of the filter unit 22 is expressed by the following equation. This relational expression is widely known in various textbooks.

Here, gm (gm1, gm2) is a mutual conductance related to the voltage-current conversion processing of the transconductance circuits 24 and 26. Β (β1, β2) is a feedback rate of the output of the buffer circuit 27 to the transconductance circuits 24 and 26, and is 1 in this embodiment. The _{Z 1} and _{Z 2} are the impedances of the capacitors C1, C2, respectively.

Further, ω ₀ and Q are each expressed by the following equations.

Here, since the filter unit 22 is a gyrator filter, the central angular frequency ω ₀ of the filter unit 22 is affected by variations in the capacitances C1 and C2, but is different from the active filter described above with reference to FIGS. The resistance value is not affected. On the other hand, when the integrated circuit is formed, the Q value varies within a range of about ± 2.5 [%] from the equation (9), and the same accuracy as the ceramic filter can be secured. Therefore, in this case, if ω ₀ can be set with high accuracy, practically sufficient characteristics can be ensured.

Therefore, in this embodiment, the control unit 23 controls the filter unit 22 to set ω ₀ to a desired angular frequency. For this reason, the control unit 23 performs frequency-current conversion on the external clock signal CLK, and generates a control current _ICNT according to the frequency of the external clock signal CLK. The variable transconductance of the transconductance circuit 24 by the control current _{I CNT.}

  That is, the control unit 23 inputs the clock signal CLK to the frequency voltage conversion unit 31 and inputs it to the triangular wave generation circuit 32. The clock signal CLK is output from a controller or the like of an electronic device in which the filter 21 is mounted, for example, at a frequency according to a desired pass band, and various methods can be applied as a generation method thereof. Specifically, for example, the clock signal CLK may be generated by dividing the master clock, and the division ratio may be set by measuring the characteristics of the filter unit 22 in advance.

  The triangular wave generating circuit 32 is a ramp circuit, and converts a rectangular wave clock signal CLK into a triangular wave signal and outputs the triangular wave signal. The comparison circuit 33 determines the signal level of the triangular wave signal with a predetermined reference voltage Vref, and outputs a rectangular wave signal based on the determination result. The smoothing circuit 34 smoothes the rectangular wave signal output from the comparison circuit 33. As a result, the frequency voltage conversion unit 31 outputs a frequency voltage conversion result with a signal level proportional to the frequency of the clock signal CLK.

The conversion circuit 35 performs a voltage-current conversion process on the frequency-voltage conversion result output from the frequency-voltage conversion unit 31, thereby outputting a control current _ICNT expressed by the following equation. Here, f _CLK is the frequency of the clock signal CLK. CR is a time constant related to the smoothing process in the smoothing circuit 34.

  Here, the mutual conductance gm of the transconductance circuits 24 and 26 can be expressed by the following equation. Here, R is the internal resistance (emitter resistance) of the differential amplifier circuit related to the mutual conductance gm of the transconductance circuits 24 and 26.

  By substituting the equation (10) into the equation (11), the following relational expression can be obtained, and the mutual conductance gm can be defined as a function proportional to the capacitance.

Substituting this relational expression (12) into the expression (8) cancels out the capacitance element with the denominator of the expression (8), and, as shown by the following expression, the frequency f of the clock signal CLK input from the outside. The central angular frequency ω0 of the passband can be set so as to depend only on _CLK .

As a result, as shown in FIG. 2, even if the internal resistance R of the transistor constituting the filter unit 2 and the capacitances of the capacitors C1 and C2 vary, the frequency f _CLK of the clock signal CLK is maintained at a constant frequency, thereby maintaining the angular frequency. ω0 can be held at a constant frequency. Furthermore, even when the capacitance C in the filter unit 2 varies, the variation in the angular frequency ω0 due to this variation can be corrected. Note that FIG. 2 is a characteristic curve diagram in which the change in the angular frequency ω0 due to the change in the CR product, which is the product of the resistor R and the capacitance C, is measured. In FIG. 2, the CR product is shown normalized by the design center value.

Here, FIG. 3 is a characteristic curve diagram showing a change in Q by comparison with FIG. FIG. 4 is a characteristic curve diagram showing a change in the angular frequency ω0 with respect to the frequency f _CLK of the clock signal CLK.

  If gm1 = gm2 = gm0, C1 = C2 = C0, and β1 = 1, (8) and (9) can be expressed by the following equations.

  Thereby, in this case, Q can be normalized to 1 without changing the central angular frequency ω0. Further, when gm1 = kgm0, gm2 = gm0 (k> 1), (8) and (9) can be expressed by the following equations.

  Thereby, it is understood that Q can be variously set by controlling the mutual conductance gm by comparing the equations (14) and (15). Further, when the mutual conductance gm is varied in this way, the central angular frequency ω0 also changes, so that Q can be varied while maintaining the central angular frequency ω0 at a constant value by controlling the feedback factor β1. Specifically, the feedback rate β1 is set to a value represented by the following equation by comparing the equations (14) and (15).

Then, the central angular frequency ω ₀ can be expressed by the following equation, and by using this relational expression, the feedback factor β 1 and the mutual conductance gm can be controlled to vary Q.

  According to the above configuration, the filter characteristic can be made to depend only on the clock frequency without depending on the resistance and the capacitance value, and thereby the characteristic variation can be remarkably reduced as compared with the conventional case.

  In addition, without depending on the resistance and capacitance values, various advantages can be achieved by actively utilizing the fact that the filter characteristics depend only on the clock frequency. That is, for example, when applied to an intermediate frequency filter of a communication apparatus, as shown in FIG. 5A, when an interference wave by an adjacent station exists in the pass band, as shown by an arrow A in FIG. The influence of the adjacent station can be reduced by moving the pass band of the intermediate frequency filter. In this case, the frequency of the clock signal CLK may be set so as to be manually variable, and the interference wave may be set outside the band. Further, for example, the interference wave may be automatically set out of the band by comparing the reception result of the target station by synchronous detection or the like with the reception result including the interference wave by envelope detection or the like.

Accordingly, in this embodiment, it is possible to set the center angular frequency ω ₀ with the accuracy of the clock signal by absorbing the dispersion of CR inside the semiconductor. Therefore, when the clock signal is created by a ceramic vibrator, the center angular frequency ω ₀ can be set to the accuracy of the ceramic vibrator, and when the clock signal is created by a crystal vibrator, The center angular frequency ω ₀ can be set with the accuracy of the crystal resonator. Further, it is possible to absorb fluctuations in resistance value and capacitance value due to temperature characteristics of the semiconductor, changes over time, and the like, and furthermore, it is possible to control the filter by changing the frequency of the clock signal. The clock signal can also be branched from a microcomputer or the like. Further, in this embodiment, it is possible to improve the frequency characteristics of the filter without generating spurious.

(2) Second Embodiment FIG. 6 is a diagram showing a filter according to a second embodiment of the present invention. In FIG. 6, the same configuration as the filter 21 described above with reference to FIG. 1 is denoted by the corresponding reference numeral, and redundant description is omitted. The filter circuit 41 is entirely formed on the same wafer by an integrated circuit. In the filter circuit 41, the filter unit 42 limits the band of a desired input signal, and the control unit 43 controls the characteristics of the filter unit 42.

  The filter unit 42 includes three filter circuits 42A, 42B, and 42C, and corrects the frequency characteristics of the input signal Vin based on the total characteristics of the three filter circuits 42A, 42B, and 42C. Here, each of the three filter circuits 42A, 42B, and 42C is a secondary filter using a gyrator. In this embodiment, the three filter circuits 42A, 42B, and 42C are connected in series, and the second-stage filter circuit 42B is set as a bandpass filter. Further, the filter circuit 42A in the preceding stage is assigned to a band elimination filter that emphasizes the attenuation characteristic on the low band side of the bandpass filter by the filter circuit 42B, and the filter circuit 42C in the latter stage has a high band in the bandpass filter by the filter circuit 42B. Is assigned to a band elimination filter that emphasizes the attenuation characteristics of the side.

  For this reason, the second stage filter circuit 42B is provided with feedback circuits 44B and 45B having feedback ratios β1 and β2, respectively, in the feedback loop that feeds back the output of the buffer circuit 27 in the final stage. The non-inverting input terminal is configured in the same manner as the filter unit 22 of the first embodiment except that the non-inverting input terminal is grounded. In the feedback circuits 44B and 45B, feedback rates β1 and β2 are variably controlled by a controller related to the control of the control unit 43 described later.

  The first-stage filter circuit 42A and the latter-stage filter circuit 42C are configured in the same manner as the filter circuit 42B, except for the configuration related to the band cutoff filter. Specifically, in the filter circuits 42A and 42C, an input signal is input to the non-inverting input terminal of the transconductance circuit 24 instead of the input to the buffer circuit 25 via the capacitor C1, and the capacitor C1 is grounded. The same configuration as that of the filter circuit 42B except that the input signal is input to the buffer circuit 27 via the capacitor C2 instead of the ground via the capacitor C2.

  The filter unit 42 inputs the input signal Vin to the series circuit of the filter circuits 42A, 42B, and 42C through the buffer 46, and outputs the output signal from the series circuit of the filter circuits 42A, 42B, and 42C through the buffer 47. .

The control unit 43 varies the central angular frequency ω ₀ of the plurality of filter circuits 42A, 42B, and 42C in the same direction, thereby changing the central angular frequency ω ₀ of the filter unit 42 and the frequency adjustment control unit 48A. , A bandwidth adjustment control unit 48B that varies the bandwidth (pass bandwidth in this embodiment) according to the overall characteristics of the plurality of filter circuits 42A, 42B, and 42C and varies the Q of the filter unit 42; And a variation correction control unit 48C for correcting variations in the center frequency in the plurality of filter circuits 42A, 42B, and 42C.

  The control units 48A, 48B, and 48C output six control currents corresponding to the transconductance circuits 24 and 26 of the filter circuits 42A, 42B, and 42C, respectively, and the control unit 43 outputs the six control currents to the control currents. The wiring patterns are connected to each other, and the control current is added for each system and output to the corresponding transconductance circuits 24 and 26 of the filter circuits 42A, 42B, and 42C.

That is, the frequency adjustment control unit 48A generates six control currents by the voltage-current conversion unit 49A. When the control terminal for controlling the control current is not set to any voltage, the frequency adjustment control unit 48A outputs a control current corresponding to the reference voltage Vref to each system. On the other hand, when a control signal is input to the control terminal and this control signal is higher than the reference voltage Vref, the control current is higher than the reference voltage Vref as much as the control signal voltage is higher than the reference voltage Vref. Increase the current value. On the other hand, when the control signal is lower than the reference voltage Vref, the current value of the control current is decreased by the amount that the control signal is lower than the reference voltage Vref as compared with the case of using the reference voltage Vref. Thus, the frequency adjustment control unit 48A outputs a control signal proportional to the voltage of the control signal to the filter circuits 42A, 42B, and 42C, and varies the center angular frequency ω ₀ of the filter unit 42 according to the control signal.

Similarly, the band adjustment control unit 48B generates six systems of control current by the voltage-current conversion unit 49B. Voltage-current conversion unit 49B in accordance with the voltage of the control signal, the filter circuit 42A relative to the center angular frequency omega ₀ of the filter circuit 42B is a band-pass filter, the center angular frequency omega ₀ of 42C to the variable, thereby controlling The band of the filter unit 42 is varied by the signal. In conjunction with this, the Q of the filter circuit 42B is varied. That is, when the control terminal is not set to any voltage, the voltage / current converter 49B outputs a control current corresponding to the reference voltage Vref to each system. On the other hand, when a control signal is input to the control terminal and this control signal is higher than the reference voltage Vref, the filter circuit 42A has a higher control signal voltage than the reference voltage Vref, compared to the case where the reference voltage Vref is used. Thus, the current value of the control current is increased. On the other hand, in the filter circuit 42C, the current value of the control current is decreased by the amount that the control signal voltage is higher than the reference voltage Vref as compared with the case of using the reference voltage Vref. In conjunction with this, the method described using the equations (14) to (19) is applied, and the Q of the filter circuit 42B is varied under the control of the host controller.

  Conversely, when a control signal is input to the control terminal of the voltage / current converter 49B and this control signal is lower than the reference voltage Vref, the voltage of the control signal is lower than the reference voltage Vref for the filter circuit 42A. The current value of the control current is decreased by an amount corresponding to the reference voltage Vref. On the other hand, in the filter circuit 42C, the current value of the control current is increased by an amount corresponding to the control signal voltage lower than the reference voltage Vref as compared with the case of using the reference voltage Vref. In conjunction with this, the method described using the equations (14) to (19) is applied, and the Q of the filter circuit 42B is varied under the control of the host controller.

  The variation correction control unit 48C is a frequency-current conversion circuit that outputs a control current according to the frequency of the clock signal CLK. The frequency-voltage conversion unit 31 generates a control signal with a voltage according to the frequency of the clock signal CLK, The control signal is converted into a control current by the voltage / current converter 35. In the correction control unit 48C, the smoothing capacitor C constituting the smoothing circuit is externally attached.

  FIG. 7 is a characteristic curve diagram showing changes in characteristics when the center frequency is changed in units of 5 [kHz] under the control of the frequency adjustment control unit 48A. FIG. 7 shows that the center frequency can be varied while maintaining the characteristics of the passband. FIG. 8 is a diagram showing the frequency characteristics of the filter 41 according to the second embodiment in comparison with FIG. According to FIG. 8, it can be seen that the filter characteristic is remarkably improved without generating spurious.

  According to the above configuration, even when the filter unit is configured by a plurality of filters, the same effects as those of the first embodiment can be obtained.

  In addition, a frequency adjustment control unit, a band adjustment control unit, and a variation correction control unit for adjusting the overall characteristics are provided so as to be configured by a plurality of filters, and control currents from these control units are added. Thus, by outputting to the corresponding filter unit, the center frequency, Q, and variation in the overall characteristics can be individually adjusted, and the configuration relating to filter control can be simplified.

  More specifically, the plurality of filters include a band-pass filter circuit, a high-frequency band elimination filter that emphasizes attenuation characteristics on the high-frequency side of the band-pass filter circuit, and a low-frequency side of the band-pass filter circuit. By being a low-frequency band elimination filter that emphasizes attenuation characteristics, a side lobe with a sufficient attenuation factor can be secured to form a band-pass filter, and the center frequency of this band-pass filter can be adjusted accurately. it can.

(3) Other Embodiments In the above-described embodiment, the preferred embodiment of the present invention has been described above. However, the present invention can be variously modified in the above-described embodiment without departing from the spirit of the present invention. The added form can be used.

That is, in the above-described first embodiment, the case where the present invention is applied to the bandpass filter has been described. However, the present invention is not limited to this, and the present invention is not limited to this. Can be widely applied. When applied to these low-pass filters, high-pass filters, etc., the cut-off frequency is applied instead of the above-mentioned center angular frequency ω ₀ as a reference on the frequency axis of the characteristics.

  In the second embodiment described above, a case is described in which a filter unit is configured by one band pass filter and two band elimination filters, and the total characteristics of these three filters connected in series are controlled. However, the present invention is not limited to this, and can be widely applied to the case of controlling the overall characteristics in which a plurality of filters are connected in various forms. Specifically, the present invention can be widely applied to a case where a pass band having a bimodal characteristic is secured by connecting two band pass filters in series.

  In the above-described embodiment, the case where the present invention is applied to the second-order filter according to the gyrator method has been described. it can.

  In the above-described embodiment, the case where the control signal is generated by performing the current conversion after the frequency conversion of the clock signal has been described, but the present invention is not limited to this, and the clock signal is directly converted into the frequency current. A control signal may be created.

1, 6, 11, 15, 21, 41, 42A, 42B, 42C Filter 2, 7 Operational amplifier 16 Buffer amplifier 22, 42 Filter unit 23, 43 Control unit 24, 26 Transconductance circuit 25, 27 Buffer circuit 28 Signal source 31 Frequency voltage conversion unit 35, 49A, 49B Voltage current conversion unit 48A Frequency adjustment control unit 48B Band adjustment control unit 48C Variation correction control unit

Claims (4)

A filter unit that corrects and outputs the frequency characteristics of the input signal;
A control unit for controlling the characteristics of the filter unit,
The filter unit is
The frequency characteristic of the input signal is corrected by a gyrator type filter circuit using a transconductance circuit that outputs a current according to an input voltage by changing a mutual conductance by a control current,
The controller is
A filter comprising: a frequency-current conversion circuit that generates a control current by performing frequency-current conversion on a control clock signal. The filter unit is
The filter according to claim 1, comprising a plurality of the gyrator type filter circuits. The controller is
A frequency adjustment control unit that outputs a control current for frequency adjustment that varies the center frequency or cut-off frequency of the plurality of filter circuits in the same direction;
A band adjustment control unit that outputs to each filter circuit a control current for band adjustment that varies the band according to the overall characteristics of the plurality of filter circuits;
A variation correction control unit that outputs a control current for variation correction of a center frequency or a cutoff frequency in the plurality of filter circuits to each filter circuit;
The filter according to claim 2, wherein the control current for frequency adjustment, the control current for band adjustment, and the control current for variation correction are added for each filter circuit and supplied to the corresponding filter circuit. The plurality of filter circuits are
A bandpass filter circuit;
A high-frequency band elimination filter that emphasizes attenuation characteristics on the high-frequency side of the band-pass filter circuit;
The filter according to claim 2 or 3, wherein the filter is a low-frequency band elimination filter that emphasizes a low-frequency attenuation characteristic of the band-pass filter circuit.

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