JP2011091563A - Receiver - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To highly accurately estimate the size of noise components regardless of the presence/absence of pilot signals. <P>SOLUTION: From received OFDM signals, first and second subcarrier signals by two subcarriers adjacent in a frequency direction are extracted. On an IQ plane, a distance between the signal point (pi0, pq0) of the first subcarrier signal and the signal point (si0, sq0) of the second subcarrier signal is obtained. In the meantime, distances between the signal points (si1, sq1), (si2, sq2) and (si3, sq3) for which the signal point (si0, sq0) of the second subcarrier signal is rotated clockwise by 90°, 180° and 270° and the signal point (pi0, pq0) of the first subcarrier signal are obtained. From the minimum value of the four obtained distances, the size of the noise components included in reception signals is estimated. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a receiving apparatus that receives a signal, and more particularly, to a receiving apparatus that receives a signal transmitted using orthogonal frequency division multiplexing (hereinafter also referred to as OFDM).

  An OFDM system that multiplexes many subcarriers orthogonal to each other is used in various communication systems. In a type of communication system using the OFDM system, a reference signal having a known fixed pattern is inserted into an OFDM signal, such as ISDB-T (Integrated Services Digital Broadcasting-Terrestrial), which is a standard adopted by Japanese terrestrial digital broadcasting. Sometimes. This reference signal is generally called a pilot signal.

  When a pilot signal is inserted in the OFDM signal, the magnitude of the noise component included in the received signal can be easily estimated by comparing the reference signal received as the pilot signal with the original signal (known fixed pattern). be able to. If estimation of the size of the noise component is made, it is possible to switch the demodulation method to an optimum one according to the estimated value.

  However, there are systems where a fixed pattern reference signal is not provided, such as digital audio broadcasting in Europe, etc., and it is possible to estimate the size of the noise component in the receiver of such a system. Have difficulty.

  There is also a system in which a reference signal having a known fixed pattern is included in a header portion at the beginning of a transmission frame (see, for example, Patent Document 1 below). Even in such a system, the magnitude of the noise component can be estimated by comparing the signal in the header part with the original signal (known fixed pattern). However, in principle, the estimated time interval cannot be shorter than the time interval between the header portions adjacent in time. Therefore, when the time interval between the adjacent header portions is long, the fluctuation in the size of the noise component increases between the adjacent header portions, which may cause a problem in estimation accuracy.

  Also, there are various demodulation methods for demodulating the modulated signal, but each demodulation method has its pros / cons. Therefore, the technology for switching the demodulation method to the optimum one according to the reception environment is naturally beneficial.

JP 2004-282207 A

  Accordingly, an object of the present invention is to provide a receiving apparatus that can perform noise estimation regardless of the presence or absence of a pilot signal. It is another object of the present invention to provide a receiving apparatus having a function of switching a demodulation method to a more suitable one.

  A receiving apparatus according to the present invention, in a receiving apparatus that receives a signal modulated using a method of multiplexing subcarrier groups orthogonal to each other, includes a plurality of subcarriers included in a received signal of the receiving apparatus. A noise estimation unit that estimates the magnitude of the noise component of the received signal by comparing subcarrier signals, and the plurality of subcarriers is a plurality of subcarriers different from each other in the frequency direction or the time direction. Features.

  This makes it possible to estimate the magnitude of the noise component regardless of the presence or absence of a pilot signal.

  For example, the receiving apparatus further includes a demodulation processing unit that demodulates the received signal, and the demodulation processing unit switches a demodulation method for demodulating the received signal according to an estimation result of the noise estimation unit. You may do it.

  As a result, the received signal is demodulated using a demodulation method corresponding to the magnitude of the noise component, and an improvement in reception characteristics is expected.

  Specifically, for example, when the estimated noise component size is relatively small, the demodulation processing unit demodulates the received signal by a demodulation method using delay detection, and the estimated noise component size Is relatively large, the received signal is demodulated by a demodulation method using synchronous detection.

  Also, specifically, for example, the plurality of subcarrier signals include first and second subcarrier signals, and the noise estimator includes the signal point of the first subcarrier signal on the IQ plane and the second subcarrier signal. The magnitude of the noise component may be estimated through comparison with the signal point of the subcarrier signal.

  More specifically, for example, the ideal position of the signal point of the second subcarrier signal on the IQ plane is any one of the first to Nth positions (N is an integer of 2 or more), and the noise The estimation unit obtains an amount based on a positional difference between the signal point of the first subcarrier signal and the signal point of the second subcarrier signal, and the signal point of the first subcarrier signal; The amount based on the position difference between the signal point obtained by applying the process of moving the position of 1 to the nth position on the signal point of the second subcarrier signal, n = 2, 3,... , N, and the magnitude of the noise component may be estimated using the smallest amount among the obtained amounts.

  Alternatively, for example, the ideal position of the signal point of the first subcarrier signal on the IQ plane is one of a plurality of reference positions including a specific reference position, and the second subcarrier signal on the IQ plane The ideal position of the signal point is any one of the first to Nth positions different from the respective reference positions (N is an integer of 2 or more), and the noise estimator is a signal point of the first subcarrier signal. And an amount based on the position difference between the signal point of the second subcarrier signal and the signal point obtained by performing the process of moving the nth position to the specific reference position, n = 1, 2, .., N may be obtained, and the magnitude of the noise component may be estimated using the smallest amount among the obtained amounts.

  Another receiving apparatus according to the present invention includes a demodulation processing unit that demodulates a received signal of the receiving apparatus in a receiving apparatus that receives a signal modulated using a method of multiplexing subcarrier groups orthogonal to each other, The demodulation processing unit switches a demodulation method for demodulating the reception signal based on a comparison result of a plurality of subcarrier signals by a plurality of subcarriers included in the reception signal.

  By comparing a plurality of subcarrier signals, it is possible to evaluate the reception environment (for example, the magnitude of noise). According to the other receiving apparatus, the received signal is demodulated using a demodulation method corresponding to the receiving environment, and an improvement in receiving characteristics is expected.

  According to the present invention, it is possible to provide a receiving apparatus capable of noise estimation regardless of the presence or absence of a pilot signal. In addition, it is possible to provide a receiving apparatus having a function of switching the demodulation method to a more suitable one.

  The significance or effect of the present invention will become more apparent from the following description of embodiments. However, the following embodiment is merely one embodiment of the present invention, and the meaning of the term of the present invention or each constituent element is not limited to that described in the following embodiment. .

It is a figure which shows the signal point of a QPSK signal. It is a figure which shows the signal point of a DQPSK signal. It is a schematic block diagram of the receiver which concerns on embodiment of this invention. It is a figure which shows the OFDM symbol sequence represented by the output signal of the FFT process part of FIG. FIG. 7A shows an example of signal point distribution of a received signal when the noise component is relatively small and the corresponding reference point is the first reference point, and the reference point corresponding to the relatively large noise component and the first reference point is the first reference point. It is a figure (b) which shows the example of signal point distribution of a received signal in the case of being a reference point. FIG. 7A shows an example of signal point distribution of a received signal when the noise component is relatively small and the corresponding reference point is the second reference point; and FIG. It is a figure (b) which shows the example of signal point distribution of a received signal in the case of being a reference point. It is a figure which shows the signal point referred by the magnitude | size estimation of a noise component. It is a figure which shows the signal point referred by the magnitude | size estimation of a noise component. It is a figure which shows the pair of a subcarrier signal compared for calculation of a noise evaluation value (1st example). It is a figure for demonstrating that the noise evaluation value (dif) for a fixed time is used for the magnitude | size estimation of the noise component in attention time. It is a figure which shows the relationship between CN ratio in a receiver, and the estimated value of the magnitude | size of a noise component. FIG. 4A is a diagram illustrating a correspondence relationship between reception characteristics using synchronous detection and a reception environment, and FIG. 5B is a diagram illustrating a correspondence relationship between reception characteristics using delay detection and the reception environment. It is a figure for demonstrating the significance of the phase angle of an attention signal point. It is a figure which shows the pair of the subcarrier signal compared for calculation of a noise evaluation value (2nd example). It is a figure which shows the pair of the subcarrier signal compared for calculation of a noise evaluation value (3rd example).

  Hereinafter, embodiments of the present invention will be specifically described with reference to the drawings. In each of the drawings to be referred to, the same part is denoted by the same reference numeral, and redundant description regarding the same part is omitted in principle. In the present specification, for the sake of simplification of description, a name corresponding to the code or symbol may be omitted or simplified by describing the code or symbol. For example, a subcarrier signal e (t, l) described later may be simply expressed as a signal e (t, l) or e (t, l).

  In communication according to the present embodiment, a DQPSK (Differential Quadrature Phase Shift Keying) system is used as a digital signal modulation system.

  DQPSK will be described while first explaining QPSK (Quadrature Phase Shift Keying) which is the basis of DQPSK.

When the two pieces of information data at the discrete time k are represented by a [k] and b [k], in QPSK, the QPSK signal represented by the following equation (A1) with respect to the information data a [k] and b [k] y [k] is assigned. Each of a [k] and b [k] is 1-bit data that takes 1 or (−1). j is an imaginary unit. √2 represents the positive square root of 2.
y [k] = (a [k] + j · b [k]) / √2 (A1)

  a and b represent first and second information data to be transmitted from the transmission device to the reception device, respectively, and y represents a QPSK signal. a [k], b [k], and y [k] represent the first information data, the second information data, and the QPSK signal, respectively, at the discrete time k. The time at which the unit time has elapsed from the discrete time (k−1) is the discrete time k, and the time at which the unit time has elapsed from the discrete time k is the discrete time (k + 1).

The signal z [k] modulated by DQPSK is expressed by the following equation (A2). A signal modulated by DQPSK is called a DQPSK modulated signal. z [k-1] is a DQPSK modulated signal at discrete time (k-1), and z [k] is a DQPSK modulated signal at discrete time k.
z [k] = z [k−1] · y [k] (A2)

The signal point of the complex signal QPSK signal y [k] is one of the following four signal points as shown in FIG.
(-1 / √2, 1 / √2), (1 / √2, 1 / √2),
(-1 / √2, -1 / √2), (1 / √2, -1 / √2)

  In this specification, when the real part and the imaginary part of an arbitrary complex signal are represented by Re and Im, respectively, the complex signal may be represented by “Re + j · Im” or (Re, Im). In a complex signal, the real part is called an in-phase component and the imaginary part is called a quadrature component. Further, as adopted in general digital modulation / demodulation considerations, the in-phase component and the quadrature component are assigned the I-axis and the Q-axis, respectively, and the complex plane is referred to as the IQ plane. The Q axis is obtained by rotating the I axis counterclockwise by 90 °. A point where a complex signal represented by “Re + j · Im” should be arranged on the IQ plane is represented as a signal point (Re, Im). Unless otherwise specified, signal points in the following description refer to signal points on the IQ plane.

The process of multiplying an arbitrary signal point on the IQ plane by the QPSK signal y [k] is equivalent to the process of rotating the position of the signal point by 315 °, 45 °, 135 °, and 225 °. The relationship between the four types of QPSK signals y [k] and the rotation angle is shown below. For example, when a complex signal whose signal point is (1, 0) is multiplied by y [k] = (1 / √2, 1 / √2), the signal point of the complex signal is 315 on the IQ plane. The position of the signal point changes from (1, 0) to (1 / √2, 1 / √2). In the present embodiment, rotation of a signal or signal point means rotation in the clockwise direction about the origin of the IQ plane unless otherwise specified.
(1 / √2, 1 / √2) ・ ・ ・ 315 °
(1 / √2, -1 / √2) ・ ・ ・ 45 °
(-1 / √2, -1 / √2) ... 135 °
(-1 / √2, 1 / √2) 225 °

Therefore, as can be seen from the equation (A2), the signal point of the DQPSK modulated signal is either one of the four first signal points shown in FIG. 2A or the four second signal points shown in FIG. Thus, a state where the signal point of the DQPSK modulation signal becomes the first signal point and a state where the signal point of the DQPSK modulation signal becomes the second signal point alternately occur.
The first signal point group includes four signal points (−1 / √2, 1 / √2), (1 / √2, 1 / √2),
(-1 / √2, -1 / √2), (1 / √2, -1 / √2)
Consisting of
The second signal point group includes four signal points (1, 0), (0, 1), (-1, 0), (0, -1).
Consists of.

  The DQPSK modulated signal received by the receiving device is superimposed with noise added in the transmission path between the transmitting device and the receiving device, and the position of the signal point of the received DQPSK modulated signal is an ideal signal. It deviates from the position of the point (however, there may be a case where there is no deviation at all). An ideal signal point means a signal point that is not affected by noise.

  Hereinafter, signal points (−1 / √2, 1 / √2), (1 / √2, 1 / √2), (−1 / √2, −1 / √2), which are ideal signal points, and (1 / √2, −1 / √2) is collectively referred to as a first reference point group, and each signal point forming the first reference point group is referred to as a first reference point. Similarly, signal points (1, 0), (0, 1), (-1, 0), and (0, -1), which are ideal signal points, are collectively referred to as a second reference point group. Each signal point forming the two reference points is called a second reference point. In addition, the first and / or second reference point may be simply referred to as a reference point.

  FIG. 3 shows a schematic block diagram of a digital broadcast receiver 10 (hereinafter abbreviated as receiver 10) according to an embodiment of the present invention.

  The receiving device 10 and a transmitting device (digital broadcasting transmitting device) (not shown) constitute a digital broadcasting system, which employs an OFDM (Orthogonal Frequency Division Multiplexing) method as a broadcasting method. . The OFDM method is a method of multiplexing a number of subcarriers (hereinafter also simply referred to as carriers) that are orthogonal to each other within a band of one channel. In the present embodiment, it is assumed that the transmission device is fixed at a certain position. When the receiving device 10 is mounted on a moving body such as a vehicle, the receiving device 10 may receive a signal while moving.

  In the transmission apparatus, for each subcarrier, the subcarrier is modulated in accordance with the baseband signal to be transmitted, and an inverse fast Fourier transform (IFFT) is performed on the signal obtained by the modulation to perform OFDM. Generate a signal. The generated OFDM signal is transmitted from the transmission device as a signal to be transmitted to the reception device 10 by digital broadcasting (for example, so-called terrestrial digital broadcasting). The baseband signal includes a video signal and an audio signal to be transmitted, and is formed from a column of first information data a and a column of second information data b. As described above, in the present embodiment, it is assumed that DQPSK is used for the modulation scheme.

  The receiving device 10 receives a digital broadcast signal transmitted by this digital broadcast. The digital broadcast signal includes OFDM signals for a plurality of channels. The receiving device 10 includes a receiving antenna 11, a tuner unit 12, an FFT processing unit 13, a noise estimation unit 14, and a demodulation processing unit 15.

  The tuner unit 12 receives the above digital broadcast signal via the receiving antenna 11, and extracts and outputs an OFDM signal corresponding to the selected channel from the received digital broadcast signal. The OFDM signal output from the tuner unit 12 is sent to the FFT processing unit 13.

  The FFT processing unit 13 converts the received OFDM signal from a signal on the time domain to a signal on the frequency domain using a fast Fourier transform (FFT). The OFDM signal on the frequency domain obtained by this conversion is sent to the noise estimation unit 14 and the demodulation processing unit 15.

  The demodulation processing unit 15 performs demodulation processing for converting the OFDM signal on the frequency domain transmitted from the FFT processing unit 13 into a baseband signal. The baseband signal obtained by this demodulation processing is, for example, an MPEG (Moving Picture Experts Group) encoded signal. The MPEG encoded signal is a signal obtained by encoding a video signal or an audio signal to be transmitted by digital broadcasting. The MPEG encoded signal obtained by this demodulation processing is sent to an MPEG decoder (not shown), decoded, then sent to a display device or a speaker (both not shown), and displayed as video or output as audio. . The receiving device 10 can also include an MPEG decoder, a display device, and a speaker.

  As shown in FIG. 4, the OFDM signal output from the FFT processing unit 13 is composed of signals that are two-dimensionally arranged in the frequency direction and the time direction (hereinafter referred to as subcarrier signals). Call. The frequency direction and the time direction are also called a carrier direction and a symbol direction, respectively. Although a fixed pattern reference signal may be inserted as a pilot signal in an OFDM symbol sequence in a general communication system, a pilot signal is not inserted in the OFDM symbol sequence in the present embodiment. (However, a pilot signal may be inserted).

  A time number (symbol number) corresponding to the time direction is represented by t, and a carrier number corresponding to the frequency direction is represented by l. Here, t is an integer of 0 or more, and l is an integer of 0 or more and (L−1) or less. L is the total number of subcarriers forming the OFDM signal. The frequencies of the L subcarriers forming the OFDM signal are different from each other and the L subcarriers are orthogonal to each other. It is assumed that the frequency of the corresponding subcarrier increases as l increases. t represents a time considered with the symbol length of the OFDM signal as a unit. The time (t + 1) is the time when the time corresponding to the symbol length of the OFDM signal has elapsed from time t. A position in the OFDM symbol string that is uniquely determined by uniquely defining t and l is called a carrier position, and is represented by (t, l). Further, the subcarrier signal arranged at the carrier position (t, l) is represented by e (t, l). The subcarrier signal e (t, l) indicates a signal by the subcarrier of the carrier number l among the received signals at the time t (in other words, the symbols received at the time t).

The reference point is exchanged between the first and second reference points in symbol units. For concrete description, the reference point for the subcarrier signal e (t, l) when t is an even number is the first reference point, and the subcarrier signal e (t, l) when t is an odd number. The reference point for is a second reference point. Furthermore, introducing the variable t _O take any even value of greater than zero. It is assumed that the types of reference points are common in one symbol. That is, all the reference points for the subcarrier signals e (t _O , 0) to e (t _O , L-1) are the first reference points, and the subcarrier signals e (t _O +1,0) to e (t _O All reference points for +1, L-1) are second reference points.

In transmitting apparatus, each sub-carrier is modulated by DQPSK, subcarrier signal _{e (t O, 0) ~e} (t O, L-1) first reference point of any of a first reference point group in together they are assigned, either the second reference points forming a second reference point group is assigned to the subcarrier signal _{e (t O +1,0) ~e (} t O + 1, L-1).

  If the same reference point is assigned to two subcarriers that have the same time number and are adjacent to each other in the frequency direction, the amplitude and phase do not vary much between the received signals for the two subcarriers. Taking the difference between the received signals for the two subcarriers (ie, the two subcarrier signals for the two subcarriers) cancels the signal component and leaves the noise component. Therefore, it is possible to estimate the magnitude of the noise component from the difference.

  Actually, since there are four types of signal components of each subcarrier signal to be compared, four differences that may be caused by the types of signal components are calculated, and the minimum of the four differences is calculated. It is necessary to obtain a value and determine the minimum value as a difference to be finally obtained (details will be described later).

  By the time the signal transmitted from the transmission device is received by the reception device 10, the signal deteriorates due to various factors, and the magnitude of the noise component at the time of reception changes. When the noise component increases, the distribution of signal points of the received signal spreads with reference to the reference point.

  FIGS. 5A and 5B and FIGS. 6A and 6B show examples of signal point distributions of received signals. FIGS. 5A and 5B and FIGS. 6A and 6B show the distribution of signal points of the subcarrier signal transmitted from the FFT processing unit 13. FIG. 5A shows a signal point distribution when the noise component is relatively small and the corresponding reference point is the first reference point, and FIG. 5B shows a relatively large noise component and corresponding. The signal point distribution when the reference point is the first reference point is shown. FIG. 6A shows the signal point distribution when the noise component is relatively small and the corresponding reference point is the second reference point. FIG. 6B shows the signal point distribution when the noise component is relatively large and the corresponding reference point is the second reference point. Each subcarrier signal transmitted from the FFT processing unit 13 is a DQPSK modulated signal including a noise component, and the signal point of each subcarrier signal is shifted from the reference point (however, in some cases, it may not be shifted at all). And the deviation tends to increase as the noise component increases.

A specific method for estimating the magnitude of the noise component by the noise estimation unit 14 will be described. The noise estimation unit 14 compares two subcarrier signals that belong to the same symbol and are adjacent to each other in the frequency direction. The two subcarrier signals to be compared are represented by e (t _A , l _A ) and e (t _A , l _A +1) for convenience. l _A is an integer of 0 or more and (L-2) or less. t _A is an arbitrary integer greater than or equal to zero, but it is assumed that t _A is an even number for the sake of illustration. Accordingly, the reference point corresponding to the subcarrier signals e (t _A , l _A ) and e (t _A , l _A +1) is the first reference point.

The in-phase component and the quadrature component of the subcarrier signal e (t _A , l _A ) are represented by pi0 and pq0, respectively, and the in-phase component and the quadrature component of the subcarrier signal e (t _A , l _A +1) are si0 and sq0, respectively. Represented by Furthermore,
The in-phase component and the quadrature component of the signal obtained by rotating the subcarrier signal e (t _A , l _A +1) by 90 ° on the IQ plane are represented by si1 and sq1, respectively.
In-phase and quadrature components of a signal obtained by rotating the subcarrier signal e (t _A , l _A +1) by 180 ° on the IQ plane are represented by si2 and sq2, respectively.
In-phase and quadrature components of a signal obtained by rotating the subcarrier signal e (t _A , l _A +1) by 270 ° on the IQ plane are represented by si3 and sq3, respectively.

In FIG. 7, the reference points for the subcarrier signal e (t _A , l _A ) are (1 / √2, 1 / √2) and the reference points for the subcarrier signal e (t _A , l _A +1) are Signal points (pi0, pq0), (si0, sq0), (si1, sq1), (si2, sq2), and (si3, sq3) assuming that (1 / √2, −1 / √2). ) Is shown as an example.
The process of rotating the subcarrier signal e (t _A , l _A +1) by 90 °, 180 ° or 270 ° on the IQ plane is performed by setting the reference point of the subcarrier signal e (t _A , l _A +1) to This is equivalent to the process of moving from the first reference point to another first reference point. For example, when the reference point for the subcarrier signal e (t _A , l _A +1) is (1 / √2, −1 / √2) (see FIG. 7), the subcarrier signal e (t _A on the IQ plane). , L _A +1) are rotated by 90 °, 180 °, 270 °, the reference points of the rotated subcarrier signals e (t _A , l _A +1) are (−1 / √2, −1), respectively. / √2), (−1 / √2, 1 / √2), (1 / √2, 1 / √2).

The noise estimating unit 14, a subcarrier signal e (t _A, l _A) with one reference point for it is the first reference point of any is not known, the subcarrier signal _{e (t A, l A +1} ) criteria for It is unknown which point is the first reference point. On the other hand, in order to estimate the magnitude of the noise component, the difference in signal components between the subcarrier signals e (t _A , l _A ) and (t _A , l _A +1) (that is, the difference in reference points) is determined. It is necessary to cancel.

Therefore, the noise estimation unit 14 is based on the amount dif0 based on the positional difference between the signal points (pi0, pq0) and (si0, sq0) and the positional difference between the signal points (pi0, pq0) and (si1, sq1). The quantity dif1, the quantity dif2 based on the positional difference between the signal points (pi0, pq0) and (si2, sq2), and the quantity dif3 based on the positional difference between the signal points (pi0, pq0) and (si3, sq3) It calculates | requires separately and calculates | requires the minimum value in dif0-dif3 as noise evaluation value dif. The noise evaluation value dif is treated as an evaluation value representing the magnitude of the noise component in which the difference between the signal components between the subcarrier signals e (t _A , l _A ) and (t _A , l _A +1) is cancelled. This is because if the noise component is not abnormally large, the minimum value of dif0 to dif3 is considered to be most likely a value that reflects only the magnitude of the noise component. In the example illustrated in FIG. 7, dif3 is selected as the noise evaluation value dif (or dif3 is highly likely to be selected as the noise evaluation value dif).

Specifically, the quantities dif0 to dif3 can be obtained according to the following formulas (B0) to (B3).
dif0 = (pi0−si0) ² + (pq0−sq0) ² (B0)
dif1 = (pi0−si1) ² + (pq0−sq1) ² (B1)
dif2 = (pi0−si2) ² + (pq0−sq2) ² (B2)
dif3 = (pi0−si3) ² + (pq0−sq3) ² (B3)
Dif0, dif1, and dif2, and dif3 according to the equations (B0) to (B3) are respectively the distance between the signal points (pi0, pq0) and (si0, sq0), the signal points (pi0, pq0), and (si1, sq1). Between the signal points (pi0, pq0) and (si2, sq2), and the distance between the signal points (pi0, pq0) and (si3, sq3). Of course, the square root of each right side of the formulas (B0) to (B3) can be set to dif0 to dif3.

The method for deriving the noise evaluation value dif when t _A is an even number has been described. However, when t _A is an odd number, that is, the subcarrier signals e (t _A , l _A ) and e (t _A , l _A +1). When the reference point corresponding to) is the second reference point, the noise evaluation value dif can be obtained by the same method as described above. In FIG. 8, when t _A is an odd number, the reference point for the subcarrier signal e (t _A , l _A ) is (−1, 0) and the subcarrier signal e (t _A , l _A +1) Of the signal points (pi0, pq0), (si0, sq0), (si1, sq1), (si2, sq2), and (si3, sq3) when it is assumed that the reference point for (1, 0) is An example is shown. In the example illustrated in FIG. 8, dif2 is selected as the noise evaluation value dif (or dif2 is highly likely to be selected as the noise evaluation value dif).

In addition, the noise estimation unit 14 compares the subcarrier signal e (t _A , l with respect to the case where l _A is 0, 1, 2, ..., (L-3) and (L-2). _A ) and e (t _A , l _A +1) noise evaluation value dif can be obtained.

  FIG. 9 shows a pair of subcarrier signals for which the noise evaluation value dif is obtained. In FIG. 9, pairs of subcarrier signals for which the noise evaluation value dif is obtained are connected by arrows (the same applies to FIGS. 14 and 15 described later). Thus, the noise estimation unit 14 includes a pair of e (0,0) and e (0,1), a pair of e (0,1) and e (0,2), ..., e (0, L-2) and e (0, L-1) pairs, e (1,0) and e (1,1) pairs, e (1,1) and e (1,2) pairs, .., E (1, L-2) and e (1, L-1) pairs, e (2,0) and e (2,1) pairs,... The value dif can be determined.

  The noise estimation unit 14 stores the noise evaluation value dif for a certain time in a memory (not shown) in the receiving apparatus 10 and takes the average of the noise evaluation value dif for the certain time to thereby determine the magnitude of the noise component. Is estimated. For example, when estimating the magnitude of a noise component at a certain time of interest, as shown in FIG. 10A, Q noise evaluation values obtained between a time before the time of interest and a time of interest are obtained. The average value of dif is calculated (Q is an integer of 2 or more), and the average value is treated as an estimated value of the size of the noise component at the time of interest, or an amount corresponding to the average value of the noise component at the time of interest Treated as an estimate of size. The estimated value of the noise component increases as the average value of the Q noise evaluation values dif increases, and the estimated value of the noise component decreases as the average value of the Q noise evaluation values dif decreases. It shall be. Note that the magnitude of the noise component estimated by the noise estimation unit 14 is the magnitude of the noise component when the magnitude of the signal component is normalized with a fixed value.

In the method described above with reference to FIG. 10 (a), the magnitude of the noise component is estimated for each time, but the entire section is divided into a plurality of unit sections, and the noise component is divided for each unit section. You may make it estimate a magnitude | size. That is, for example, as shown in FIG. 10B, the unit interval between times T ₁ and T _{2 is} referred to at P ₁₂ , the unit interval between times T ₂ and T _{3 is} referred to at P ₂₃ , and time T _When the unit interval between ₃ and T _{4 is} referred to at P ₃₄ , the average value of the Q noise evaluation values dif obtained during the unit interval P ₁₂ or the amount corresponding to the average value is expressed as the unit interval P _23. the amount corresponding to the average value or the average value of the noise evaluation value dif handling as the magnitude estimate of the Q-number obtained during a unit interval P ₂₃ noise component at each time in the, in the unit section P ₃₄ May be treated as an estimated value of the magnitude of the noise component at each time (same for subsequent unit intervals). According to this method, in each unit interval, the estimated value of the noise component in the unit interval is constant, but compared with the method of estimating the noise component size for each time point, Requires less memory capacity. When i is a natural number, time T _{i + 1} is a time visited after time T _i .

  FIG. 11 is a simulation result showing the relationship between the CN ratio (Carrier to Noise Ratio) and the estimated value of the noise component in the receiving apparatus 10. In the graph of FIG. 11, the horizontal axis is the CN ratio (unit is decibel), and the vertical axis is the estimated value of the noise component. In FIG. 11, a solid curve 201 is a simulation result when only AWGN (Additive White Gaussian Noise) affects the signal in the transmission path between the transmission device and the reception device 10. Dashed curve 202 is affected by Rayleigh fading in accordance with AWGN (Additive White Gaussian Noise) and TU-6 (Typical Urban 6-path Rayleigh fading channel model) in the transmission line between the transmission device and the reception device 10. It is a simulation result in the case of doing. The Doppler frequency, which is a parameter of TU-6, was 80 Hz.

  As the CN ratio increases and decreases, the estimated value of the noise component also increases and decreases. That is, as can be seen from FIG. 11, the magnitude of the noise component can be estimated well by the above estimation method.

  The demodulation processing unit 15 in FIG. 3 can demodulate the output signal of the FFT processing unit 13 by using any one of a plurality of demodulation methods. The plurality of demodulation systems include a first demodulation system suitable for demodulating a signal having a relatively small noise component and a second demodulation suitable for demodulating a signal having a relatively large noise component. System. The first demodulation method is a demodulation method suitable for demodulating a signal having a relatively small fading effect, and the second demodulation method is a demodulation method suitable for demodulating a signal having a relatively large fading effect. It may be interpreted as. By receiving a signal while the receiving apparatus 10 is stationary, a signal that includes a relatively small influence of fading (including a signal that is not affected by fading) is received by the receiving apparatus 10, and the receiving apparatus 10 By receiving a signal in a moving state, a signal that has a relatively large influence of fading is received by the receiving device 10.

  The demodulation processing unit 15 switches the demodulation method actually used according to the estimation result of the noise estimation unit 14. For example, when the estimated value of the magnitude of the noise component obtained by the noise estimation unit 14 is smaller than a predetermined reference value, the magnitude of the noise component is considered to be relatively small, and the FFT is performed using the first demodulation method. When the output signal of the processing unit 13 is demodulated and the estimated value of the noise component obtained by the noise estimating unit 14 is equal to or larger than the reference value, the noise component is regarded as being relatively large. The output signal of the FFT processing unit 13 is demodulated using the second demodulation method. The reason why the demodulation method is switched according to the magnitude of the noise component is that there is a higher possibility that the reception characteristic is improved when the demodulation method is switched between when the noise component is small and when the noise component is large.

  The first demodulation method is, for example, a demodulation method using delayed detection, and the second demodulation method is, for example, a demodulation method using synchronous detection. The demodulation method using delay detection here is an arbitrary demodulation method using arbitrary delay detection that can be used for demodulation of the OFDM signal, and the demodulation method using synchronous detection here is an OFDM signal demodulation method. Any demodulation method using any synchronous detection that can be used for demodulation. As delay detection, synchronous detection, and demodulation methods using them, known ones can be used, or those not known can be used. For example, the demodulation method described in Japanese Patent Application No. 2009-182279 proposed by the present applicant is realized using synchronous detection, but the demodulation method described in Japanese Patent Application No. 2009-182279 is used as the second demodulation method. It is also possible to use it.

  In the demodulation method using delay detection, when demodulating the received signal of the target symbol, the received signal one symbol before the target symbol is used as a reference signal, and the reference signal and the received signal of the target symbol are compared to compare the received signal of the target symbol. Demodulation is performed by determining the phase of the received signal. That is, for example, in the demodulation system using delay detection, when demodulating the subcarrier signal e (t, l), the subcarrier signal e (t-1, l) is used as the reference signal, and the subcarrier as the reference signal is used. The phase of the subcarrier signal e (t, l) is determined by comparing the signal e (t-1, l) with the subcarrier signal e (t, l), and demodulation is performed thereby.

  In the demodulation method using the synchronous detection, the subcarrier whose phase is synchronized with that of the subcarrier used when the transmission device performs the modulation is reproduced in the reception device 10 based on the reception signal of the reception device 10 to thereby perform the phase detection. Is generated by generating a phase signal serving as a reference for the FFT and comparing the output signal of the FFT processing unit 13 with the phase signal serving as the reference.

  In general, when subcarriers that are carrier waves can be accurately reproduced, synchronous detection has better reception characteristics than when delayed detection is used. However, when fading occurs due to mobile reception or the like and the phase of the received signal rotates, the carrier wave cannot be reproduced accurately, and the reception characteristics due to synchronous detection deteriorate. Since delay detection is less susceptible to fading, delay detection generally has better reception characteristics than synchronous detection when mobile reception is performed. On the other hand, when delay detection is used, reception characteristics deteriorate rapidly when noise increases.

  FIG. 12A shows the correspondence between reception characteristics using synchronous detection and the reception environment, and FIG. 12B shows the correspondence between reception characteristics using delay detection and the reception environment. When the reception characteristic is simply binarized with good or bad, when using synchronous detection, the reception characteristic in stationary reception is good regardless of the magnitude of the noise component, and the reception characteristic in mobile reception is bad. When delay detection is used, the reception characteristics when the noise component is relatively large are poor in both stationary reception and mobile reception, and the reception characteristics when the noise component is relatively small are good. Here, “good” and “bad” do not represent absolute amounts of reception characteristics, but represent relative amounts when delay detection and synchronous detection are compared. As can be seen from FIGS. 12A and 12B, the use of delay detection is suitable when the noise component is small, while the use of synchronous detection is suitable when the noise component is large. Considering this, the demodulation processing unit 15 switches the demodulation method as described above according to the estimation result of the noise estimation unit 14.

  In the above example, the amounts dif0 to dif3 based on the positional difference between the signal points are obtained according to the above formulas (B0) to (B3). However, the amounts dif0 to dif3 are calculated from the phase angle difference between the signal points. May be obtained, or the amounts dif0 to dif3 may be obtained from other indices depending on the positional difference between signal points.

When obtaining the quantities dif0 to dif3 from the difference in phase angle between signal points, for example,
The absolute value of the difference between the phase angle of the signal point (pi0, pq0) and the phase angle of the signal point (si0, sq0) is defined as an amount dif0 based on the positional difference between the signal points (pi0, pq0) and (si0, sq0). Seeking
The absolute value of the difference between the phase angle of the signal point (pi0, pq0) and the phase angle of the signal point (si1, sq1) is defined as an amount dif1 based on the positional difference between the signal points (pi0, pq0) and (si1, sq1). Seeking
The absolute value of the difference between the phase angle of the signal point (pi0, pq0) and the phase angle of the signal point (si2, sq2) is defined as an amount dif2 based on the positional difference between the signal points (pi0, pq0) and (si2, sq2). Seeking
The absolute value of the difference between the phase angle of the signal point (pi0, pq0) and the phase angle of the signal point (si3, sq3) is defined as an amount dif3 based on the positional difference between the signal points (pi0, pq0) and (si3, sq3). Can be sought.
The phase angle of a signal point of interest is an angle formed by the line segment 211 and the line segment 212 when the line segment 212 is viewed in a counterclockwise direction from the line segment 211 as shown in FIG. A line segment 211 is a line segment extending from the origin along the I axis in the positive direction of the I axis with the origin of the IQ plane as one end point, and a line segment 212 is the origin of the IQ plane and the signal of interest It is a line segment with a point at both ends.

In the above example, when the noise evaluation value dif is calculated by comparing two subcarrier signals, the subcarrier signals e (t, l) and e (t, l + 1) in the same symbol adjacent in the frequency direction are calculated. The two subcarrier signals to be compared are treated as two subcarrier signals, but the two subcarrier signals to be compared may be separated by two or more carrier numbers in the frequency direction. That is, the noise evaluation value dif may be calculated by comparing the subcarrier signals e (t, l) and e (t, l + l _B ) in the same symbol (l _B is an integer of 2 or more). In this case, pi0 and pq0 are determined from the signal points of the subcarrier signal e (t, l), and si0 to si3 and sq0 to sq3 are determined from the signal points of the subcarrier signal e (t, l + l _B ).

FIG. 14 shows a pair of subcarrier signals to be compared when the noise evaluation value dif is obtained from the subcarrier signals e (t, l) and e (t, l + 2) in the same symbol (in this case, l _B is 2). In the case of the example illustrated in FIG. 14, the noise estimation unit 14 sets a pair of e (t, 0) and e (t, 2), e (t, 1), and e (t, 3) for an arbitrary time number t. , E (t, 2) and e (t, 4), e (t, 3) and e (t, 5), ..., e (t, L-3) and e ( The noise evaluation value dif for the pair of t, L-1) can be obtained.

Alternatively, the two subcarrier signals to be compared to calculate the noise evaluation value dif may be two subcarrier signals having different time numbers t to which they belong. That is, the noise evaluation value dif may be calculated by comparing the subcarrier signals e (t, l) and e (t + t _B , l) (t _B is an integer of 1 or more). In this case, pi0 and pq0 are determined from the signal points of the subcarrier signal e (t, l), and si0 to si3 and sq0 to sq3 are determined from the signal points of the subcarrier signal e (t + t _B , l).

FIG. 15 shows a pair of subcarrier signals to be compared when a noise evaluation value dif is obtained from two subcarrier signals e (t, l) and e (t + 1, l) adjacent in the time direction ( In this case, t _B is 1.) In the case of the example illustrated in FIG. 15, the noise estimation unit 14 performs a pair of e (0, l) and e (1, l), e (1, l) and e (2, l) for an arbitrary carrier number l. Noise evaluation value dif for the pair of e (2, l) and the pair of e (3, l),. However, in this case, since the type of the reference point is different between e (0, l) and e (1, l), the difference in the type of the reference point (difference between the first reference point and the second reference point) is canceled out. Therefore, it is necessary to include a correction for deriving the noise evaluation value dif.

Specifically, when obtaining the noise evaluation value dif of the subcarrier signals e (0, l) and e (1, l),
Substituting the in-phase and quadrature components of e (0, l) into pi0 and pq0, respectively,
Substituting the in-phase component and the quadrature component of the signal obtained by rotating e (1, l) by 45 ° on the IQ plane into si0 and sq0, respectively.
Substituting the in-phase component and the quadrature component of the signal obtained by rotating e (1, l) by 135 ° on the IQ plane into si1 and sq1, respectively.
Substituting the in-phase component and the quadrature component of the signal obtained by rotating e (1, l) by 225 ° on the IQ plane into si2 and sq2, respectively.
Substituting the in-phase component and the quadrature component of the signal obtained by rotating e (1, l) by 315 ° on the IQ plane into si3 and sq3, respectively, based on the above formulas (B0) to (B3), or The noise evaluation value dif may be obtained through the above calculation of the phase angle difference.

Any one of the four first reference points forming the first reference point group is regarded as a specific reference position. If the specific reference position is (1 / √2, 1 / √2),
The process of rotating e (1, l) by 45 ° is equivalent to the rotation process for moving the position of the second signal point (0, 1) to the specific reference position.
The process of rotating e (1, l) by 135 ° is equivalent to the rotation process for moving the position of the second signal point (−1, 0) to the specific reference position.
The process of rotating e (1, l) by 225 ° is equivalent to the rotation process for moving the position of the second signal point (0, −1) to the specific reference position.
The process of rotating e (1, l) by 315 ° is equivalent to the rotation process for moving the position of the second signal point (1, 0) to the specific reference position.

  Although the calculation method of the noise evaluation value dif for e (0, l) and e (1, l) has been described, it is similar to that for e (1, l) and e (2, l), and e (2, l) and e. The same applies to (3, l).

  According to this embodiment, even if a known signal is not multiplexed as a pilot signal, the magnitude of the noise component can be estimated well. Since known signals exist sparsely, the amount of information is small, and the noise estimation method using only known signals may reduce the estimation accuracy. However, the method of this embodiment can use all subcarriers. The estimation accuracy is increased. Also, when a reference signal having a known fixed pattern is included in the header part at the beginning of the transmission frame, the noise component size is estimated by comparing the signal in the header part with the original signal (known fixed pattern). However, when performing such estimation, in principle, the estimation time interval cannot be shorter than the time interval between header portions that are temporally adjacent. On the other hand, according to the noise estimation unit 14 of the present embodiment, it is possible to estimate the magnitude of the noise component at a time interval shorter than the time interval between adjacent header portions. Then, the reception characteristics are improved by appropriately switching the demodulation method according to the estimated value of the magnitude of the noise component.

  In addition, the amplitude of adjacent subcarrier signals in the same symbol does not vary so much. Therefore, as shown in FIG. 9, when noise estimation is performed through comparison of adjacent subcarrier signals within the same symbol, accurate noise estimation is possible. Further, when noise estimation is performed through comparison of adjacent subcarrier signals within the same symbol, the memory capacity for storing data for calculating the noise evaluation value dif, particularly compared to the method shown in FIG. Is less.

<< Deformation, etc. >>
Specific numerical values shown in the above description are merely examples, and as a matter of course, they can be changed to various numerical values. As modifications or annotations of the above-described embodiment, notes 1 to 4 are described below. The contents described in each comment can be arbitrarily combined as long as there is no contradiction.

[Note 1]
The embodiment in which the modulation scheme used in the transmission apparatus is DQPSK has been described above. However, the modulation scheme is not limited to DQPSK, and any modulation scheme can be used in the transmission apparatus. For example, QPSK or 16QAM (16 Quadrature Amplitude Modulation) can be used instead of DQPSK.

[Note 2]
It is also possible to add an equalization processing unit (not shown) for reducing transmission path distortion to the receiving apparatus 10 of FIG. This equalization processing unit reduces a transmission path distortion component included in the output signal of the FFT processing unit 13. The output signal of the FFT processing unit 13 after the reduction may be provided to the demodulation processing unit 15.

[Note 3]
In the above-described embodiment, it is assumed that the pilot signal is not inserted into the OFDM signal. However, even in the case where the pilot signal is inserted into the OFDM signal as in ISDB-T, the above-described embodiment describes. These methods are applicable.

[Note 4]
The receiving apparatus 10 in FIG. 3 can be realized by hardware or a combination of hardware and software. When the receiving apparatus 10 is configured using software, a block diagram of a part realized by software represents a functional block diagram of the part. A function realized using software may be described as a program, and the function may be realized by executing the program on a program execution device (for example, a computer).

DESCRIPTION OF SYMBOLS 10 Receiver 11 Receiving antenna 12 Tuner part 13 FFT processing part 14 Noise estimation part 15 Demodulation processing part

Claims (6)

In a receiving apparatus that receives a signal modulated using a method of multiplexing subcarrier groups orthogonal to each other,
A noise estimation unit that estimates a magnitude of a noise component of the reception signal by comparing a plurality of subcarrier signals by a plurality of subcarriers included in the reception signal of the reception device,
The receiving apparatus, wherein the plurality of subcarriers are a plurality of subcarriers different from each other in a frequency direction or a time direction. A demodulating processor for demodulating the received signal;
The receiving apparatus according to claim 1, wherein the demodulation processing unit switches a demodulation method for demodulating the reception signal in accordance with an estimation result of the noise estimation unit. The demodulation processing unit includes:
When the estimated noise component size is relatively small, the received signal is demodulated by a demodulation method using delayed detection,
The receiving apparatus according to claim 2, wherein when the estimated noise component is relatively large, the received signal is demodulated by a demodulation method using synchronous detection. The plurality of subcarrier signals include first and second subcarrier signals;
The noise estimation unit estimates the magnitude of the noise component through a comparison between a signal point of the first subcarrier signal and a signal point of the second subcarrier signal on an IQ plane. The receiving device according to any one of claims 1 to 3. The ideal position of the signal point of the second subcarrier signal on the IQ plane is any one of the first to Nth positions (N is an integer of 2 or more),
The noise estimator is
Obtaining an amount based on a positional difference between the signal point of the first subcarrier signal and the signal point of the second subcarrier signal;
Position difference between the signal point of the first subcarrier signal and the signal point obtained by applying the process of moving the first position to the nth position on the signal point of the second subcarrier signal For each of n = 2, 3,..., N,
The receiving apparatus according to claim 4, wherein the magnitude of the noise component is estimated using a minimum amount among the obtained amounts. In a receiving apparatus that receives a signal modulated using a method of multiplexing subcarrier groups orthogonal to each other,
A demodulation processing unit for demodulating the received signal of the receiving device;
The receiving apparatus, wherein the demodulation processing unit switches a demodulation method for demodulating the received signal based on a comparison result of a plurality of subcarrier signals by a plurality of subcarriers included in the received signal.

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