JP2011066570A - Semiconductor integrated circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a semiconductor integrated circuit switching a frequency band used by an amplifier while maintaining a good high frequency characteristic. <P>SOLUTION: The semiconductor integrated circuit is constituted of: a coil L11 and a coil L12 which have a coupling factor k1 and are connected parallel to each other; a coil L13 connected in series to the coil L11 and coil L12; a capacitor C11 connected parallel to the coil L11; a capacitor C12 connected parallel to the coil L12; an input terminal p1 connected to one end of the coil L11 and to one end of the capacitor C11; an input terminal n1 connected to one end of the coil L12 and one end of the capacitor C12; and an input signal supply part 200 which supplies input signals of opposite phases to the input terminal p1 and input terminal n1 respectively. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a semiconductor integrated circuit that switches a frequency band of an amplifier.

  In recent years, in a wireless communication system such as a portable terminal, in order to cope with all communication systems such as GSM, GPS, WCDMA, etc., the transmission / reception system has been widely widened and multibanded.

  If the frequency band used is several hundred MHz, and the input and output of the wireless communication system is not so wide, distortion components will be generated in the desired band by intermodulation due to in-band and out-of-band interference waves. There is.

  Therefore, it is desirable to narrow down the band in a narrow band system, that is, to use by switching the frequency band from the viewpoint of suppressing deterioration of distortion characteristics, noise characteristics, and the like.

  The wireless communication system is composed of a resonant circuit. The resonance circuit is composed of an inductor and a capacitor, and the resonance frequency is represented by 1 / {2π√ (LC)}. Here, L is the inductor value (inductive coefficient) of the inductor, and C is the capacitance value of the capacitor. The resonant frequency can be changed by changing the inductor value or the capacitance value. That is, the frequency band can be switched.

  As a technique for changing the resonance frequency, a technique of frequency tuning using a passive element such as a resistor is generally used.

  Particularly at high frequencies, an inductor or a capacitor is used as an amplifier load, a method 1 for switching the capacitance value according to a band, a method 2 for switching an inductor value, or a method for changing an inductor value by using an inductor coupling coefficient. 3 or the like is used.

  However, since the method 1 switches the capacitance value, a large capacitance value is required when it is desired to take a wide variable frequency range, and it is difficult to obtain a high Q value at a low frequency. Here, the Q value is expressed as Q = 1 / R√ (L / C), where R is the parasitic resistance of the inductor, L is the inductance of the coil, and C is the capacitance of the capacitor. The higher the value, the sharper the frequency characteristics, and the higher the frequency selectivity.

  Moreover, since the method 2 switches the inductor value itself using a switch, the number of inductors corresponding to the number of applied systems is required, and the area increases. Further, when it is desired to take a wider variable frequency range, there is a problem that a large inductor is required, the parasitic resistance is increased, and the Q value is deteriorated.

  Further, in Method 3, since the inductor value changes from (1−k) L to (1 + k) L according to the coupling coefficient k by switching between the common mode and the differential mode, the frequency range is changed according to the value. However, the resonance frequency is higher in the common mode in which the Q value is as low as (1 / R) √ {(1-k) L / C}. In particular, it is difficult to obtain good characteristics at high frequencies compared to low frequencies. In addition, analog circuit blocks such as low-noise amplifiers (LNA), power amplifiers (PA), and voltage-controlled oscillators (VCOs) load and degenerate. Since the Q value of the inductor used instead of the resistor has a large effect on the gain characteristics, maximum output power, oscillation amplitude, etc., we would like to use an element with a Q value as high as possible. There is a problem that the desired gain and noise characteristics cannot be secured due to deterioration.

Japanese Patent No. 3959371

  SUMMARY OF THE INVENTION An object of the present invention is to provide a semiconductor integrated circuit capable of switching the used frequency band of an amplifier while maintaining good high frequency characteristics.

  To achieve the above object, a semiconductor integrated circuit of one embodiment of the present invention includes a first coil and a second coil having a first coupling coefficient and connected in parallel to each other, and the first coil. And a third coil connected in series to the second coil, a first capacitor connected in parallel to one end of the first coil and one end of the third coil, and the second coil A second capacitor connected in parallel to one end of the coil and one end of the third coil, a first input terminal connected to one end of the first coil and one end of the first capacitor And a second input terminal connected to one end of the second coil, and one end of the second capacitor, the first input terminal, and the second input terminal are respectively in opposite phases. An input signal supply unit for supplying an input signal is provided. .

  According to the present invention, it is possible to provide a semiconductor integrated circuit capable of switching the used frequency band of an amplifier while maintaining good high frequency characteristics.

1 is a system block diagram of a semiconductor integrated circuit of the present invention. It is a circuit diagram of Example 1 of the present invention. It is a circuit diagram of Example 2 of the present invention. It is a circuit diagram of Example 3 of the present invention. It is a circuit diagram of Example 4 of the present invention. It is a circuit diagram of Example 5 of the present invention. It is a circuit diagram of Example 6 of the present invention. It is a circuit diagram of Example 7 of the present invention. It is a circuit diagram of Example 8 of the present invention.

  Embodiments of the present invention will be described below with reference to the drawings.

  FIG. 1 is a system block diagram of a semiconductor integrated circuit according to the present invention. The semiconductor integrated circuit according to the present invention includes an antenna 10 for transmitting and receiving data, a T / R switch unit 11 for switching transmission and reception of data, an LNA 12 (low noise amplifier), a PA 13 (power amplifier :), It is composed of MIX14 (Mixer), VCO15 (Voltage Controlled Amplifier), and LPF16 (Low Path Filter), among which LNA12 and PA13 are particularly characterized.

FIG. 2 shows the resonance circuit unit 100 of the semiconductor integrated circuit according to the first embodiment of the present invention, which is included in the LNA 12 (or PA 13) in FIG. Resonant circuit 100 of FIG. 2 (a) has a coupling coefficient k1, is connected to the coil L11 and coil L12 of the inductance _{L 1} of the inductance _{L 1} connected in parallel with one another, in series with the coil L11 and the coil L12 a coil L13 of the inductance _{L 2,} a capacitor C11 of the capacitance _{C 1} connected in parallel to the coil L11, composed of the capacitance _{C 1} of capacitor C12 Prefecture connected in parallel to the coil L12.

The input terminal p1 is connected to one end of the coil L11 and one end of the capacitor C11, the input terminal n1 is connected to one end of the coil L12 and one end of the capacitor C12, and the input terminal p1 and the input terminal n1 have an input signal supply unit. A signal is supplied from 200.

  Here, it is assumed that the coil L11 and the coil L12 are wound in a direction in which magnetic fluxes strengthen each other when signals having opposite phases flow.

When the input signal supply unit 200 supplies signals of opposite phases to the input terminal p1 and the input terminal n1, signals of opposite phases flow through the inductors composed of the coil L11 and the coil L12 to strengthen the magnetic flux. Since the magnetic force acts in the direction, the inductance of the inductor composed of the coil L11, the coil L12, and the coil L13 is L _L = L ₁ (1 + k ₁ ) where the coupling coefficient between the coil L11 and the coil L12 is k ₁ . Further, since the signal flowing through L12 is a reverse phase signal, the influence of L13 is not visible. Therefore, the resonance frequency is f _L = 1 / R {(2π√ (L _L × C ₁ )), and the Q value is Q _L = (1 / R) √ (L _L / C ₁ ), which degrades the Q value. Without causing the resonance circuit unit 100 to resonate at a low frequency. R is a parasitic resistance of each inductor (the same applies hereinafter).

2B is the same as FIG. 2A except for the phase of the signal supplied to the input terminal n1. In FIG. 2B, signals having the same phase are supplied from the input signal supply unit 200 to the input terminal n1 and the input terminal p1. When signals having the same phase are supplied to the input terminals p1 and n1, the in-phase signals flow through the inductors configured by the coil L11 and the coil L12, respectively, and the magnetic force acts in the direction to cancel the magnetic flux. The inductance of the inductor composed of the coil L12 is L ₁ (1−k ₁ ) when the coupling coefficient k ₁ between the coil 11 and the coil 12 is used. Further, since the signal flowing through L21 is an in-phase signal, the inductance of the inductor constituted by the coil L11, the coil L12, and the coil L13 is L _H = 2L ₂ + L ₁ (1−k ₁ ), and the resonance frequency is f _H = 1 / {2π√ (L _H C ₁ )}, Q value is Q _H = (1 / R) √ (L _H / C ₁ ), and the frequency is changed to a desired frequency while ensuring sufficient isolation. Furthermore, since the Q value in the high-frequency operation mode can be improved, good gain characteristics and noise characteristics can be obtained. Specifically, the Q value of the coil L11 and coil L12, the coil L21 and the coil L22 deteriorated by the same phase mode can be compensated by the coil L13 / L23, and a high Q value can be maintained even in the high frequency mode. Gain characteristics and noise characteristics can be obtained.

FIG. 3 shows a resonant circuit section 100 of a semiconductor integrated circuit according to the second embodiment of the present invention. Resonant circuit 100 of FIG. 3 (a) has a coupling coefficient k1, and the coil L11 and coil L12 of the inductance _{L 1} of the inductance _{L 1} connected in parallel with each other, similarly, has a coupling coefficient k1, parallel with each other a coil L22 of the coil L21 and the inductance _{L 1} of the inductance _{L 1} connected to the inductance of the coil L13 of the inductance _{L 2} connected in series to the coil L11 and the coil L12, is connected in series with the coil L21 and the coil L22 a coil L23 of L _2, a capacitor C11 and the capacitance _{C 1} connected in parallel to the coil L11, the capacitance _{C 1} of capacitor C12 connected in parallel to the coil L12, is connected in parallel to the coil L21 that the capacitor C21 and the capacitance _{C 1,} the electrostatic connected in parallel to the coil L22 And a capacitor C22 Metropolitan capacity _{C 1.}

Here, the coupling coefficient between the coil L13 and the coil L23 is set to _{k 2.}

Further, it is assumed that the coil L11 and the coil L12, the coil L21 and the coil L22, the coil L13 and the coil L23 are wound in directions in which magnetic fluxes are strengthened when signals having opposite phases flow.

The input terminal p1 is connected to one end of the coil L11 and one end of the capacitor C11, the input terminal n1 is connected to one end of the coil L12 and one end of the capacitor C12, and the input terminal p2 is connected to one end of the coil L21 and one end of the capacitor C21. The input terminal n2 is connected to one end of the coil L22 and one end of the capacitor C22, and a signal is supplied from the input signal supply unit 200 to the input terminals p1, n1, p2, and n2.

When signals with opposite phases are supplied to the input terminal p1 and the input terminal n1, opposite phase signals flow through the inductors configured by the coil L11 and the coil L12, respectively, and magnetic force acts in the direction of strengthening the magnetic flux. The inductance of the inductor composed of the coil L11, the coil L12, and the coil L13 is L _L = L ₁ (1 + k ₁ ) where the coupling coefficient between the coil L11 and the coil L12 is k ₁ . Similarly, the coil L21 and the coil L22, the coupling coefficient of the coil L21 and the coil L22 and _{k 1,} a _{L L = L 1 (1 +} k 1).

In addition, since the signals flowing through L12 and L22 are antiphase signals, the influence of L13 and L23 is invisible. Therefore, the resonance frequency is f _L = 1 / R {(2π√ (L _L × C ₁ )), and the Q value is Q _L = (1 / R) √ (L _L / C ₁ ), which degrades the Q value. Without causing the resonance circuit unit 100 to resonate at a low frequency.

  The resonance circuit unit 100 in FIG. 3B is the same as that in FIG. 3A except for the phase of the signal supplied to the input terminals n1 and p2. In FIG. 3B, signals having the same phase are supplied to the input terminal p1 and the input terminal n1, and to the input terminal p2 and the input terminal n2. Further, the phase of the signal input to the input terminal p1 and the input terminal n1 and the phase of the signal input to the input terminal p2 and the input terminal n2 are opposite to each other.

When signals having the same phase are supplied to the input terminals p1 and n1, and p2 and n2, signals having the same phase flow through the coils L11 and L12, and the coils L21 and L22, respectively, in a direction that cancels the magnetic flux. Since the magnetic force works, the inductance of the inductor composed of the coil L11 and the coil L12, the coil L21 and the coil L22 is L ₁ (when the coupling coefficient of the coil L11 and the coil L12 and the coil L21 and the coil L22 is k ₁ , respectively. 1-k ₁ ).

In addition, since the signals flowing in the coils L13 and L23 are opposite in phase, a magnetic force acts on the coils L13 and L23 in the direction of strengthening the magnetic flux, so that the inductance of the inductor constituted by the coils L13 and L23 Is L ₂ (1 + k ₂ ), where k ₂ is the coupling coefficient between the coil L13 and the coil L23.

From the above, the inductance of the inductor constituted by the coil L11, the coil L12, and the coil L13 is L _H = 2L ₂ (1 + k ₂ ) + L ₁ (1−k ₁ ), and the resonance frequency is f _H = 1 / {2π√ (L _H C)}, the Q value is Q _H = (1 / R) √ (L _H / C), and the resonance circuit unit 100 can resonate at a high frequency without degrading the Q value. Become.

Here, if there is no coil L13 and coil L23 having a coupling coefficient _{k 2} is the inductance _{L H '= L 1 (1} -k 1) , and the better the Q value in the present embodiment, √ _{(L H} It can be seen that / L _H ′) times higher. That is, the resonance circuit unit 100 can be resonated at a high frequency without deteriorating the Q value.

  FIG. 4 shows a semiconductor integrated circuit according to Embodiment 3 of the present invention. The present embodiment is configured from the resonance circuit unit 100 of the second embodiment and switches M21 to M26 that selectively switch the signals flowing through the coil L11, the coil L12, and the coil L21 and the coil L22 to the same phase / antiphase. A differential amplifier circuit having a cascode configuration configured with the input signal supply unit 200.

  When it is desired to resonate the resonance circuit unit 100 at a low frequency, M21 and M23, M24 and M26 are turned ON (H), and M22 and M25 are turned OFF (L) to supply a signal having a reverse phase to the resonator.

As with the second embodiment, the inductance L _L , resonance frequency f _{L, and} Q value are as follows: L _L = L ₁ (1 + k ₁ ), f _L = 1 / {2π√ (L _L C)}, Q _L = (1 / R) √ (L _L / C).

  When the resonance circuit unit 100 is desired to resonate at a high frequency, M21 and M22, M24 and M25 are turned on (H), and M23 and M26 are turned off (L) to supply in-phase signals to the resonator.

As with the second embodiment, the inductance L _H , resonance frequency f _{H, and} Q value are L _H = 2L ₂ (1 + k ₂ ) + L ₁ (1−k ₁ ), f _H = 1 / {2π√ (L _H C)}, Q _H = (1 / R) √ (L _H / C).

  As described above, in addition to the effects of the second embodiment, it is possible to selectively switch the signal flowing through each coil to the same phase / antiphase.

  FIG. 5 shows a circuit according to a fourth embodiment of the present invention. In the circuit of this embodiment, the input signal supply unit 200 according to the third embodiment is further provided with a variable capacitor C31 in parallel with the coil L31 and a variable capacitor C32 in parallel with the coil L32. An LC resonance circuit is configured.

  By changing the capacitances of the variable capacitor C31 and the variable capacitor C32, in addition to the effects of the first to third embodiments, it is possible to appropriately change to the high frequency operation mode or the low frequency operation mode. Note that the impedance of the LC resonant circuit is configured to be sufficiently high at a desired operating frequency.

  FIG. 6 shows a circuit according to a fifth embodiment of the present invention. The circuit of the present embodiment includes two resonance circuits 100 of the second embodiment, and switches MN21 to MN26 and MP21 to selectively switch the signals flowing through the coils L11 and L12 and the coils L21 and L22 to the same phase / antiphase. This is a complementary cascode-connected gate-grounded differential amplifier circuit including an input signal supply unit 200 composed of MP26. The resonant circuit unit 100 performs the same control as that of the third embodiment, and the control signals of the switches MP21 to MP26 are controlled by the inverted signals with respect to the control signals of the switches MN21 to MN26. Since the output signals VOUT_P1 and VOUT_P2 are in-phase signals, the currents are added and output.

  FIG. 7 shows a circuit according to a sixth embodiment of the present invention. The circuit of the present embodiment includes a resonance circuit unit 100 similar to that of the second embodiment, and includes switches M21 to M26 that selectively switch the signals flowing through the coil L11, the coil L12, the coil L21, and the coil L22 to the same phase / antiphase. The input signal supply unit 200 is configured, and M11 and M12 are amplifiers that operate as a common source amplifier, and M31 and M32 operate as a common gate amplifier. Since the grounded source circuit and the grounded gate circuit are inverting amplification and non-inverting amplification, respectively, by arranging the transconductances Gm of M11 and M12, M31 and M32, signals having the same amplitude and opposite phase flow through the resonance circuit unit 100. Further, I1 and I2 are added as DC current sources of M31 and M32.

  When it is desired to resonate the resonator circuit unit 100 at a low frequency, M21 / M22 and M24 / M25 are turned on (H), and M23 / M26 is turned off (L), so that signals having opposite phases are supplied to the resonator. A negative phase signal flows through each of the inductors constituted by the coil L11, the coil L12, the coil L21, and the coil L22, and resonates at a low frequency because it works in the direction of strengthening the magnetic flux.

When the resonance circuit unit 100 is desired to resonate at a high frequency, M21 / M23 and M24 / M26 are turned on (H), and M22 / M25 is turned off (L), so that an in-phase signal is supplied to the resonator. An in-phase signal flows through each of the inductors constituted by L11 / L12 and L13 / L14, and resonates at a high frequency because it works in a direction to cancel the magnetic flux.

  The circuit in FIG. 8 shows a circuit according to the seventh embodiment of the present invention. The resonance circuit unit 100 is the same as that of the second embodiment, and includes switches M21 to M26 that selectively switch the signals flowing through the coil L11 and the coil L12, the coil L21 and the coil L22 to the same phase / antiphase, and M11 / M12 is a drain. The amplifier circuit operates as a grounded amplifier circuit. The above LC resonator performs the same control as in the first embodiment, and is mainly used as a buffer circuit.

The circuit of FIG. 9 shows the circuit of the eighth embodiment of the present invention. A resonance circuit unit 100 similar to that of the second embodiment is provided, and an input signal supply unit 200 is configured to selectively switch signals flowing through the coils L11 / L12 and L13 / L14 to the same phase / opposite phase, and the current source I2 / I3 and the buffer element M21 to M24 are provided, and I1 operates as a current source that supplies current to M11 / M12.

Since the voltage signals input to the gates of M21 to 24 appear as inverted signals at the drains of the respective transistors, they are input to the gates of M22 and M23, and M21 and M24 by selectively turning on and off the current sources I2 and I3. The switched signal can switch the voltage signal at the drains of M11 and M12 to in-phase / anti-phase. As a result, the frequency of the output signal can be changed as in the second embodiment.
(Other examples)

  The capacitors C11 and C12, and the capacitors C21 and C22 of the first to seventh embodiments may be variable capacitors. Thereby, the frequency can be changed, and the Q values of the coil L11 and the coil L12, the coil L21 and the coil L22 deteriorated by the same phase mode are changed to the coil L13 and the coil L23, the variable capacitor C11 and the variable capacitor C12, and variable. Compensating with the capacitance capacitor C21 and the variable capacitance capacitor C22 makes it possible to finely adjust the frequency band while ensuring a wide variable frequency range, and also to maintain a high Q value even in the high frequency mode, and has good gain characteristics and noise characteristics. Can be obtained.

  In each of the above-described embodiments, when the inductors composed of L11 / L12 and L13 / L14 are used in the same phase, the signal level can be made larger than the one-side output by adding the output signals. .

In addition, the amplifier including the resonance circuit unit 100 can be formed of a bipolar circuit in addition to the FET, and can also be realized in a PMOS having a reversed polarity.

10 Antenna
11 T / R switch
12 LNA
13 PA
14 MIX
15 VCO
16 LPF
100 Resonant circuit unit 200 Input signal supply unit L11, L12, L13, L21, L22, L23, L31, L32 Coil
C11, C12, C13, C21, C22, C23 capacitors
C31, C32 variable capacitor
M11, M12, M21 to M26, MN21 to MN26, MP21 to MP26 switch

Claims (5)

A first coil and a second coil having a first coupling coefficient and connected in parallel to each other;
A third coil connected in series to the first coil and the second coil;
A first capacitor connected in parallel to one end of the first coil and one end of the third coil;
A second capacitor connected in parallel to one end of the second coil and one end of the third coil;
A first input terminal connected to one end of the first coil and one end of the first capacitor;
A second input terminal connected to one end of the second coil and one end of the second capacitor;
A semiconductor integrated circuit comprising: an input signal supply unit that supplies input signals having opposite phases to the first input terminal and the second input terminal, respectively.   The semiconductor integrated circuit, wherein the input signal supply unit supplies input signals having the same phase to the first input terminal and the second input terminal, respectively. A fourth coil and a fifth coil having the first coupling coefficient and connected in parallel to each other;
A sixth coil connected in series to the fourth coil and the fifth coil and having a second coupling coefficient with the third coil;
A third capacitor connected in parallel to one end of the fourth coil and one end of the sixth coil;
A fourth capacitor connected in parallel to one end of the fifth coil and one end of the sixth coil;
A third input terminal connected to one end of the fourth coil and one end of the third capacitor;
A fourth input terminal connected to one end of the fifth coil and one end of the fourth capacitor;
Inputs that supply input signals of opposite phases to the first input terminal and the second input terminal, respectively, and supply input signals of opposite phases to the third input terminal and the fourth input terminal, respectively. The semiconductor integrated circuit according to claim 1, further comprising: a signal supply unit. The input signal supply unit supplies input signals having the same phase to the first input terminal and the second input terminal,
The input signal supply unit supplies input signals having the same phase to the third input terminal and the fourth input terminal,
The input signals of the first input terminal and the second input terminal, and the input signals of the third input terminal and the fourth input terminal are in opposite phases, respectively. Semiconductor integrated circuit.   The said input signal supply part is comprised from the switch which selectively switches the input signal input into the said 1st thru | or 4th input terminal to the same phase or an opposite phase. Semiconductor integrated circuit.

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