JP2009301747A - Large high frequency power device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a large high frequency power device capable of preventing destruction of a last stage power amplifier due to a reflected wave power without deteriorating a high frequency power outputted from an antenna. <P>SOLUTION: The large high frequency power device includes an amplifying part for amplifying a high frequency power to output the same, a variable impedance matching part provided on an output side of the amplifying part and changing an input impedance based on a controlling signal to output a high frequency power, an antenna for outputting the high frequency power, a quadrature detection part outputting a common mode and a quadrature detection output signal by conducting a quadrature detecting of the reflected wave power from the antenna by the high frequency power input to the amplifying part, and a controlling part generating a controlling signal in response to the common mode and the quadrature detection output signal and sending out it to the variable impedance matching part. The controlling part, based on phase information of the reflected wave power obtained from the common mode and the quadrature detection output signal, generates a controlling signal for changing a phase shift volume in a transmission passage of the variable impedance matching part so that the phase of the reflected wave power may stay in a predetermined range. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a high-power, high-frequency output device that operates from the RF band to the microwave band. More specifically, the present invention relates to a technique for efficiently protecting a high frequency power amplifier in a high power high frequency output device.

  In recent years, global environmental protection has become a global problem, and power semiconductors using materials with high-speed operation, high breakdown voltage, and high power density, such as silicon carbide (SiC) and gallium nitride (GaN), It is attracting attention as an energy-saving device for equipment that handles high power.

  Cooking equipment such as microwave ovens that heat foods by irradiating high-frequency power of several hundreds of watts, including deployment to power amplifiers for transmission in base stations of communication systems that use high frequencies from the RF band to the microwave band The use of power semiconductors is also being developed.

  In these devices, in order to achieve both high power and high efficiency, the power amplifier closest to the antenna, that is, the final stage power amplifier is generally used in a non-linear operation state close to saturation. Therefore, in order for the power amplifier to operate stably, matching between the input impedance of the circuit connected to the next stage of the power amplifier and the output impedance of the power amplifier is very important.

  If the impedance matching is deviated from the optimum state, the output power is reduced and the efficiency is lowered. In the worst case, the power amplifier may be destroyed.

  Also, even if all impedance matching from the power amplifier to the antenna is optimized, the antenna impedance matching is disrupted due to changes in the environment of use of the equipment and the surrounding environment of the antenna, causing reflection at the antenna, and the reflected wave When the power flows backward and is input to the power amplifier, the output power and efficiency of the power amplifier may be reduced, and in the worst case, destruction may occur. In particular, in the case of cooking equipment such as a microwave oven, the antenna impedance state becomes very complicated due to the very small space around the antenna and the shape and physical properties of the food to be cooked, and the probability that large reflected wave power will be generated. Will be very expensive.

  Conventionally, in order to ensure stable operation of the power amplifier even when the impedance matching of the antenna is broken, an isolator is arranged on the output side of the power amplifier to prevent the reflected wave power from the antenna from flowing back to the power amplifier. Is commonly used. The configuration and operation of this conventional example will be described with reference to FIG.

  FIG. 17 is a block diagram showing a configuration of a conventional high-frequency power output device 300. In the high-frequency power output device 300, a high-frequency power oscillation unit 301 that oscillates high-frequency power, a high-frequency power amplifier 302, an impedance matching unit 303, and an isolator 304 are connected in series, and an antenna 305 is connected to the output of the isolator 304. The output power of the high frequency power oscillation unit 301 is amplified by the high frequency power amplifier 302 and is emitted from the antenna 305 via the impedance matching unit 303 and the isolator 304. If the impedance matching of the antenna is disrupted due to changes in the environment of use of the device or the environment around the antenna, reflection occurs at the antenna, and a backflow of the reflected wave power occurs. However, the impedance matching arranged at the output of the high-frequency power amplifier 302 Since the isolator 304 is arranged in series between the unit 303 and the antenna 305, most of the reflected wave power flowing back from the antenna is absorbed by the isolator 304, so that the high-frequency power amplifier 302 receives the reflected power. The input reflected wave power becomes very small, and the high frequency power amplifier 302 can avoid destruction without lowering the output power.

  In addition, as a conventional technique for preventing destruction of the power amplifier due to the reflected wave power from the antenna without using an isolator, the reflected wave power is detected, and the power corresponding to the magnitude of the reflected wave power is detected. There is a method of preventing the destruction of the power amplifier by increasing the attenuation of the variable attenuator arranged between the amplifier and the load (antenna) and reducing the amount of reflected wave power injected into the power amplifier (for example, Patent Document 1).

In addition, as a conventional technique for preventing destruction of the power amplifier due to reflected wave power from the antenna without using an isolator, the current consumed by the power amplifier (transistor) is monitored, There is a method of preventing the overcurrent from flowing through the power amplifier and preventing the breakdown by controlling the bias of the power amplifier (transistor) so as not to exceed a predetermined value (for example, refer to Patent Document 2). .
JP 2006-166153 A JP 2002-076791 A

  However, the shape of the isolator increases as the power handled increases, resulting in a significant increase in cost. In the method of Patent Document 1, an attenuator is disposed between the power amplifier output and the antenna, and the amount of attenuation is increased to suppress the reflected wave power to the power amplifier. Electric power will also decrease.

  Further, in the method of Patent Document 2, when the reflected wave power flows backward and is input to the power amplifier, the increase in the current of the power amplifier is detected and the current is decreased. Similar to the method, the high-frequency power output from the antenna is reduced.

  An object of the present invention is to provide a high-power high-frequency output device that can prevent destruction of a final-stage power amplifier due to reflected wave power without reducing high-frequency power output from an antenna.

  A high-power high-frequency output device according to the present invention is provided on the output side of a high-frequency power amplification unit that receives, amplifies and outputs high-frequency power, and changes input impedance based on an input control signal. A variable impedance matching unit that outputs the high-frequency power, an antenna that outputs the high-frequency power output from the variable impedance matching unit, and reflected wave power from the antenna is input to the high-frequency power amplification unit A quadrature detection unit that performs quadrature detection with the high-frequency power and outputs an in-phase detection output signal and a quadrature detection output signal; and generates the control signal according to the in-phase detection output signal and the quadrature detection output signal; and the variable impedance A control unit for sending to the matching unit, and the control unit includes the in-phase detection output signal and the quadrature detection output. Based on the phase information of the reflected wave power obtained from the signal, the phase shift amount of the transmission path of the variable impedance matching unit is set so that the impedance of the reflected wave power falls within a range where the high frequency power amplifier is not destroyed. A control signal to be changed is generated.

  The impedance of the reflected wave power at the input of the variable impedance matching unit viewed from the output of the high-frequency power amplification unit can be classified into one of a value of a first impedance region or a value of a second impedance region, and The impedance region includes an impedance that destroys the high-frequency power amplification unit when a current flowing through the high-frequency power amplification unit enters an overcurrent state, and the second impedance region includes a current of a current flowing through the high-frequency power amplification unit. The control unit generates the control signal such that the impedance of the reflected wave power falls within the second impedance region, the impedance including the impedance at which the high frequency power amplifying unit is not destroyed because the value does not exceed a breakdown resistance limit. And changing the phase shift amount of the transmission line of the variable impedance matching unit Good.

  The control unit includes a converter that converts the analog in-phase detection output signal and the quadrature detection output signal into a digital format, respectively, and a storage unit that holds a control table for specifying the second impedance region. The control signal may be generated based on the in-phase detection output signal in the digital format, the quadrature detection output signal, and the control table.

  The storage unit may be a semiconductor memory.

  The control unit may convert the control signal into a digital signal of 1 bit or more and send it.

  The control unit may convert the control signal into an analog signal and send it out.

  The control unit may convert the control signal into a pulse width modulation (PWM) signal and send it out.

  The high-power high-frequency output device may further include a high-frequency power oscillation unit that oscillates the high-frequency signal and inputs the high-frequency signal to the high-frequency power amplification unit and the quadrature detection unit.

  The high-power high-frequency output device is provided between the output of the variable impedance matching unit and the antenna, sends the output power of the variable impedance matching unit to the antenna, and the reflected wave power from the antenna There may be further provided a directional coupler for taking out.

  According to the high frequency power output device of the present invention, the reflected wave power from the antenna is orthogonally detected, and the variable frequency impedance is prevented from being destroyed by the reflected wave impedance at the input of the variable impedance matching unit viewed from the high frequency power amplifier output. Adjust at the matching section. As a result, the high frequency power amplifier is destroyed by the reflected wave power from the antenna due to the impedance mismatch of the antenna without using expensive parts such as an isolator and reducing the high frequency output power of the high frequency power amplifier. Can be prevented.

  Hereinafter, the concept of a high-power, high-frequency output device according to the present invention will be described with reference to the accompanying drawings. Thereafter, an embodiment of a high-power high-frequency output device will be described.

  FIG. 1 is a block diagram showing a basic configuration of a high-power, high-frequency output device 100 of the present invention. A high frequency power oscillation unit 101 that oscillates high frequency power, a high frequency power amplifier 102, a variable impedance matching unit 103, and a directional coupler 104 are connected in series, and an antenna 107 is connected to the output of the directional coupler 104. Further, a quadrature detection unit 105 and a control unit 106 are provided, and the reverse power output of the directional coupler 104 and the output of the high frequency power oscillation unit 101 are connected to the input of the quadrature detection unit 105, respectively, and the quadrature detection unit. The detection output 105 is connected to the input of the control unit 106. The output of the control unit is connected to the impedance control signal input of the variable impedance matching unit 103.

  The output power of the high frequency power oscillation unit 101 is amplified by the high frequency power amplifier 102 and emitted from the antenna 107 through the variable impedance matching unit 103 and the directional coupler 104. When the impedance matching of the antenna is lost due to the use state of the device or the environment surrounding the antenna, reflection occurs at the antenna, and the reflected wave power flows backward from the antenna 107 via the directional coupler 104 and the variable impedance matching unit 103. And input to the high-frequency power amplifier 102.

  On the other hand, the reflected wave power from the antenna 107 is output from the reverse power output of the directional coupler 104 and input to the quadrature detection unit 105. The quadrature detection unit 105 performs quadrature detection on the reflected wave power input from the directional coupler 104 using the output signal of the high-frequency power oscillation unit 101. At this time, since the frequency of the output signal of the high-frequency power oscillation unit 101 and the frequency of the reflected wave power are the same, the quadrature detection in the quadrature detection unit 105 is completely synchronous detection. An in-phase detection output signal and a quadrature detection output signal are output from the quadrature detection unit 105 and input to the control unit 106.

  The control unit 106 detects the phase information of the reflected wave power from the in-phase detection output signal and the quadrature detection output signal input from the quadrature detection unit 105, and changes the phase shift amount of the transmission path of the variable impedance matching unit 103. . “Phase shift amount” means the amount of phase change of a high-frequency signal.

  At this time, the control unit 106 changes the input impedance of the variable impedance matching unit 103 as seen from the output of the high-frequency power amplifier 102 so that the phase information of the detected reflected wave power is within a predetermined range. The impedance control signal to the impedance matching unit 103 is controlled. “Within a predetermined range” means within an impedance range in which the high-frequency power amplifier 102 is not destroyed.

  According to this configuration, the reflected wave power flowing backward from the antenna to the power amplifier is taken out, subjected to quadrature detection, and based on the in-phase detection output signal and the quadrature detection output signal, the reflected wave at the input of the variable impedance matching unit viewed from the power amplifier output Therefore, the high frequency power amplifier can be prevented from being destroyed by the reflected wave power from the antenna without reducing the high frequency output power of the high frequency power amplifier.

  Hereinafter, embodiments of the high-power high-frequency output device will be described.

  FIG. 2 is a block diagram of the high-power high-frequency output device 200 according to the present embodiment.

  The high power high frequency output device 200 includes a high frequency power oscillation unit 201, a high frequency power buffer amplifier 208, a final high frequency power amplifier 202, a variable impedance matching unit 203, a directional coupler 204, a quadrature detection unit 205, a control unit 206, a reflected wave. A low-pass filter 209, a reflected wave buffer amplifier 210, an oscillation signal low-pass filter 211, and a power distributor 220 are included. Further, the quadrature detection unit 205 includes a π / 2 phase shifter 212, an in-phase (I) side mixer 213, and a quadrature (Q) side mixer 214, and the control unit 206 includes an in-phase (I) side analog-digital converter. 215, a quadrature (Q) analog-digital converter 216, an arithmetic processing unit 217, a data storage device 218, and an impedance control signal output unit 219.

  The high-frequency power oscillation unit 201 is usually composed of a voltage controlled oscillator (VCO) and a phase locked loop (PLL), or a digital controlled oscillator (DCO) and a digital phase locked loop (DPLL). The present invention is not limited to this, and other configurations may be used as long as a stable oscillation frequency can be obtained.

  The high-frequency power signal output from the high-frequency power oscillation unit 201 is amplified by the high-frequency power buffer amplifier 208, and then further amplified by the final-stage high-frequency power amplifier 202 via the power distributor 220. The light is emitted from the antenna 207 through the coupler 204.

  The directional coupler 204 is a circuit using inductive coupling by two parallel microstrip lines 204a and 204b. When the microstrip lines 204a and 204b are used, a directional coupler with a low pass loss can be obtained with a very simple configuration.

  The directional coupler 204 functions as follows. That is, the signal input from the input terminal P1 is output from the output terminal P2 through the microstrip line 204a, and at the same time, output from the forward output terminal P4 by inductive coupling between the microstrip line 204a and the microstrip line 204b. However, it is not output from the reverse direction output terminal P3. Similarly, a signal input from the output terminal P2 is output from the input terminal P1 via the microstrip line 204a, and at the same time, from the reverse output terminal P3 due to inductive coupling between the microstrip line 204a and the microstrip line 204b. Although it is output, it is not output from the forward direction output terminal P4.

  The passing power loss from the input terminal P1 to the output terminal P2 and from the output terminal P2 to the input terminal P1, and the magnitude of the power output from the reverse output terminal P3 and the forward output terminal P4 are as follows. It depends on the size of the inductive coupling of the strip line 204b. If the coupling is increased, the power output from the reverse and forward output terminals P3 and P4 increases, but the passing power loss also increases. Conversely, if the coupling is reduced, the power output from the reverse and forward output terminals P3 and P4 is reduced, but the passing power loss can be reduced. In general, the size of the coupling is determined by the required dynamic range width of the signal to be extracted and the allowable power loss.

  In the present embodiment, the power output from the forward output terminal P4 is consumed by the termination resistor 221 and a part of the reflected wave power is extracted from the reverse output terminal P3. Therefore, by using the directional coupler 204, it is possible to extract a part of the reflected wave that passes the high-frequency power signal from the impedance matching unit 203 to the antenna 207 and flows backward from the antenna 207 to the impedance matching unit 203.

  Usually, the output impedance of a high-frequency power amplifier that outputs a large power signal is very low and is generally several Ω or less. The input impedance of the variable impedance matching unit 203 is configured to be optimally matched with the output impedance of the final stage high frequency power amplifier 202.

  When the impedance matching of the antenna 207 is lost due to the use state of the device or the environment change around the antenna, the antenna 207 reflects a high-frequency power signal, and the reflected wave power flows back to the output terminal P2 of the directional coupler 204. To do. The reflected wave power is input to the output of the final-stage high-frequency power amplifier 202 via the directional coupler 204 and the variable impedance matching unit 203.

  A part of the reflected wave signal from the antenna 207 is output to the quadrature detector 205 via the backward output terminal P3 of the directional coupler 204 and input to the reflected wave low-pass filter 209. The cut-off frequency of the reflected wave low-pass filter 209 is determined so as to suppress harmonics of the frequency output from the high-frequency power oscillation unit 201. The reflected wave power whose harmonic components are suppressed by the reflected wave low-pass filter 209 is amplified by the reflected wave buffer amplifier 210 to a power sufficient for direct detection by the direct detection unit 205 at the subsequent stage, and then quadrature detection is performed. Input to the unit 205. Here, it goes without saying that the reflected wave buffer amplifier 210 may be omitted when the reflected wave power is sufficient for quadrature detection and amplification is not necessary. The reflected wave power input to the quadrature detection unit 205 is divided into two and input to the in-phase (I) side mixer 213 and the quadrature (Q) side mixer 214, respectively.

  On the other hand, part of the output power of the high-frequency power buffer amplifier 208 distributed by the power distributor 220 is input to the oscillation signal low-pass filter 211. This filter has the same characteristics as the above-described reflected wave low-pass filter 209, and the cutoff frequency is determined so as to suppress the harmonics of the frequency output from the high-frequency power oscillation unit 201. The oscillation signal power whose harmonic components are suppressed by the oscillation signal low-pass filter 211 is input to the quadrature detection unit 205. The oscillation signal power input to the quadrature detection unit 205 is input to the π / 2 phase shifter 212, and is orthogonal to the input oscillating signal power in phase with the in-phase in-phase oscillation signal power by π / 2. Oscillation signal power is output. The in-phase oscillation signal power is input to the in-phase (I) side mixer 213, and the quadrature oscillation signal power is input to the quadrature (Q) side mixer 214.

  The in-phase (I) side mixer 213 performs detection by integrating the reflected wave power with the in-phase oscillation signal power input from the π / 2 phase shifter 212, that is, synchronously detects the reflected wave power with the in-phase oscillation signal power. As a result of multiplication of two input signals, an in-phase (I) component detection signal is output and input to the control unit 206. Similarly, the quadrature (Q) mixer 214 synchronously detects the reflected wave power with the quadrature oscillation signal power input from the π / 2 phase shifter 212, and outputs a quadrature (Q) component detection signal. 206 is input. At this time, although not shown in FIG. 2, in order to improve the S / N ratio of the in-phase (I) component detection signal and the quadrature (Q) component detection signal, the in-phase (I) side mixer 213 and the quadrature (Q) A low-pass filter may be arranged at each output of the side mixer 214.

  The in-phase (I) component detection signal and the quadrature (Q) component detection signal output from the in-phase (I) side mixer 213 and the quadrature (Q) side mixer 214 of the quadrature detection unit 205 and input to the control unit 206 are respectively in phase. (I) The signal is input to the analog-digital converter 215 and the quadrature (Q) analog-digital converter 216, converted into a digital signal, and input to the arithmetic processing unit 217.

  The arithmetic processing unit 217 calculates the impedance of the reflected wave power by digital calculation from the in-phase (I) component detection signal and the quadrature (Q) component detection signal obtained by quadrature synchronous detection of the reflected wave power with the oscillation signal power. Based on the calculation result, the impedance control signal is determined by referring to the impedance control signal data stored in the data storage device 218, and is output to the variable impedance control unit 203 via the impedance control signal output unit 219.

  The impedance control signal output unit 219 converts the impedance control signal input from the arithmetic processing unit 217 into a signal format that matches the element value control method of the variable element of the variable impedance matching unit 203, and sends it to the variable impedance matching unit 203. Send it out. The signal format described here includes a digital data format of 1 bit or more, an analog voltage format, and a pulse width modulation (PWM) signal format.

  According to the above configuration, since the in-phase detection output signal and the quadrature detection output signal obtained by quadrature detection of the reflected wave power are converted into digital signals, high-speed digital arithmetic processing is possible, and the impedance of the reflected wave power is reduced. It can be calculated accurately at high speed and easily. Therefore, an accurate impedance control signal can be sent to the variable impedance matching unit 203 at high speed, and the variable impedance matching unit 203 can be controlled. As a result, it is possible to prevent the high frequency power amplifier 202 from being damaged by the reflected wave power from the antenna without reducing the high frequency output power of the high frequency power amplifier 202.

  FIG. 3 shows a circuit configuration of an output matching circuit of a general high frequency power amplifier.

  A microstrip line 403 having a certain electrical length and a DC cut capacitor 404 are connected in series to the output (drain) of the high frequency power amplifier 401, and the DC cut capacitor 404 is connected to the output terminal 405 of the output matching circuit. And connected to the next stage circuit. One end of the shunt capacitor 402 is connected immediately to the output of the high-frequency power amplifier 401, and the other end of the shunt capacitor 402 is grounded. Furthermore, one end of an inductor 406 is connected between the output of the high-frequency power amplifier 401 and the microstrip line 403. One end of a bypass capacitor 407 is connected to the other end of the inductor 406, and the other end of the bypass capacitor 407 is grounded. Has been. The connection point between the inductor 406 and the bypass capacitor 407 is connected to the DC power supply terminal 408, and the driving power (Vdd) of the high frequency power amplifier 401 is supplied from the DC power supply terminal. Note that the description of the circuit at the input of the high-frequency power amplifier 401 is not particularly relevant to the present invention. Therefore, explanation is omitted.

  Inductor 406 has an inductance value selected so that the impedance is sufficiently large at the operating frequency so as not to affect the output impedance matching of high-frequency power amplifier 401. In FIG. 3, the inductor 406 is connected between the high frequency power amplifier 401 and the microstrip line 403, but may be connected between the microstrip line 403 and the DC cut capacitor 404.

  Usually, the output impedance of the high-frequency power amplifier 401 is matched by the shunt capacitor 402 and the microstrip line 403, but the DC cut capacitor 404 may be used as an impedance matching capacitor.

  FIG. 4 is a diagram schematically showing a change in impedance when the electrical length of the microstrip line 403 is changed. When the initial impedance is point A, the impedance changes to point B when the electrical length of the microstrip line 403 is increased, and conversely to point C when the electrical length is shortened. Therefore, the phase shift amount of the transmission line can be changed by changing the electrical length of the microstrip line 403.

  In addition, when the DC cut capacitor 404 is used as an impedance matching variable element and the capacitance of the capacitor is changed, assuming that the initial impedance is a point A shown in FIG. When the capacitance is increased, the impedance changes to point B. Conversely, when the capacitance is reduced, the impedance changes to point C. Therefore, the phase shift amount of the transmission line can be changed by changing the capacitance of the DC cut capacitor 404.

  The variable impedance control unit 203 is a variable element such as the microstrip line 403 or the DC cut capacitor 404 described above according to the impedance control signal converted into an appropriate signal format from the impedance control signal output unit 219 of the control unit 206 and input. By changing the element value, the phase shift amount of the signal transmission line is changed, and the input impedance of the variable impedance matching unit 203 viewed from the output of the final-stage high-frequency power amplifier 202 is changed.

  That is, based on the in-phase detection output signal and the quadrature detection output signal, the impedance of the reflected wave at the input of the variable impedance matching unit 203 viewed from the output of the power amplifier 202 is controlled to an impedance region where the power amplifier 202 is not destroyed. A high-power high-frequency output device 200 that can prevent destruction of the high-frequency power amplifier due to reflected wave power from the antenna without reducing the output power can be realized.

  In this embodiment, a directional coupler is used to detect the impedance of the reflected wave. But this is an example. It can be realized by other configurations.

  Here, an impedance region where the power amplifier 202 is not destroyed will be described with reference to FIGS.

  FIG. 5 shows an example of drain current (Id) load pull data of the power amplifier 202. The distribution of the drain current with respect to the impedance of the amplifier output can be easily measured by a commonly used load pull method.

  For example, the data storage device 218 of the control unit 206 holds the drain current (Id) load pull data shown in FIG. 5 in advance. The computing unit 217 detects the impedance of the reflected wave, refers to the load pull data held using the detected impedance, determines whether or not to adjust the impedance, and if necessary, further determines the load pull. Adjust the impedance based on the data.

  In this embodiment, it is assumed that the amplifier is destroyed by an overcurrent at a drain current (Id) = 500 mA. This current value can also be specified by experiments or the like.

  A region where Id> 500 mA, that is, a region surrounded by the drain current (Id) = 500 mA in FIG. 5 is an impedance region where the power amplifier 202 is destroyed by an overcurrent. This area is hereinafter referred to as “destructive area”.

  Therefore, the “region where the power amplifier is not destroyed” means a region outside the destruction region surrounded by the drain current (Id) = 500 mA in FIG. The control unit 206 generates an impedance control signal and controls the variable impedance matching unit 203 so that the impedance of the reflected wave power enters a region where the power amplifier 202 is not destroyed. As is clear from the figure, the impedance of the reflected wave power can be classified as “destructive region” or “region other than the destructive region”, and the subsequent control method differs depending on which of them belongs. ing.

  Next, an impedance control method according to the detected impedance of the reflected wave power will be described with reference to FIG. FIG. 6 is a diagram for explaining the direction in which the phase is rotated.

  First, when the impedance of the reflected wave power is outside the destruction region, for example, when it is located at point D in FIG. 6, it is not necessary to perform phase shift control. Since the control unit 206 does not output an impedance control signal for changing the phase, the phase shift amount of the variable impedance matching unit 203 does not change.

  Next, when the impedance of the reflected wave power is within the breakdown region, for example, when it is located at points A, B, and C in FIG. 6, the amount of phase shift is controlled as follows.

  In principle, in order to change the impedance out of the breakdown region, the control unit 206 shifts the phase in the direction in which the amount of phase shift is small. When the phase shift amount is the same in any direction due to the impedance being substantially at the center of the breakdown region, the phase shifts in a direction approaching the real axis.

  Description will be made by taking points A, B, and C in FIG. 6 as examples.

  When the impedance of the reflected wave power is located at point A in FIG. 6, the impedance is controlled so that the impedance is shifted in the direction of A → A ′, which is the direction of the smallest transfer amount. In order to realize this transfer, as described later with reference to FIGS. 8 to 10, the inductance value of the series inductor L may be increased, or the capacitance of the series capacitor C may be decreased.

  Further, when the impedance of the reflected wave power is located at the point B in FIG. 6, the impedance is controlled so that the impedance is shifted in the direction of B → B ′, which is the direction of the smallest transfer amount. In order to realize this transfer, as will be described later with reference to FIGS. 8 to 10, the inductance value of the series inductor L may be reduced, or the capacitance of the series capacitor C may be increased.

  On the other hand, when the impedance of the reflected wave power is located at the point C in FIG. 6, the impedance is controlled so that the impedance is shifted in the direction of C → C ′ because the impedance is substantially at the center of the destruction region. In order to realize this transfer, as will be described later with reference to FIGS. 8 to 10, the inductance value of the series inductor L may be reduced, or the capacitance of the series capacitor C may be increased.

  Note that “impedance is approximately in the center of the breakdown region” means that the phase angle that must be turned to move the impedance out of the breakdown region, that is, the angle A → A ′ and B → B in FIG. It means that the angle of 'becomes substantially the same.

  Next, the initial operation of the high-power high-frequency output device 200 will be described with reference to FIG. The reason for taking the initial operation in particular is to make it operate more safely when the system is started. Since the impedance of the reflected wave power cannot be controlled at the time of starting the system, if the power amplifier 202 is operated at a high output in that state, the power amplifier 202 may be destroyed. Therefore, at the time of starting the system, the power amplifier 202 is operated at a low output, the impedance of the reflected wave is detected, and after the control of the reflected wave impedance outside the destruction region is completed, the power amplifier 202 is operated at a normal high output. It is preferable to control it.

  FIG. 7 is a flowchart showing an algorithm of an initial operation of the high-power high-frequency output device 200.

  In step S1, the power amplifier 202 starts operating at a low output. In the next step S2, the control unit 206 detects the impedance of the reflected wave. In step S3, it is determined whether or not the impedance of the reflected wave is within the destruction region. If it is within the destruction region, the process proceeds to step S4, and the impedance phase shift amount is controlled by the method described with reference to FIG. On the other hand, if it is not in the destruction area, the process proceeds to step S5.

  In step S5, since the phase is controlled, the power amplifier 202 operates at a high output. Thereafter, the normal high output operation is performed, but the impedance of the reflected wave is monitored constantly or intermittently (step S6), and the processes after step S2 are performed as necessary.

  By performing the initial control at the time of starting the system in this way, it is possible to prevent the amplifier from being suddenly destroyed when the reflected wave impedance at the time of starting the system is within the destruction region.

  Next, a configuration example of the variable impedance matching unit 203 according to the present embodiment will be described with reference to FIGS.

  As shown in FIG. 8, the output (drain) of the high-frequency power amplifier 651 is connected in series in the order of the first DC cut capacitor 602, the microstrip line 603, the second DC cut capacitor 604, and the output terminal 606. ing. A shunt capacitor 601 is connected between the output of the high-frequency power amplifier 651 and GND. One end of a resistor 605 is connected between the microstrip line 603 and the second DC cut capacitor 604, and the other end of the resistor 605 is connected to a control voltage input terminal 607. In addition, one end of an inductor 652 is connected to the output of the high frequency power amplifier 651, and the other end of the inductor 652 is connected to one end of a DC power supply terminal 654. Further, one end of a bypass capacitor 653 is connected between the inductor 652 and the DC power supply terminal 654 and between GND.

  At this time, the microstrip line 603 is formed on a ferroelectric substrate using a ferroelectric as a dielectric. Further, the first DC cut capacitor 602 and the second DC cut capacitor 604 have respective capacitor capacities so that the impedance becomes sufficiently small at the frequency at which the circuit operates so as not to affect the impedance matching. A value is selected. Further, the inductor 652 has an inductance value selected so that the impedance becomes sufficiently large at the frequency at which the circuit operates so as not to affect the impedance matching.

  The resistor 605 normally uses a fixed resistor having a resistance value of several hundred Ω to several kΩ, but may be an inductor having an inductance value with which the impedance becomes sufficiently large at the frequency at which the circuit operates. Good.

  In the configuration of FIG. 8, when a DC voltage is applied to the control voltage input terminal 607, the circuit is OPEN in terms of DC, so that the microstrip line 603 is also applied to the control voltage input terminal 607 via the resistor 605. The same voltage as the DC voltage thus applied is applied, and the same voltage is applied to the ferroelectric substrate forming the microstrip line 603. A ferroelectric is a material whose dielectric constant changes when an applied voltage is changed. Therefore, by applying a DC voltage to the microstrip line formed on the ferroelectric substrate and changing the voltage, the dielectric constant between the transmission line and GND changes, and as a result, the electrical length of the microstrip line is reduced. Change. Therefore, by changing the DC voltage applied to the control voltage input terminal 607, the electrical length of the microstrip line 603 can be changed, and as a result, the phase shift amount can be changed.

  In the case of the configuration of FIG. 8, the impedance control signal output unit 219 that supplies a DC voltage to the control voltage input terminal 607 converts the impedance control signal input from the arithmetic processing unit 217 into a DC voltage, and the control voltage input terminal 607 is configured by a digital-analog converter, or an integrator (not shown) is arranged between the impedance control signal output unit 219 and the control voltage input terminal 607 to perform pulse width modulation (PWM ) It is preferable to configure with signal output, but it is not limited thereto.

  Subsequently, as shown in FIG. 9, one end of each of a shunt capacitor 701, a first DC cut capacitor 702, and an inductor 752 is connected to the output (drain) of the high-frequency power amplifier 751. The other end of the first DC cut capacitor 702 is connected to the common terminal of the first high frequency switch 706, and the other end of the inductor 752 is connected to the bypass capacitor 753 and the DC power supply terminal 754, respectively. The other end of the bypass capacitor 753 is connected to GND.

  The first high-frequency switch 706 has its (a) side terminal connected to one end of the first microstrip line 703 and (b) side terminal connected to one end of the second microstrip line 704, respectively. The other end of the strip line 703 is connected to the (a) side terminal of the second high frequency switch 707, and the other end of the second microstrip line 704 is connected to the (b) side terminal of the second high frequency switch 707. . The common terminal of the second high frequency switch 707 is connected to one end of the second DC cut capacitor 705, and the other end of the second DC cut capacitor 705 is connected to the output terminal 708.

  Furthermore, the control terminal of the first high-frequency switch 706 and the control terminal of the second high-frequency switch 707 are connected, and are connected to the control signal input terminal 709. The first high-frequency switch 706 and the second high-frequency switch 707 are operated simultaneously by the control signal input to the control signal input terminal 709, and both the first high-frequency switch 706 and the second high-frequency switch 707 are connected to the terminal (a). (B) a state in which both the first high frequency switch 706 and the second high frequency switch 707 are connected to the terminal side (hereinafter, this state is referred to as the first state). The two states (the state is referred to as the second state) are selectively controlled.

  Further, the first DC cut capacitor 702 and the second DC cut capacitor 705 have respective capacitor capacities so that the impedance is sufficiently small at the frequency at which the circuit operates so as not to affect the impedance matching. A value is selected. Further, the inductor 752 has an inductance value selected so that the impedance becomes sufficiently large at the frequency at which the circuit operates so as not to affect the impedance matching.

  Here, the first microstrip line 703 and the second microstrip line 704 are designed to have different electrical lengths, and the first high frequency switch 706 and the second high frequency switch 707 are connected to the control signal input terminal. By switching the control signal input to 709 and switching the path to the first state and the second state, the electrical length of the microstrip line portion is selectively set to two preset electrical lengths. As a result, two kinds of phase shift amounts can be selectively switched.

  In the case of the configuration of FIG. 9, two types of microstrip lines having different electrical lengths are switched to switch two types of phase shift amounts. However, the configuration is not limited to this, and three or more types of microstrip lines are switched. The strip line may be switched.

  9, the impedance control signal output unit 219 that supplies a control signal to the control signal input terminal 709 converts the impedance control signal input from the arithmetic processing unit 217 into a binary signal of 1 bit or more. It is preferable that the signal is converted and output to the control signal input terminal 709.

  As shown in FIG. 10, one end of each of a shunt capacitor 801, a microstrip line 802, and an inductor 852 is connected to the output (drain) of the high-frequency power amplifier 851, and the other end of the shunt capacitor 801 is connected to GND. The other end of the inductor 852 is connected to one end of a bypass capacitor 853 and a DC power supply terminal 854, and the other end of the bypass capacitor 853 is connected to GND. The other end of the microstrip line 802 is connected to one end of the first DC cut capacitor 803, and the other end of the first DC cut capacitor 803 is connected to the inductor 807 and the anode terminal of the varactor diode 804. The other end of the inductor 807 is connected to GND.

  The cathode terminal of the varactor diode 804 is connected to one end of each of the inductor 806 and the second DC cut capacitor 805, and the other end of the inductor 806 is connected to the control voltage input terminal 809 for the second DC cut use. The other end of the capacitor 805 is connected to the output terminal 808.

  Further, the first DC cut capacitor 803 and the second DC cut capacitor 805 have capacities of the respective capacitors so that the impedance becomes sufficiently small at the frequency at which the circuit operates so as not to affect the impedance matching. A value is selected. Inductors 852, 804, and 806 are selected so that the impedance of each inductor is sufficiently large at the frequency at which the circuit operates so that impedance matching is not affected.

  In FIG. 10, the inductor 852 is connected between the output of the high-frequency power amplifier 851 and the microstrip line 802, but may be connected between the microstrip line 802 and the first DC cut capacitor 803. .

  In the configuration of FIG. 10, when a DC voltage is applied to the control voltage input terminal 809, a DC circuit is a series circuit of an inductor 806, a varactor diode 804, and an inductor 807. A reverse bias voltage is applied. The varactor diode has a characteristic that the capacitance between the two terminals is changed by changing the reverse bias voltage between the cathode terminal and the anode terminal. Therefore, by changing the reverse bias voltage between the cathode terminal and the anode terminal of the varactor diode, the capacitance between both terminals changes, and as a result, the capacitance of the series capacitor in the transmission line changes. Therefore, by changing the DC voltage applied to the control voltage input terminal 809, the capacitance of the series capacitor of the transmission line can be changed, and as a result, the phase shift amount of the transmission line can be changed.

  In the case of the configuration of FIG. 10, the impedance control signal output unit 219 that supplies a DC voltage to the control voltage input terminal 809 converts the impedance control signal input from the arithmetic processing unit 217 into a DC voltage, and the control voltage input terminal 809 is configured by a digital-analog converter, or an integrator (not shown) is arranged between the impedance control signal output unit 219 and the control voltage input terminal 809 to perform pulse width modulation (PWM ) It is preferable to configure with signal output, but it is not limited thereto.

  As described above, according to the present embodiment, even when reflection occurs in the antenna due to the impedance matching of the antenna being broken due to the usage state of the device or the environmental change around the antenna, the antenna flows back to the power amplifier. Takes out reflected wave power, performs quadrature detection, and controls the impedance of the reflected wave at the input of the variable impedance matching unit viewed from the power amplifier output to an impedance region where the power amplifier is not destroyed based on the in-phase detection output signal and the quadrature detection output signal Thus, it is possible to realize a high-power high-frequency output device that can prevent the high-frequency power amplifier from being destroyed by reflected wave power from the antenna without reducing the high-frequency output power of the high-frequency power amplifier.

  FIG. 11 is an external view of the microwave oven 901. The microwave oven 901 includes a high-power high-frequency output device 200, and can heat food by high-power high-frequency output from the antenna 207 of the high-power high-frequency output device 200.

  FIG. 12 is an external view of a radio tower 902 of a base station such as a mobile phone. The radio tower 902 includes a high-power high-frequency output device 200, and communication can be performed by the high-power high-frequency output from the antenna 207 of the high-power high-frequency output device 200.

  Both the microwave oven 901 shown in FIG. 11 and the radio wave tower 902 shown in FIG. 12 do not require expensive parts such as an isolator. Further, it is possible to prevent the high frequency power amplifier from being destroyed by the reflected wave power from the antenna due to the impedance mismatch of the antenna without reducing the high frequency output power of the high frequency power amplifier.

  So far, it has been described that a part of the reflected wave signal is extracted using a directional coupler. However, the directional coupler is an example, and other configurations can be utilized. Hereinafter, a description will be given with reference to FIGS.

  FIG. 13 shows a partial configuration of a high-power high-frequency output device 200 using a circulator 253 instead of the directional coupler.

  The circulator 253 is a high-frequency passive element using a magnet that is used when a transmission circuit and a reception circuit that operate at the same frequency are connected to one antenna. A signal input to a certain port is output only from a specific next port. According to FIG. 13, a signal passes only in the direction of terminal P1, terminal P2, terminal P2, terminal P3, terminal P3, and terminal P1.

  Therefore, the circulator 253 can receive the high frequency power signal from the variable impedance matching unit 203 at the terminal P 1 and pass it from the terminal P 2 toward the antenna 207. Further, the reflected wave signal from the antenna 207 can be received at the terminal P2 and demultiplexed from the terminal P3 toward the quadrature detector 209.

  Compared to the directional coupler, the demultiplexing level of the reflected wave is increased in the circulator 253, and therefore the gain of the amplifier (the reflected wave buffer amplifier 210 in FIG. 2) before the quadrature detector 206 can be reduced or omitted. be able to.

  FIG. 14 shows a partial configuration of a high-power high-frequency output device using a hybrid coupler 263 instead of the directional coupler.

  The hybrid coupler 263 is a circuit element that obtains an output similar to that of a directional coupler by phase operation using a microstrip line. A signal input to a certain port of the hybrid coupler 263 is output only from a specific next port. Specifically, a signal from the terminal P1 to the terminal P2 is not output to the terminal P3. On the other hand, the signal input to the terminal P2 is output to the terminal P3.

  Therefore, the hybrid coupler 263 can receive the high frequency power signal from the variable impedance matching unit 203 at the terminal P 1 and pass it from the terminal P 2 toward the antenna 207. Further, the reflected wave signal from the antenna 207 can be received at the terminal P2 and demultiplexed from the terminal P3 toward the quadrature detector 209.

  Compared to the directional coupler 214, the hybrid coupler 263 can increase the degree of coupling, and can also increase the demultiplexing level of the reflected wave. However, since the passage loss becomes larger, it is necessary to optimize the loss and the degree of coupling.

  In the embodiment described above, the direction in which the phase is rotated (phase shift control method) has been described with reference to FIG. In the method of FIG. 6, the transfer control is performed in the direction in which the turning angle becomes smaller.

  However, after the transfer control is performed once, the reflection state is changed again, and when the transfer control is performed again, the initial impedance as much as possible, that is, the impedance of the variable element becomes an impedance that is near the center of the controllable range. It is preferable to control so that it may approach. Therefore, the following transfer control may be performed. As described above, the “variable element” corresponds to the microstrip line 403, the DC cut capacitor 404 (FIG. 3), or the like.

  Hereinafter, different phase control methods will be described with reference to FIGS. 15 and 16.

  FIG. 15 is a diagram for explaining a direction in which the phase is rotated by a method different from that in FIG. 6. FIG. 16 is a diagram showing the relationship between the control voltage and the element value related to the control element. Points A to F described in FIGS. 15 and 16 correspond to each other.

  As shown in FIG. 16, the variable range of the element value of the variable element is finite. However, the width of the variable range and the sensitivity of the element differ depending on the type of element used as the phase variable element. When the vicinity of the center of the variable range is compared with the vicinity of the upper limit or the lower limit, the change amount (sensitivity) of the element value differs with respect to the change amount of the control voltage. At the upper or lower limit, the sensitivity is lower than near the center of the variable range. Therefore, it is desirable to employ an element value corresponding to the central region of the variable range as much as possible.

  Reference is now made to FIG. Let the initial impedance of the control element be point A. As shown in FIG. 15, control is not performed because point A is outside the destruction region. Further, as shown in FIG. 16, the point A is a substantially central region of the control range of the control element.

  Next, it is assumed that the state of the load has changed and the impedance has changed to point B. According to FIG. 16, the state (element value) of the control element is the same at point A and point B. In order to change the impedance out of the destruction region, the phase is controlled so that the impedance moves to the point C. The control element operates at point C. Up to this point, the control method is the same as that shown in FIG.

  Next, it is assumed that the state of the load has changed again and the impedance has changed to point D. The state of the control element remains at point C. Here, according to the above description, control is performed so that the impedance moves to the point E.

  However, in this modification, control is performed so that the impedance moves to point F. When moving to point E, the amount of phase rotation can be smaller than when moving to point F. However, since the control element approaches the variable upper limit, the impedance is moved to the F point so as to approach the median value of the variable width. Thereby, it becomes easy to cope with the next change.

  In this embodiment, when adjusting the impedance, the calculation unit 217 (FIG. 2) detects the impedance of the reflected wave, and refers to the load pull data held in the data storage device 218 using the detected impedance. The necessity of adjusting the impedance is determined, and if necessary, the impedance is further adjusted based on the load pull data.

  Since such processing generally requires the use of a microcomputer, there may be a problem with the circuit scale and processing time.

  Therefore, as another example, information regarding a control signal corresponding to data indicating the direction and amount of rotating the phase may be stored in the data storage device 218 and output as necessary. Based on the load pull information acquired in advance through experiments or the like, the direction and amount of phase rotation are obtained for the impedance value of the reflected wave, and the corresponding signal information to the impedance control signal output unit 219 is stored in the storage device. Keep it.

  However, in this case, as the resolution of the impedance value is increased, that is, the memory capacity increases as the impedance value is more accurately controlled, it is necessary to individually determine how much resolution is practically required.

  According to the high-power high-frequency output device of the present invention, it can be easily applied to cooking equipment using a high-frequency such as a communication base station amplifier and a microwave oven, and the industrial applicability is extremely large.

It is a block diagram which shows the basic composition of the high electric power high frequency output device 100 of this invention. 1 is a block configuration diagram of a high power high frequency output device 200 according to an embodiment of the present invention. FIG. It is a figure which shows the circuit structure of the output matching circuit of a general high frequency power amplifier. It is the figure which showed typically the change of the impedance when the electrical length of the microstrip line 403 changes. It is a figure which shows an example of the drain current (Id) load pull data of the power amplifier 202. It is a figure for demonstrating the direction which rotates a phase. 5 is a flowchart showing an algorithm of an initial operation of the high-power high-frequency output device 200. It is a figure which shows the 1st structural example of a variable impedance matching part. It is a figure which shows the 2nd structural example of a variable impedance matching part. It is a figure which shows the 3rd structural example of a variable impedance matching part. 1 is an external view of a microwave oven 901. FIG. It is an external view of the radio tower 902 of a base station such as a mobile phone. It is a figure which shows a partial structure of the high power high frequency output device 200 using a circulator 253 instead of the directional coupler. It is a figure which shows a part of structure of the high electric power high frequency output device which replaced with the directional coupler and used the hybrid coupler 263. FIG. It is a figure for demonstrating the direction which rotates a phase by the method different from FIG. It is a figure which shows the relationship between the control voltage regarding a control element, and element value. It is a block diagram which shows the structure of the high frequency electric power output apparatus 300 of a prior art example.

Explanation of symbols

DESCRIPTION OF SYMBOLS 100 High power high frequency output device of this invention 101 High frequency power oscillation part 102 High frequency power amplifier 103 Variable impedance matching part 104 Directional coupler 105 Orthogonal detection part 106 Control part 107 Antenna 200 High power high frequency output apparatus 201 Example of this invention Power oscillation unit 202 Final stage high frequency power amplifier 203 Variable impedance matching unit 204 Directional coupler 205 Quadrature detection unit 206 Control unit 207 Antenna 208 High frequency power buffer amplifier 209 Reflected wave low-pass filter 210 Reflected wave buffer amplifier 211 Oscillating signal low band Pass filter 212 π / 2 phase shifter 213 In-phase (I) side mixer 214 Quadrature (Q) side mixer 215 In-phase (I) side analog-digital converter 216 Quadrature (Q) side analog-digital converter 217 Arithmetic processing unit 18 data storage device 219 a digital - analog converter 220 power divider

Claims (9)

A high-frequency power amplifier that receives, amplifies and outputs high-frequency power;
A variable impedance matching unit that is provided on the output side of the high-frequency power amplification unit and outputs the high-frequency power by changing an input impedance based on an input control signal;
An antenna that outputs the high-frequency power output from the variable impedance matching unit;
The quadrature detection unit that quadrature-detects the reflected wave power from the antenna with the high-frequency power input to the high-frequency power amplification unit, and outputs an in-phase detection output signal and a quadrature detection output signal;
A high-power high-frequency output device comprising: a control unit that generates the control signal according to the in-phase detection output signal and the quadrature detection output signal and sends the control signal to the variable impedance matching unit;
Based on the phase information of the reflected wave power obtained from the in-phase detection output signal and the quadrature detection output signal, the control unit causes the impedance of the reflected wave power to fall within a range where the high-frequency power amplification unit is not destroyed. A high-power, high-frequency output device that generates a control signal that changes the phase shift amount of the transmission line of the variable impedance matching unit. The impedance of the reflected wave power at the input of the variable impedance matching unit viewed from the output of the high-frequency power amplification unit can be classified into one of the value of the first impedance region or the value of the second impedance region,
The first impedance region includes an impedance at which the high-frequency power amplification unit is destroyed due to an overcurrent state of a current flowing through the high-frequency power amplification unit.
The second impedance region includes an impedance in which the high-frequency power amplification unit is not destroyed because a current value of a current flowing through the high-frequency power amplification unit does not exceed a breakdown resistance limit.
2. The control unit according to claim 1, wherein the control unit generates the control signal such that an impedance of the reflected wave power falls within the second impedance region, and changes a phase shift amount of a transmission path of the variable impedance matching unit. The high power high frequency output device described. The control unit includes a converter that converts the analog in-phase detection output signal and the quadrature detection output signal into a digital format, respectively, and a storage unit that holds a control table for specifying the second impedance region. And
The high-power, high-frequency output device according to claim 2, wherein the control signal is generated based on the digital in-phase detection output signal, the quadrature detection output signal, and the control table.   The high-power high-frequency output device according to claim 3, wherein the storage unit is a semiconductor memory.   The high-power high-frequency output device according to claim 3, wherein the control unit converts the control signal into a digital signal of 1 bit or more and transmits the digital signal.   The high-power high-frequency output device according to claim 3, wherein the control unit converts the control signal into an analog signal and transmits the analog signal.   The high-power, high-frequency output device according to claim 3, wherein the control unit converts the control signal into a pulse width modulation (PWM) signal and transmits the signal.   The high-power high-frequency output device according to claim 1, further comprising a high-frequency power oscillation unit that oscillates the high-frequency signal and inputs the high-frequency signal to the high-frequency power amplification unit and the quadrature detection unit.   A directional coupler that is provided between the output of the variable impedance matching unit and the antenna, transmits the output power of the variable impedance matching unit to the antenna, and extracts the reflected wave power from the antenna; The high-power high-frequency output device according to claim 8 provided.

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