JP2009194888A - Mixer circuit and radio communication device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To obtain high linearity of a base band signal to a current in a voltage-current conversion circuit for carrying out voltage-current conversion requiring high linearity, and to reduce the cost by realizing reduction in a channel width of a transistor subjected to cascode connection and reduction in a chip size of the entirety of the circuit. <P>SOLUTION: The device is constituted of a carrier wave modulating part 100 and a voltage-current conversion part 1. The voltage-current conversion part 1 has a transistor M1, which operates in a linear region, and a transistor M2 which is subjected to cascode connection to the transistor M1 and whose drain side is connected to a carrier wave modulating part 100. A negative feed back amplifier A1, whose feed back gain is A, is added between the drain of the transistor M1 and the gate of the transistor M2. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a mixer circuit typified by a Gilbert cell mixer and the like, and a wireless communication apparatus having the mixer circuit, and relates to linearity of frequency components of an output signal with respect to an input signal.

  In Patent Document 1, giving priority to the peak of the input third-order intercept point IIP3 sacrifices the conversion voltage gain to some extent, and accordingly, the noise figure NF is lowered. It cannot be said that it is drawing out to the maximum. Therefore, a first current source, an amplification stage, a switch stage, and an output load are connected in series between the power supply lines, and an output obtained by mixing the first and second signals input to the amplification stage and the switch stage is the switch. In a mixer circuit configured to output from between a stage and an output load, by supplying a bias current between the amplification stage and the switch stage, the operating currents of the amplification stage and the switch stage are individually set. There is disclosed a mixer circuit that includes a second current source and sets a value of a bypass current by the second current source so that an output third-order intercept point OIP3 is maximized, that is, linearity is improved.

Further, in Patent Document 2, an output of an operational amplifier circuit having an inverting terminal as an input, not a cascode, is given to the gate of a PMOS transistor, and the drain side of the PMOS transistor is grounded via a resistor, and the source side Is connected to the power supply. In this circuit, the same voltage as that applied to the input terminal is applied to both ends of the resistor to generate a current at the output terminal. A PLL circuit is disclosed in which the current flowing through the PMOS transistor can be taken out as being proportional to the input voltage.
JP 2004-104515 A JP 2006-33197 A

In a mixer circuit that converts the frequency of a baseband signal that transmits information in telecommunications to a high frequency side (hereinafter referred to as “up-conversion”), when the carrier wave (high frequency) is multiplied by the baseband signal, the harmonics of the carrier wave Components containing can be easily removed by LPF. However, since the frequency of the baseband signal is set lower than that of the carrier wave, components related to the harmonics of the baseband signal centered on the reference carrier frequency cannot be easily removed by a filter or the like. Thus, there is a risk of entering the intermediate frequency band of the mixer circuit.
Ω ₁ shown in FIG. 6 is the carrier angular frequency, and ω ₂ is the angular frequency of the baseband signal. It can be seen from FIG. 6 that the unnecessary signal surrounded by the broken line is close to the desired signal.

In addition, when a carrier wave (high frequency) is multiplied by a baseband signal in a mixer circuit that performs frequency conversion of the band of the baseband signal to the low frequency side (hereinafter referred to as “down conversion”), harmonics of the carrier wave and the baseband signal are also used. Can be easily removed by LPF. However, since the component related to the harmonics of the carrier wave-baseband signal is not significantly different from that of the intermediate frequency, it cannot be easily removed by LPF or the like, and further, as shown in FIG. There is a risk of entering the intermediate frequency band.
Ω ₁ shown in FIG. 7 is the carrier angular frequency, and ω ₂ is the angular frequency of the baseband signal. It can be seen from FIG. 7 that the unnecessary signal surrounded by the broken line is close to the desired signal.
From the above, the generation of harmonic components in the baseband signal leads to transmission / reception of interference signals in both up-conversion and down-conversion. For this reason, it is important as transmission / reception performance to increase the interference elimination capability so that transmission / reception of a target signal is not hindered by an unintended signal.

Here, FIG. 8 shows an example of a mixer circuit that can solve the above problem.
The mixer circuit shown in FIG. 8 includes a carrier wave modulation unit 100 and a baseband signal voltage-current conversion unit 110. The in-phase component and the differential component of the carrier signal are input to the nodes LO + and LO− of the carrier modulation unit 100, respectively, and the modulated signals are output as differential outputs OUT1 and OUT2 by the load.
This mixer circuit uses an NMOS transistor and has a single-phase baseband signal. Incidentally, the same applies to a mixer in which the baseband signal is a differential input and various active mixers.

The voltage-current conversion unit 110 transmits the baseband signal to the carrier modulation unit in the form of current. Therefore, it is desirable that the MOS transistor M1 that performs voltage-current conversion of the baseband signal operates in a linear region so that harmonic components of the baseband signal are not generated. However, if a change occurs in the source-drain voltage (source or drain potential when one of them is a fixed potential) in the linear region, it affects the drain current in a proportional relationship, resulting in a nonlinear element. In particular, in the mixer circuit shown in FIG. 8, when the signal input to the gate of the transistor M1 changes, the drain current changes, and as a result, the drain potential (source-drain voltage) changes.
Further, since the circuit to which the carrier wave is input is connected to the drain side of the transistor M1, it is easily affected by the carrier wave. Furthermore, since it operates in the linear region, the gain is lower than that in the saturation region.
Conventionally, the transistor M2 is cascode-connected in order to suppress the fluctuation of the drain of the transistor M1 and obtain a higher gain when performing voltage-current conversion of the baseband signal to avoid the influence. There are also technologies (for example, Behzad Razavi / translated by Tadahiro Kuroda; analog CMOS integrated circuit design basics, 7th edition, Maruzen Co., Ltd., refer to Section 3.5). Note that the gate of the transistor M2 is fixed by a bias voltage.

ただし、式内のβは、式（２）のようにチャネル幅Ｗ、チャネル長Ｌ及びプロセスによって決まる定数β₀で表される。以降βの値は各トランジスタによって決められるものとする。

式（１）の電流Ｉ₀が流れている状態からＶ_inが△Ｖ_inだけ変化し、それに伴いＶ₁が△Ｖ₁、Ｉ₀が△Ｉ₀だけ変化したものとすると、式（１）から式（３）が得られる。

トランジスタＭ１に関する式（１）及び（３）から△Ｉ₀に関する式（４）が求められる。

Equation (1) shows an equation regarding the current I ₀ flowing through the drains of the transistors M1 and M2 operating in the linear region in this circuit.

However, β in the equation is represented by a constant β ₀ determined by the channel width W, the channel length L, and the process as in equation (2). Hereinafter, the value of β is determined by each transistor.

Assuming that V _in changes by ΔV _in from the state where the current I _{0 in} equation (1) flows, V ₁ changes by ΔV ₁ and I ₀ changes by ΔI ₀ accordingly, equation (1) From which equation (3) is obtained.

Equation (4) relating to ΔI ₀ is obtained from equations (1) and (3) relating to transistor M1.

式（５）からトランジスタＭ２のトランスコンダクタンスによって、△Ｖ_inに対するトランジスタＭ１のドレイン電位の変動△Ｖ₁が抑えられていることが分かる。
この式（５）を上述の式（４）に代入すると式（６）が得られ、その式（６）の△Ｖ_inの二乗に関する項が非線形項となる。

式（６）からトランジスタＭ１のトランスコンダクタンスをより低く、トランジスタＭ２のトランスコンダクタンスをより高く設定することが望ましいことが分かる。そのため、トランジスタＭ２は飽和領域で動作させることが良いことも分かる。 Here, since ΔI ₀ flowing through the transistors M1 and M2 is equal, Expression (5) is established using the transconductances g _m1 and g _m2 of the transistors M1 and M2.

From the equation (5), it can be seen that the variation ΔV ₁ of the drain potential of the transistor M1 with respect to ΔV _in is suppressed by the transconductance of the transistor M2.
By substituting this equation (5) into the above equation (4), equation (6) is obtained, and the term relating to the square of ΔV _{in in} the equation (6) becomes a nonlinear term.

From equation (6), it can be seen that it is desirable to set the transconductance of the transistor M1 lower and the transconductance of the transistor M2 higher. Therefore, it can be seen that the transistor M2 is preferably operated in the saturation region.

However, since the gain of the entire mixer circuit is also required to some extent, the gain of the transistor M1 cannot be made zero. As a result, the transconductance g _m2 of the transistor M2 is also increased, the channel width is remarkably increased, and the current value flowing through the drains of the transistors M1 and M2 and the power consumption of the entire mixer circuit are increased.
Further, according to the circuit described in Patent Document 1, there are problems that the number of elements constituting the circuit is larger than that of the cascode circuit, resulting in an increase in chip area / power consumption and complicated design.
Further, in Patent Document 2, when the nonlinear component of the input voltage with respect to the current is set to be low, the linear component also becomes low, and a sufficient gain cannot be obtained.
The present invention has been made in view of the above points, and in the cascode circuit that performs voltage-current conversion that requires high linearity as described above, high linearity of the baseband signal with respect to current is obtained. Another object of the present invention is to provide a mixer circuit capable of reducing costs by reducing the channel width of a transistor to be cascode-connected and reducing the chip size of the entire circuit, and a wireless communication apparatus having the mixer circuit.

  To achieve the above object, the invention according to claim 1 is a mixer circuit that performs frequency conversion by multiplying two input signals, and an input stage having a first MOS transistor that operates in at least one linear region, and A voltage-current conversion circuit that has at least one second MOS transistor that is cascode-connected to the first MOS transistor, converts one of the two input signals to a voltage-current, and a drain of the first MOS transistor; And a feedback circuit provided between the gate of the second MOS transistor and negatively feeding back the drain potential of the first MOS transistor to the gate of the second MOS transistor.

According to a second aspect of the present invention, in the mixer circuit according to the first aspect, the feedback circuit includes a regulating amplifier, an inverting input terminal is connected to a drain of the first MOS transistor, and an output terminal is the first circuit. It is connected to the gate of a 2MOS transistor.
According to a third aspect of the present invention, in the mixer circuit according to the second aspect, the regulating amplifier includes a common-source or common-emitter amplifier circuit using a third MOS transistor, and the drain of the first MOS transistor is the third MOS transistor. The third MOS transistor is connected to the gate or base of the transistor, and the drain or emitter of the third MOS transistor is connected to the gate of the second MOS transistor.
According to a fourth aspect of the present invention, in the mixer circuit according to any one of the first to third aspects, the signal input to the voltage-current conversion circuit is single-phase.
According to a fifth aspect of the present invention, in the mixer circuit according to any one of the first to third aspects, two sets of the voltage-current conversion circuits are provided, and a differential input signal is provided to the two sets of voltage-current conversion circuits. Are inputted respectively.

According to a sixth aspect of the present invention, in the mixer circuit according to any of the first to fifth aspects, a frequency band of a signal input to the voltage-current conversion circuit is frequency-converted to a high frequency side.
According to a seventh aspect of the present invention, in the mixer circuit according to any one of the first to fifth aspects, a band of a signal input to the voltage-current conversion circuit is frequency-converted to a low frequency side. .
The invention according to claim 8 has two sets of the mixer circuits according to any one of claims 1 to 7, and inputs the input signals to the voltage-current conversion circuit separately into in-phase components and quadrature components, respectively. The carrier wave, which is the other input signal, is divided into an in-phase component and a quadrature component, respectively, and input to the mixer circuits.
The invention described in claim 9 is a wireless communication apparatus that requires high linearity with respect to a signal for transmitting information, and includes the mixer circuit according to any one of claims 1 to 8.

According to the mixer circuit of the present invention, the input stage has a first MOS transistor that operates in a linear region and converts an input signal to voltage-current, and a second MOS transistor that is cascode-connected to the first MOS transistor, and operates in the linear region. By negatively feeding back the drain of the first MOS transistor or the source of the second MOS transistor to the gate of the second MOS transistor connected in cascode, the linearity of the input signal to the output signal is improved. Furthermore, by reducing the linearity, the cost can be reduced by reducing the channel width of the cascode-connected second MOS transistor and reducing the current consumption associated therewith.
Further, according to the wireless communication apparatus of the present invention, high linearity can be obtained, and cost reduction such as reduction in channel width of the second MOS transistor connected in cascode and reduction in current consumption associated therewith can be realized. Further, it is possible to suppress intermodulation due to various frequency components contained in a narrow band of wireless communication and nonlinearity of the mixer circuit. In order to cancel or avoid the influence of interfering signals due to intermodulation, sophisticated techniques are required for amplification, filtering, etc., but these costs can be reduced and the complexity of the design can be reduced.

Embodiments of the present invention will be described below with reference to the drawings.
FIG. 1 is a diagram illustrating a mixer circuit according to a first embodiment of the present invention, and the mixer circuit according to the first embodiment includes a carrier wave modulation unit 100 and a voltage-current conversion unit 1. . The in-phase component and the differential component of the carrier signal are input to the nodes LO + and LO− of the carrier modulation unit 100, respectively, and the modulated signals are output as differential outputs OUT1 and OUT2 by the load.
The voltage-current conversion unit 1 includes a transistor (first MOS transistor) M1 that operates in a linear region, and a transistor (second MOS transistor) M2 that is cascode-connected to the transistor M1 and that has a drain side connected to the carrier wave modulation unit 100. have.
In the first embodiment, as a feedback circuit that performs negative feedback between the drain of the transistor M1 (or the source of the transistor M2) and the gate of the transistor M2, a negative feedback amplifier (regulating amplifier) having a feedback gain of A. ) There is a feature in that A1 is added.
The negative feedback amplifier A1 has an inverting input terminal connected to the drain of the transistor M1, an output terminal connected to the gate of the transistor M2, and functions as a circuit that negatively feeds back fluctuations in the drain potential of the transistor M1.

式（７）の電流Ｉ₀が流れている状態からＶ_inが△Ｖ_inだけ変化し、それに伴いＶ₁が△Ｖ₁、Ｖ₂がΔＶ₂、Ｉ₀が△Ｉ₀だけ変化したものとすると式（７）から式（８）が得られる。

トランジスタＭ１に関する式（７）及び（８）から△Ｉ₀に関する式（９）が求められる。

This will be specifically described using equations. Expression (7) shows an expression relating to the current I ₀ flowing through the drain of the transistor M1 operating in the linear region in this circuit.

V _in the state in which the current I ₀ of the formula (7) is flowing is changed by △ V _in, and that the accompanying V ₁ is △ V _1, V ₂ it changes [Delta] V _2, I ₀ only △ I ₀ Then, Expression (8) is obtained from Expression (7).

Expression (9) regarding ΔI ₀ is obtained from Expressions (7) and (8) regarding the transistor M1.

さらに、ネガティブフィードバック増幅器Ａ１において、式（１１）のような関係式が成り立つので、この式（１１）を上述の式（１０）に代入すると式（１２）が得られる。

Here, since ΔI ₀ flowing through the transistors M1 and M2 is equal, Expression (10) is established using the transconductances g _m1 and g _m2 of the transistors M1 and M2.

Further, in the negative feedback amplifier A1, since a relational expression such as Expression (11) is established, Expression (12) is obtained by substituting Expression (11) into Expression (10) described above.

式（１３）からトランジスタＭ１のトランスコンダクタンスをより低く、トランジスタＭ２のトランスコンダクタンスをより高く設定することが望ましいことが分かる。そのため、トランジスタＭ２は飽和領域で動作させることが良いことも分かる。
以上説明したように、第１の実施形態のミキサ回路によれば、電圧−電流変換部１にネガティブフィードバック増幅器Ａ１を付加したことにより、トランジスタＭ２のトランスコンダクタンスが（１＋Ａ）倍された形となり、非線形項がより低く調整できるため線形性を高めることが出来る。また、チャネル幅と線形性がトレードオフの関係となり、同じ非線形性を持つ場合においては、チップサイズの縮小及び消費電力の面からも効果的である。 Comparing equations (12) and obtained expression in a conventional manner (5), the denominator of this equation is (1 + A) is multiplied, the variation △ V ₁ of the drain potential of the transistor M1 for correspondingly △ V _in It can be seen that the potency of suppression has increased.
By substituting this equation (12) into the above equation (9), equation (13) is obtained, and the term relating to the square of ΔV _{in in} the equation (13) becomes a nonlinear term.

From equation (13), it can be seen that it is desirable to set the transconductance of the transistor M1 lower and the transconductance of the transistor M2 higher. Therefore, it can be seen that the transistor M2 is preferably operated in the saturation region.
As described above, according to the mixer circuit of the first embodiment, by adding the negative feedback amplifier A1 to the voltage-current converter 1, the transconductance of the transistor M2 is multiplied by (1 + A), Since the nonlinear term can be adjusted lower, the linearity can be improved. In addition, the channel width and linearity are in a trade-off relationship, and in the case of the same nonlinearity, it is effective from the viewpoint of chip size reduction and power consumption.

以上説明したように、第２の実施形態のミキサ回路によれば、フィードバック回路としてソース接地増幅器１１を付加したことによりトランジスタＭ２のトランスコンダクタンスが（１＋ｇ_m3Ｚ）倍された形となり、非線形項がより低く調整できるため線形性を高めることが出来る。特許文献１の構成と比べても比較的少量の素子数で非線形性に対して対応が出来る。また、チャネル幅と線形性がトレードオフの関係となり、同じ非線形性を持つ場合においては、チップサイズの縮小及び消費電力の面からも効果的である。 Next, a mixer circuit according to a second embodiment of the present invention will be described.
FIG. 2 is a diagram showing a mixer circuit according to the second embodiment. The mixer circuit according to the second embodiment shown in FIG. 2 includes a transistor ( The third MOS transistor) is characterized in that a common source amplifier 11 composed of M3 and a load Z is added. The grounded source amplifier 11 functions as a circuit that feeds back the fluctuation of the drain potential of the transistor M1.
The drain potential V ₁ of the transistor M1 is connected to the gate of the transistor M3. The fluctuation of the drain potential V ₁ is propagated to the current I ₁ through the transistor M3. A signal amplified by the load Z appears at the potential V ₂ .
Specifically, it is calculated using an equation. In the common source amplifier 11 composed of the transistor M3 and the load Z, if the transconductance g _m3 of the transistor M3 is used, the relational expression as shown in the equation (14) is established. Therefore, from the first embodiment, g _m3 Z is a gain. It can be seen that this corresponds to A.

As described above, according to the mixer circuit of the second embodiment, by adding the common source amplifier 11 as a feedback circuit, the transconductance of the transistor M2 is multiplied by (1 + g _m3 Z), and the nonlinear term is Since it can be adjusted lower, the linearity can be increased. Compared with the configuration of Patent Document 1, it is possible to cope with nonlinearity with a relatively small number of elements. In addition, the channel width and linearity are in a trade-off relationship, and in the case of the same nonlinearity, it is effective from the viewpoint of chip size reduction and power consumption.

Next, a mixer circuit according to a third embodiment of the present invention will be described.
In the mixer circuit of the third embodiment, voltage-current conversion is performed by a single-phase baseband signal in the mixer circuit of the first or second embodiment shown in FIGS. Called).
In this mixer circuit, the feedthrough from the carrier wave to the output has a drawback. However, in the baseband input stage, the baseband signal is supplied to the modulation unit with the carrier wave by voltage-current conversion. The generated harmonic component of the baseband signal always causes an interference wave at the output of the mixer circuit.
As described above, according to the mixer circuit according to the third embodiment, since the linearity of the baseband signal with respect to the current is important even in the single balance mixer, as in the first or second embodiment. It is important to improve the linearity by performing voltage-current conversion with a cascode circuit to which a negative feedback amplifier is added. Furthermore, interference signals can be suppressed, the chip size can be reduced, and power consumption can be improved.

Next, a mixer circuit according to a fourth embodiment of the present invention will be described.
FIG. 3 is a diagram showing a mixer circuit according to the fourth embodiment. The mixer circuit shown in FIG. 3 includes a carrier wave modulation unit 20 and a voltage-current conversion unit 30, and the carrier wave modulation unit 20 and the voltage-current conversion circuit. The conversion unit 30 has a configuration in which two mixer circuits of the first embodiment are arranged. As a feature, both the carrier wave signal and the baseband signal are differentially input, which is called double balance. FIG. 3 shows the form of the Gilbert cell which is an example of the active type.
Even in the fourth embodiment, the basic frequency conversion method as a mixer does not change. The baseband signal is voltage-to-current converted at the input stage, and the in-phase component and differential component of the carrier signal are input to the nodes LO + and LO-, respectively, and the modulated signals are output as differential outputs OUT1 and OUT2 by the load. The

Compared with a single-balance mixer, a double-balance mixer has a differential pair of transistors M5-M6 and transistors M7-M8 of the carrier modulation unit 20, so that by combining the carrier signals of opposite phases, Provide a good offset. However, the baseband signal conversion unit is not greatly different from the single balance mixer, and its harmonic component directly affects the output.
As described above, according to the mixer circuit of the fourth embodiment, the linearity of the baseband signal with respect to the current is important even in the double balance mixer, but by adding the negative feedback amplifiers A1 and A2, The transconductance of the transistor M2 and the transistor M4 is multiplied by (1 + A), and the nonlinearity can be adjusted lower, so that the linearity can be improved. In addition, the channel width and linearity are in a trade-off relationship, and in the case of the same nonlinearity, it is effective from the viewpoint of chip size reduction and power consumption.

Next, a mixer circuit according to a fifth embodiment of the present invention will be described.
The mixer circuit according to the fifth embodiment is the mixer circuit according to the first to fourth embodiments, and includes a mixer circuit that converts the frequency band of the baseband signal to the high frequency side.
Since this mixer circuit performs up-conversion, it is called an up-conversion mixer. When up-converting, components including harmonics of the carrier can be easily removed by the LPF. However, since the frequency of the baseband signal is set lower than that of the carrier wave, the components related to the harmonics of the baseband signal around the reference carrier frequency cannot be easily removed by a filter or the like. In the intermediate frequency band (see FIG. 6). As described above, ω ₁ in FIG. 6 is the carrier angular frequency, and ω ₂ is the angular frequency of the baseband signal. It can be seen from FIG. 6 that the unnecessary signal surrounded by the broken line is close to the desired signal.
From the above, generation of harmonic components of the baseband signal leads to transmission / reception of unnecessary signals. For this reason, it is important as transmission / reception performance to increase the interference elimination capability so that transmission / reception of a target signal is not hindered by an unintended signal.
As described above, according to the mixer circuit according to the fifth embodiment, since the linearity of the baseband signal with respect to the current is important even in the upconversion mixer, as in the first to fourth embodiments. It is important to improve the linearity by performing voltage-current conversion with a cascode circuit to which a negative feedback amplifier is added. Furthermore, interference signals can be suppressed, the chip size can be reduced, and power consumption can be improved.

Next, a mixer circuit according to a sixth embodiment of the present invention will be described.
The mixer circuit according to the sixth embodiment is the mixer circuit of the first to fourth embodiments, and includes a mixer circuit that converts the frequency of the baseband signal to the low frequency side.
This mixer circuit is called a down conversion mixer because it performs down conversion. When down-converting, components related to the harmonics of the carrier wave and the baseband signal can be easily removed by the LPF. However, the component related to the harmonics of the carrier-baseband signal is not significantly different from that of the intermediate frequency, so it cannot be easily removed by LPF or the like, and further enters the intermediate frequency band of the mixer circuit. There is a risk (see FIG. 7). As described above, ω ₁ in FIG. 7 is the carrier angular frequency, and ω ₂ is the angular frequency of the baseband signal. It can be seen from FIG. 7 that the unnecessary signal surrounded by the broken line is close to the desired signal.
From the above, generation of harmonic components of the baseband signal leads to transmission / reception of unnecessary signals. For this reason, it is important as transmission / reception performance to increase the interference elimination capability so that transmission / reception of a target signal is not hindered by an unintended signal.

As described above, according to the mixer circuit of the sixth embodiment, since the linearity of the baseband signal with respect to the current is important even in the down-conversion mixer, the negative circuit as in the first to fourth embodiments is used. It is important to improve the linearity by performing voltage-current conversion with a cascode circuit to which a feedback amplifier is added.
Furthermore, interference signals can be suppressed, the chip size can be reduced, and power consumption can be improved. However, in this embodiment, since the baseband signal and the carrier wave signal are relatively close to each other, the baseband signal also has a relatively high frequency. Therefore, if the frequency band exceeds the cutoff frequency of the feedback amplifier, the gain reduction or phase delay in the feedback amplifier becomes significant, and the conversion circuit performance cannot be sufficiently achieved. Care should be taken as it can be a bad thing if you cut the performance.

Next, a mixer circuit according to a seventh embodiment of the present invention will be described.
FIG. 4 is a block diagram of a mixer circuit according to the seventh embodiment. In FIG. 4, the first multiplication symbol MIX1 and the second multiplication symbol MIX2 represent the mixer circuits in the first to sixth embodiments. Show. Therefore, in the seventh embodiment, two mixer circuits are arranged. There are roughly four input signals. The carrier signal (ω ₁ ) and the baseband signal (ω ₂ ) and the signals obtained by shifting the respective signals by 90 ° are input to the mixers MIX1 and MIX2.
When each in-phase signal and quadrature signal are input, a relationship such as Equation (15) or Equation (16) is established, and a single sideband (single sideband) is generated. Upper (frequency sum) or lower (frequency difference) is determined by the combination of the carrier wave signal and the baseband signal.

Here, each mixer circuit is frequency-converted by the same conversion method as the above-mentioned mixer, but the mixer circuit of the seventh embodiment includes two mixers. Since the harmonic signals of the carrier signal (ω ₁ ) and the baseband signal (ω ₂ ) are present at the input of each mixer, each is mixed there to produce several harmonics. Finally, the sum of the outputs of the mixer circuits becomes the overall output.
Therefore, in a mixer circuit that outputs a frequency difference, if harmonic components of the baseband signal are generated for both down and up conversion, an interference wave may enter near the desired wave, which is further equivalent to two mixers. Appears as a size.
Similarly, even in a mixer circuit that outputs the sum of frequencies, when the frequency band of the baseband signal is smaller than that of the carrier wave signal, if a harmonic component of the baseband signal is generated, an interference wave may enter the vicinity of the desired wave. , It appears as the size of two more mixers.
Thus, it is understood that the linearity of the baseband signal with respect to the current is very important even in a mixer that generates a single sideband (single sideband).

In particular, in the seventh embodiment, since the sum of the outputs of the two mixer circuits is used as the entire output, sufficient attention must be paid to the generation of harmonics (nonlinearity) due to the baseband signal.
Furthermore, by adding two mixer circuits, the baseband signal input stage in the circuit is doubled. In addition, in order to construct an input stage with sufficient linearity only by a cascode circuit having two transistors, the channel width of the transistor is remarkably increased, and the size (cost) and power consumption of the entire chip are increased. It leads to big deterioration.
As described above, according to the mixer circuit of the seventh embodiment, the linearity of the baseband signal with respect to the current is essential even in a mixer that generates a single sideband. Further, since two mixer circuits are provided, the interference wave and the increase in chip size due to the nonlinearity of the mixer greatly affect the performance of the entire circuit as compared with a single mixer. Therefore, it is important to improve the linearity by performing voltage-current conversion by a cascode circuit to which a negative feedback amplifier is added as in the first to sixth embodiments. Furthermore, the interference signal can be suppressed, the chip size can be reduced, and the power consumption can be improved. The effect is greater than that of a single mixer.

Next, a wireless communication apparatus according to the eighth embodiment of the present invention is described.
The wireless communication apparatus according to the eighth embodiment of the present invention includes the mixer circuit according to the first to seventh embodiments, and is characterized by having high linearity with respect to a signal for transmitting information in telecommunication. is there.
The environment surrounding the wireless communication system is particularly bad in urban areas, and severe restrictions are imposed on the design of wireless terminals. The most severe limitation is that the frequency band assigned to each user is very narrow.
The narrow frequency assigned to each user greatly affects the design of the baseband signal input stage. The transmitter side must perform modulation with less distortion, and the receiver side must be able to remove a strong interference signal even if it is adjacent to the desired signal. Therefore, the linearity of the mixer is very important.

In the first to seventh embodiments, the baseband signal has been described as having only one frequency. However, in the actual environment as described above, signals having various frequencies are transmitted and received in a very narrow frequency band. Sometimes. In such an environment, when two very close frequencies are input as signals when the mixer circuit has nonlinearity, odd-order distortion occurs in the vicinity thereof. This is because a signal that does not originally exist is generated without permission in the circuit, and if there is a desired reception / transmission signal in the vicinity of the interference signal, it is prevented. This is called intermodulation (see FIG. 5).
Therefore, when many frequencies are included in the baseband signal, the linearity of the mixer circuit becomes a big problem in practice. The part for converting the baseband signal into voltage-current is a particularly important device.
As described above, according to the eighth embodiment of the present invention, in a wireless communication apparatus that requires high linearity with respect to a signal for transmitting information in telecommunications, the problem of intermodulation and the like are added, and the voltage − The ability to perform linear operation in current conversion becomes very important. Therefore, it is important to improve the linearity by performing voltage-current conversion with a cascode circuit to which a negative feedback amplifier is added as in the first to seventh embodiments. Furthermore, it is possible to suppress interference signals, reduce the chip size, and increase the efficiency in terms of power consumption.

1 is a diagram illustrating a mixer circuit according to a first embodiment of the present invention. It is the figure which showed the mixer circuit which concerns on the 2nd Embodiment of this invention. It is the figure which showed the mixer circuit which concerns on the 4th Embodiment of this invention. It is a block diagram of the mixer circuit which concerns on the 7th Embodiment of this invention. It is explanatory drawing of the intermodulation in a mixer circuit. It is a figure explaining the problem of the conventional mixer circuit. It is a figure explaining the problem of the conventional mixer circuit. It is the figure which showed an example of the conventional mixer circuit.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1, 10, 30 ... Current conversion part, 11 ... Common source amplifier, 20, 100 ... Carrier wave modulation part, A1, A2 ... Negative feedback amplifier, M1-M8 ... Transistor, MIX ... Mixer

Claims (9)

In a mixer circuit that performs frequency conversion by multiplying two input signals,
An input stage having a first MOS transistor operating in at least one linear region, and at least one second MOS transistor cascode-connected to the first MOS transistor, wherein one of the two input signals is a voltage A voltage-current conversion circuit for current conversion;
A feedback circuit provided between the drain of the first MOS transistor and the gate of the second MOS transistor and negatively feeding back the drain potential of the first MOS transistor to the gate of the second MOS transistor;
A mixer circuit comprising: The mixer circuit according to claim 1,
The feedback circuit includes a regulating amplifier, an inverting input terminal is connected to a drain of the first MOS transistor, and an output terminal is connected to a gate of the second MOS transistor. The mixer circuit according to claim 2, wherein
The regulating amplifier includes a grounded source or grounded emitter amplifier circuit using a third MOS transistor, the drain of the first MOS transistor is connected to the gate or base of the third MOS transistor, and the drain or emitter of the third MOS transistor is the A mixer circuit connected to the gate of a second MOS transistor. The mixer circuit according to any one of claims 1 to 3,
A mixer circuit characterized in that a signal input to the voltage-current conversion circuit has a single phase. The mixer circuit according to any one of claims 1 to 3,
2. A mixer circuit, wherein two sets of the voltage-current conversion circuits are provided, and differential input signals are respectively input to the two sets of voltage-current conversion circuits. The mixer circuit according to any one of claims 1 to 5,
A mixer circuit, wherein a frequency band of a signal input to the voltage-current conversion circuit is converted to a high frequency side. The mixer circuit according to any one of claims 1 to 5,
A mixer circuit, wherein a frequency band of a signal input to the voltage-current conversion circuit is converted to a low frequency side.   8. A carrier wave that has two sets of mixer circuits according to claim 1 and that inputs an input signal to the voltage-current conversion circuit separately into an in-phase component and a quadrature component, and is another input signal. Is divided into an in-phase component and a quadrature component, respectively, and is input to each mixer circuit.   A wireless communication apparatus that requires high linearity with respect to a signal for transmitting information, comprising the mixer circuit according to claim 1.

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