JP2009100545A - Adjustment method for motor controller and adjustment device thereof - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a method and a device for adjusting a motor controller in order to correctly acquire the rotation position of a rotor without depending on the difference in gain of current detecting means. <P>SOLUTION: A current is supplied to a three-phase stator winding, thereby directing the pole of a rotor in a direction of forming an electric angle of 90° for a U-phase (S2). A high-frequency voltage vector is applied in this state (S3), a magnitude M<SB>U</SB>of current response when the high frequency voltage vector is directed to the U-phase is obtained (S4). Then, a current is applied to the three-phase stator winding, thereby directing the pole of the rotor to a direction forming an electric angle of 90° for a V-phase (S5). A high-frequency voltage vector is applied in this state (S6), and a magnitude M<SB>V</SB>of current response when the high-frequency voltage vector is directed to the V-phase is obtained (S7). A correction coefficient χ<SB>V</SB>is obtained on the basis of the magnitudes M<SB>U</SB>, M<SB>V</SB>of the current response (S8), and the obtained correction coefficient is written into the motor controller (S9). <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a method and apparatus for adjusting a motor control device for sensorless driving of a brushless motor. The brushless motor is used, for example, as a source for generating a steering assist force in an electric power steering apparatus.

  A motor control device for driving and controlling a brushless DC motor is generally configured to control the supply of motor current according to the output of a position sensor for detecting the rotational position of the rotor. However, the environmental resistance of the position sensor becomes a problem, and the expensive position sensor and the wiring associated therewith hinder the cost reduction and the size reduction. Therefore, a sensorless driving method for driving a brushless DC motor without using a position sensor has been proposed. The sensorless driving method is a method for estimating the phase of the magnetic pole (electrical angle of the rotor) by estimating the induced voltage accompanying the rotation of the rotor.

  Since the induced voltage cannot be estimated when the rotor is stopped and when rotating at a very low speed, the phase of the magnetic pole is estimated by another method. Specifically, as shown in FIG. 4A, a high-frequency voltage vector that rotates along the rotation direction of the rotor 50 around the origin of αβ coordinates that are two-phase fixed coordinates with the rotation center of the rotor 50 as the origin. A high frequency exploration voltage is applied to the U, V, W phase stator windings 51, 52, 53 so that a constant size is formed. The high-frequency voltage vector is a voltage vector that rotates at a sufficiently high speed with respect to the rotation speed of the rotor 50. With the application of the high-frequency voltage vector, a current flows through the U, V, and W-phase stator windings 51, 52, and 53. A current vector representing the magnitude and direction of the three-phase current on the αβ coordinate rotates around the origin.

  The inductance of the rotor 50 takes different values for the d-axis, which is the magnetic pole axis along the magnetic flux direction, and the q-axis (axis along the torque direction) orthogonal thereto. For this reason, the magnitude of the current vector is large in the direction close to the d-axis, and is small in the direction close to the q-axis. As a result, as shown in FIG. 4B, the end point of the current vector draws an elliptical locus 55 having the major axis in the d-axis direction of the rotor 50 on the αβ coordinates.

  Therefore, the magnitude of the current vector has a maximum value in the N-pole direction and the S-pole direction of the rotor 50. That is, as shown in FIG. 5A, the magnitude of the current vector has two maximum values in one cycle. In this case, if the magnitude of the voltage vector is sufficiently large, the inductance of the N pole side of the rotor 50 becomes smaller than that of the S pole side due to the influence of the magnetic saturation of the stator, and the magnitude of the current vector in the N pole direction. Takes the maximum value (see curve L1).

Therefore, a sufficiently large high-frequency voltage vector is applied to specify the maximum of the current vector corresponding to the N pole, and thereafter, a high-frequency voltage vector having a reduced size is applied, based on the maximum value of the current vector. The phase of the rotor 50 can be estimated. More specifically, the phase angle (electrical angle) θ of the rotor 50 becomes θ = Tan ⁻¹ (I by the α-axis component I _α and the β-axis component I _β of the current vector when the magnitude is maximum. _β / I _α ).

In order to obtain the current vector, a U-phase current sensor and a V-phase current sensor are provided for detecting a U-phase current and a V-phase current flowing in two phases of the brushless motor, for example, the U-phase and V-phase stator windings, respectively. The W-phase current can be obtained by calculation from the U-phase current and the V-phase current.
JP 2007-97263 A

However, since the U-phase current sensor and the V-phase current sensor are separate elements, there is a difference in gain between these sensors. This difference in gain affects the major axis direction of the ellipse drawn by the end point of the current vector on the αβ coordinate plane. This problem will be described with reference to FIG.
First, it is assumed that a high frequency voltage vector is applied without the rotor 50. Since the magnitude of the high-frequency voltage vector is constant, the locus of the end point draws a circle 80 centered on the origin on the αβ coordinate plane. In the situation where the rotor 50 does not exist, since it is not affected by the saliency, the magnitude of the current vector is constant, and the locus of the end point should draw a circle centered on the origin on the αβ coordinate plane. However, in reality, the end point of the current vector will draw an ellipse 81 due to the gain difference between the U-phase current sensor and the V-phase current sensor.

On the other hand, when the rotor 50 is present and there is no difference in gain between the U-phase current sensor and the V-phase current sensor, it is assumed that the end point of the current vector draws an ellipse 82. The major axis direction of the ellipse 82 corresponds to the true pole position of the rotor 50.
The ellipse 83 is obtained by multiplying the ellipse 82 (the true current vector locus) by the ellipse 81 (distortion by the current sensor). This ellipse 83 corresponds to the locus of the end point of the actually obtained current vector. In this case, the major axis direction of the ellipse 83 is shifted from the major axis direction of the ellipse 82 due to the influence of the gain difference between the U-phase current sensor and the V-phase current sensor. The major axis direction of the ellipse 83 is calculated as an incorrect pole position.

As described above, there is a problem that the rotational position of the rotor 50 cannot be obtained accurately due to the difference in gain between the two-phase current sensors.
SUMMARY OF THE INVENTION An object of the present invention is to provide a method and apparatus for adjusting a motor control device so that the rotational position of a rotor can be obtained accurately regardless of the difference in gain of current detection means.

  In order to achieve the above object, an invention according to claim 1 is an electric motor comprising a rotor (50) as a field magnet and three-phase stator windings (51, 52, 53) of U phase, V phase and W phase. A method for adjusting a motor control device (5) for driving a motor (3), wherein the motor control device supplies a current of a first phase and a second phase, which are two phases of the three phases, respectively. The electric motor is driven based on the outputs of the first and second current detecting means (9U, 9V) to be detected, and the method includes a rotor in a direction that forms an electrical angle of 90 degrees with respect to the first phase. A first rotor position control step (S2) for supplying current to the three-phase stator winding so that the poles of the three-phase stator are directed, and after the first rotor position control step, are applied to the three-phase stator windings, respectively. Voltage vector represented by the voltage During the execution of the first exploration voltage application step (S3) for applying an exploration voltage to the stator winding so as to rotate at a predetermined cycle while keeping the size constant, and the first exploration voltage application step Obtaining a first current detection value output by the first current detection means when the voltage vector is directed to the first phase (S4); and an electrical angle of 90 degrees with respect to the second phase. A second rotor position control step (S5) for supplying a current to the three-phase stator winding so that the rotor poles are oriented in the direction formed, and after the second rotor position control step, the three-phase stator winding A second search voltage applying step (S6) for applying a search voltage to the stator winding so that the voltage vectors represented by the voltages respectively applied to are rotated at a predetermined period while maintaining a constant magnitude. And said (2) acquiring a second current detection value output by the second current detection means when the voltage vector is directed to the second phase during execution of the two exploration voltage application steps; A correction coefficient setting step (S8) for determining a correction coefficient for correcting at least one of the outputs of the first and second current detection means so as to equalize the second current detection value, and setting the correction coefficient in the motor control device. , S9), and a method for adjusting the motor control device. The alphanumeric characters in parentheses indicate corresponding components in the embodiments described later. The same applies hereinafter.

  The invention described in claim 2 is for driving an electric motor (3) including a rotor (50) as a magnetic field and three-phase stator windings (51, 52, 53) of U phase, V phase and W phase. The apparatus (60) for adjusting the motor control device (5) of the first, wherein the motor control device detects first and second phase currents which are two phases of the three phases, respectively. The electric motor is driven based on the output of the second current detection means (9U, 9V), and the apparatus is arranged such that the rotor poles are oriented in a direction that forms an electrical angle of 90 degrees with respect to the first phase. First rotor position control means (S2) for supplying current to the three-phase stator windings, and with the rotor poles oriented in a direction forming an electrical angle of 90 degrees with respect to the first phase, Represented by the voltage applied to each of the three-phase stator windings The first exploration voltage applying means (S3) for applying the exploration voltage to the stator winding and the first exploration voltage applying means so that the pressure vector keeps its magnitude constant and rotates at a predetermined cycle. Means (S4) for obtaining a first current detection value output by the first current detection means when the voltage vector is directed to the first phase in a state in which the exploration voltage is applied; Second rotor position control means (S5) for supplying current to the three-phase stator winding so that the rotor poles are oriented in a direction that makes an electrical angle of 90 degrees with respect to the phase; The voltage vector represented by the voltage applied to each of the three-phase stator windings in a state where the rotor poles are oriented in the direction of 90 degrees in electrical angle, the magnitude of the voltage vector is kept constant and a predetermined period The stator windings to rotate at The second exploration voltage applying means (S6) for applying the exploration voltage and the second exploration voltage applying means applying the exploration voltage to the second phase when the voltage vector is directed to the second phase. At least one of the outputs of the first and second current detection means so that the second current detection value output by the two current detection means (S7) and the first and second current detection values are equal. An adjustment device for a motor control device, including correction coefficient setting means (S8, S9) for determining a correction coefficient for correcting one of the two and setting the correction coefficient in the motor control device.

  According to these inventions, by supplying current to the three-phase stator winding at a certain phase, the rotor poles are directed in a direction that forms an electrical angle of 90 degrees with respect to the first phase. In this state, a search voltage is applied to the stator winding. Then, the output of the first current detection means when the voltage vector faces the first phase is acquired as the first current detection value. On the other hand, by supplying a current having a different fixed phase to the three-phase stator winding, the rotor poles are oriented in a direction that makes an electrical angle of 90 degrees with respect to the second phase. In this state, a search voltage is applied to the stator winding. Then, the output of the second current detection means when the voltage vector faces the second phase is acquired as the second current detection value.

  In an ideal state where there is no difference between the gains of the first and second current detection means, the first and second current detection values should match, but in reality, since there is a difference in gain, the first and second current detection values are the same. The two current detection values are inconsistent. Therefore, if a correction coefficient is determined so that the first and second current detection values are equal, the gain difference is compensated by applying this correction coefficient to the output of the first and / or second current detection means. be able to.

As a result, by using the output signals of the first and second current detection means corrected by the correction coefficient, the motor control device can accurately determine the rotational position of the rotor.
Since it is necessary to supply current to the stator winding and direct the rotor pole in a desired direction, the adjustment as described above is performed in a motor load state (preferably with the rotor direction substantially matching the current phase). Is preferably performed with the rotor free.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.
FIG. 1 is a block diagram for explaining an electrical configuration of an electric power steering apparatus including a motor control apparatus to which an adjustment method and an adjustment apparatus according to an embodiment of the present invention are applied. The electric power steering apparatus includes a torque sensor 1 that detects a steering torque applied to a steering wheel of a vehicle, an electric motor 3 that applies a steering assist force to a steering mechanism 2 of the vehicle, and a motor control that drives and controls the electric motor 3. And a device 5. The motor control device 5 drives the electric motor 3 in accordance with the steering torque detected by the torque sensor 1, thereby realizing appropriate steering assistance according to the steering situation.

  In this embodiment, the electric motor 3 is a three-phase brushless DC motor. As shown schematically in FIG. 2, the electric motor 3 has a rotor 50 as a field, and stator windings 51 of U phase, V phase and W phase, 52, 53. The electric motor 3 may be an inner rotor type in which a stator is disposed outside the rotor, or may be an outer rotor type in which a stator is disposed inside a cylindrical rotor.

The motor control device 5 includes a microcomputer 7, a drive circuit (inverter circuit) 8 that is controlled by the microcomputer 7 and supplies electric power to the electric motor 3, and U-phase and V-phase stator windings of the electric motor 3. A U-phase current sensor 9U and a V-phase current sensor 9V that detect currents flowing in 51 and 52 are provided.
The microcomputer 7 includes a CPU and a memory (such as a ROM and a RAM), and functions as a plurality of function processing units by executing a predetermined program. The plurality of function processing units include a current command value generation unit 11, a PI (proportional integration) control unit 12, an instruction voltage generation unit 13, a γδ / αβ coordinate conversion unit 14, and an αβ / UVW coordinate conversion unit 15. A PWM control unit 16, a UVW / αβ coordinate conversion unit 17, an αβ / γδ coordinate conversion unit 18, a deviation calculation unit 19, a position estimation unit 20, a sensing signal generation unit 21, an addition unit 22, A gain correction unit 40 and a W-phase current calculation unit 41 are included.

The current command value generation unit 11 generates a command value I _d ^* of a d-axis current component along the rotor magnetic pole direction of the electric motor 3 and a command value I _q ^* of a q-axis current component orthogonal to the d axis. Hereinafter, these are collectively referred to as “current command value I _dq ”. However, the dq coordinate plane is a plane along the rotation direction of the rotor 50, and the d axis and the q axis define a two-phase rotation coordinate system (dq) that rotates together with the rotor 50 (see FIG. 2).

When the current command value I ^* representing the amplitude of the current (sine wave current) to be applied to the U phase, V phase and W phase of the electric motor 3 is used, the d axis current command value I _d ^* and the q axis current command value I _q ^* Is expressed by the following equations (1) and (2).
I _d ^* = 0 (1)
I _q ^* = − (3/2) ^1/2・ I ^* (2)
Therefore, the current command value generation unit 11 generates the d-axis current command value I _d ^* = 0, while generating the q-axis current command value I _q ^* corresponding to the steering torque detected by the torque sensor 1. More specifically, the q-axis current command value I _q ^* may be generated using a map (table) that stores the q-axis current command value I _q ^* corresponding to the steering torque. Since the torque generated by the electric motor 3 corresponds to the motor current, the current command value I _dq can be rephrased as a “torque command value” for commanding the torque to be generated from the electric motor 3.

A U-phase current sensor 9U and a V-phase current sensor 9V are provided to detect the phase currents flowing in the U-phase and V-phase stator windings 51 and 52, respectively. The output signal of the U-phase current sensor 9U is input to the A / D conversion port 10U of the microcomputer 7 and converted into a U-phase detection current value I _U represented by a digital signal. The output signal of the V-phase current sensor 9V is input to the A / D conversion port 10V of the microcomputer 7 and converted into a V-phase detection current value I _V represented by a digital signal.

The gain correction unit 40 multiplies the V-phase detection current value I _V from the A / D conversion port 10V by the correction coefficient χ _V. Thereby, the gain correction unit 40 outputs the corrected V-phase detection current value I _V. The correction coefficient χ _V is a coefficient for compensating for a gain difference caused by an individual difference between the two current sensors 9U and 9V. That is, by multiplying the V-phase detection current value I _V by the correction coefficient χ _V, a detection result equivalent to that when each current of the U-phase and the V-phase is detected by a current sensor having an equal gain is obtained. . The correction coefficient χ _V is stored in a memory 7M (rewritable memory, for example, a rewritable nonvolatile memory) provided in the microcomputer 7.

W-phase current calculator 41, based on the U-phase detected current values I _U and corrected by the gain correction unit 40 V-phase detected current value I _V from the A / D conversion port 10 U, W-phase detected current value I _W (= 0−I _U −I _V ) is obtained. Thus, the three-phase detection current values I _U , I _V , and I _W are obtained (hereinafter referred to as “three-phase detection current value I _UVW ” when collectively referred to). The three-phase detection current value I _UVW is input to the UVW / αβ coordinate conversion unit 17.

The UVW / αβ coordinate conversion unit 17 converts the three-phase detection current value I _UVW into the currents I _α and I _β on the two-phase fixed coordinate system (α-β) (hereinafter referred to as “two-phase detection current”). The coordinates are converted to “I _αβ ”. The two-phase fixed coordinate system (α-β) is a fixed coordinate system in which the rotation center of the rotor 50 is the origin and the α axis and the β axis orthogonal to the rotation axis of the rotor 50 are defined (see FIG. 2). ). The coordinate-converted two-phase detection current I _αβ is given to the αβ / γδ coordinate conversion unit 18.

The αβ / γδ coordinate conversion unit 18 converts the two-phase detection current I _αβ into a two-phase rotation coordinate system (hereinafter referred to as “estimated rotation position θ ^”) estimated by the position estimation unit 20. Coordinates are converted into currents I _γ and I _δ on γ−δ) (hereinafter, collectively referred to as “two-phase detection current I _γδ ”). The two-phase rotational coordinate system (γ−δ) is a rotational coordinate system defined by a γ axis along the rotor magnetic pole direction and a δ axis orthogonal to the γ axis when the rotor 50 is at the estimated rotational position θ ^. (See FIG. 2). When the estimated rotational position θ ^ has no error and coincides with the actual rotor rotational position, the two-phase rotational coordinate system (dq) and the two-phase rotational coordinate system (γ-δ) coincide.

The two-phase detection current I _γδ is supplied to the deviation calculation unit 19. The deviation calculator 19 calculates a deviation of the γ-axis current I _{γ with} respect to the d-axis current command value I _d ^* and a deviation of the δ-axis current I _{δ with} respect to the q-axis current command value I _q ^* . These deviations are given to the PI control unit 12 and are each subjected to PI calculation processing. Then, in accordance with these calculation results, the command voltage generator 13 causes the γ-axis command voltage V _γ ^* and the δ-axis command voltage V _δ ^* (hereinafter referred to as “two-phase command voltage V _γδ ” when collectively referred to). ) Is generated and provided to the γδ / αβ coordinate converter 14.

The γδ / αβ coordinate converter 14 converts the γ-axis command voltage V _γ ^* and the δ-axis command voltage V _δ ^* into α-axis command voltages V _α ^* and β that are command voltages of the two-phase fixed coordinate system (α-β). Coordinates are converted to an axis command voltage V _β ^* (hereinafter, collectively referred to as “two-phase command voltage V _αβ ”). The two-phase instruction voltage V _αβ is given to the αβ / UVW coordinate conversion unit 15.
The αβ / UVW coordinate converter 15 converts the α-axis command voltage V _α ^* and the β-axis command voltage V _β ^* into the command voltages of the three-phase fixed coordinate system, that is, the command voltages V _U ^{* of the} U phase, the V phase, and the W phase ^. , V _V ^* , V _W ^* (hereinafter collectively referred to as “three-phase indicating voltage V _UVW ”).

The PWM control unit 16 generates a drive signal having a duty ratio controlled according to the three-phase instruction voltages V _U ^* , V _V ^* , and V _W ^* and supplies the drive signal to the drive circuit 8. Thus, a voltage is applied to each phase of the electric motor 3 with a duty ratio corresponding to the instruction voltages V _U ^* , V _V ^* , and V _W ^* of the corresponding phase.
With such a configuration, when a steering torque is applied to a steering wheel (not shown) as an operation member coupled to the steering mechanism 2, this is detected by the torque sensor 1. Then, the current command value generation unit 11 generates a current command value I _dq corresponding to the detected steering torque. A deviation between the current command value I _dq and the two-phase detection current I _γδ is obtained by the deviation calculation unit 19, and PI calculation is performed by the PI control unit 12 so as to lead this deviation to zero. A two-phase command voltage V _γδ corresponding to the calculation result is generated by the command voltage generation unit 13, and this is converted into a three-phase command voltage V _UVW through the coordinate conversion units 14 and 15. Then, the drive circuit 8 operates with a duty ratio corresponding to the three-phase instruction voltage V _UVW by the action of the PWM control unit 16, whereby the electric motor 3 is driven, and an assist torque corresponding to the current command value I _dq is generated. The steering mechanism 2 will be given. Thus, steering assistance can be performed according to the steering torque. The three-phase detection current I _UVW is converted into a two-phase detection current I _γδ expressed by a two-phase rotation coordinate system (γ-δ) so as to correspond to the current command value I _dq through the coordinate conversion units 17 and 18. Is then given to the deviation calculator 19.

In order to perform coordinate conversion between the rotating coordinate system and the fixed coordinate system, a phase angle (electrical angle) θ representing the rotational position of the rotor 50 is required. An estimated rotational position θ ^ representing this phase angle is generated by the position estimation unit 20 and is provided to the γδ / αβ coordinate conversion unit 14 and the αβ / γδ coordinate conversion unit 18.
The sensing signal generator 21 functions as a search voltage application unit that applies a search voltage to the electric motor 3 in order to estimate the phase angle θ of the rotor 50. The sensing signal generation unit 21 uses a high-frequency sine voltage (see FIG. 5B) having a sufficiently high frequency (for example, 400 Hz) as compared with the rated frequency of the electric motor 3 as a search voltage, and the U of the electric motor 3. Sensing signals to be applied to the phase, V-phase, and W-phase stator windings 51, 52, 53 are generated. More specifically, V-W phase energization, W-U phase energization, and U-V phase energization are sequentially repeated by applying a high-frequency voltage having a duty ratio that does not cause the rotor 50 to rotate. A sensing signal that can apply a high-frequency voltage vector that rotates spatially around the rotation center of the rotor 50 is generated. This high-frequency voltage vector is a voltage vector having a constant magnitude that rotates at a constant speed around the origin of αβ coordinates, which are fixed coordinates with the rotation center of the rotor 50 as the origin (see FIG. 4A).

In this embodiment, the sensing signal generation unit 21 generates a sensing signal to be superimposed on the two-phase instruction voltage V _αβ . This sensing signal is superimposed on the two-phase instruction voltage V _αβ by the adder 22.
The position estimation unit 20 estimates the rotational position of the rotor 50 based on the two-phase detection current I _αβ and generates an estimated rotational position θ ^.

FIG. 3 is a block diagram for explaining the configuration of the position estimation unit 20. The position estimation unit 20 includes a current value calculation unit 26, an inverse notch filter 27, a current peak timing extraction unit 28, a current synchronization extraction unit 29, and a rotor position calculation unit 30.
The current value calculator 26 obtains the magnitude of the current vector, that is, the magnitude I of the current. More specifically, the current value calculation unit 26 calculates the current magnitude I based on the two-phase detection current I _αβ given from the UVW / αβ coordinate conversion unit 17. For example, the current magnitude i is calculated according to the following equation (3).

I = {I _α ² + I _β ² } ^1/2 (3)
The inverse notch filter 27 removes noise components from the output of the current value calculator 26. The current magnitude I has two local maxima during one period of the high-frequency voltage vector. Therefore, the frequency of the significant signal component in the output of the current value calculation unit 26 has a frequency about twice the frequency of the sensing signal generated by the sensing signal generation unit 21. Therefore, the reverse notch filter 27 has a characteristic (reverse notch filter characteristic) that allows a signal in a predetermined band having a center frequency fc to be twice the frequency of the sensing signal to pass, and removes signals outside this range. Yes.

  Based on the output of the inverse notch filter 27, the current peak timing extraction unit 28 detects the timing (maximum value timing) at which the current magnitude I obtained by the current value calculation unit 26 takes the maximum value (peak value). More specifically, the current peak timing extraction unit 28 performs peak hold processing for each predetermined sampling period, and extracts a maximum value timing at which the current magnitude I takes a maximum value. This local maximum timing corresponds to the rotational position (phase) of the rotor 50.

When the current peak timing extraction unit 28 generates the extremum timing t, the current synchronization extraction unit 29 responds to the two-phase detection current I _αβ (t) at the timing t from the UVW / αβ coordinate conversion unit 17. Import and output.
The rotor position calculation unit 30 uses the extremum timing two-phase detection current I _αβ (t) given from the current synchronization extraction unit 29 to calculate the estimated rotation position (rotation angle) θ ^ of the rotor according to the following equation (4). Ask.

θ = Tan ⁻¹ (I _β / I _α ) (4)
4A shows a high-frequency voltage vector corresponding to the rotation search voltage generated by the sensing signal generator 21 and applied to the stator windings 51, 52, and 53, and FIG. 4B shows the high-frequency voltage vector. Shows the response of the current vector to. The high-frequency voltage vector formed by applying the rotation exploration voltage has a constant magnitude and rotates at a constant speed around the origin of the αβ coordinate. At this time, the magnitude of the current vector changes according to the pole position of the rotor 50. More specifically, the magnitude of the current vector, that is, the magnitude I of the current takes a maximum value at positions corresponding to the N pole and the S pole of the rotor 50, and the electrical angle with respect to them differs by two by 90 degrees. The current magnitude I takes a minimum value at the position. As a result, the end point of the current vector forms an elliptical locus 55 around the origin of the αβ coordinate. The ellipse has a major axis direction corresponding to the N pole and S pole of the rotor 50.

  While the rotor 50 is stopped or rotating at a low speed, as shown in FIG. 5 (a), the current waveform (the waveform indicating the time change of the current magnitude I. The curve L2 in FIG. 5) shows the high-frequency voltage vector. In one period, local maximum points P1 and P2 corresponding to two major axis directions of an elliptical locus 55 (see FIG. 4B) formed by the end point of the current vector appear. If the process (N pole determination process) for specifying the maximum point corresponding to the N pole of the rotor 50 among the two maximum points P1 and P2 is performed in advance, the timing of the specified maximum point is extracted. Thus, the rotational position of the rotor 50 can be obtained. A known method can be applied to the N pole determination. For example, when a sufficiently large high-frequency voltage vector is applied, due to the magnetic saturation of the stator, the inductance on the N pole side of the rotor 50 is smaller than that on the S pole side, and the magnitude of the current vector in the N pole direction is maximum. The value is taken (see curve L1 in FIG. 5 (a)). By utilizing this, one of the two maximum points P1, P2 of the current waveform can be specified as the maximum point corresponding to the N pole.

In the production line for the electric power steering apparatus, adjustment for obtaining the correction coefficient χ _V is performed in a free state in which the electric motor 3 is not substantially loaded. The free state means a state where the phase of the rotor 50 substantially matches the current phase when a current is passed through the stator windings 51, 52, 53. For example, the electric motor 3 is coupled to the steering mechanism 2. It is the previous state.

As shown in FIG. 1, an adjustment device 60 for obtaining the correction coefficient χ _V can be connected to the motor control device 5. The adjusting device 60 has a basic configuration as a computer. The adjustment device 60 can give a voltage command value to the PWM control unit 16 via the line 61, can control the sensing signal generation unit 21 via the line 62, and can be connected to the A / D via the line 63. Capable of acquiring U-phase detection current value I _U and V-phase detection current value I _V from conversion ports 10U, 10V, and further writing correction coefficient χ _V into memory 7M via line 64 It is.

FIG. 6 is a flowchart for explaining the procedure for setting the correction coefficient χ _V by the adjustment device 60.
First, as described above, the electric motor 3 is set to a free state (step S1).
In this state, the adjusting device 60 sets a voltage command value for causing a current to flow through the stator windings 51, 52, and 53 so that the N pole of the rotor 50 is oriented in a direction that forms an electrical angle of 90 degrees with respect to the U phase. Generate (step S2). Thereby, the N pole of the rotor 50 is oriented in a direction that forms an electrical angle of 90 degrees with respect to the U phase, and the rotor 50 is stationary in that state. After the rotor 50 is stationary, the adjusting device 60 stops generating the voltage command value.

Next, a sensing signal generation command is given from the adjustment device 60 to the sensing signal generation unit 21. As a result, a voltage for forming a high-frequency voltage vector (constant magnitude) that rotates around the rotation center of the rotor 50 is applied to the stator windings 51, 52, and 53 (step S3).
In this state, the adjustment device 60 monitors the outputs of the A / D conversion ports 10U and 10V (that is, the raw outputs of the current sensors 9U and 9V), and the current response when the high-frequency voltage vector faces the U phase. The size _MU is acquired (step S4). Size M _U of current response, as represented by the following formula (5), corresponding to the U-phase current. Therefore, instead of obtaining the magnitude M _U of current response may be determined the magnitude of the U-phase detected current I _U.

M _U = {I _U ² + I _V ² + I _W ² } ^1/2 = {I _U ² + I _V ² + (0−I _U −I _V ) ² } ^1/2
= {I _U ² + (0−I _U ) ² } ^1/2 ∵I _V = 0
= √2 | I _U | …… (5)
Next, the adjustment device 60 stops the generation of the sensing signal from the sensing signal generation unit 21 and the stator winding 51 so that the N pole of the rotor 50 is oriented in a direction that forms an electrical angle of 90 degrees with respect to the V phase. , 52, and 53, a voltage command value for causing current to flow is generated (step S5). Thereby, the N pole of the rotor 50 is oriented in a direction that forms an electrical angle of 90 degrees with respect to the V phase, and the rotor 50 is stationary in this state. After the rotor 50 is stationary, the adjusting device 60 stops generating the voltage command value.

Next, a sensing signal generation command is given from the adjustment device 60 to the sensing signal generation unit 21. As a result, a voltage for forming a high-frequency voltage vector (constant magnitude) rotating around the rotation center of the rotor 50 is applied to the stator windings 51, 52, 53 (step S6).
In this state, the adjustment device 60 monitors the outputs of the A / D conversion ports 10U and 10V (that is, the raw outputs of the current sensors 9U and 9V), and the magnitude of the current response when the voltage vector faces the V phase. It is to obtain the M _V (step S7). Size M _V of the current response, as represented by the following formula (6), corresponding to the V phase current. Therefore, instead of obtaining the magnitude M _V of the current response, the magnitude of the V-phase detection current I _V may be obtained.

M _V = {I _U ² + I _V ² + I _W ² } ^1/2 = {I _U ² + I _V ² + (0−I _U −I _V ) ² } ^1/2
= {I _V ² + (0−I _V ) ² } ^1/2 ∵I _U = 0
= √2 | I _V | …… (6)
Next, the adjusting device 60 calculates the correction coefficient χ _V according to the following equation (7) (step S8).
χ _V = M _U / M _V = | I _U | / | I _V | (7)
Then, the adjusting device 60 writes the obtained correction value χ _V in the memory 7M of the microcomputer 7 (step S9).

When the adjustment is thus completed, the adjustment device 60 is removed from the motor control device 5.
Two current sensors 9U, in an ideal state gain equal 9V, since the size M _U and M _V of the current response are equal, the correction coefficient χ _V = 1. When the gains of the two current sensors 9U and 9V are different, the correction coefficient χ _V ≠ 1. In either case, M _U = χ _V · M _V.

Accordingly, by using the chi _V · I _V multiplied by the correction coefficient chi _V to V-phase detected current I _V as V-phase detected current I _V after correction, two current sensors 9U, compensate for the difference in gain of 9V can do. That is, when a sensing signal is generated from the sensing signal generator 21 and a rotating high-frequency voltage vector is applied, if there is no rotor 50, the current vector circulates around the coordinate origin with a constant magnitude.

Therefore, by using the corrected V-phase detection current I _V , the position calculation of the rotor 50 by the position estimation unit 20 can be performed accurately, and as a result, the control accuracy of the electric motor 3 is improved, This can contribute to an improvement in steering feeling.
As mentioned above, although one Embodiment of this invention was described, this invention can also be implemented with another form. For example, in the above-described embodiment, the V-phase detection current I _V is corrected with the correction coefficient χ _V , but the U-phase detection current is corrected with the correction coefficient χ _U (= 1 / χ _V ). also, to obtain the same effect, also, as multiplied by χ _U · M _U = correction factor determined such that χ _V · M _V χ _U, detected current values chi _V respectively I _U and I _V The same effect can be obtained. Of course, the combination of the two phases for detecting the phase current may be a combination of the U phase and the W phase, or a combination of the V phase and the W phase.

Further, instead of calculating the W-phase current I _W by the W-phase current calculation unit 41, a W-phase current sensor for detecting the W-phase current I _W may be provided. In this case, the adjusting device 60 uses a voltage command value for causing a current to flow through the stator windings 51, 52, and 53 so that the N pole of the rotor 50 is oriented in a direction that forms an electrical angle of 90 degrees with respect to the W phase. Is generated. As a result, the N pole of the rotor 50 is oriented in a direction that forms an electrical angle of 90 degrees with respect to the W phase, and the rotor 50 is stationary in this state. After the rotor 50 is stationary, the adjusting device 60 stops generating the voltage command value. Thereafter, a sensing signal generation command is given from the adjustment device 60 to the sensing signal generation unit 21. As a result, a voltage for forming a high-frequency voltage vector (constant magnitude) that rotates around the rotation center of the rotor 50 is applied to the stator windings 51, 52, and 53. In this state, the adjusting device 60 monitors the output of the W-phase current sensor, and acquires the magnitude M _W of current response when the voltage vector facing W phase. Size M _W of the current response becomes as the following equation (8).

M _W = √2 | I _W | (8)
Therefore, adjustment device 60 obtains correction coefficient χ _W for the W-phase detection current according to the following equation (9).
χ _W = M _U / M _W (9)
If this correction coefficient χ _W is multiplied by the W-phase detection current I _W , the gain difference between the U-phase current sensor and the W-phase current sensor can be compensated. Of _{course, χ U · M U = χ} V · M V = χ W · M W satisfies the correction coefficient χ _U, χ _V, χ _W in each phase of the detected current value may be corrected respectively.

In the motor control device 5 described above, the rotational position θ of the rotor 50 is obtained using the two-phase detection current I _αβ when the current magnitude I takes the maximum value. However, the current magnitude I is the maximum. The estimated rotational position θ ^ may be obtained according to the following equation (10) using the two-phase indicating voltages V _α and V _β when taking the values (see the line 23 in FIG. 1).
θ ^ = Tan ⁻¹ (V _β / V _α ) (10)
Since the magnitude of the voltage vector is constant, the phase can be easily calculated as compared with the current vector in which distortion occurs. Therefore, the calculation process can be simplified by applying the expression (10).

  In order to obtain the phase of the voltage vector when the current magnitude I takes the maximum value, a counter that performs a counting operation in synchronization with the application of the high-frequency voltage vector is used instead of performing the calculation according to the above equation (10). You may do it. Specifically, a counter that is repeatedly operated so as to be initialized and to start a counting operation when the high-frequency voltage vector is along the α-axis (coincident with the U-phase direction) (that is, when the phase of the high-frequency voltage vector is zero). Is provided. For example, this counter counts up every cycle T / n obtained by dividing the cycle T of the high-frequency voltage vector into n equal parts (n is the number of samplings per cycle. For example, n = 360). Represents the phase of the vector. Therefore, as shown in FIG. 5 (c), when the count value of the counter is referred to when the extreme value of the current magnitude I is detected, this count value is obtained as the magnetic pole position of the rotor 50 (the magnitude of the current vector). Represents the phase angle of the high-frequency voltage vector when is maximum.

In the above-described embodiment, the example in which the present invention is applied to the electric motor 3 as a drive source of the electric power steering apparatus has been described. However, the present invention controls an electric motor for uses other than the electric power steering apparatus. The present invention can also be applied to control device adjustment.
In addition, various design changes can be made within the scope of matters described in the claims.

It is a block diagram for demonstrating the electrical structure of the electric power steering apparatus provided with the motor control apparatus with which the adjustment method and adjustment apparatus which concern on one Embodiment of this invention are applied. It is an illustration figure for demonstrating the structure of an electric motor. It is a block diagram for demonstrating the structure of a position estimation part. FIG. 4A shows a high-frequency voltage vector corresponding to the rotation exploration voltage, and FIG. 4B shows a current vector response to the high-frequency voltage vector. FIG. 5A shows an example of a current waveform, FIG. 5B shows an example of a voltage waveform, and FIG. 5C shows a change in the count value of the counter for detecting the phase of the voltage vector. It is a flowchart for explaining the procedure for setting the correction coefficient chi _V. It is a figure for demonstrating that a position detection error arises when the gains of a U-phase current sensor and a V-phase current sensor are different.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 5 ... Motor control apparatus, 7 ... Microcomputer, 9U ... U-phase current sensor, 9V ... V-phase current sensor, 50 ... Rotor, 51, 52, 53 ... Stator winding

Claims (2)

A method of adjusting a motor control device for driving an electric motor including a rotor as a field and a three-phase stator winding of U phase, V phase and W phase,
The motor control device drives an electric motor based on outputs of first and second current detecting means for detecting currents of first and second phases, which are two phases of the three phases, respectively. ,
The method
A first rotor position control step for supplying current to the three-phase stator winding so that the rotor poles are oriented in a direction that forms an electrical angle of 90 degrees with respect to the first phase;
After the first rotor position control step, the stator is rotated such that voltage vectors represented by voltages applied to the three-phase stator windings rotate at a predetermined period while maintaining a constant magnitude. A first exploration voltage application step for applying an exploration voltage to the winding;
Obtaining a first current detection value output by the first current detection means when the voltage vector is directed to the first phase during the execution of the first exploration voltage application step;
A second rotor position control step for supplying current to the three-phase stator winding so that the rotor poles are oriented in a direction that forms an electrical angle of 90 degrees with respect to the second phase;
After the second rotor position control step, the stator is rotated such that voltage vectors represented by voltages applied to the three-phase stator windings rotate at a predetermined cycle while maintaining a constant magnitude. A second exploration voltage application step for applying an exploration voltage to the winding;
Obtaining a second current detection value output by the second current detection means when the voltage vector is directed to the second phase during the execution of the second exploration voltage application step;
A correction coefficient for correcting at least one of the outputs of the first and second current detection means so as to make the first and second current detection values equal, and a correction coefficient set in the motor control device A method for adjusting a motor control device, comprising a setting step. An apparatus for adjusting a motor control device for driving an electric motor including a rotor as a field and a three-phase stator winding of U phase, V phase and W phase,
The motor control device drives an electric motor based on outputs of first and second current detecting means for detecting currents of first and second phases, which are two phases of the three phases, respectively. ,
The device is
First rotor position control means for supplying current to the three-phase stator winding so that the rotor poles are oriented in a direction that forms an electrical angle of 90 degrees with respect to the first phase;
The voltage vector represented by the voltage applied to each of the three-phase stator windings in a state where the rotor poles are oriented in a direction that forms an electrical angle of 90 degrees with respect to the first phase, A first exploration voltage applying means for applying an exploration voltage to the stator winding so as to be held constant and rotate at a predetermined period;
Means for acquiring a first current detection value output by the first current detection means when the voltage vector is directed to the first phase in a state where the search voltage is applied by the first search voltage application means. When,
Second rotor position control means for supplying current to the three-phase stator winding so that the rotor poles are oriented in a direction that forms an electrical angle of 90 degrees with respect to the second phase;
The voltage vector represented by the voltage applied to each of the three-phase stator windings in a state where the rotor poles are oriented in a direction that forms an electrical angle of 90 degrees with respect to the second phase, Second exploration voltage application means for applying an exploration voltage to the stator winding so as to be held constant and rotate at a predetermined period;
Means for acquiring a second current detection value output by the second current detection means when the voltage vector is directed to the second phase in a state where the search voltage is applied by the second search voltage application means. When,
A correction coefficient for correcting at least one of the outputs of the first and second current detection means so as to make the first and second current detection values equal, and a correction coefficient set in the motor control device An adjustment device for a motor control device, comprising setting means.

Images