JP2008187503A - Reception device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To improve reception characteristics even in the environment of low CN ratio. <P>SOLUTION: A UW correlation unit 32 holds a UW in advance, and performs correlation processing between a reception signal output from a reception band limiting filter 30 and the UW. A symbol data extracting unit 34 detects the peak of sequentially input correlation values as burst timing. A modulated component removing unit 36 removes the component of the UW from a series in a section containing the UW among symbols extracted by the symbol data extracting unit 34. A frequency offset correcting unit 38 uses unmodulated UW symbols to estimate a frequency offset. A carrier wave regenerating unit 40 regenerates a carrier wave on the basis of the unmodulated UW symbols corrected by the frequency offset correcting unit 38. A demodulating unit 60 demodulates the extracted symbols corrected by the frequency offset correcting unit 38 with the carrier wave regenerated by the carrier regenerating unit 40. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a receiving device, and more particularly to a receiving device that processes a received signal.

When receiving a signal, a digital wireless communication receiving apparatus generally performs timing detection, frequency offset estimation, carrier wave recovery, and the like from the received signal. In addition, the receiving apparatus extracts data included in the received signal by performing symbol identification based on the detected timing, frequency offset correction, demodulation using the reproduced carrier wave, and the like (for example, Patent Document 1, Non-Patent Document 1). Patent Document 1).
Japanese Patent Laid-Open No. 11-145921 Heiichi Yamamoto, Shuzo Kato, TDMA Communication, Japan, IEICE, April 1989, p. 76-89

  Reception characteristics in the receiving apparatus are determined by timing detection accuracy, frequency offset estimation accuracy, carrier wave reproduction accuracy, and the like. When digital wireless communication is applied to satellite communication, the CN ratio of a signal received by a receiving apparatus is generally low, and in an environment with a large noise component, timing detection, frequency offset estimation, carrier wave recovery Etc. are executed. Therefore, even in an environment with a low CN ratio, timing detection accuracy, frequency offset estimation accuracy, carrier wave reproduction accuracy, and the like are required to be high.

  The present invention has been made in view of such circumstances, and an object of the present invention is to provide a receiving apparatus that improves reception characteristics even in an environment where the CN ratio is low.

  In order to solve the above-described problem, a receiving apparatus according to an aspect of the present invention includes an input unit that inputs a sequence of received signals that include a known signal in a part of a section, and a sequence of received signals that are input in the input unit. Among these, the phase difference between the removal unit that removes the modulation component of the known signal from the sequence of the section including the known signal, the sequence from which the modulation component of the known signal is removed by the removal unit, and the sequence obtained by delaying the sequence Estimate the frequency offset included in the sequence of received signals input at the input unit by combining the derivation unit that derives multiple types while changing the delay amount and the multiple types of phase differences derived by the derivation unit while weighting them An estimation unit.

  According to this aspect, a plurality of types of phase differences are derived while changing the delay amount, and the frequency offset is estimated by combining the derived types of phase differences, thereby improving reception characteristics in a low CN ratio environment. it can.

  The estimation unit may increase the size of the weighting factor when weighting the phase difference of the delay amount close to the already estimated frequency offset amount. In this case, since the weighting factor is controlled based on the already estimated frequency offset amount, the accuracy of the weighting factor can be improved.

  You may further provide the measurement part which measures the intensity | strength of the series of the received signal input in the input part. The estimation unit may adjust the distribution of the magnitudes of weighting factors over a plurality of types of phase differences when weighting a plurality of types of phase differences according to the intensity measured by the measurement unit. In this case, according to the measured intensity, the distribution of weight coefficient sizes over a plurality of types of phase differences is adjusted, so that a weight coefficient suitable for the measured intensity can be determined.

  It should be noted that any combination of the above-described constituent elements and a conversion of the expression of the present invention between a method, an apparatus, a system, a recording medium, a computer program, etc. are also effective as an aspect of the present invention.

  According to the present invention, reception characteristics can be improved even in an environment where the CN ratio is low.

  Before describing the present invention in detail, an outline will be described. Embodiments of the present invention relate to a communication system including a base station device, a satellite relay device, and a terminal device. In such a configuration, the Forward Link signal is transmitted from the base station apparatus and received by the terminal apparatus via the satellite relay apparatus. On the other hand, the Return Link signal is transmitted from the terminal device and received by the base station device via the satellite relay device. In order to improve reception characteristics, it is necessary to improve symbol timing estimation accuracy, frequency offset estimation accuracy, and carrier wave reproduction accuracy. On the other hand, in the communication system as described above, since the strength of the received signal is generally small, it is difficult to improve accuracy. In response to such a problem, the receiving unit (hereinafter referred to as “receiving device”) of the base station device or terminal device according to the present embodiment executes the following processing.

  The received signal includes a unique word (UW) as a known signal in the front part. The symbol timing is estimated based on the peak detected for the correlation value between the received signal and UW. Here, not only the correlation value between the received signal and UW, but also the correlation value between the differential value of the received signal and the delayed signal of the signal and the differential value of the delayed signal of UW and UW is derived. Further, a plurality of types of correlation values are derived while changing the delay amount. The receiving apparatus combines these correlation values to increase the signal strength, and then detects the peak of the combined correlation value.

  The frequency offset is estimated based on the differential value between the received signal and the delayed signal of the signal. Here, the signal strength is increased by deriving a plurality of types of differential values while changing the delay amount and combining them. Note that the receiving apparatus adjusts the weighting coefficient when combining.

  The carrier wave is estimated by the loop filter so that the phase error included in the received signal is reduced. Here, the received signal is stored, phase error control is executed along the forward direction of the received signal, and then phase error control is executed along the reverse direction of the received signal. By such processing, the control period can be lengthened and the bandwidth of the loop filter can be narrowed, so that the phase error estimation accuracy is improved.

  In addition to the above features, the present embodiment improves the reception characteristics of the terminal device when the receiving device is applied to the terminal device. Note that the receiving apparatus can also be applied to a base station apparatus. A frame is formed by a plurality of the aforementioned burst signals, and such frames are continuously arranged. Further, the terminal device must receive a predetermined burst signal (hereinafter referred to as “target burst signal”) among the plurality of burst signals. Here, the receiving apparatus may reproduce the carrier wave only in the target burst signal, or may reproduce the carrier wave in the entire frame including burst signals other than the target burst signal. In the latter, since the observation period for carrier wave reproduction becomes long, the influence of noise can be reduced. On the other hand, when a sudden change in the transmission path characteristics occurs, the latter cannot be followed because the past transmission path characteristics are reflected. However, the former can follow a rapid fluctuation in transmission path characteristics, but the effect of reducing the influence of noise is small. Therefore, the terminal apparatus according to the present embodiment switches the carrier wave regeneration period according to the degree of fluctuation of the transmission path characteristics.

  In order to clarify the description of the present embodiment, here, 1. Overall configuration, 2. 2. timing detection; 3. frequency offset correction; 4. carrier wave regeneration; Operation, 6. A description will be given in the order of processing in the terminal device.

1. Overall Configuration FIG. 1 shows a configuration of a communication system 100 according to an embodiment of the present invention. The communication system 100 includes a base station apparatus 10, a satellite relay apparatus 12, and a first terminal apparatus 14a and a second terminal apparatus 14b collectively referred to as a terminal apparatus 14.

  The base station device 10 and the terminal device 14 perform communication via the satellite relay device 12. As described above, the route from the base station device 10 to the terminal device 14 via the satellite relay device 12 is Forward Link, and the route from the terminal device 14 to the base station device 10 via the satellite relay device 12 is Return Link. is there. In addition, since Forward Link and Return Link use different frequencies, FDD (Frequency Division Duplex) is used in the communication system 100. Furthermore, the radio link between the base station device 10 and the satellite relay device 12 is called a feeder link, and the radio link between the satellite relay device 12 and the terminal device 14 is called a service link. Also, different frequencies are used for both. In addition, the base station apparatus 10 performs communication with the plurality of terminal apparatuses 14 by assigning time slots to each of the plurality of terminal apparatuses 14.

  That is, in the communication system 100, TDMA (Time Division Multiple Access) is used. The satellite relay device 12 relays communication between the base station device 10 and the terminal device 14. That is, the satellite relay apparatus 12 performs amplification and frequency conversion on the signal received from the base station apparatus 10 and then transmits the signal to the terminal apparatus 14. Further, the satellite relay device 12 performs amplification and frequency conversion on the signal received from the terminal device 14, and then transmits the signal to the base station device 10. In such a communication system 100, since the distance of the wireless transmission path is long, the strength of the signal received by the base station device 10 and the terminal device 14 becomes small.

  FIGS. 2A to 2D show the configuration of frames used in the communication system 100. FIG. FIG. 2A shows a configuration of a plurality of frames in the forward link and the return link. A plurality of frames are continuously arranged as shown in “Frame # n−1”, “Frame #n”, and “Frame # n + 1”. FIG. 2B shows the configuration of one frame. A plurality of time slots are continuously arranged as shown in “Slot # 0”, “Slot # 1”, and “Slot # m−1”. That is, the frame is composed of a plurality of time slots.

  FIG. 2C shows a slot configuration in the case of Forward Link. “UW” is placed at the head of the slot, followed by “information data”. FIG. 2D shows a slot configuration in the case of Return Link. “Guard Time” and “UW” are arranged at the head portion of the slot, and a burst signal in which “information data” is arranged subsequent thereto. That is, no preamble for timing synchronization or frequency synchronization is arranged. Although not shown, a “PILOT” symbol is inserted in the “information data” area of FIGS. 2C and 2D. One “PILOT” symbol is inserted per fixed symbol period.

  FIG. 3 shows a configuration of the receiving device 20 according to the embodiment of the present invention. The receiving device 20 includes an RF unit 22, a baseband unit 24, and a control unit 26. The receiving device 20 corresponds to a receiving function portion provided in the base station device 10 and the terminal device 14.

  The RF unit 22 receives a transmission signal from a base station device 10 (not shown) relayed by a satellite relay device 12 (not shown) via an antenna. Here, the slot configuration of the received signal is shown in FIG. 2 (c) or (d). The RF unit 22 converts the frequency of the received signal from a radio frequency to an intermediate frequency. The RF unit 22 generates a baseband signal by performing quadrature detection on the intermediate frequency signal. In general, baseband signals include in-phase and quadrature components and should therefore be represented by two signal lines, but here they are represented by a single signal line for the sake of clarity. And The RF unit 22 also performs analog-digital conversion on the baseband signal and outputs the converted digital signal to the baseband unit 24. Hereinafter, for convenience of explanation, the converted digital signal is also referred to as a received signal.

  The baseband unit 24 performs demodulation and decoding on the reception signal received from the RF unit 22. The baseband unit 24 also executes processing associated with demodulation and decoding. In the following, timing detection, frequency offset correction, and carrier wave reproduction will be described later as accompanying processes. The carrier wave reproduction process in the terminal device 14 will also be described later. The control unit 26 controls timing and the like in the receiving device 20.

  FIG. 4 shows the configuration of the baseband unit 24. The baseband unit 24 includes a reception band limiting filter 30, a UW correlation unit 32, a symbol data extraction unit 34, a modulation component removal unit 36, a frequency offset correction unit 38, a carrier wave recovery unit 40, a UW identification unit 42, and a frame synchronization processing unit 44. A symbol soft decision unit 46, a Turbo decoding unit 48, a descrambling unit 50, a CRC check unit 54, a CNR measurement unit 58, and a demodulation unit 60.

  As shown in FIGS. 2C and 2D, the reception band limiting filter 30 receives a signal including UW as a known signal in a part of the section. Here, the reception band limiting filter 30 has a low-pass characteristic, and reduces a component having a frequency higher than the cutoff frequency of the low-pass characteristic. The UW correlator 32 holds a UW pattern in advance and executes a correlation process between the reception signal output from the reception band limiting filter 30 and the UW. The UW correlation unit 32 has, for example, a matched filter configuration. The UW correlation unit 32 sequentially outputs the correlation values derived by the correlation process.

  On the one hand, the symbol data extraction unit 34 sequentially receives the correlation values from the UW correlation unit 32 and, on the other hand, the signals output from the reception band limiting filter 30. Further, the symbol data extraction unit 34 detects the peak of the correlation value sequentially input as the slot timing. Here, the slot timing corresponds to the head timing of the UW shown in FIGS. 2 (c) and 2 (d). In addition, the symbol data extraction unit 34 extracts symbols from the signal output from the reception band limiting filter 30 based on the detected burst timing. That is, the symbol data extraction unit 34 extracts symbols from the signal input to the baseband unit 24 at the timing detected based on the correlation value derived by the UW correlation unit 32.

  The modulation component removal unit 36 performs reverse modulation with a UW pattern stored in advance from a series of sections that are symbols extracted by the symbol data extraction unit 34 and include UW in the received signal. Remove the modulation component. That is, the modulation component removal unit 36 removes the modulation component from the UW portion of the received signal. Here, it is assumed that UW is modulated by π / 4 shift QPSK. Therefore, the modulation component removal unit 36 removes the QPSK modulation component after removing the π / 4 shift from the symbol. With the above processing, the signal output from the modulation component removal unit 36 corresponds to a phase error component.

  The frequency offset correction unit 38 estimates the frequency offset by using the UW symbol unmodulated in the modulation component removal unit 36 (hereinafter referred to as “unmodulated UW symbol”). Details of the frequency offset estimation method will be described later. The frequency offset correction unit 38 corrects the unmodulated UW symbol from the modulation component removal unit 36 and corrects the symbol extracted by the symbol data extraction unit 34 (hereinafter referred to as “extracted symbol”) based on the estimated frequency offset. To do. Here, the symbols extracted by the symbol data extraction unit 34 are intended for the entire slot. The frequency offset correction unit 38 outputs the corrected unmodulated UW symbol to the carrier wave recovery unit 40 and outputs the corrected extracted symbol to the demodulation unit 60.

  The carrier recovery unit 40 recovers the carrier based on the unmodulated UW symbol corrected by the frequency offset correction unit 38. Although details of the carrier wave recovery method will be described later, the carrier wave recovery unit 40 includes a loop filter. Also, the carrier wave recovery unit 40 reproduces the carrier wave using the PILOT symbol after the end of the UW period. The carrier wave reproduction unit 40 outputs the reproduced carrier wave to the demodulation unit 60. The demodulating unit 60 demodulates the extracted symbol corrected by the frequency offset correcting unit 38 using the carrier wave reproduced by the carrier wave reproducing unit 40. Since a known technique may be used for demodulation, description thereof is omitted here. The demodulator 60 outputs the demodulated symbols to the UW identification unit 42 and the symbol soft decision unit 46.

  The UW identifying unit 42 identifies the UW part in the received signal from the demodulated symbols from the demodulating unit 60. When the UW part is identified by the UW identification unit 42, the frame synchronization processing unit 44 outputs the timing as a frame synchronization timing to the control unit 26 (not shown). The control unit 26 uses the frame synchronization timing for upper layer processing. The symbol soft decision unit 46 makes a soft decision on the demodulated symbol from the demodulation unit 60, and the Turbo decoding unit 48 decodes using the soft decision symbol. Here, it is assumed that Turbo encoding is performed in a transmission device (not shown). The descrambling unit 50 descrambles the decoded symbols. The descrambling unit 50 outputs the descrambling result to the control unit 26 (not shown).

  The CNR measurement unit 58 measures the CNR based on the symbols soft-determined by the symbol soft-decision unit 46. The CNR measurement unit 58 outputs the measured CNR to the control unit 26 (not shown). The CRC check unit 54 performs a CRC check on the symbols decoded by the Turbo decoding unit 48.

  This configuration can be realized in terms of hardware by a CPU, memory, or other LSI of any computer, and in terms of software, it is realized by a program having a communication function loaded in the memory. Describes functional blocks realized by collaboration. Accordingly, those skilled in the art will understand that these functional blocks can be realized in various forms by hardware only, software only, or a combination thereof.

2. Timing Detection As described above, the baseband unit 24 detects the symbol timing in the symbol data extraction unit 34 based on the correlation value derived in the UW correlation unit 32. Such a deterioration factor of the timing detection accuracy is that an error is included in the correlation value due to noise and frequency offset. Therefore, an object of the present invention is to improve the timing detection accuracy at the time of low C / N reception with a frequency offset.

  FIG. 5 shows the configuration of the UW correlation unit 32. The UW correlation unit 32 includes a CNR measurement unit 80, a frequency offset estimation unit 82, a first basic correlator 84a, a second basic correlator 84b, a Z basic correlation unit 84z, an integration unit 86, A first differential correlator 88a, a second differential correlator 88b, a Kth differential correlator 88k, and a combining unit 90, which are collectively referred to as a differential correlator 88, are included.

  The basic correlator 84 derives a correlation value by executing a correlation process between a reception signal from a reception band limiting filter 30 (not shown) and a UW pattern stored in advance. In general, the correlation process is performed on the entire UW, and the timing detection is realized by comparing the maximum correlation value with a threshold value. However, in the present embodiment, a section including UW is divided into a plurality of partial sections, and a plurality of basic correlators 84 are arranged so as to correspond to each of the plurality of partial sections. That is, each basic correlator 84 derives a partial correlation value between the received signal from the reception band limiting filter 30 (not shown) and the UW in the corresponding partial section. Here, when the number of UW symbols is “N” and the “N” symbol is divided into “D” partial sections, the correlation length of each divided partial section is assumed to be “L” symbols.

  The accumulating unit 86 derives a correlation value by accumulating the partial correlation values derived in each of the plurality of basic correlators 84. In general, when the correlation length is long as in the configuration of the correlator, the multiplication result of the received signal and the UW symbol is phase-shifted over the correlation length due to the influence of the frequency offset, resulting in a decrease in the correlation value due to the phase rotation. . On the other hand, since the amount of phase rotation per correlation length can be reduced by shortening the correlation length in one basic correlator 84 as in the configuration of this embodiment, the correlation value per correlation length is reduced. Can be suppressed.

  The differential correlator 88 derives a differential value (hereinafter referred to as “differential signal”) between a reception signal from a reception band limiting filter 30 (not shown) and a signal obtained by delaying the reception signal. The differential correlator 88 holds in advance a differential value between the UW and the delayed UW (hereinafter referred to as “differential UW”). Here, the delay amounts of the differential signal and the differential UW are the same. The differential correlator 88 derives a correlation value between the differential signal and the differential UW. Here, as shown in the figure, K differential correlators 88 are provided from the first differential correlator 88a to the Kth differential correlator 88k, and the delay amount of each differential correlator 88 is different from each other. ing. For example, the first differential correlator 88a has a one symbol delay, while the second differential correlator 88b has a two symbol delay.

  That is, the first differential correlator 88a sequentially stores the received signal in the shift register after one-symbol differential detection, and executes correlation processing with the result of UW one-symbol differential detection. As described above, the plurality of differential correlators 88 derive a plurality of types of correlation values while changing the delay amount. Thus, since the correlation processing by the differential correlator 88 performs integration of the differential detection result, it is possible to suppress a decrease in the correlation value due to the frequency offset.

  The combining unit 90 combines a plurality of types of correlation values derived by the differential correlator 88 and the correlation values derived by the basic correlator 84 and the integrating unit 86. That is, the UW correlation unit 32 increases the signal strength by combining a plurality of correlation values. As a result, even when the C / N of the received signal is low, the error included in the correlation value is reduced. In addition, since the correlation value in the synthesizing unit 90 is obtained as a sum of different correlation characteristics, even in the case of a UW symbol pattern having an autocorrelation characteristic that generates a high correlation value other than the correlation peak position, It becomes possible to suppress the false detection probability other than the position. The synthesizer 90 outputs the synthesized correlation value to a symbol data extractor 34 (not shown).

  The frequency offset estimation unit 82 estimates a frequency offset with respect to a reception signal from a reception band limiting filter 30 (not shown). Since a known technique or a technique described later may be used for estimating the frequency offset, the description thereof is omitted here. In addition, the basic correlator 84 adjusts the length of the partial section, that is, the correlation length in the correlation process, according to the value of the frequency offset estimated by the frequency offset estimation unit 82. That is, the basic correlator 84 variably controls the division number D in order to provide frequency offset tolerance corresponding to the estimated frequency offset amount. For example, if the frequency offset value increases, the division number D increases, and if the frequency offset value decreases, the division number D decreases.

  The CNR measurement unit 80 measures the strength of the reception signal from the reception band limiting filter 30 (not shown). Since a known technique may be used for measuring the signal intensity, the description is omitted here. The combining unit 90 adjusts the number of correlation values to be combined among the correlation values from the integrating unit 86 and the differential correlator 88 according to the intensity value measured by the CNR measuring unit 80. For example, if the measured signal strength is large, the synthesis unit 90 selects only the correlation value from the integration unit 86. Further, as the measured signal intensity decreases, the synthesis unit 90 increases the number of correlation values to be synthesized.

  FIG. 6 shows the configuration of the basic correlator 84. The basic correlator 84 includes a reception signal delay unit 140, a multiplication unit 142, a UW holding unit 144, an addition unit 146, and a conversion unit 148. Here, the number of samples per symbol, that is, the sampling rate is “M”.

  The reception signal delay unit 140 is a shift register that delays a reception signal from a reception band limiting filter 30 (not shown). As described above, since the sampling rate of the received signal is “M”, the received signal delay unit 140 is arranged to correspond to the sampling rate “M”. The UW holding unit 144 stores the UW part in the corresponding partial period. Multiplier 142 performs complex multiplication of the symbol delayed by reception signal delay section 140 and the UW symbol held in UW holding section 144. The adder 146 adds the multiplication results. Specifically, the conversion unit 148 derives the magnitude of the addition result from the addition unit 146 as a partial correlation value. Specifically, the conversion unit 148 calculates the square value of the in-phase component of the addition result and the square value of the quadrature component of the addition result. Is added.

  FIG. 7 shows the configuration of the first differential correlator 88a. The first differential correlator 88a includes a differential delay unit 110, a UW delay unit 112, a multiplication unit 114, a multiplication unit 116, a reception signal delay unit 118, a differential UW holding unit 120, a multiplication unit 122, an addition unit 124, A conversion unit 126 is included. The other differential correlator 88 has the same configuration as that shown in FIG. 7, although the delay amount when generating the differential signal and the differential UW is different.

  The differential delay unit 110 delays a reception signal from a reception band limiting filter 30 (not shown). Here, M samples are delayed. Since the sampling rate is faster than the symbol rate, a plurality of samples that should correspond to one symbol are delayed by the differential delay unit 110. Multiplier 114 performs complex multiplication of the reception signal delayed by differential delay section 110 and the reception signal from reception band limiting filter 30 (not shown). At this time, since the complex conjugate operation is performed on the reception signal delayed in the differential delay unit 110, the multiplication in the multiplication unit 114 is equivalent to deriving the difference between the two. To do. The difference here is the aforementioned differential signal. The reception signal delay unit 118 delays the differential signal derived by the multiplication unit 114. Here, the reception signal delay unit 118 performs a delay over M (N−1) samples.

  The UW delay unit 112 delays the UW by one symbol. Multiplier 116 performs complex multiplication of UW and UW delayed in UW delay section 112. At that time, since the multiplication is executed after the complex conjugate operation is performed on the UW delayed in the UW delay unit 112, the multiplication in the multiplication unit 116 is equivalent to deriving the difference between the two. The difference here is the aforementioned differential UW. The differential UW holding unit 120 delays the differential UW derived in the multiplication unit 116. Here, the differential UW holding unit 120 performs a delay over N−1 symbols. Multiplier 122 performs complex multiplication of the differential signal delayed by reception signal delay unit 118 and the differential UW held in differential UW holding unit 120. The adding unit 124 adds the multiplication results. Specifically, the conversion unit 126 derives the magnitude of the addition result in the addition unit 124 as a partial correlation value. Specifically, the conversion unit 126 calculates the square value of the in-phase component of the addition result and the square value of the orthogonal component of the addition result. Is added.

3. Frequency offset correction The frequency offset correction unit 38 derives a frequency offset. Generally, when the C / N of the received signal is low, the influence of noise increases, so the frequency offset estimation accuracy decreases. Further, when the frequency offset estimation accuracy is lowered, the reception characteristic of the reception device 20 is deteriorated. Therefore, the purpose here is to improve the frequency offset estimation performance at the time of low C / N reception.

  FIG. 8 shows the configuration of the frequency offset correction unit 38. The frequency offset correction unit 38 includes a first offset estimation unit 230, a second offset estimation unit 232, a j + 1 offset estimation unit 234, a J offset estimation unit 236, a CNR measurement unit 200, a weight control unit 202, a synthesis unit 204, a rotation Unit 206, multiplication unit 208, and multiplication unit 210. The first offset estimation unit 230 includes a first delay unit 160, a first multiplication unit 162, a first addition unit 164, a first conversion unit 166, and a first adjustment unit 168. The second offset estimation unit 232 includes: It includes a second delay unit 170, a second multiplication unit 172, a second addition unit 174, a second conversion unit 176, and a second adjustment unit 178. The j + 1 offset estimation unit 234 includes a j + 1 delay unit 180, a j + 1 multiplication unit. 182, j + 1 adder 184, j + 1 converter 186, j + 1 adjuster 188, J offset estimator 236 includes J delay 190, J multiplier 192, J adder 194, J A conversion unit 196 and a Jth adjustment unit 198 are included.

  The first offset estimation unit 230 receives an unmodulated UW symbol from a modulation component removal unit 36 (not shown). First delay section 160 delays the received unmodulated UW symbol by one symbol. The first multiplication unit 162 performs complex multiplication of the received unmodulated UW symbol and the unmodulated UW symbol delayed by the first delay unit 160. At that time, a complex conjugate operation of the delayed unmodulated UW symbol is executed. By the multiplication in the first multiplication unit 162, a phase difference over one symbol is derived. The first addition unit 164 adds the phase difference. Since the addition here is vector addition, it corresponds to the derivation of the average value of the phase differences. The first conversion unit 166 converts the addition result in the first addition unit 164 from a complex value to a phase value. The first adjustment unit 168 converts the phase value converted by the first conversion unit 166 into a phase rotation amount over one symbol. The phase rotation amount corresponds to the frequency offset amount.

  The second offset estimator 232, the j + 1th offset estimator 234, and the Jth offset estimator 236 perform the same processing as the first offset estimator 230. However, while the delay amount in the first offset estimation unit 230 is one symbol, the delay amount in each of the second offset estimation unit 232, the j + 1th offset estimation unit 234, and the Jth offset estimation unit 236 is 2 symbols, j + 1 symbol, and J symbol. That is, the first offset estimation unit 230 to the Jth offset estimation unit 236 derive a plurality of types of phase differences while changing the delay amount.

  Note that the value of “J” should be appropriately set from the value assumed to be the maximum frequency offset amount with respect to the symbol rate. The second delay unit 170, the j + 1th delay unit 180, and the Jth delay unit 190 correspond to the first delay unit 160, and the second multiplication unit 172, the j + 1th multiplication unit 182 and the Jth multiplication unit 192 are the first multiplication. The second addition unit 174, the j + 1th addition unit 184, and the Jth addition unit 194 correspond to the first addition unit 164, and correspond to the second conversion unit 176, the j + 1th conversion unit 186, and the Jth conversion unit. 196 corresponds to the first conversion unit 166, and the second adjustment unit 178, the j + 1th adjustment unit 188, and the Jth adjustment unit 198 correspond to the first adjustment unit 168.

  The combining unit 204 combines the plurality of types of frequency offset amounts derived from the first offset estimating unit 230 by the Mth offset estimating unit 236 while weighting them. Here, the weighting factor for weighting is received from the weight control unit 202 described later. Through the above processing, the synthesis unit 204 estimates the frequency offset included in the received signal. That is, the first offset estimation unit 230 to the M-th offset estimation unit 236 estimate a plurality of types of frequency offsets while setting a plurality of symbol intervals for detecting the amount of phase rotation, and the combining unit 204 estimates Multiply multiple frequency offsets by weighting factor.

  As a result of the above processing, a final frequency offset is estimated. By combining multiple frequency offsets, a final frequency offset is derived. If the delay time for detecting the amount of phase rotation is different, the behavior of the random component due to noise is different, so that each estimation accuracy is improved. The effect is particularly great when the C / N of the received signal is low.

  The CNR measurement unit 200 measures the intensity of the unmodulated UW symbol from the modulation component removal unit 36 (not shown). Here, since a known technique may be used for the measurement of the signal intensity, the description is omitted here. Note that when the receiving apparatus 20 includes a component having the same function as the CNR measurement unit 200, the CNR measurement unit 200 may not be provided.

  The weight control unit 202 derives a weighting factor for weighting the plurality of frequency offsets output from the first offset estimation unit 230 to the Mth offset estimation unit 236. Here, the weight control unit 202 derives a weighting factor by a combination of the following two processing results. For the first eye process, the weight control unit 202 receives the frequency offset amount already estimated by the synthesis unit 204. Further, the weight control unit 202 identifies an offset correction unit having a delay amount close to the received frequency offset amount. Furthermore, the weight control unit 202 increases the size of the weighting coefficient when weighting the frequency offset output from the specified offset correction unit.

  Here, the already estimated frequency offset amount corresponds to, for example, the frequency offset amount estimated for the UW portion of the previous slot. When there is no already estimated frequency offset amount as in the first reception slot, the weight control unit 202 equalizes the weight coefficients for all frequency offsets. As a second process, the weight control unit 202 adjusts the distribution of the weighting factor sizes over a plurality of types of frequency offsets according to the intensity measured by the CNR measurement unit 200.

  For example, if the C / N measured by the CNR measurement unit 200 is large, the number of frequency offsets to be combined may be small, so that the weight control unit 202 increases only the size of the weight coefficient for one frequency offset. . On the other hand, if the C / N measured by the CNR measurement unit 200 is small, the number of frequency offsets to be combined needs to be increased, so that the weight control unit 202 increases the size of the weighting coefficient for a plurality of frequency offsets. .

  The rotation unit 206 generates a signal that rotates in phase according to the frequency offset derived by the synthesis unit 204. Note that the rotation unit 206 generates a signal so as to cancel the frequency offset included in the received signal, that is, reverse rotation. Multiplying section 208 rotates the phase of the unmodulated UW symbol from modulation component removing section 36 (not shown) by the signal generated in rotating section 206. This phase rotation corresponds to correction of the frequency offset. Further, the multiplication unit 210 rotates the phase of the extracted symbol from the symbol data extraction unit (not shown) by the signal generated by the rotation unit 206.

4). Carrier wave reproduction As shown in FIGS. 2 (c) to (d), since the slot configuration does not include a preamble, in order to prevent the reception characteristics of the information data part from deteriorating, it is sufficient for carrier wave phase estimation in the UW part in carrier wave reproduction. Accuracy is required. However, in the carrier wave recovery so far, phase errors are sequentially detected with respect to input data, and an NCO (Numerically Controlled Oscillator) is controlled in accordance with the errors. Therefore, the accuracy of the regenerated carrier wave is high at the time when the processing has progressed to some extent, but in the first slot at the start of the Forward Link terminal reception processing or every slot in the Return Link base station reception processing The phase estimation accuracy of the regenerated carrier wave is not high at the head portion of the burst signal. Here, an object is to improve the accuracy of carrier wave reproduction at the head portion of the received signal at the time of low C / N reception.

  FIG. 9 shows the configuration of the carrier recovery unit 40. The carrier recovery unit 40 includes a storage unit 220, a multiplication unit 222, a conversion unit 224, a loop filter 226, and an NCO 228. The storage unit 220 receives the unmodulated UW symbol corrected by the frequency offset correction unit 38 (not shown) and stores it.

  Multiplier 222 performs complex multiplication of the unmodulated UW symbol stored in storage unit 220 and the reproduced carrier wave from NCO 228. Here, a complex conjugate operation is performed on the carrier wave from the NCO 228 before multiplication. The multiplier 222 outputs the multiplication result. The multiplication result corresponds to a phase error included in the carrier wave. The conversion unit 224 converts the multiplication result output from the multiplication unit 222 from a complex value to a phase value. The loop filter 226 has an LPF function and plays a role of smoothing the signal from the conversion unit 224. The amount of phase rotation of the NCO 228 is controlled according to the magnitude of the phase error signal smoothed by the loop filter 226. This is equivalent to controlling the phase error included in the unmodulated UW symbol stored in the storage unit 220 to be small. The NCO 228 regenerates and outputs a carrier wave according to the control of the loop filter 226.

  The control unit 26 (not shown) outputs a command so that the unmodulated UW symbol stored in the storage unit 220 is used in the forward direction and then used in the reverse direction at the time of carrier wave reproduction. Here, the forward direction is a direction from the head portion toward the rear portion in the slot configuration of FIGS. 2C to 2D, and the reverse direction is a direction opposite to the forward direction. Specifically, the control unit 26 reads the unmodulated UW symbol stored in the storage unit 220 in the forward direction from the head, and outputs it to the multiplication unit 222. When reading to the end of the unmodulated UW symbol, the control unit 26 reads in the reverse direction from the end of the unmodulated UW symbol. By the above processing, the length of the unmodulated UW symbol that can be used for reproducing the carrier wave is increased.

  FIG. 10 shows an outline of the operation of the carrier wave reproducing unit 40. The vertical axis represents the amount of phase error contained in the regenerated carrier wave, and the horizontal axis corresponds to a sequence of unmodulated UW symbols. The left side is close to the beginning of the unmodulated UW symbol, and the right side is close to the end of the unmodulated UW symbol. That is, the direction from left to right is the forward direction, and the direction from right to left is the reverse direction. First, it is assumed that the phase error θe0 at the beginning time of UW becomes θe1 at the time of the last symbol of UW by the forward processing. Next, the residual frequency offset estimation value is also held while the phase of the NCO 228 is held at θe1, and the process is executed from the last symbol of the UW to the head of the UW. By executing this time-repetitive carrier recovery process, the carrier wave at the head of the UW is estimated.

  Here, another operation content of the carrier wave reproduction processing is also described. When the C / N of the received signal is sufficiently high, the carrier wave with sufficient accuracy can be reproduced once the above processing is executed. However, when the C / N of the received signal is low, sufficient estimation accuracy cannot be obtained unless the band of the loop filter 226 is sufficiently narrowed. In this case, it is conceivable that the predetermined number of UW symbols is insufficient to obtain sufficient estimation accuracy.

  The shortage of UW symbols due to the narrowing of the band of the loop filter 226 is compensated by increasing the number of iterations. If the phase of the carrier wave is not sufficiently estimated, phase error information corresponding to the estimation error can be obtained. Therefore, the band of the loop filter 226 is sufficiently narrowed to extract highly reliable phase error information. If the iterative process is repeated, the carrier phase at the head of the UW is estimated with high accuracy even at low C / N. That is, after using the unmodulated UW symbol stored in the storage unit 220 in the forward direction, the control unit 26 repeatedly executes a process of using the unmodulated UW symbol in the reverse direction a plurality of times.

  FIG. 11 shows another operation outline of the carrier wave reproducing unit 40. FIG. 11 is shown in the same manner as FIG. 10, but the iterative process is executed a plurality of times. Further, the phase error is gradually reduced in the order of θe0, θe1, θe2, and θe3 in accordance with the number of executions of the iterative process.

5. Operation The operation of the receiving apparatus 20 configured as above will be described. When a signal transmitted from the transmission side is received, the UW correlator 32 performs correlation processing between the received signal and UW. At that time, the UW correlator 32 derives the correlation value in the basic correlator 84 and the accumulator 86, and also derives the correlation value in the differential correlator 88, and synthesizes them in the synthesizer 90 to obtain the final value. Correlation value. Note that the correlation length in the basic correlator 84 and the number of synthesis in the synthesis unit 90 are adjusted based on the processing results in the CNR measurement unit 80 and the frequency offset estimation unit 82.

  The symbol data extraction unit 34 detects the peak of the correlation value sequentially input from the UW correlation unit 32 as the UW head timing. After detecting the UW head timing, the modulation component removal unit 36 removes the UW modulation component and outputs an unmodulated UW symbol. The frequency offset correction unit 38 derives a frequency offset based on the unmodulated UW symbol. At that time, the frequency offset correction unit 38 derives the frequency offset from each of the first offset estimation unit 230 in each of the J-th offset estimation units 236 and synthesizes them while weighting them in the synthesis unit 204, thereby obtaining the frequency offset value. Is estimated. Further, the weight coefficient is set by the CNR measurement unit 200 and the weight control unit 202. Further, the frequency offset correction unit 38 corrects the unmodulated UW symbol and the received signal symbol data with the estimated frequency offset.

  The carrier recovery unit 40 stores the unmodulated UW symbol in the storage unit 220 and causes the loop filter 226 and the NCO 228 to recover the carrier by using the stored unmodulated UW symbol in the forward direction and the reverse direction. The demodulating unit 60 demodulates the received signal corrected by the frequency offset correcting unit 38 using the carrier wave reproduced by the carrier wave reproducing unit 40.

6). Processing in Terminal Device The communication system 100 is executing TDMA communication, and exclusively uses time slots among a plurality of terminal devices 14 as shown in FIG. Each terminal device 14 continuously receives transmission signals from the base station device 10. Using this property, the terminal device 14 uses the UW of the entire frame including the allocated time slot and other time slots to increase the frequency offset in order to lengthen the processing period in the loop filter 226. Then, the carrier wave recovery process by the loop filter 226 is executed.

  In such a case, the frequency and phase of the NCO 228 are inherited between time slots. The time constant of the loop filter 226 is set to a very large value. As a result, when the carrier is out of synchronization due to fading, noise, or the like, the time for detecting the out of synchronization or the period required for resynchronization becomes long. Here, the purpose is to improve the reception performance of Forward Link in TDMA communication. In other words, not only the carrier recovery process for the entire frame, but also the carrier recovery process limited to the time slot allows early detection and resynchronization when synchronization is lost due to fading, etc. The

  As in the past, the receiving device 20 has a continuous frame formed by a plurality of time slots as shown in FIGS. 2A to 2D, and receives the assigned time slot. The control unit 26 in FIG. 3 measures the degree of change in the wireless transmission path based on the received slot signal. The control unit 26 measures the Doppler frequency as the degree of change of the wireless transmission path. A known technique may be used for measuring the Doppler frequency. Note that the radio transmission path may be estimated in an unassigned time slot.

  The carrier wave reproducing unit 40 reproduces the carrier wave in the assigned time slot. Since the carrier wave reproduction may be performed as described with reference to FIGS. 9 to 11, the description thereof is omitted here. Note that the carrier recovery unit 40 newly estimates the carrier at the slot start timing. Note that the frequency offset correction unit 38 may perform the same processing. That is, here, of the frequency offset estimation and carrier phase estimation results obtained in the time slot, only the estimation result obtained in the assigned time slot is used as the initial value of the carrier recovery unit 40 and the frequency offset correction unit 38. .

  Therefore, the loop filter 226 executes the carrier wave recovery process without taking over the past frequency and the past phase information in the NCO 228. As a result, detection of loss of synchronization is immediately performed by checking the CRC for each time slot. In addition, since the out-of-synchronization effect does not propagate to subsequent time slots, resynchronization is possible in the next assigned time slot. Hereinafter, for convenience of explanation, such processing is referred to as “slot processing”.

  Further, the carrier wave reproducing unit 40 reproduces the carrier wave in a frame including other than the assigned time slot. Here, the loop filter 226 executes the carrier wave reproduction process while taking over the past frequency and phase information in the NCO 228. Hereinafter, for convenience of explanation, such processing is referred to as “frame processing”.

  The control unit 26 selects “slot processing” or “frame processing” for reception as processing in the carrier recovery unit 40 in accordance with the measured degree of change, for example, the Doppler frequency. Here, the control unit 26 holds a threshold value in advance, and compares the threshold value with the Doppler frequency. For example, if the Doppler frequency is greater than the threshold, the control unit 26 selects “slot processing”, and if the Doppler frequency is equal to or less than the threshold, the control unit 26 selects “frame processing”.

  According to the embodiment of the present invention, a plurality of types of correlation values are derived while changing the combination, and the derived plurality of types of correlation values are combined to generate a final correlation value. Reception characteristics can be improved. Further, since the correlation value is derived by synthesizing the partial correlation values with a short correlation length, the influence of the frequency offset on the correlation value can be reduced. Further, since the correlation value for the differential signal is derived, the influence of the frequency offset on the correlation value can be reduced. Moreover, since the length of the partial section is adjusted according to the value of the frequency offset, the error included in the correlation value can be reduced.

  Also, if the frequency offset value is large, the length of the partial section is shortened, so that the influence of the frequency offset can be reduced. Also, if the frequency offset value is small, the length of the partial section is lengthened, so that a more reliable correlation result can be obtained. Further, since the number of correlation values to be combined is adjusted according to the measured intensity value, errors included in the correlation value can be reduced in the measured intensity value. Also, if the measured intensity value is small, the number of correlation values to be combined is increased, so that the reliability of the correlator output can be improved.

  In addition, since the frequency offset is estimated by deriving a plurality of types of phase differences while changing the delay amount, and synthesizing the derived types of phase differences, it is possible to improve reception characteristics in a low CN ratio environment. Further, since a plurality of types of phase differences are combined, the signal intensity can be increased. Further, since the weighting coefficient corresponding to the delay time close to the already estimated frequency offset amount is increased, it is possible to increase the weighting of the likely frequency offset. Moreover, since the weighting of the likely frequency offset is increased, the accuracy of the weighting coefficient can be improved. In addition, since the distribution of the weight coefficient sizes over a plurality of types of phase differences is adjusted according to the measured intensity, a weight coefficient suitable for the measured intensity can be determined. In addition, if the measured intensity is small, the weighting factor is controlled so as to use a large number of phase differences, so that the signal intensity can be increased.

  In addition, after storing the series from which the modulation component of the known signal is removed, the control for reducing the phase error is executed while using the series in the forward direction and the reverse direction, so that the control for reducing the phase error is performed. The reception period under a low CN ratio environment can be improved. In addition, since the process of using the sequence in the forward direction and the backward direction is repeated a plurality of times, the control period for reducing the phase error can be further extended. In addition, since “frame processing” or “slot processing” can be selected as a period for reproducing a carrier wave, and flexible processing can be performed, reception characteristics in a low CN ratio environment can be improved. Further, since the period for reproducing the carrier wave is changed according to the measured degree of change, processing suitable for the degree of change can be executed. Further, if the Doppler frequency is large, “slot processing” is used, so that resynchronization can be easily executed even when synchronization is lost. Further, if the Doppler frequency is small, “frame processing” is used, so that the processing period can be extended.

  In the above, this invention was demonstrated based on the Example. This embodiment is an exemplification, and it will be understood by those skilled in the art that various modifications can be made to the combination of each component and each processing process, and such modifications are also within the scope of the present invention. .

  In the embodiment of the present invention, the basic correlator 84 performs control to change the number of divisions according to the frequency offset estimated by the frequency offset estimation unit 82. However, the present invention is not limited to this. For example, the basic correlator 84 may change the number of divisions based on a period that has elapsed since the start of the reception process. For example, control is performed such that the number of divisions decreases. When the frequency synchronization process is executed immediately after the start of the reception process in the reception device 20 in the communication system 100, the frequency offset amount is expected to be smaller than that at the start of the reception process after a predetermined period has elapsed. The When the frequency offset amount is reduced, the phase rotation amount resulting therefrom is reduced. Therefore, the basic correlator 84 makes the division number smaller than that immediately after the start of the reception process. According to this modification, the number of divisions is automatically changed, so that the processing can be simplified.

  In the embodiment of the present invention, the carrier recovery unit 40 uses the loop filter 226 to perform control for reducing the phase error, and forwards the unmodulated UW symbol stored in the storage unit 220 in the forward direction. After being used, the processing used for the reverse direction is executed. However, the present invention is not limited to this. For example, the band of the loop filter 226 may be changed according to the number of times the unmodulated UW symbol is used in the forward direction and the backward direction. For example, the carrier recovery unit 40 performs control so as to narrow the band of the loop filter 226 as the number of times increases. In addition, the time constant of the loop filter 226 is switched for each of (1) UW unit repetitive processing, (2) forward carrier recovery processing in the UW unit after repetitive processing, and (3) PILOT symbol unit. Also good. According to the present modification, the band of the loop filter is changed according to the number of times the unmodulated UW symbol is used in the forward direction and the reverse direction, so that a band suitable for the degree of convergence of the phase error can be used. .

  In the embodiment of the present invention, the receiving apparatus 20 includes a combination of the UW correlation unit 32, the frequency offset correction unit 38, and the carrier wave recovery unit 40. However, the present invention is not limited thereto, and for example, any one or any combination of the UW correlation unit 32, the frequency offset correction unit 38, and the carrier wave recovery unit 40 may be provided in the reception device 20. In that case, a well-known technique should just be used with respect to the part which is not provided. According to this modification, the freedom degree of the structure of the receiver 20 can be improved.

The features of the invention described in the embodiments may be defined by the following items.
(Item 4-1)
An input unit for inputting a series of received signals including a known signal in some sections;
A removing unit that removes a modulation component of a known signal from a series of sections in which a known signal is included in a series of received signals input in the input unit;
A storage unit for storing a sequence obtained by removing a modulation component of a known signal in the removing unit;
A carrier wave reproducing unit for reproducing a carrier wave by controlling to reduce a phase error included in the sequence stored in the storage unit;
A demodulator that demodulates a sequence of received signals input at the input unit, using the carrier recovered at the carrier recovery unit;
The receiving apparatus according to claim 1, wherein the carrier recovery unit performs control for reducing a phase error while using the sequence stored in the storage unit in the forward direction and then using the sequence in the forward direction.

(Item 4-2)
The receiving apparatus according to (Item 4-1), wherein the carrier wave reproduction unit repeatedly uses a sequence stored in the storage unit in the forward direction and then repeatedly performs a process that uses the sequence in the reverse direction a plurality of times. .
(Item 4-3)
The carrier recovery unit performs control for reducing phase error by using a loop filter, and uses the sequence stored in the storage unit in the forward direction and then uses it in the reverse direction. The receiving device according to (Item 4-2), wherein the band of the loop filter is changed according to the number of times.

(Item 2-1)
An input unit for inputting a series of received signals including a known signal in some sections;
A first correlation processing unit for deriving a correlation value by executing a correlation process between a series of received signals input at the input unit and a known signal;
While changing the amount of delay, the correlation processing of the differential value between the received signal sequence inputted in the input unit and the differential value obtained by delaying the relevant sequence, and the differential value between the known signal and the delayed known signal is performed. A second correlation processing unit for deriving a plurality of types of different correlation values by executing,
A synthesis unit that synthesizes a plurality of different correlation values derived in the second correlation processing unit and the correlation values derived in the first correlation processing unit;
A processing unit that processes a series of received signals input in the input unit at a timing detected based on the correlation value combined in the combining unit;
When the first correlation processing unit divides a section including a known signal into a plurality of partial sections, a received signal sequence and a known signal corresponding to the partial section in each of the plurality of partial sections. A correlation apparatus is derived by integrating the derived partial correlation values after deriving the partial correlation values.

(Item 2-2)
An estimation unit for estimating a frequency offset with respect to a series of received signals input in the input unit;
The receiving apparatus according to (Item 2-1), wherein the first correlation processing unit adjusts the length of the partial section according to the value of the frequency offset estimated by the estimation unit.
(Item 2-3)
A measuring unit that measures the intensity of the series of received signals input at the input unit;
The combining unit adjusts the number of correlation values to be combined among the correlation value and a plurality of different correlation values according to the intensity value measured by the measuring unit (item) 2-1) or the receiving device according to (Item 2-2).

(Item 6-1)
A reception device for receiving a signal of an assigned time slot, in which frames formed by a plurality of time slots are continuous,
A first carrier recovery unit for recovering a carrier in an assigned time slot;
A second carrier reproducing unit for reproducing a carrier wave in a frame including other than the assigned time slot;
A selection unit that selects, for reception, a carrier wave reproduced in the second carrier wave reproduction unit or a carrier wave reproduced in the first carrier wave reproduction unit;
A receiving apparatus comprising:

(Item 6-2)
It further comprises a measurement unit that measures the degree of change in the wireless transmission path,
The receiving device according to (Item 6-1), wherein the selection unit performs selection according to a degree of change measured in the measurement unit.

It is a figure which shows the structure of the communication system which concerns on the Example of this invention. FIGS. 2A to 2D are diagrams showing a frame configuration used in the communication system of FIG. It is a figure which shows the structure of the receiver which concerns on the Example of this invention. It is a figure which shows the structure of the baseband part of FIG. It is a figure which shows the structure of the UW correlation part of FIG. It is a figure which shows the structure of the basic correlator of FIG. It is a figure which shows the structure of the 1st differential correlator of FIG. It is a figure which shows the structure of the frequency offset correction | amendment part of FIG. FIG. 5 is a diagram illustrating a configuration of a reproduction unit in FIG. 4. FIG. 10 is a diagram showing an outline of the operation of the playback unit in FIG. 9. FIG. 10 is a diagram illustrating another operation outline of the reproduction unit in FIG. 9.

Explanation of symbols

  10 base station apparatus, 12 satellite relay apparatus, 14 terminal apparatus, 20 receiving apparatus, 22 RF section, 24 baseband section, 26 control section, 30 reception band limiting filter, 32 UW correlation section, 34 symbol data extracting section, 36 modulation Component removal unit, 38 frequency offset correction unit, 40 carrier recovery unit, 42 UW identification unit, 44 frame synchronization processing unit, 46 symbol soft decision unit, 48 Turbo decoding unit, 50 descrambling unit, 54 CRC check unit, 58 CNR measurement Unit, 60 demodulator, 100 communication system.

Claims (3)

An input unit for inputting a series of received signals including a known signal in some sections;
A removing unit that removes a modulation component of a known signal from a series of sections in which a known signal is included in a series of received signals input in the input unit;
A deriving unit for deriving a plurality of types of phase differences between a sequence in which the modulation component of the known signal is removed in the removing unit and a sequence in which the sequence is delayed;
An estimation unit that estimates a frequency offset included in a series of received signals input in the input unit by combining the plurality of types of phase differences derived in the deriving unit while weighting;
A receiving apparatus comprising:   The receiving apparatus according to claim 1, wherein the estimation unit increases a magnitude of a weighting factor in weighting with respect to a phase difference of a delay amount close to an already estimated frequency offset amount. A measuring unit that measures the intensity of the series of received signals input at the input unit;
2. The estimation unit adjusts a distribution of weight coefficient sizes over a plurality of types of phase differences when weighting a plurality of types of phase differences according to the intensity measured by the measurement unit. Or the receiving apparatus of 2.

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