JP2008145648A - Display device and method of driving the same - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To suppress luminance unevenness due to variation in source voltage of a driving transistor caused by a driving current while simplifying a pixel circuit capable of correcting variation in luminance due to variation in element characteristic. <P>SOLUTION: A correcting transistor 512 is cascade-connected between a drain end D of the driving transistor 121 and a power supply line 105DSL, and a control signal generating section 520 which generates a control signal Vcont1 for controlling a gate end G is provided in common to each of pixel circuits P. On the basis of a driving current flowing to the driving transistor 121, a voltage drop is generated using operating resistance R512 of the correcting transistor 512, and a voltage V512D at the drain end D is varied based upon a driving current corresponding to a signal potential Vin. Influence of a voltage drop from a first potential Vcc_H of a p-type transistor 402 of an output circuit 400 due to the signal potential Vin is reduced at the drain end D of the driving transistor 121 to suppress lateral crosstalk during black window pattern display. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a display device having a pixel array portion in which pixel circuits (also referred to as pixels) having electro-optical elements (also referred to as display elements and light-emitting elements) are arranged in a matrix and a driving method thereof. More specifically, pixel circuits having electro-optic elements whose luminance changes depending on the magnitude of the drive signal as display elements are arranged in a matrix, each pixel circuit has an active element, and the active element is used for each pixel. The present invention relates to an active matrix display device in which display driving is performed and a driving method thereof.

  As a display element of a pixel, there is a display device using an electro-optical element whose luminance changes depending on an applied voltage or a flowing current. For example, a liquid crystal display element is a typical example of an electro-optical element whose luminance changes depending on an applied voltage, and an organic electroluminescence (Organic Electro Luminescence, Organic EL, Organic) (Light Emitting Diode, OLED; hereinafter referred to as “organic EL”) A typical example is an element. The organic EL display device using the latter organic EL element is a so-called self-luminous display device using an electro-optic element which is a self-luminous element as a pixel display element.

  An organic EL element is an electro-optical element utilizing a phenomenon that light is emitted when an electric field is applied to an organic thin film. Since the organic EL element can be driven with a relatively low applied voltage (for example, 10 V or less), the power consumption is low. Further, since the organic EL element is a self-luminous element that emits light by itself, an auxiliary illumination member such as a backlight that is required in a liquid crystal display device is not required, and the weight and thickness can be easily reduced. Furthermore, since the response speed of the organic EL element is very high (for example, about several μs), no afterimage occurs when displaying a moving image. Because of these advantages, development of flat self-luminous display devices using organic EL elements as electro-optical elements has been actively performed in recent years.

  By the way, in a display device using an electro-optic element such as a liquid crystal display device using a liquid crystal display element and an organic EL display device using an organic EL element, a simple (passive) matrix method and an active device are used as the driving method. A matrix method can be adopted. However, a simple matrix display device has problems such as a simple structure and a difficulty in realizing a large and high-definition display device.

  Therefore, in recent years, a pixel signal supplied to a light emitting element in a pixel has been converted into an active element, for example, an insulated gate field effect transistor (generally a thin film transistor (TFT)) as a switching transistor. Active matrix systems that are used and controlled have been actively developed.

  Here, when the electro-optic element in the pixel circuit is caused to emit light, an input image signal supplied via the video signal line is switched by a switching transistor and a storage capacitor (control input terminal) provided at the gate end (control input terminal) of the drive transistor. The drive signal corresponding to the input image signal is supplied to the electro-optical element.

  In a liquid crystal display device using a liquid crystal display element as an electro-optical element, the liquid crystal display element is a voltage-driven element, and thus the liquid crystal display element is driven with a voltage signal itself corresponding to an input image signal taken into the storage capacitor. On the other hand, in an organic EL display device using an organic EL element as an electro-optical element, the organic EL element is a current-driven element, and therefore, a drive signal (voltage signal) corresponding to an input image signal taken into the storage capacitor. ) Is converted into a current signal by the driving transistor, and the driving current is supplied to the organic EL element.

  In a current-driven electro-optical element, typically an organic EL element, the light emission luminance varies depending on the drive current value. Therefore, in order to emit light with stable luminance, it is important to supply a stable drive current to the electro-optical element. For example, driving methods for supplying a driving current to the organic EL element can be broadly classified into a constant current driving method and a constant voltage driving method (this is a well-known technique, and publicly known literature is not presented here).

  Since the voltage-current characteristic of the organic EL element has a large inclination, when constant voltage driving is performed, a slight voltage variation or a variation in element characteristics causes a large current variation, resulting in a large luminance variation. Therefore, generally, constant current driving using a driving transistor in a saturation region is used. Of course, even with constant current driving, if there is a current variation, luminance variations will be caused, but if the current variation is small, only small luminance variations will occur.

  In other words, even in the constant current driving method, the driving signal written and held in the holding capacitor according to the input image signal may be constant because the light emission luminance of the electro-optic element is unchanged. It becomes important. For example, in order that the light emission luminance of the organic EL element remains unchanged, it is important that the drive current corresponding to the input image signal is constant.

  However, the threshold voltage and mobility of an active element (driving transistor) that drives the electro-optical element vary due to process variations. In addition, characteristics of electro-optical elements such as organic EL elements vary with time. If there is such a variation in characteristics of the active element for driving or a characteristic variation of the electro-optical element, even the constant current driving method affects the light emission luminance.

  Therefore, in order to uniformly control the light emission luminance over the entire screen of the display device, a mechanism for correcting the luminance variation caused by the characteristic variation of the driving active element and the electro-optical element described above in each pixel circuit. Various studies have been made.

JP 2006-215213 A

  For example, in the mechanism described in Patent Document 1, as a pixel circuit for an organic EL element, a threshold correction function for making the drive current constant even when the threshold voltage of the drive transistor varies or changes over time, In order to keep the driving current constant even when the mobility-correction function for making the driving current constant even when the mobility of the organic EL element varies or changes with time, or when the current-voltage characteristic of the organic EL element changes with time A bootstrap function has been proposed.

  However, the mechanism described in Patent Document 1 requires a wiring for supplying a correction potential, a correction switching transistor, and a switching pulse for driving the wiring. A 5TR drive configuration using two transistors is employed, and the configuration of the pixel circuit is complicated. Since there are many components of a pixel circuit, it becomes a hindrance to high definition of a display apparatus. As a result, the 5TR drive configuration makes it difficult to apply to a display device used in a small electronic device such as a portable device (mobile device).

  For this reason, there is a demand for development of a method for suppressing luminance change due to variation in element characteristics while simplifying the pixel circuit. At this time, it should be taken into consideration that a new problem that does not occur in the configuration of the 5TR drive does not occur with the simplification.

  The present invention has been made in view of the above circumstances, and it is a general object of the present invention to provide a display device and a driving method thereof capable of increasing the definition of the display device by simplifying the pixel circuit.

  It is particularly desirable to provide a mechanism that can reduce the influence of the operation of driving the pixel circuit on the image quality (especially suppressing luminance unevenness) while simplifying the pixel circuit.

  Another object of the present invention is to provide a mechanism capable of suppressing a change in luminance due to variations in element characteristics when the pixel circuit is simplified.

  One embodiment of a display device according to the present invention is a display device that emits electro-optic elements in a pixel circuit based on a video signal. First, in a pixel circuit arranged in a matrix in a pixel array unit, At least a drive transistor for generating a drive current, a holding capacitor connected between a control input terminal (gate terminal is a typical example) and an output terminal (source terminal is a typical example) of the drive transistor, and a drive transistor connected to an output terminal of the drive transistor And a sampling transistor for writing information corresponding to the signal potential in the video signal to the storage capacitor. In this pixel circuit, the electro-optic element is caused to emit light by generating a drive current based on information held in the holding capacitor by the drive transistor and flowing it through the electro-optic element.

  Since the sampling transistor writes information corresponding to the signal potential to the holding capacitor, the sampling transistor takes in the signal potential at its input terminal (source terminal is a typical example) and is connected to its output terminal (drain terminal is a typical example) Information corresponding to the signal potential is written in the storage capacitor. Of course, the output terminal of the sampling transistor is also connected to the control input terminal of the drive transistor.

  Note that the connection configuration of the pixel circuit shown here is the most basic configuration, and the pixel circuit only needs to include at least each of the above-described components. May be included. Further, the “connection” is not limited to being directly connected, but may be connected via other components.

  For example, a change such as interposing a switching transistor or a functional unit having a certain function may be added between the connections as necessary. Typically, in order to dynamically control the display period (in other words, non-light emission time), a switching transistor is provided between the output terminal of the driving transistor and the electro-optical element, or the power supply terminal of the driving transistor. There is a case where it is arranged between a power supply line which is a wiring for power supply (a drain end is a typical example).

  Even in a pixel circuit having such a modified mode, as long as the configuration and operation described in this section (means for solving the problem) can be realized, these modified modes are also displayed according to the present invention. 1 is a pixel circuit that implements an embodiment of an apparatus.

  Further, in the peripheral portion for driving the pixel circuit, for example, the pixel circuit is line-sequentially scanned by sequentially controlling the sampling transistors in the horizontal period, and the signal potential of the video signal is set to each holding capacitor for one row. Write scanning unit for writing the corresponding information, and the first potential for use in flowing a driving current to the electro-optic element to each pixel circuit for one row in accordance with the line sequential scanning in the writing scanning unit A control unit including a drive scanning unit having an output circuit that switches and outputs a first potential from a supply end and a second potential from a supply end for a second potential different from the first potential is provided.

  More preferably, the control unit includes a horizontal driving unit that controls the video signal that is switched between the reference potential and the signal potential within each horizontal period in accordance with the line sequential scanning in the writing scanning unit to be supplied to the sampling transistor. Provide.

  Here, it is preferable that the control unit supplies a voltage corresponding to the first potential used to flow the drive current to the electro-optic element to the power supply end of the drive transistor, and sets the reference potential in the video signal to the sampling transistor. Control is performed so as to perform a threshold correction operation for holding a voltage corresponding to the threshold voltage of the driving transistor in the holding capacitor by turning on the sampling transistor in the supplied time zone.

  This threshold value correcting operation may be repeatedly executed at a plurality of horizontal periods preceding the writing of the signal potential to the storage capacitor as necessary. Here, “as necessary” means a case where a voltage corresponding to the threshold voltage of the driving transistor cannot be sufficiently held in the storage capacitor in the threshold correction period within one horizontal cycle. By executing the threshold correction operation a plurality of times, a voltage corresponding to the threshold voltage of the drive transistor is reliably held in the holding capacitor.

  More preferably, prior to the threshold value correcting operation, the control unit supplies a voltage corresponding to the second potential to the power supply end of the drive transistor and a reference potential at the input end (source end is a typical example) of the sampling transistor. The threshold transistor correction preparation operation (discharge operation or initialization operation) in which the sampling transistor is turned on in the time zone in which is supplied to set the control input terminal of the drive transistor to the reference potential and the output terminal to the second potential. Control to execute.

  More preferably, after the threshold correction operation, the control unit supplies the voltage corresponding to the first potential to the driving transistor and causes the sampling transistor to conduct in a time zone in which the reference potential is supplied to the sampling transistor, thereby holding the storage capacitor. When the signal potential information is written to the signal, the correction for the mobility of the driving transistor is controlled to be added to the signal written to the storage capacitor.

  At this time, the sampling transistor may be turned on at a predetermined position within a time zone in which the reference potential is supplied to the sampling transistor for a period shorter than the time zone.

  More preferably, when the signal potential is written to the storage capacitor, the control unit makes the sampling transistor non-conductive to stop the supply of the video signal to the control input terminal of the drive transistor, and the potential of the output terminal of the drive transistor The bootstrap operation in which the potential of the control input terminal is interlocked with the fluctuation and the voltage at the control input terminal and the output terminal can be maintained constant, and the temporal variation correction operation of the electro-optic element is realized.

  Here, as a characteristic matter in the embodiment of the display device according to the present invention, the influence of the voltage drop in the output circuit due to the signal potential between the supply terminal for the first potential and the power supply terminal of the driving transistor. Is provided at the power supply end of the driving transistor. The power supply potential fluctuation suppressing unit employs a configuration that uses a voltage drop caused by the operating current of the driving transistor flowing through its own operating resistance. The power supply potential fluctuation suppressing unit generates a voltage drop based on the drive current flowing through the drive transistor, and adjusts the potential of the power supply end of the drive transistor by this voltage drop.

  For example, for each row, an output circuit and a power supply line for connecting the output circuit and the pixel circuit for one row are provided. As the power supply potential fluctuation suppressing unit, a current-voltage conversion unit is provided between the supply terminal for the first potential and the power supply terminal of the driving transistor.

  By generating a voltage drop by the product of the operating resistance of the current-voltage converter and the drive current flowing in the drive transistor, the voltage at the power supply end of the drive transistor changes according to the drive current corresponding to the signal potential. Thus, the influence of the voltage drop of the first potential in the output circuit due to the signal potential is suppressed at the power supply end of the drive transistor.

  Preferably, the current-voltage conversion unit is configured by providing a correction transistor arranged in a cascode connection between the power supply line and the power supply end of the drive transistor.

  A constant voltage is supplied to the gate terminal of the correction transistor regardless of the signal potential.

  Alternatively, a control signal generation unit that generates a control signal for controlling the operating resistance of the current-voltage conversion unit is provided as the power supply potential fluctuation suppression unit. For example, a control signal is supplied to the gate terminal of the correction transistor to set the operating resistance of the correction transistor to an appropriate value.

  Alternatively, a control signal generation unit that generates a control signal for controlling the operating resistance of the current-voltage conversion unit based on the signal potential is provided as the power supply potential fluctuation suppression unit. For example, a control signal is supplied to the gate terminal of the correction transistor, and the operating resistance of the correction transistor is set to an appropriate value based on the signal potential. In this case, the control signal generation unit may generate a control signal for controlling the operating resistance of the current-voltage conversion unit based on the signal potential (for example, average value, median value, etc.) for one row.

  According to the embodiment of the present invention, the influence of the voltage drop in the output circuit due to the signal potential between the first potential supply end and the power supply end of the drive transistor is suppressed at the power supply end of the drive transistor. A power supply potential fluctuation suppressing part is provided.

  By causing the power supply potential fluctuation suppressing unit to generate a voltage drop based on the drive current flowing in the drive transistor, the voltage at the power supply end of the drive transistor is changed based on the drive current corresponding to the signal potential. Thereby, the influence of the voltage drop of the first potential in the output circuit due to the signal potential is suppressed at the power supply end of the drive transistor.

  The voltage at the power supply end of the drive transistor is a potential that is lower than the first potential at the supply end for the first potential by the voltage drop caused by the power supply potential fluctuation suppressing unit. Furthermore, the amount of voltage drop is large when the drive current flowing through the drive transistor (effective light emission current of the organic EL element) is large, and the amount of voltage drop is small when the drive current is small. As a result, regardless of the presence or absence of the window pattern, the light emission luminance is adjusted, and horizontal crosstalk, which is an example of luminance unevenness, can be prevented and an image with good image quality can be obtained.

  Further, in an active matrix display device using a current-driven electro-optic element such as an organic EL element in a pixel circuit, if each pixel circuit has at least a threshold correction function of a drive transistor, the threshold voltage varies. Therefore, a display device with good image quality can be realized. Desirably, a higher quality image can be obtained by providing a mobility correction function of the drive transistor and a temporal variation correction function (bootstrap operation) of the electro-optic element.

  By correcting the threshold fluctuation of the driving transistor with the threshold correction function or correcting the mobility fluctuation of the driving transistor with the mobility correction function, the light emission luminance is kept constant without being affected by these fluctuations and variations. Because it can. Also, even if the current-voltage characteristics of the electro-optic element change with time due to the bootstrap operation of the storage capacitor during light emission, the potential difference between the control input terminal and the output terminal of the drive transistor is kept constant by the bootstrap storage capacitor. This is because a constant light emission luminance can always be maintained.

  Here, in realizing the threshold correction function and the threshold correction preparation function (initialization function) preceding it, the power supply end of the drive transistor is transitioned between the first potential and the second potential, that is, the power supply voltage is switched to the switching pulse. Use effectively as a function. That is, if the power supply voltage supplied to the drive transistor of each pixel circuit is used as a switching pulse in order to incorporate the threshold correction function, a switching transistor for threshold correction and a scanning line for controlling the control input terminal thereof become unnecessary.

  As a result, the number of constituent elements and the number of wirings of the pixel circuit can be greatly reduced, the pixel array portion can be reduced, and high definition of the display device can be easily achieved. While simplifying the pixel circuit, it is possible to realize a function of correcting a luminance change due to a variation in element characteristics.

  Since the number of elements and the number of wirings are small, it is suitable for high definition, and a small display device that requires high definition display can be easily realized.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

<Overview of display device>
FIG. 1 is a block diagram showing an outline of a configuration of an active matrix display device which is an embodiment of a display device according to the present invention. In this embodiment, for example, an organic EL element is used as a display element (electro-optic element, light emitting element) of a pixel, a polysilicon thin film transistor (TFT) is used as an active element, and an organic film is formed on a semiconductor substrate on which a thin film transistor is formed. A case where the present invention is applied to an active matrix organic EL display (hereinafter referred to as “organic EL display device”) formed with EL elements will be described as an example.

  In the following, an organic EL element will be specifically described as an example of a pixel display element. However, this is merely an example, and the target display element is not limited to an organic EL element. In general, all embodiments described later can be applied to all display elements that emit light by current drive.

  As shown in FIG. 1, the organic EL display device 1 has an aspect ratio in which a pixel circuit (also referred to as a pixel) 110 having a plurality of organic EL elements (not shown) as display elements has a display aspect ratio. A display panel unit 100 arranged so as to constitute an effective video area of X: Y (for example, 9:16), and a drive that is an example of a panel control unit that generates various pulse signals for driving and controlling the display panel unit 100 A signal generation unit 200 and a video signal processing unit 300 are provided. The drive signal generation unit 200 and the video signal processing unit 300 are built in a one-chip IC (Integrated Circuit).

  As shown in the figure, the product form is provided as an organic EL display device 1 in the form of a module (composite part) including all of the display panel unit 100, the drive signal generation unit 200, and the video signal processing unit 300. For example, the organic EL display device 1 can be provided only by the display panel unit 100. Such an organic EL display device 1 is used in a display unit of a portable music player or other electronic device using a recording medium such as a semiconductor memory, a mini disk (MD), or a cassette tape.

  The display panel unit 100 includes a pixel array unit 102 in which pixel circuits P are arranged in a matrix of n rows × m columns on a substrate 101, a vertical drive unit 103 that scans the pixel circuits P in the vertical direction, and pixels A horizontal driving unit (also referred to as a horizontal selector or a data line driving unit) 106 that scans the circuit P in the horizontal direction, a terminal unit (pad unit) 108 for external connection, and the like are integrated. That is, peripheral drive circuits such as the vertical drive unit 103 and the horizontal drive unit 106 are formed on the same substrate 101 as the pixel array unit 102.

  The vertical driving unit 103 (the writing scanning unit 104 and the driving scanning unit 105) and the horizontal driving unit 106 control writing of the signal potential to the holding capacitor, threshold correction operation, mobility correction operation, and bootstrap operation. A control unit 109 is configured.

  The vertical drive unit 103 includes, for example, a write scan unit (write scanner WS; Write Scan) 104 and a drive scan unit (drive scanner DS; Drive Scan) 105 that functions as a power supply scanner having power supply capability.

  For example, the pixel array unit 102 is driven by the writing scanning unit 104 and the driving scanning unit 105 from one side or both sides in the horizontal direction shown in the figure, and driven by the horizontal driving unit 106 from one side or both sides in the vertical direction shown in the figure. It has come to be.

  Various pulse signals are supplied to the terminal unit 108 from the drive signal generation unit 200 arranged outside the organic EL display device 1. Similarly, the video signal Vsig is supplied from the video signal processing unit 300.

  As an example, necessary pulse signals such as shift start pulses SPDS, SPWS and vertical scanning clocks CKDS, CKWS, which are examples of vertical write start pulses, are supplied as pulse signals for vertical driving. In addition, necessary pulse signals such as a horizontal start pulse SPH and a horizontal scanning clock CKH, which are examples of horizontal write start pulses, are supplied as pulse signals for horizontal driving.

  Each terminal of the terminal unit 108 is connected to the vertical driving unit 103 and the horizontal driving unit 106 via a wiring 109. For example, each pulse supplied to the terminal unit 108 is internally adjusted to a voltage level by a level shifter unit (not shown) as necessary, and then supplied to each unit of the vertical driving unit 103 and the horizontal driving unit 106 via a buffer. Supplied.

  Although the pixel array unit 102 is not shown in the drawing (details will be described later), pixel circuits P in which pixel transistors are provided with respect to an organic EL element as a display element are two-dimensionally arranged in a matrix form. On the other hand, scanning lines are wired for each row, and signal lines are wired for each column.

  For example, in the pixel array portion 102, a scanning line (gate line) 104WS and a video signal line (data line) 106HS are formed. An organic EL element (not shown) and a thin film transistor (TFT) for driving the organic EL element are formed at the intersection of the two. A pixel circuit P is configured by a combination of an organic EL element and a thin film transistor.

  Specifically, for each pixel circuit P arranged in a matrix, the write scanning lines 104WS_1 to 104WS_n for n rows driven by the write scanning unit 104 with the write drive pulse WS and the drive scanning unit Power supply lines 105DSL_1 to 105DSL_n for n rows driven by the power supply drive pulse DSL by 105 are wired for each pixel row.

  The writing scanning unit 104 and the driving scanning unit 105 sequentially select the pixel circuits P via the writing scanning line 104WS and the power supply line 105DSL based on the vertical driving system pulse signal supplied from the driving signal generation unit 200. To do. The horizontal driving unit 106 samples a predetermined potential in the video signal Vsig to the selected pixel circuit P via the video signal line 106HS based on the horizontal driving system pulse signal supplied from the driving signal generation unit 200. To write to the holding capacity.

  In the organic EL display device 1 of the present embodiment, only line-sequential driving is possible, and the writing scanning unit 104 and the driving scanning unit 105 of the vertical driving unit 103 are pixel-sequentially (that is, in units of rows). The horizontal drive unit 106 scans the unit 102 and writes the image signal to the pixel array unit 102 simultaneously for one horizontal line in synchronization with the scanning.

  For example, the horizontal drive unit 106 is configured to include a driver circuit that simultaneously turns on the switches that are omitted from the illustration provided on the video signal lines 106HS of all the columns in order to support line-sequential driving, and the video signal processing unit A switch that omits the illustration provided on the video signal lines 106HS of all the columns in order to simultaneously write the pixel signals input from 300 to all the pixel circuits P for one line of the row selected by the vertical driving unit 103. Turn on all at once.

  Each unit of the vertical driving unit 103 is configured by a combination of logic gates (including latches) in order to support line sequential driving, and selects each pixel circuit P of the pixel array unit 102 in units of rows. FIG. 1 shows a configuration in which the vertical drive unit 103 is disposed only on one side of the pixel array unit 102. However, a configuration in which the vertical drive unit 103 is disposed on both the left and right sides with the pixel array unit 102 interposed therebetween is employed. Is also possible.

  Similarly, FIG. 1 shows a configuration in which the horizontal drive unit 106 is disposed only on one side of the pixel array unit 102, but a configuration in which the horizontal drive unit 106 is disposed on both upper and lower sides with the pixel array unit 102 interposed therebetween is employed. It is also possible.

<Pixel circuit>
FIG. 2 is a diagram showing a first comparative example for the pixel circuit P of the present embodiment. Note that a vertical driving unit 103 and a horizontal driving unit 106 provided on the periphery of the pixel circuit P on the substrate 101 of the display panel unit 100 are also shown. FIG. 3 is a timing chart for explaining the operation of the pixel circuit P of the first comparative example shown in FIG. FIG. 4 is a diagram for explaining the influence of characteristic variations of the organic EL element 127 and the drive transistor 121 on the drive current Ids, and FIG. 4A is a diagram for explaining the concept of the improvement method.

  FIG. 5 is a diagram illustrating a second comparative example for the pixel circuit P of the present embodiment. Note that a vertical driving unit 103 and a horizontal driving unit 106 provided on the periphery of the pixel circuit P on the substrate 101 of the display panel unit 100 are also shown.

  Although details will be described later, the pixel circuit P of the present embodiment basically adopts the same mechanism as that of the second comparative example shown in FIG. A variation on the supply format is added. A basic control operation for driving a pixel circuit P of the present embodiment to be described later is the same as that of the second comparative example. Therefore, first, the configuration and operation of the pixel circuit P of the second comparative example serving as the base of the pixel circuit P of the present embodiment will be described in detail.

  The pixel circuit P of the second comparative example is characterized in that a drive transistor is basically composed of an n-channel thin film field effect transistor. In addition, a circuit for suppressing fluctuations in the drive current Ids to the organic EL element due to deterioration over time of the organic EL element, that is, driving by correcting a change in current-voltage characteristics of the organic EL element which is an example of an electro-optical element The present invention is characterized in that a drive signal stabilizing circuit for maintaining the current Ids constant is provided. In addition, the organic EL element is characterized in that it has a function of making the drive current constant even when the current-voltage characteristics of the organic EL element change with time.

  If a driving transistor can be formed of an n-channel transistor instead of a p-channel transistor, a conventional amorphous silicon (a-Si) process can be used in transistor formation. Thereby, the cost of the transistor substrate can be reduced, and the development of the pixel circuit P having such a configuration is expected.

  MOS transistors are used as the transistors including the drive transistor. In this case, for the drive transistor, the gate end is handled as the control input end, and either the source end or the drain end (here, the source end) is handled as the output end, and the other is the power supply end (here, the drain end). ).

<Pixel Circuit of Comparative Example: First Example>
First, as a comparative example for explaining the characteristics of the pixel circuit P of the second comparative example, the pixel circuit P of the first comparative example shown in FIG. 2 will be described. The organic EL display device 1 including the pixel circuit P of the first comparative example in the pixel array unit 102 is referred to as an organic EL display device 1 of the first comparative example.

  The pixel circuit P of the first comparative example is the same as that of the present embodiment in that the drive transistor is basically composed of an n-channel thin film field effect transistor, but the drive current due to deterioration with time of the organic EL element 127. There is no drive signal stabilization circuit for preventing the influence on Ids.

  Specifically, the pixel circuit P includes an n-channel driving transistor 121 and a sampling transistor 125, and an organic EL element 127 that is an example of an electro-optical element that emits light when a current flows. In general, since the organic EL element 127 has a rectifying property, it is represented by a diode symbol. The organic EL element 127 has a parasitic capacitance Cel. In the figure, this parasitic capacitance Cel is shown in parallel with the organic EL element 127.

  The drive transistor 121 has a drain end D connected to the power supply line DSL that supplies the first power supply potential, a source end (output end) S connected to the anode end A of the organic EL element 127, and the organic EL element 127. The cathode terminal K is connected to a ground wiring Vcath (GND) common to all pixels for supplying a reference potential.

  The sampling transistor 125 has a source terminal S connected to the video signal line HS, a drain terminal (power supply terminal) D connected to a gate terminal (control input terminal) G of the driving transistor 121, and the connection point and the second power supply potential. A storage capacitor 120 is provided between the reference line and the reference line. The sampling transistor 125 may have a connection mode in which the source terminal S and the drain terminal D are reversed. In this configuration, the reference line for supplying the second power supply potential is the same as the ground wiring Vcath for supplying the reference potential for the organic EL element 127 as shown in the figure, but may be a wiring for supplying another potential. .

  Although not shown, in the case of a 3TR type in which a light emission control transistor for controlling the light emission period is added, for example, the source end of the drive transistor 121 is connected to the drain end D of the n-channel type light emission control transistor to emit light. The source terminal S of the control transistor is connected to the anode terminal of the organic EL element 127.

  In such a pixel circuit P, regardless of whether or not a light emission control transistor is provided, when driving the organic EL element 127, the drain end D side of the drive transistor 121 is connected to the first power supply potential, and the source end S is organic. By connecting to the anode end A side of the EL element 127, a source follower circuit is formed as a whole.

  The timing chart of FIG. 3 for explaining the operation of the pixel circuit P of the first comparative example shown in FIG. 2 is effective in the potential of the video signal Vsig supplied from the signal line HS (hereinafter also referred to as video signal line potential). This represents an operation of sampling a potential (referred to as a signal potential) in a period and setting an organic EL element 127 which is an example of a light emitting element to a light emitting state.

  In the time zone (t1 to t4) in which the video signal line 106HS is at the signal potential that is the effective period of the video signal Vsig, the potential of the write scanning line WS transitions to a high level (t2), thereby performing n-channel sampling. The transistor 125 is turned on and charges the storage capacitor 120 with the video signal line potential supplied from the signal line HS. As a result, the potential of the gate terminal G (gate potential Vg) of the drive transistor 121 starts to rise and starts to flow a drain current. Therefore, the anode potential of the organic EL element 127 rises and light emission starts.

  Thereafter, when the write drive pulse WS transitions to a low level (t3), the holding capacitor 120 holds the video signal line potential at that time, that is, the potential (signal potential) in the effective period within the potential of the video signal Vsig. Is done. As a result, the gate potential Vg of the drive transistor 121 becomes constant, and the light emission luminance is kept constant until the next frame (or field). Timing t2 to t3 is a sampling period of the video signal Vsig, and timing after timing t3 is a holding period.

  Here, in the pixel circuit P of the first comparative example, the potential of the source terminal S of the drive transistor 121 (source potential Vs) is determined by the operating point of the drive transistor 121 and the organic EL element 127, and the voltage value is determined by the drive transistor. The gate potential Vg of 121 has a different value.

  Here, generally, the drive transistor 121 is driven in a saturation region. Therefore, the current flowing between the drain end and the source of the transistor operating in the saturation region is Ids, the mobility is μ, the channel width (gate width) is W, the channel length (gate length) is L, and the gate capacitance (per unit area). When the gate oxide film capacitance) is Cox and the threshold voltage of the transistor is Vth, the drive transistor 121 is a constant current source having a value represented by the following equation (1). As apparent from the equation (1), the drain current Ids of the transistor is controlled by the gate-source voltage Vgs in the saturation region.

<IV characteristics of organic EL element>
Here, in the current-voltage (IV) characteristics of the organic EL element shown in FIG. 4 (1), the curve indicated by the solid line indicates the characteristic in the initial state, and the curve indicated by the broken line indicates the characteristic after change with time. ing. Generally, the IV characteristic of an organic EL element deteriorates with time as shown in the graph.

  In the pixel circuit P of the first comparative example, the operating point changes due to deterioration with time, and the source potential Vs of the drive transistor 121 changes even when the same gate potential Vg is applied. As a result, the gate-source voltage Vgs of the drive transistor 121 changes. As is apparent from the characteristic equation (1), when the gate-source voltage Vgs varies, the drive current Ids varies even if the gate potential Vg is constant, and the current value flowing through the organic EL element 127 also varies. . When the IV characteristic of the organic EL element 127 changes in this way, the emission luminance of the organic EL element 127 changes over time in the pixel circuit P of the first comparative example having the source follower configuration shown in FIG. End up.

  In a simple circuit using an n-channel type as the driving transistor 121, the source terminal S is connected to the organic EL element 127 side, so that the gate-source voltage Vgs changes as the organic EL element 127 changes over time. As a result, the amount of current flowing through the organic EL element 127 changes, and as a result, the light emission luminance changes.

  A variation in the anode potential of the organic EL element 127 due to a change in characteristics of the organic EL element 127, which is an example of the light emitting element, appears as a variation in the gate-source voltage Vgs of the driving transistor 121, and the drain current (driving current Ids). Cause fluctuations. Variations in the drive current due to this cause appear as variations in light emission luminance for each pixel circuit P, resulting in degradation of image quality.

  On the other hand, as will be described in detail later, a circuit configuration and a driving timing for realizing a bootstrap function in which the potential Vg of the gate terminal G is interlocked with the fluctuation of the potential Vs of the source terminal S of the driving transistor 121. Thus, even if there is an anode potential fluctuation (that is, a source potential fluctuation) of the organic EL element 127 due to a change in characteristics of the organic EL element 127 with time, the gate potential Vg is changed so as to cancel the fluctuation, thereby reducing the screen luminance. Uniformity can be ensured. The bootstrap function can improve the deterioration correction capability of a current-driven light emitting element typified by an organic EL element.

  Of course, in the bootstrap function, the light emission current Iel begins to flow through the organic EL element 127 at the start of light emission, and as a result, the anode-cathode voltage Vel rises until it becomes stable. It also functions when the source potential Vs of the drive transistor 121 varies with the variation of the voltage Vel.

<Vgs-Ids characteristics of drive transistor>
In addition, due to variations in the manufacturing process of the drive transistor 121, there are variations in characteristics such as threshold voltage and mobility for each pixel circuit P. Even when the driving transistor 121 is driven in the saturation region, even if the same gate potential is applied to the driving transistor 121 due to this characteristic variation, the drain current (driving current Ids) varies for each pixel circuit P, and the emission luminance is reduced. Appears as variations.

  For example, FIG. 4B is a diagram illustrating the voltage-current (Vgs-Ids) characteristics focusing on the threshold variation of the drive transistor 121. A characteristic curve is given for each of the two drive transistors 121 having different threshold voltages of Vth1 and Vth2.

  As described above, the drain current Ids when the driving transistor 121 operates in the saturation region is expressed by the characteristic formula (1). As apparent from the characteristic equation (1), when the threshold voltage Vth varies, the drain current Ids varies even if the gate-source voltage Vgs is constant. In other words, if no countermeasure is taken against the variation in the threshold voltage Vth, the drive current corresponding to Vgs becomes Ids1 when the threshold voltage is Vth1, as shown in FIG. The drive current Ids2 corresponding to the same gate voltage Vgs when is Vth2 is different from Ids1.

  FIG. 4 (3) is a diagram showing voltage-current (Vgs-Ids) characteristics focusing on the mobility variation of the drive transistor 121. Characteristic curves are given for two drive transistors 121 having different mobility in μ1 and μ2.

  As apparent from the characteristic equation (1), when the mobility μ varies, the drain current Ids varies even when the gate-source voltage Vgs is constant. That is, if no countermeasure is taken against the variation in mobility μ, the drive current corresponding to Vgs becomes Ids1 when the mobility is μ1, as shown in FIG. When I is μ2, the drive current Ids2 corresponding to the same gate voltage Vgs is different from Ids1.

  On the other hand, by setting the drive timing (details will be described later) to realize the threshold value correction function and the mobility correction function, as understood from each diagram of FIG. Uniformity of screen brightness can be ensured.

<Concept of threshold correction and mobility correction>
Although details will be described later in the threshold value correcting operation and the mobility correcting operation of the present embodiment, the drain-source current can be expressed by expressing the gate-source voltage Vgs at the time of light emission as “Vin + Vth−ΔV”. Ids is not dependent on variations or fluctuations in the threshold voltage Vth, and is not dependent on variations or fluctuations in the mobility μ. As a result, even if the threshold voltage Vth and the mobility μ vary depending on the manufacturing process, the drive current Ids does not vary, and the light emission luminance of the organic EL element 127 does not vary.

  For example, in each diagram of FIG. 4A, the current-voltage characteristics of the drive transistor 121, the signal potential Vin on the horizontal axis, and the drive current Ids on the vertical axis, the threshold voltage Vth is relatively low and the mobility μ is compared. Pixel circuit Pa (solid curve) composed of a relatively large drive transistor 121 and, conversely, pixel circuit Pb (dotted curve) composed of a drive transistor 121 having a relatively high threshold voltage Vth and a relatively low mobility μ. For each, the characteristic curves are listed.

  FIG. 4A (1) shows a case where neither threshold correction nor mobility correction is executed. At this time, since the threshold voltage Vth and the mobility μ are not corrected at all in the pixel circuit Pa and the pixel circuit Pb, the difference in the threshold voltage Vth and the mobility μ causes a large difference in Vin-Ids characteristics. Therefore, even if the same signal potential Vin is applied, the drive current Ids, that is, the light emission luminance differs, and the uniformity of the screen luminance cannot be obtained.

  FIG. 4A (2) shows a case where threshold correction is performed while mobility correction is not performed. At this time, the difference in threshold voltage Vth between the pixel circuit Pa and the pixel circuit Pb is cancelled. However, the difference in mobility μ appears as it is. Therefore, a difference in mobility μ appears remarkably in a region where the signal potential Vin is high (that is, a region where the luminance is high), and the luminance is different even in the same gradation. Specifically, at the same gradation (same signal potential Vin), the luminance (driving current Ids) of the pixel circuit Pa having a high mobility μ is high, and the luminance of the pixel circuit Pb having a low mobility μ is low.

  FIG. 4A (3) shows a case where both threshold value correction and mobility correction are executed. The difference between the threshold voltage Vth and the mobility μ is completely corrected. As a result, the Vin-Ids characteristics of the pixel circuit Pa and the pixel circuit Pb match. Therefore, the luminance (Ids) becomes the same level in all the gradations (signal potential Vin), and the uniformity of the screen luminance (uniformity) is remarkably improved.

  FIG. 4A (4) shows a case where threshold value correction and mobility correction are both performed, but the threshold voltage Vth is not sufficiently corrected. For example, a case where a voltage corresponding to the threshold voltage Vth of the drive transistor 121 cannot be sufficiently held in the storage capacitor 120 in one threshold correction operation is an example. At this time, since the difference in threshold voltage Vth is not removed, there is a difference in luminance (drive current Ids) in the low gradation region between the pixel circuit Pa and the pixel circuit Pb. Therefore, when the correction of the threshold voltage Vth is insufficient, luminance unevenness appears at a low gradation and the image quality is impaired.

<Pixel Circuit of Comparative Example: Second Example>
In the pixel circuit P of the first comparative example shown in FIG. 2, a circuit (bootstrap circuit) for preventing a drive current fluctuation due to deterioration with time of the organic EL element 127 is mounted, and a characteristic fluctuation (threshold voltage fluctuation or mobility) of the drive transistor 121 is mounted. The pixel circuit P of the second comparative example shown in FIG. 5 employs a driving method that prevents fluctuations in the driving current due to variation. The organic EL display device 1 including the pixel circuit P of the second comparative example in the pixel array unit 102 is referred to as an organic EL display device 1 of the second comparative example.

  The pixel circuit P of the second comparative example adopts a 2TR drive configuration that uses one switching transistor (sampling transistor 125) for scanning in addition to the drive transistor 121, and includes a power supply drive pulse DSL for controlling each switching transistor and By setting the on / off timing of the write drive pulse WS, the influence of the deterioration of the organic EL element 127 over time and the change in characteristics of the drive transistor 121 (for example, variations and fluctuations in threshold voltage, mobility, etc.) on the drive current Ids are prevented. Characterized by points.

  In addition, since it has a 2TR drive configuration and the number of elements and wirings is small, high definition can be achieved, and in addition, sampling can be performed without deterioration of the video signal Vsig, so that good image quality can be obtained.

  A major difference in configuration with respect to the first comparative example shown in FIG. 2 is that the connection mode of the storage capacitor 120 is modified, and the drive signal is constant as a circuit that prevents fluctuations in the drive current due to deterioration over time of the organic EL element 127. This is in the configuration of a bootstrap circuit which is an example of a circuit. As a method of suppressing the influence on the drive current Ids due to the characteristic variation of the drive transistor 121 (for example, variation or fluctuation in threshold voltage, mobility, etc.), this is dealt with by devising the drive timing of each of the transistors 121 and 125.

  Specifically, the pixel circuit P of the second comparative example includes a storage capacitor 120, an n-channel driving transistor 121, and an n-channel sampling transistor 125 to which an active H (high) write driving pulse WS is supplied. And an organic EL element 127 which is an example of an electro-optical element (light emitting element) that emits light when a current flows.

  The storage capacitor 120 is connected between the gate terminal G (node ND122) of the driving transistor 121 and the source terminal S, and the source terminal S of the driving transistor 121 is directly connected to the anode terminal A of the organic EL element 127. The cathode terminal K of the organic EL element 127 is set to a cathode potential Vcath as a reference potential. This cathode potential Vcath is connected to the ground wiring Vcath (GND) common to all the pixels for supplying the reference potential as in the first comparative example shown in FIG.

  The drain terminal D of the drive transistor 121 is connected to a power supply line 105DSL from the drive scanning unit 105 that functions as a power scanner. The power supply line 105DSL is characterized in that the power supply line 105DSL itself has a power supply capability to the drive transistor 121.

  Specifically, the drive scanning unit 105 supplies the drain terminal D of the drive transistor 121 by switching between the first voltage Vcc_H on the high voltage side corresponding to the power supply voltage and the second voltage Vcc_L on the low voltage side. A power supply voltage switching circuit is provided. The second potential Vcc_L is a potential that is sufficiently lower than the reference potential Vo of the video signal Vsig in the video signal line 106HS.

  Specifically, the gate-source voltage Vgs of the driving transistor 121 (difference between the gate potential Vg and the source potential Vs) is larger than the threshold voltage Vth of the driving transistor 121. Two potential Vcc_L is set. The reference potential Vo is used for an initialization operation prior to the threshold correction operation and also used for precharging the video signal line 106HS in advance.

  The sampling transistor 125 has a gate terminal G connected to the writing scanning line 104WS from the writing scanning unit 104, a source terminal S connected to the video signal line 106HS, and a drain terminal D connected to the gate terminal G (node) of the driving transistor 121. ND122). The gate terminal G is supplied with an active H write drive pulse WS from the write scanning unit 104. The sampling transistor 125 may have a connection mode in which the source terminal S and the drain terminal D are reversed.

<Operation of Pixel Circuit of Second Comparative Example>
In the pixel circuit P of the second comparative example (in fact, the same applies to the pixel circuit P of the present embodiment which will be described later; the same applies to the driving timing below), as the driving timing, first, the sampling transistor 125 starts from the write scanning line 104WS. Conduction is performed according to the supplied write drive pulse WS, and the video signal Vsig supplied from the video signal line 106HS is sampled and held in the holding capacitor 120. This is basically the same as the case of driving the pixel circuit P of the first comparative example shown in FIG.

  Note that the drive timing in the pixel circuit P of the second comparative example is such that when the information of the signal potential Vin of the video signal Vsig is written into the holding capacitor 120, the video signals for one row are simultaneously applied to each column from the viewpoint of sequential scanning. Line-sequential driving to be transmitted to the video signal line 106HS.

  The driving transistor 121 is supplied with a current from the power supply line 105DSL at the first potential (high potential side), and corresponds to the signal potential held in the holding capacitor 120 (potential corresponding to the potential of the video signal Vsig during the effective period). Then, the drive current Ids is passed through the organic EL element 127.

  The vertical drive unit 103 writes the control signal for making the sampling transistor 125 conductive in a time zone in which the power supply line 105DSL is at the first potential and the video signal line 106HS is at the reference potential Vo which is the ineffective period of the video signal Vsig. A driving pulse WS is output, and a voltage corresponding to the threshold voltage Vth of the driving transistor 121 is held in the holding capacitor 120. This operation realizes a threshold correction function. By this threshold value correction function, it is possible to cancel the influence of the threshold voltage Vth of the drive transistor 121 that varies for each pixel circuit P.

  As the driving timing in the pixel circuit P of the second comparative example, the vertical driving unit 103 repeatedly drives the threshold correction operation repeatedly in a plurality of horizontal periods preceding the sampling of the signal potential Vin in the video signal Vsig. A voltage corresponding to the threshold voltage Vth of the transistor 121 is held in the holding capacitor 120.

  Thus, in the pixel circuit P of the second comparative example, a sufficiently long writing time is ensured by executing the threshold value correction operation a plurality of times. In this way, a voltage corresponding to the threshold voltage Vth of the drive transistor 121 can be reliably held in advance in the storage capacitor 120.

  The voltage corresponding to the held threshold voltage Vth is used to cancel the threshold voltage Vth of the drive transistor 121. Therefore, even if the threshold voltage Vth of the drive transistor 121 varies for each pixel circuit P, it is completely canceled for each pixel circuit P. Therefore, the uniformity of the image, that is, the uniformity of the light emission luminance over the entire screen of the display device is achieved. Rise. In particular, luminance unevenness that tends to appear when the signal potential is low gradation can be prevented.

  Preferably, prior to the threshold correction operation, the vertical drive unit 103 is in a time zone in which the power supply line 105DSL is at the second potential and the video signal line 106HS is at the reference potential Vo, which is the ineffective period of the video signal Vsig. The write drive pulse WS is made active (H level in this example) to turn on the sampling transistor 125, and then the power supply line 105DSL is set to the first potential while the write drive pulse WS remains active H.

  As a result, the threshold value correcting operation is started after the gate terminal G of the driving transistor 121 is set to the reference potential Vo and the source terminal S is set to the second potential. By such a reset operation (initialization operation) of the gate potential and the source potential, it is possible to reliably execute the subsequent threshold value correction operation.

  The pixel circuit P of the second comparative example has a mobility correction function in addition to the threshold value correction function. That is, the vertical drive unit 103 writes the write drive supplied to the write scan line 104WS in order to bring the sampling transistor 125 into a conductive state in a time zone in which the video signal line 106HS is at the signal potential Vin that is the effective period of the video signal Vsig. The pulse WS is made active (H level in this example) only for a period shorter than the above-described time zone. By appropriately setting the active period (which is both a sampling period and a mobility correction period) of the write drive pulse WS, when the signal potential Vsig is held in the storage capacitor 120, the drive transistor 121 is simultaneously controlled with respect to the mobility μ. Correction is applied to the signal potential Vsig.

  In particular, at the drive timing in the pixel circuit P of the second comparative example, the write drive pulse WS is within the time zone in which the power supply line 105DSL is at the first potential on the high potential side and the video signal Vsig is in the valid period. Is active. That is, as a result, the mobility correction time (also the sampling period) includes the time width in which the potential of the video signal line 106HS is at the potential (signal line potential) of the effective period of the video signal Vsig and the active period of the write drive pulse WS. It is determined in the range where both of these overlap. In particular, in this embodiment, since the active period width of the write drive pulse WS is determined so that the video signal line 106HS falls within the time width at the signal potential, the mobility correction time is consequently written. Is determined by the drive pulse WS.

  To be precise, the mobility correction time (also the sampling period) is the time from when the write drive pulse WS rises and the sampling transistor 125 is turned on until the write drive pulse WS falls and the sampling transistor 125 is turned off. It becomes.

  Here, when considering the horizontal direction of the screen, a detailed explanatory diagram is omitted, but the write drive pulse WS is commonly supplied from the write scanning unit 104 to all the pixel circuits P in one row. The pixel circuit P (near side) is closer to the writing scanning unit 104 in the pixel circuit P (referred to as a far-side pixel) farther from the writing scanning unit 104 due to the influence of wiring capacitance and wiring resistance on the waveform of the writing drive pulse WS. The waveform becomes duller than the pixel). On the other hand, regarding the video signal line potential, there is no difference in waveform because the distance from the horizontal drive unit 106 as the signal source is the same for both the far-side pixel and the near-side pixel.

  Therefore, in the far-side pixel where the waveform of the write drive pulse WS is greatly dull and deteriorates, the on-timing of the sampling transistor 125 is shifted backward as compared with the near-side pixel, but the off-timing is also shifted backward. Therefore, the mobility correction time determined by the difference between them is not much different from the mobility correction time of the near side pixel after all.

  Further, the signal potential (sampling potential) finally sampled in the storage capacitor 120 by the sampling transistor 125 is given by the video signal line potential when the sampling transistor 125 is turned off. The sampling potential becomes the signal potential Vin in both the near side pixel and the far side pixel, and no difference occurs.

  Thus, at the drive timing in the pixel circuit P of the second comparative example, there is almost no difference in the video signal potential sampled between the far side pixel and the near side pixel. Further, the mobility correction time is almost negligible between the far side pixel and the near side pixel. Thereby, the organic EL display device 1 of the present embodiment can realize a display device with good image quality in which no luminance difference appears on the left and right sides of the screen and shading is suppressed.

  When considering the vertical direction of the screen, the write drive pulse WS is the same for the pixel circuit P (referred to as the upper pixel) on the upper side of the screen and the pixel circuit P (referred to as the lower pixel) on the lower side of the screen. Since the position is taken, there is no difference in the waveform (scan line potential waveform) of the write drive pulse WS. On the other hand, since the video signal Vsig is commonly supplied from the horizontal driving unit 106 via the video signal line 106HS to all the pixel circuits P in one column, the horizontal driving unit 106 is affected by the wiring capacitance and wiring resistance. The far side pixels far from the video signal voltage have a larger delay amount than the near side pixels closer to the horizontal drive unit 106.

  However, even if the signal potential waveform appearing on the video signal line 106HS is delayed, as long as the write drive pulse WS is included in a time width in which the video signal line 106HS is at the signal potential (potential of the effective period of the video signal Vsig), There is almost no difference in sampling potential and mobility correction time. As a result, the sampled video signal potentials are substantially equal on the lower and upper sides of the screen, and the mobility correction time is also substantially equal. Thereby, the luminance difference between the upper side and the lower side of the screen is suppressed, and a display device with good image quality can be realized.

  The pixel circuit P of the second comparative example also has a bootstrap function. That is, the writing scanning unit 104 cancels the application of the writing driving pulse WS to the writing scanning line 104WS (ie, inactive L (low)) when the signal potential Vin of the video signal Vsig is held in the holding capacitor 120. The sampling transistor 125 is turned off, and the gate terminal G of the drive transistor 121 is electrically disconnected from the video signal line 106HS.

  A storage capacitor 120 is connected between the gate terminal G and the source terminal S of the driving transistor 121, and the gate potential Vg is interlocked with the variation of the source potential Vs of the driving transistor 121 due to the effect of the storage capacitor 120. Thus, the gate-source voltage Vgs can be kept constant.

<Timing chart>
FIG. 6 is a timing chart for explaining the operation when the information of the signal potential Vin is written in the storage capacitor 120 by the line sequential method as an example of the driving timing related to the pixel circuit P of the second comparative example shown in FIG. 6B to 6I are diagrams illustrating an equivalent circuit and an operation state in each period of the timing chart illustrated in FIG.

  In FIG. 6, the change in the potential of the write scanning line 104WS, the change in the potential of the power supply line 105DSL, and the change in the potential of the video signal line 106HS are shown with a common time axis. In parallel with these potential changes, changes in the gate potential Vg and source potential Vs of the drive transistor 121 are also shown for one row (the first row in the figure).

  Basically, the same driving is performed for each row of the write scanning line 104WS and the power supply line 105DSL with a delay of one horizontal scanning period. Each timing and signal in FIG. 6 are indicated by the same timing and signal as the timing and signal of the first row regardless of the processing target row. When distinction is required in the description, the processing target row is indicated by a reference with “_” in the timing and signal.

  Further, at the drive timing in the pixel circuit P of the second comparative example, the period in which the video signal Vsig is at the reference potential Vo, which is the ineffective period, is the first half of one horizontal period, and the period in the signal potential Vin, which is the effective period. The second half of one horizontal period.

  Here, a case where the threshold correction operation is executed only once will be described, but this is not essential. The threshold correction operation may be repeated a plurality of times with one horizontal period as a processing cycle.

  Note that, when the threshold correction operation is performed a plurality of times, the processing period of the threshold correction operation is one horizontal period before the sampling transistor 125 samples the signal potential Vin information in the storage capacitor 120 for each row. Prior to the threshold correction operation, the potential of the power supply line 105DSL is set to the second potential Vcc_L, the gate of the driving transistor 121 is set to the reference potential Vin, and the source potential is set to the second potential Vcc_L. After the operation, the sampling transistor 125 is turned on to correspond to the threshold voltage Vth of the driving transistor 121 in a time zone in which the potential of the power supply line 105DSL is at the first potential Vcc_H and the video signal line 106HS is at the reference potential Vo. This is because a threshold value correction operation for holding the voltage to be held in the holding capacitor 120 is performed.

  Inevitably, the threshold correction period is shorter than one horizontal period. Accordingly, due to the magnitude relationship between the capacity Cs and the second potential Vcc_L of the storage capacitor 120 and other factors, an accurate voltage corresponding to the threshold voltage Vth is stored in the storage capacitor 120 in this short threshold correction operation period. There may be no cases. It is preferable to execute the threshold correction operation a plurality of times for this purpose. That is, the voltage corresponding to the threshold voltage Vth of the drive transistor 121 is surely obtained by repeatedly executing the threshold correction operation in a plurality of horizontal periods preceding sampling (signal writing) of the signal potential Vin to the storage capacitor 120. 120.

  For a certain row (here, the first row), in the light emission period B of the previous field before timing t11, the write drive pulse WS is inactive L and the sampling transistor 125 is in a non-conducting state, while power supply drive The pulse DSL is at the first potential Vcc_H which is the high potential power supply voltage side.

  Therefore, as shown in FIG. 6B, regardless of the potential of the video signal line 106HS, the organic state depends on the voltage state (the gate-source voltage Vgs of the driving transistor 121) held in the holding capacitor 120 by the operation of the previous field. The drive current Ids is supplied from the drive transistor 121 to the EL element 127 and flows into the ground wiring Vcath (GND) common to all the pixels, whereby the organic EL element 127 is in a light emitting state.

  Thereafter, a new field of line sequential scanning is entered. First, the drive scanning unit 105 supplies a power drive pulse DSL_1 to be supplied to the power supply line 105DSL_1 in the first row in a state where the write drive pulse WS is inactive L. The first potential Vcc_H on the high / low potential side is switched to the second potential Vcc_L on the low potential side (t11_1: see FIG. 6C).

  This timing (t11_1) is within a period in which the video signal Vsig is at the signal potential Vin in the effective period in the embodiment shown in FIG. For example, the first row is within the range of timings t15V to t13V. However, this is not essential, and may be performed when the video signal Vsig is at the ineffective period reference potential Vo. The first row may be within the range of timing t13V to t15V.

  Next, the write scanning unit 104 switches the write drive pulse WS to active H while the power supply line 105DSL_1 is at the second potential Vcc_L (t13W). At this timing (t13W), the video signal Vsig in the immediately preceding horizontal period is switched from the reference potential Vo in the ineffective period to the signal potential Vin in the effective period (t15V), and then the effective period of the video signal Vsig in the horizontal period. The timing is the same as or slightly delayed from the timing (t13V) at which the signal potential Vin is switched from the signal potential Vin to the reference potential Vo which is an ineffective period. Thereafter, the timing (t15W) at which the write drive pulse WS is switched to inactive L is the same as or more than the timing (t15V) at which the video signal Vsig is switched from the reference potential Vo in the ineffective period to the signal potential Vin in the effective period. Set a little earlier.

  That is, preferably, the period (t13W to t15W) in which the write drive pulse WS is active H is within the time period (t13V to t15V) in which the video signal Vsig is at the reference potential Vo which is the ineffective period. This is because when the power supply line 105DSL is at the first potential Vcc_H and the video signal Vsig is at the signal potential Vin and the write drive pulse WS is set to active H, the signal potential Vin is sampled into the holding capacitor 120. This is because (signal potential writing operation) is performed, which is inconvenient as a threshold correction operation.

  At timings t11_1 to t13W (referred to as a discharge period C), the potential of the power supply line 105DSL is discharged to the second potential Vcc_L, and the source potential Vs of the driving transistor 121 changes to a potential close to the second potential Vcc_L. Further, a storage capacitor 120 is connected between the gate terminal G and the source terminal S of the drive transistor 121, and the gate potential Vg is linked to the variation of the source potential Vs of the drive transistor 121 due to the effect of the storage capacitor 120. To do.

  When the wiring capacity of the power supply line 105DSL is large, the power supply line 105DSL may be switched from the high potential Vcc_H to the low potential Vcc_L at a relatively early timing. By ensuring a sufficient discharge period C (t11_1 to t13W), it is prevented from being affected by wiring capacitance and other pixel parasitic capacitances.

  When the write drive pulse WS is switched to active H (t13W) while the power supply drive pulse DSL is kept at the second potential Vcc_L on the low potential side, the sampling transistor 125 becomes conductive as shown in FIG. 6D.

  At this time, the video signal line 106HS is at the reference potential Vo. Therefore, the gate potential Vg of the drive transistor 121 becomes the reference potential Vo of the video signal line 106HS through the conducting sampling transistor 125. At the same time, when the drive transistor 121 is turned on, the source potential Vs of the drive transistor 121 is immediately fixed to the second potential Vcc_L on the low potential side.

  That is, when the potential of the power supply line 105DSL is from the first potential Vcc_H on the high potential side to the second potential Vcc_L that is sufficiently lower than the reference potential Vo of the video signal line 106HS, the source potential Vs of the drive transistor 121 is changed to the video signal line 106HS. Is initialized (reset) to a second potential Vcc_L that is sufficiently lower than the reference potential Vo. In this way, by initializing the gate potential Vg and the source potential Vs of the drive transistor 121, the preparation for the threshold correction operation is completed. Next, a period (t13W to t14_1) until the power supply driving pulse DSL is set to the first potential Vcc_H on the high potential side is an initialization period D. Note that the discharge period C and the initialization period D are also collectively referred to as a threshold correction preparation period in which the gate potential Vg and the source potential Vs of the drive transistor 121 are initialized.

  Next, the power supply drive pulse DSL applied to the power supply line 105DSL is switched to the first potential Vcc_H while the write drive pulse WS remains active H (t14_1). Thereafter, the drive scanning unit 105 keeps the potential of the power supply line 105DSL at the first potential Vcc_H until the next frame (or field) processing.

  As a result, the drain current flows into the storage capacitor 120 and enters a threshold correction period E in which the threshold voltage Vth of the drive transistor 121 is corrected (cancelled). This threshold value correction period E continues until the timing (t15W) when the write drive pulse WS is made inactive L.

  In the threshold correction period E after the timing (t14_1), as shown in FIG. 6E, the potential of the power supply line 105DSL is changed from the second potential Vcc_L on the low potential side to the first potential Vcc_H on the high potential side, thereby driving. The source potential Vs of the transistor 121 starts to rise.

  That is, the gate terminal G of the drive transistor 121 is held at the reference potential Vo of the video signal Vsig, and the drain current flows until the potential Vs of the source terminal S of the drive transistor 121 rises and the drive transistor 121 is cut off. And When cut off, the source potential Vs of the driving transistor 121 becomes “Vo−Vth”.

  In the threshold correction period E, the drain current flows exclusively to the storage capacitor 120 side (when Cs << Cel) and does not flow to the organic EL element 127 side, so that the organic EL element 127 is cut off. Is set to the potential Vcath of the common ground wiring cath.

  Since the equivalent circuit of the organic EL element 127 is represented by a parallel circuit of a diode and a parasitic capacitance Cel, as long as “Vel ≦ Vcath + VthEL”, that is, the leakage current of the organic EL element 127 is considerably larger than the current flowing through the drive transistor 121. As long as the current is small, the current of the drive transistor 121 is used to charge the storage capacitor 120 and the parasitic capacitor Cel.

  As a result, when the current path of the drain current flowing through the drive transistor 121 is interrupted, the voltage Vel at the anode end A of the organic EL element 127, that is, the potential of the node ND121 increases with time. Then, when the potential difference between the potential of the node ND121 (source potential Vs) and the voltage of the node ND122 (gate potential Vg) is just the threshold voltage Vth, the driving transistor 121 is turned off from the on state, and the drain current does not flow. The threshold correction period ends. That is, after a certain time has elapsed, the gate-source voltage Vgs of the drive transistor 121 takes a value called the threshold voltage Vth.

  Here, actually, a voltage corresponding to the threshold voltage Vth is written in the storage capacitor 120 connected between the gate terminal G and the source terminal S of the driving transistor 121. However, during the threshold correction period E, the timing at which the write drive pulse WS is changed to inactive L from the timing at which the write drive pulse WS is set to active H (t13W) (specifically, the time t14 at which the power supply drive pulse DSL is subsequently returned to the first potential Vcc_H). t15W), and when this period is not sufficiently secured, the process ends before that time. In order to solve this problem, it is preferable to repeat the threshold correction operation a plurality of times. Here, illustration of the timing is omitted.

  Next, the drive scanning unit 105 switches the write drive pulse WS to inactive L in the second half of one horizontal period (t15W), and the horizontal drive unit 106 further changes the potential of the video signal line 106HS from the reference potential Vo. Switching to the signal potential Vin (t15V). As a result, at timings t15W to t15V, as shown in FIG. 6F, the potential of the write scanning line 104WS (write drive pulse WS) is at a low level while the video signal line 106HS is at the reference potential Vo.

  After that, the signal potential Vin of the video signal Vsig is actually supplied to the video signal line 106HS by the horizontal drive unit 106, and the signal potential Vin is written to the storage capacitor 120 during the period when the write drive pulse WS is set to active H. A period (also referred to as a sampling period). This signal potential Vin is held in the form of adding to the threshold voltage Vth of the drive transistor 121.

  As a result, fluctuations in the threshold voltage Vth of the drive transistor 121 are always canceled, and threshold correction is performed. By this threshold correction, the gate-source voltage Vgs held in the holding capacitor 120 becomes “Vsig + Vth” = “Vin + Vth”. At the same time, mobility correction is executed during this sampling period. That is, at the drive timing in the pixel circuit P of the second comparative example, the sampling period also serves as the mobility correction period.

  Specifically, first, after the write drive pulse WS is switched to inactive L (t15W), the horizontal drive unit 106 further switches the potential of the video signal line 106HS from the reference potential Vo to the signal potential Vin (t15V). . By doing so, as shown in FIG. 6G, preparation for the next sampling operation and mobility correction operation is completed in a state where the sampling transistor 125 is in a non-conductive (off) state. Next, a period until the timing (t16_1) when the write drive pulse WS is set to active H is referred to as a write & mobility correction preparation period G.

  Next, the write scanning unit 104 sets the write drive pulse WS to active H while keeping the potential of the power supply line 105DSL at the first potential Vcc_H and holding the potential of the video signal line 106HS at the signal potential Vin. Switching (t16_1), at an appropriate timing until the horizontal driving unit 106 switches the potential of the video signal line 106HS from the signal potential Vin to the reference potential Vo (t18_1), that is, the video signal line 106HS is set to the signal potential Vin. At an appropriate time in a certain time zone, it is switched to inactive L (t17_1). A period (t16_1 to t17_1) in which the write drive pulse WS is active H is referred to as a sampling period & mobility correction period H.

  As a result, as shown in FIG. 6H, the sampling transistor 125 is turned on (on) while the gate potential Vg of the drive transistor 121 is at the signal potential Vin. Therefore, in the sampling period & mobility correction period H, the drive current Ids flows through the drive transistor 121 while the gate terminal G of the drive transistor 121 is fixed to the signal potential Vin of the video signal Vsig.

  Here, when the threshold voltage of the organic EL element 127 is set to VthEL, by setting “Vo−Vth <VthEL”, the organic EL element 127 is placed in a reverse bias state and is in a cutoff state (high impedance). In this state, no light is emitted, and simple capacitance characteristics are shown instead of diode characteristics. Therefore, the drain current (drive current Ids) flowing through the drive transistor 121 is a capacitance “C = Cs + Cel” obtained by combining both the capacitance value Cs of the storage capacitor 120 and the capacitance value Cel of the parasitic capacitance (equivalent capacitance) Cel of the organic EL element 127. It will be written. As a result, the drain current of the driving transistor 121 flows into the parasitic capacitance Cel of the organic EL element 127 and starts charging. As a result, the source potential Vs of the drive transistor 121 increases.

  In the timing chart of FIG. 6, this increase is represented by ΔV. This increase, that is, the negative feedback amount ΔV, which is a mobility correction parameter, is subtracted from the gate-source voltage “Vgs = Vin + Vth” held in the holding capacitor 120 by the threshold correction, and “Vgs = Vin−ΔV + Vth”. Therefore, negative feedback is applied. At this time, the source potential Vs of the driving transistor 121 becomes “−Vth + ΔV” obtained by subtracting the voltage “Vgs = Vin−ΔV + Vth” held in the holding capacitor from the gate potential Vg (= Vin).

  In this manner, at the drive timing in the pixel circuit P of the second comparative example, the sampling of the signal potential Vin and the mobility μ in the video signal Vsig are corrected in the sampling period & mobility correction period H (t16 to t17). The feedback amount (mobility correction parameter) ΔV is adjusted. The writing scanning unit 104 can adjust the time width of the sampling period & mobility correction period H, thereby optimizing the negative feedback amount of the drive current Ids for the storage capacitor 120.

  Here, “optimizing the negative feedback amount” means that the mobility correction can be appropriately performed at any level in the range from the black level to the white level of the video signal potential. The amount of negative feedback applied to the gate-source voltage Vgs depends on the drain current Ids extraction time, that is, the sampling period & mobility correction period H. The longer this period, the larger the negative feedback amount. The negative feedback amount ΔV is ΔV = Ids · Cel / t.

  As is apparent from this equation, the negative feedback amount ΔV increases as the drive current Ids, which is the drain-source current of the drive transistor 121, increases. Conversely, when the drive current Ids of the drive transistor 121 is small, the negative feedback amount ΔV is small. Thus, the negative feedback amount ΔV is determined according to the drive current Ids.

  Further, as the signal potential Vin increases, the drive current Ids increases and the absolute value of the negative feedback amount ΔV also increases. Therefore, mobility correction according to the light emission luminance level can be realized. At that time, the sampling period & mobility correction period H does not necessarily have to be constant, and conversely, it may be preferable to adjust according to the drive current Ids. For example, when the drive current Ids is large, the mobility correction period t should be short, and conversely, when the drive current Ids is small, the sampling period & mobility correction period H should be set long.

  Further, the negative feedback amount ΔV is Ids · Cel / t, and even if the drive current Ids varies due to variations in the mobility μ for each pixel circuit P, the negative feedback amount ΔV corresponds to each. Variations in mobility μ for each pixel circuit P can be corrected. That is, when the signal potential Vin is constant, the absolute value of the negative feedback amount ΔV increases as the mobility μ of the drive transistor 121 increases. In other words, since the negative feedback amount ΔV increases as the mobility μ increases, the variation in mobility μ for each pixel circuit P can be removed.

  In this way, at the drive timing in the pixel circuit P of the second comparative example, the negative feedback amount ΔV for correcting the sampling of the signal potential Vin and the variation in the mobility μ in the sampling period & mobility correction period H. Adjustments are made simultaneously. Of course, the negative feedback amount ΔV can be optimized by adjusting the time width of the sampling period & mobility correction period H.

  Next, the write scanning unit 104 switches the write drive pulse WS to inactive L in a state where the potential of the video signal line 106HS is at the signal potential Vin (t17_1). As a result, as shown in FIG. 6I, the sampling transistor 125 enters a non-conduction (off) state and proceeds to the light emission period I. The horizontal driving unit 106 stops the supply of the signal potential Vin of the video signal Vsig to the video signal line 106HS at an appropriate time thereafter, and returns it to the reference potential Vo (t18_1). Thereafter, the process proceeds to the next frame (or field), and the threshold correction preparation operation, the threshold correction operation, the mobility correction operation, and the light emission operation are repeated again.

  As a result, the gate terminal G of the drive transistor 121 is disconnected from the video signal line 106HS. Since the application of the signal potential Vin to the gate terminal G of the drive transistor 121 is released, the gate potential Vg of the drive transistor 121 can be increased.

  At this time, the drive current Ids flowing through the drive transistor 121 flows through the organic EL element 127, and the anode potential of the organic EL element 127 rises according to the drive current Ids. Let this increase be Vel. Eventually, as the source potential Vs rises, the reverse bias state of the organic EL element 127 is canceled, so that the organic EL element 127 actually starts to emit light by the inflow of the drive current Ids. The rise (Vel) of the anode potential of the organic EL element 127 at this time is nothing but the rise of the source potential Vs of the drive transistor 121, and the source potential Vs of the drive transistor 121 becomes “−Vth + ΔV + Vel”.

  The relationship between the drive current Ids and the gate voltage Vgs can be expressed as in Expression (2) by substituting “Vin−ΔV + Vth” into Vgs in Expression (1) representing the previous transistor characteristics. In formula (2), k = (1/2) (W / L) Cox.

  From this equation (2), it can be seen that the term of the threshold voltage Vth is canceled and the drive current Ids supplied to the organic EL element 127 does not depend on the threshold voltage Vth of the drive transistor 121. Basically, the drive current Ids is determined by the signal potential Vin of the video signal Vsig. In other words, the organic EL element 127 emits light with a luminance corresponding to the signal potential Vin.

  At that time, the signal potential Vin is corrected by the feedback amount ΔV. This correction amount ΔV works so as to cancel the effect of the mobility μ located in the coefficient part of the equation (2). Therefore, the drive current Ids substantially depends only on the signal potential Vin. Since the drive current Ids does not depend on the threshold voltage Vth, even if the threshold voltage Vth varies depending on the manufacturing process, the drain-source drive current Ids does not vary, and the light emission luminance of the organic EL element 127 does not vary.

  In addition, a storage capacitor 120 is connected between the gate terminal G and the source terminal S of the drive transistor 121. Due to the effect of the storage capacitor 120, a bootstrap operation is performed at the beginning of the light emission period. The gate potential Vg and the source potential Vs of the drive transistor 121 rise while maintaining the gate-source voltage “Vgs = Vin−ΔV + Vth” at a constant. When the source potential Vs of the driving transistor 121 becomes “−Vth + ΔV + Vel”, the gate potential Vg becomes “Vin + Vel”.

  At this time, since the gate-source voltage Vgs of the drive transistor 121 is constant, the drive transistor 121 passes a constant current (drive current Ids) to the organic EL element 127. As a result, a voltage drop occurs, and the potential Vel at the anode end A of the organic EL element 127 (= potential at the node ND121) rises to a voltage at which a driving current Ids in a saturated state can flow through the organic EL element 127.

  Here, the organic EL element 127 has its IV characteristic changed as the light emission time becomes longer. Therefore, the potential of the node ND121 also changes with time. However, even if the anode potential fluctuates due to such deterioration of the organic EL element 127 with time, the gate-source voltage Vgs held in the holding capacitor 120 is always kept constant at “Vin−ΔV + Vth”.

  Since the drive transistor 121 operates as a constant current source, the IV characteristic of the organic EL element 127 changes with time, and even if the source potential Vs of the drive transistor 121 changes accordingly, the drive transistor 121 drives the drive transistor 121. Since the gate-source potential Vgs 121 is kept constant (≈Vin−ΔV + Vth), the current flowing through the organic EL element 127 does not change, and thus the emission luminance of the organic EL element 127 is also kept constant.

  Regardless of the characteristic variation of the organic EL element 127, an operation for correction (operation based on the effect of the storage capacitor 120) for maintaining the gate-source voltage of the driving transistor 121 constant and maintaining the luminance constant is performed. This is called a bootstrap operation. By this bootstrap operation, even if the IV characteristic of the organic EL element 127 changes with time, it is possible to display an image without luminance deterioration associated therewith.

  That is, in the pixel circuit P of the second comparative example and the driving timing for driving the pixel circuit P, the driving signal for correcting the change in the current-voltage characteristic of the organic EL element 127 which is an example of the electro-optical element and maintaining the driving current constant. A bootstrap circuit, which is an example of a stabilizing circuit, is configured so that the bootstrap operation functions. Therefore, even if the IV characteristic of the organic EL element 127 deteriorates, the constant current Ids always flows, so that the organic EL element 127 continues to emit light with the luminance according to the pixel signal Vsig, and the luminance changes. Absent.

  Further, in the pixel circuit P of the second comparative example and the drive timing for driving the pixel circuit P, a threshold correction circuit which is an example of a drive signal stabilization circuit that corrects the threshold voltage Vth of the drive transistor 121 and maintains the drive current constant is provided. The threshold correction operation is configured and functions. As the gate-source potential Vgs reflecting the threshold voltage Vth of the drive transistor 121, a constant current Ids that is not affected by variations in the threshold voltage Vth can be passed.

  In particular, although not shown in the figure, if the processing cycle of one threshold correction operation is one horizontal period and the threshold correction operation is repeated a plurality of times, the threshold voltage Vth is reliably supplied to the storage capacitor 120. Can be retained. The inter-pixel difference of the threshold voltage Vth is reliably removed, and luminance unevenness due to the variation of the threshold voltage Vth can be suppressed regardless of the gradation.

  On the other hand, when the threshold voltage Vth is not sufficiently corrected, for example, when the threshold correction operation is performed once, that is, when the threshold voltage Vth is not held in the holding capacitor 120, the pixel circuits P are different. In the low gradation region, there is a difference in luminance (driving current Ids). Therefore, when the correction of the threshold voltage is insufficient, luminance unevenness appears at a low gradation and the image quality is impaired.

  In addition, at the driving timing in the pixel circuit P of the second comparative example, the driving current is made constant by correcting the mobility μ of the driving transistor 121 in conjunction with the writing operation of the signal potential Vin to the holding capacitor 120 by the sampling transistor 125. A mobility correction circuit, which is an example of a drive signal stabilization circuit to be maintained, is configured so that the mobility correction operation functions. As the gate-source potential Vgs reflecting the carrier mobility μ of the driving transistor 121, a constant current Ids that is not affected by variations in the carrier mobility μ can be passed.

  That is, in the pixel circuit P of the second comparative example, a threshold correction circuit and a mobility correction circuit are automatically configured by devising drive timing, and characteristic variations of the drive transistor 121 (threshold voltage Vth and carrier in this example). In order to prevent the influence on the drive current Ids due to the variation in mobility μ), it functions as a drive signal stabilization circuit that maintains the drive current constant by correcting the influence of the threshold voltage Vth and the carrier mobility μ. It is.

  Since not only the bootstrap operation but also the threshold correction operation and the mobility correction operation are performed, the gate-source voltage Vgs maintained in the bootstrap operation is a voltage corresponding to the threshold voltage Vth and for mobility correction. Therefore, the light emission luminance of the organic EL element 127 is not affected by variations in the threshold voltage Vth and mobility μ of the driving transistor 121, and is also affected by deterioration with time of the organic EL element 127. I do not receive it. A stable gradation corresponding to the input signal potential Vin can be displayed, and a high-quality image can be obtained.

  Further, since the pixel circuit P of the second comparative example can be configured by a source follower circuit using the n-channel type driving transistor 121, even if the current organic EL elements of the anode and cathode electrodes are used as they are, The organic EL element 127 can be driven.

  In addition, the pixel circuit P can be configured using only n-channel transistors including the driving transistor 121 and the sampling transistor 125 in the periphery thereof, and an amorphous silicon (a-Si) process is also used in TFT fabrication. Therefore, the cost of the TFT substrate can be reduced.

<< Control method using power supply drive pulse DSL >>
Next, the organic EL display of this embodiment is driven from the side of the control method in which the drain terminal D of the driving transistor 121 is driven by the power supply driving pulse DSL, that is, driven by the power supply voltage switched between the first potential Vcc_H and the second potential Vcc_L. The device 1 will be described.

<Output circuit of drive scanning unit>
FIG. 7 is a diagram illustrating an output circuit of the drive scanning unit 105 that functions as a drive scanner. As shown in the figure, the drive scanning unit 105 switches the first potential Vcc_H and the second potential Vcc_L to the power supply line 105DSL of each row so as to perform display while performing the threshold correction operation, and thereby the drain end D of the drive transistor 121. Two kinds of power supply voltages are switched and supplied. For this reason, an output circuit 400 having sufficient driving capability is provided in a portion connected to the power supply line 105DSL. Although only the output circuit 400 for one row is shown in the figure, an output circuit 400 is provided for each power supply line 105DSL. Although not shown, the drive scanning unit 105 is provided outside the display panel unit 100 and is supplied with the first potential Vcc_H and the second potential Vcc_L from a power supply circuit having a sufficiently small output impedance. It has become.

  As an example, the output circuit 400 includes a p-channel transistor (p-type transistor) 402 and an n-channel transistor (n-type transistor) 404 for supplying the first potential Vcc_H and the second potential Vcc_L. The supply end 400L is arranged in series. The source terminal S of the p-type transistor 402 is connected to the supply terminal 400H for the first potential Vcc_H, and the source terminal S of the n-type transistor 404 is connected to the supply terminal 400L for the second potential Vcc_L. The drain terminals D of the p-type transistor 402 and the n-type transistor 404 are connected in common, and the connection point is connected to the power supply line 105DSL. As a whole, a CMOS inverter is configured. The gate terminals G of the p-type transistor 402 and the n-type transistor 404 are connected in common, and an active-L power supply scanning pulse NDS is supplied to the connection point.

  When the power scan pulse NDS is active L, the n-type transistor 404 is turned off and the p-type transistor 402 is turned on, so that the first potential Vcc_H is supplied to the power supply line 105DSL, while when the power scan pulse NDS is inactive H Since the p-type transistor 402 is turned off and the n-type transistor 404 is turned on, the second potential Vcc_L is supplied to the power supply line 105DSL. As can be seen from this operation, the output circuit 400 functions as a power supply voltage switching circuit. Unlike the power supply circuit that can be configured with individual components, the output circuit 400 is configured as an integrated circuit, and the output impedance of the output circuit 400 (p-type transistor 402 and n-type transistor 404) is There are circumstances that cannot be reduced sufficiently.

<Operation of drive scanning unit output circuit and pixel circuit during light emission>
FIG. 8 is a diagram for explaining the operation of the output circuit of the drive scanning unit 105 and the pixel circuit P during the light emission period of the organic EL element 127.

  During the light emission period of the organic EL element 127, the power supply scanning pulse NDS becomes active L, the n-type transistor 404 is turned off and the p-type transistor 402 is turned on, so that the first potential Vcc_H is supplied to the power supply line 105DSL. As a result, the drive current Ids (= the light emission current Iel of the organic EL element 127) flowing in the drive transistors 121 of all the pixel circuits P in the row flows in the p-type transistor 402.

  When considering the vertical direction (column direction) of the screen, the actual potential V105DSL of the power supply line 105DSL is not the first potential Vcc_H but the first potential Von_P due to the on-resistance Ron_P of the p-type transistor 402 constituting the output circuit 400. The potential is lower than Vcc_H. If the current sums Ids_Sum and Iel_Sum differ from row to row according to the display pattern, the voltage V105DSL of the power supply line 105DSL differs from row to row due to the influence of the output impedance of the output circuit 400 (in the previous example, the on-resistance Ron_P of the p-type transistor 402). Lateral crosstalk, which is an example of uneven brightness, occurs.

  Specifically, as shown in Equation (3), the sum Ids_Sum (= Ids_1 + Ids_2 +... + Ids_m) of the drive currents Ids flowing through the drive transistors 121 (organic EL elements 127) in each row and the sum Iel_Sum (= Iel_1 + Iel_2 +. + Iel_m) and the on-resistance Ron_P of the p-type transistor 402 are subtracted from the first potential Vcc_H. As a result, the operating point of the drive transistor 121 varies, and even if the signal potential Vin is the same in the column direction, the drive current Ids is different, resulting in luminance unevenness similar to shading in the column direction, and the window pattern. At the time of display, it is visually recognized as horizontal crosstalk. This point will be described in detail with reference to FIGS.

<Problem 1: When displaying a black window pattern>
FIG. 9 shows luminance unevenness that can occur when the drain terminal D of the drive transistor 121 is driven by the power supply drive pulse DSL, that is, when the control method is driven by the power supply voltage switched between the first potential Vcc_H and the second potential Vcc_L (particularly, It is FIG. (1) explaining the problem of a horizontal crosstalk concretely.

  Here, the operation for each screen and line when the black window pattern BW is displayed is described. Specifically, lines with all white displayed at the top and bottom of the screen are arranged, and lines with black at the center of the screen are arranged on the black display period at the center in one horizontal period and on the left and right. The case where 50% display in which the white display period is 1: 1 is shown.

  In this case, the value of the current flowing through the p-type transistor 402 for each line differs between a line where all white is displayed at the top and bottom of the screen and a line where black is displayed at the center of the screen. In contrast, as a result of the output impedance of the output circuit 400 of the drive scanning unit 105 being not sufficiently small, the actual potential V105DSL of the power supply line 105DSL is also displayed in a row where all white is displayed at the top and bottom of the screen (hereinafter, a row of 100% display) And a line where white and black are displayed at the center of the screen (hereinafter referred to as a 50% display line).

  Specifically, as shown in Expression (4-1), there is a double difference between the sum Iel_Sum100 of the light emission current Iel in the 100% display row and the sum Iel_Sum50 of the light emission current Iel in the 50% display row. Therefore, as shown in the equations (4-2) and (4-3), the current difference between the potential V105DSL100 of the 100% display row and the potential V105DSL50 of the 50% display row depends on the current difference. A difference in voltage drop occurs, and as a result, the actual first potential Vcc_H (here, V105DSL) of the power supply line 105DSL differs depending on whether the line is a 100% display line or a 50% display line. It will be. The actual voltage V121D at the drain terminal D of the drive transistor 121 varies in units of rows in accordance with the display pattern.

<Problem 2: When displaying black window pattern>
FIG. 10 shows a driving transistor caused by the difference between the voltage V105DSL100 and the voltage V105DSL50 of the power supply line 105DSL between the 100% display row and the 50% display row (that is, depending on the display pattern), as shown in FIG. FIG. 6 is a diagram (part 2) for specifically explaining problems (particularly lateral crosstalk) given to the drive current Ids of 121 and the light emission current Iel of the organic EL element 127;

  Since the pixel circuit P normally operates the n-type driving transistor 121 in a saturated state to drive the organic EL element 127 at a constant current, the power supply voltage supplied to the drain terminal D of the driving transistor 121 varies. However, as long as the drive transistor 121 operates in the saturation region, the drive current Ids should not be affected.

  However, if the voltage Vds between the drain and the source of the p-type transistor 402 is different in the column direction due to the difference in the actual light emission currents Iel_Sum100 and Iel_Sum50, the voltage V121D of the drain terminal D of the driving transistor 121 is different for each row. Even if the driving transistor 121 is operating in the saturation region, actually, as shown in FIG. 10A, due to the Early effect, there is a slight difference between the 100% display row and the 50% display row. There is a difference between the drive currents Ids_100 and Ids_50 (Ids_100 <Ids_50).

  The first potential Vcc_H differs between V105DSL50 and V105DSL100 in the 50% display line (line in which the black window is displayed) and the 100% display line (line in which the black window is not displayed). As a result, 50% In the display row, the drive current Ids slightly varies between the black display portion, that is, the column displaying black, and the white display portion, ie, the column not displaying black. There is a slight difference between the drive current IdsW_100 of the white display portion in the 50% display row and the drive current IdsW_50 of the white display portion in the 50% display row.

  Specifically, as shown in FIG. 10B, the sum Iel_Sum100 of the light emission current Iel in the 100% display row is almost twice as large as that in the 50% display row. The voltage drop with respect to Vcc_H is larger than the 50% display row. As a result, the drive currents Ids_1,..., Ids_m and the light emission currents Iel_1,. However, the degree of the decrease is larger than that of the 50% display line. Of course, the totals Ids_Sum100 and Iel_Sum100 are also slightly decreased, and the degree of the decrease is larger than that of the 50% display row.

  Therefore, as shown in FIG. 10C, even in the white display portion based on the signal potential Vin of the same magnitude, the light emission luminance at the ideal operating current is slightly in the 100% display row. It becomes darker and the degree of brightness reduction is larger than that of the 50% display line. Since the white display portion of the 100% display row has a greater brightness reduction and darker than the white display portion displayed on the left and right of the black display portion of the 50% display row, , Visually recognized as crosstalk. In FIG. 10C, the difference in luminance reduction is indicated by the difference in density of black dots in each white display portion.

<Improvement Method: First Embodiment>
FIG. 11 is a diagram showing a first embodiment of an organic EL display device that solves the problems described in FIG. Similar to the pixel circuit P of the second comparative example shown in FIG. 5, the organic EL display device 1 of the present embodiment is equipped with a circuit (bootstrap circuit) that prevents fluctuations in the drive current due to deterioration over time of the organic EL element 127. Further, the present invention is characterized in that a driving method that prevents a drive current fluctuation due to a characteristic fluctuation (threshold voltage fluctuation or mobility fluctuation) of the driving transistor 121 is employed. Therefore, the same drive timing as that of the second comparative example shown in FIGS. 6 to 6I is applied.

  In addition, in the first embodiment of the pixel circuit P, the influence that the voltage V105DSL of the power supply line 105DSL fluctuates in the column direction depending on the display pattern is reduced by the voltage V125D of the drain terminal D of the driving transistor 121. Thus, it has a feature in that a mechanism for suppressing lateral crosstalk is adopted.

  That is, a mechanism for preventing display luminance fluctuation caused by the fact that the actual voltage V121D at the drain terminal D of the driving transistor 121 fluctuates in units of rows according to the display pattern, which is the problem described in FIG. 9 and FIG. . Specifically, the pixel circuit P of the present embodiment has a drive current Ids (that is, a light emission current Iel) for each pixel circuit P with respect to the supply format of the first potential Vcc_H and the second potential Vcc_L to the drain terminal D of the drive transistor 121. This is characterized in that a voltage drop corresponding to is applied to the drain terminal D of the drive transistor 121.

  Here, the organic EL display device 1 according to the first embodiment is an example of a semiconductor element when a voltage drop corresponding to the drive current Ids or the light emission current Iel is given to the drain terminal D of the drive transistor 121 for each pixel circuit P. It is characterized in that the operating resistance of the transistor is used.

  Specifically, the organic EL display device 1 of the first embodiment is provided in the form of a cascode connection between the power supply line 105DSL from the drive scanning unit 105 and the drain end D of the drive transistor 121 of each pixel circuit P. A voltage drop generation unit 510 including the n-channel transistor (correction transistor 512), and a control signal generation unit 520 that generates a control signal Vcnt1 having a predetermined magnitude (a constant value). The control signal generation unit 520 may be provided in common for all the pixel circuits P, and may be connected to the gate terminal G with a row wiring.

  By the voltage drop generation unit 510 and the control signal generation unit 520, the influence of the voltage drop in the output circuit 400 due to the signal potential Vin is suppressed at the drain terminal D as the power supply terminal of the drive transistor 121. A first example of the suppression unit 107 is configured.

  The magnitude of the control signal Vcnt1 is such that the actual voltage V121D at the drain end D of the drive transistor 121 varies in units of rows according to the display pattern, depending on the voltage drop Vds between the drain and source caused by the operating resistance of the correction transistor 512. It is preferable to adopt a variable configuration that can be adjusted to an optimum value so that the display luminance fluctuation caused by the above can be more reliably suppressed. In the first embodiment, a constant control signal Vcnt1 is supplied to the gate terminal G of the correction transistor 512 regardless of the signal potential Vin regardless of whether or not it is variable.

  The correction transistor 512 has a drain terminal D connected to the power supply line 105 DSL and a source terminal S connected to the drain terminal D of the driving transistor 121. As a feature of the first embodiment, the control signal Vcnt1 is supplied from the control signal generator 520 to the gate terminal G of the correction transistor 512.

  As described above, the correction transistor 512 functions as a current-voltage conversion unit and is supplied to the gate terminal G by cascode-connecting the correction transistor 512 between the power supply line 105DSL and the drain terminal D of the drive transistor 121. The voltage drop Vds (= R512 * Ids = R512 * Iel) corresponding to the product of the operating resistance R512 (= f (Vcnt0)) corresponding to the control signal Vcnt1 and the drive current Ids (= light emission current Iel) 121 is given to the drain end D of 121.

  As a result, for each pixel circuit P, a voltage drop Vds corresponding to the drive current Ids (= light emission current Iel) occurs between the drain and source of the correction transistor 512. Specifically, the voltage V121D of the drain terminal D of the driving transistor 121 is lower than the voltage V105DSL of the power supply line 105DSL by the voltage drop Vds (= R512 * Ids = R512 * Iel) due to the correction transistor 512, V121D = V105DSL−Vds ”. The voltage V121D at the drain terminal D of the drive transistor 121 is a voltage lower than the voltage V105DSL supplied to the power supply line 105DSL.

  Of course, the larger the drive current Ids (= light emission current Iel), the greater the voltage drop Vds due to the correction transistor 512, and the lower the voltage V121D at the drain terminal D of the drive transistor 121, while the drive current Ids (= light emission current Iel). Is smaller, the voltage drop Vds due to the correction transistor 512 is smaller, and the decrease in the voltage V121D at the drain terminal D of the drive transistor 121 is smaller.

  As described above, in the organic EL display device 1 according to the first embodiment, the correction transistor 512 is caused to function as a current-voltage conversion unit, whereby the voltage drop amount Vds due to the correction transistor 512 is calculated for each pixel circuit P as the drive current Ids (= light emission). It was automatically adjusted according to the current Iel). Since the drive current Ids (= the light emission current Iel) varies depending on the display pattern for each pixel circuit P, the voltage V121D of the drain terminal D of the drive transistor 121 is adjusted for each pixel circuit P.

  In order to make the circuit configuration compact, it is conceivable to adopt a configuration in which one correction transistor 512 is shared by a plurality of pixel circuits P in a row. However, in this case, since it is affected by the currents Ids and Iel of the other pixel circuits P, a sufficient correction effect cannot be obtained and an appropriate correction operation is not performed. As a result, the sharpness at the edge portion is reduced. It is not a good idea because it causes new problems such as lowering.

  The correction transistor 512 is disposed between the supply terminal 400H for the first potential Vcc_H and the drain terminal D of the driving transistor 121, and the gate terminal G is controlled by the control signal Vcnt1 optimized by the control signal generator 520. As a result, the influence of the voltage drop in the p-type transistor 402 of the output circuit 400 due to the signal potential Vin is suppressed at the drain terminal D as the power supply terminal of the driving transistor 121. As a result, the light emission luminance is automatically adjusted for each pixel circuit P, so that horizontal crosstalk that can occur, for example, when a window pattern is displayed is improved. Hereinafter, the principle of this point will be described in detail.

<Improvement method: operation of the first embodiment>
12 and 13 are diagrams for explaining the operation and effect of the voltage drop generation unit 510 in the organic EL display device 1 of the first embodiment shown in FIG.

  Although described with reference to FIG. 10, even if the drive transistor 121 operates in the saturation region even if the drive transistor 121 operates in the saturation region, a difference occurs in the drain-source voltage Vds of the p-type transistor 402 due to the difference in the actual light emission currents Iel_Sum100 and Iel_Sum50. As shown in FIG. 12A, due to the Early effect, there is a slight difference between the drive currents Ids_100 and Ids_50 between the 100% display row and the 50% display row (Ids_100 <Ids_50).

  In addition, in the organic EL display device 1 according to the first embodiment shown in FIG. 11, a correction transistor 512 is provided between the power supply line 105 DSL and the drain end D of the drive transistor 121 for each pixel circuit P. Therefore, as shown in FIGS. 12A and 13, first, at the drain end D of the driving transistor 121 in the 50% display row, the white display pixel is corrected more than the voltage V105DSL50 of the power supply line 105DSL in that row. The voltage drop by the transistor 512 is further reduced by Vds_50W (= R512 * Ids_50W) to become “V121D_50W = V105DSL50−R512 * Ids_50W”, whereas in the black display pixel, it may be considered that Ids_50B≈0 “V121D_50B = V105DSL50” It becomes.

  The influence that the voltage V105DSL of the power supply line 105DSL fluctuates depending on the display pattern acts in such a direction that the voltage V125D at the drain terminal D of the driving transistor 121 is mitigated. When the correction transistor 512 is not provided, the actual voltage V121D at the drain terminal D of the drive transistor 121 varies in units of rows according to the display pattern. However, by providing the correction transistor 512 for each pixel circuit P, the drive current Ids ( That is, a voltage drop corresponding to the light emission current Iel) occurs between the drain and the source, so that the voltage difference at the drain end D of the drive transistor 121 due to the display pattern becomes small.

  By setting the control signal Vcnt1 to an optimum value, the voltage at the drain terminal D of the drive transistor 121 is maintained constant regardless of the window pattern, and the display luminance fluctuation caused by the display pattern is suppressed. Thereby, the light emission luminance is adjusted regardless of the presence or absence of the window pattern, and the horizontal crosstalk can be prevented.

  For example, as shown in FIG. 12A, the actual drive current Ids_50W of the white display pixels in the 50% display row is further reduced than the operating current Ids_50 in V105DSL50 (Ids_50W <Ids_50). As a result, as shown in FIGS. 12B and 12C, in the 50% display row, the degree of luminance reduction of the white display pixels is slightly increased as compared with the case where the correction transistor 512 is not provided. Become.

  If the brightness of the white display pixels in the 100% display row is the same as that in the case where the correction transistor 512 is not provided, the degree of the brightness reduction of the white display pixels in the 50% display row is 100% display. It means approaching the degree of brightness reduction of white display pixels in a row. As a result, by further reducing the light emission luminance of the white display pixels in the 50% display row, the difference from the light emission luminance of the white display pixels in the 100% display row can be reduced, and lateral crosstalk can be prevented. It becomes.

  Here, since the correction transistor 512 is also provided in the 100% display row, it is necessary to consider this point. That is, all of the 100% display rows are white display pixels. At the drain terminal D of the drive transistor 121 in this white display pixel, as shown in FIGS. 12A and 13, the voltage drop Vds_100W (= R512 *) by the correction transistor 512 is higher than the voltage V105DSL100 of the power supply line 105DSL in that row. Ids_100W) further decreases to “V121D_100W = V105DSL100−R512 * Ids_100W”. As a result, as shown in FIG. 12A, the actual drive current Ids_100W of the white display pixel in the 100% display row is further reduced than the operating current Ids_100 in V105DSL100 (Ids_100W <Ids_100).

  As a result, in the 100% display row, the luminance of the white display pixel is lower than that in the case where the correction transistor 512 is not provided. In this case, the effect of preventing horizontal crosstalk by further reducing the light emission luminance of the white display pixels in the 50% display row is offset.

  However, if the luminance reduction due to Vds (= R512 * Ids_100W) of the correction transistor 512 in this 100% display row is smaller than the luminance reduction due to Vds (= R512 * Ids_50W) of the correction transistor 512 in the 50% display row, The effect of preventing lateral crosstalk can be obtained.

  In the organic EL display device 1 of the first embodiment, the correction transistor 512 functioning as a current-voltage conversion unit is used as the voltage drop generation unit 510. However, this is only an example, and a predetermined operating resistance is used. Various elements (semiconductor elements) as long as the voltage drop Vds (= R512 * Ids = R512 * Iel) corresponding to the product of the operating resistance and the drive current Ids (= light emission current Iel) can be generated. Can be used). As an example, a diode element can be used. This diode element may have a configuration in which the gate terminal G and the drain terminal D of the transistor are connected.

  Moreover, not only semiconductor elements, such as a transistor and a diode element, you may use a resistive element. When a resistance element is used, the resistance value is set so that the display luminance fluctuation caused by the actual voltage V121D at the drain terminal D of the driving transistor 121 fluctuating in units of rows according to the display pattern can be more reliably suppressed. Set to the optimal value. However, in terms of process, a semiconductor element is easier to manufacture than a resistance element.

  In addition, when a diode element is used, an operating resistance defined by the slope of the voltage-current characteristic is used. In order to more surely suppress the display luminance fluctuation caused by the fluctuation of the actual voltage V121D at the drain end D of the driving transistor 121 according to the display pattern in units of rows, the inclination is optimized in the process. Will be adjusted to the correct value.

  On the other hand, if the correction transistor 512 is used as in the organic EL display device 1 of the first embodiment, the control signal generator 520 is provided to optimize the control signal Vcnt1 supplied to the gate terminal G. There are advantages that can be made. It is also possible to adjust dynamically based on the signal potential Vin as necessary. In the organic EL display device 1 of the second embodiment to be described later, this dynamic adjustment is applied.

<Improvement Method: Second Embodiment>
FIG. 14 is a diagram showing a second embodiment of an organic EL display device that solves the problem described in FIG. In the organic EL display device 1 according to the second embodiment, when a voltage drop corresponding to the drive current Ids and the light emission current Iel is applied to the drain terminal D of the drive transistor 121 for each pixel circuit P, the operation of a transistor as an example of a semiconductor element is performed. It has a feature in that a resistor is used (same as the pixel circuit P of the first embodiment), and also has a feature in that its gate terminal G is dynamically controlled according to the video signal Vsig (especially the signal potential Vin). .

  Specifically, a control signal generation unit 522 that generates the control signal Vcnt2 is provided instead of the control signal generation unit 520 of the first embodiment. The control signal generator 522 is supplied with the video signal Vsig. The control signal generation unit 522 may be provided in common for all the pixel circuits P, and may be connected to the gate terminal G of each correction transistor 512 for each row by row wiring.

  The voltage drop generation unit 510 and the control signal generation unit 522 suppress the influence of the voltage drop in the output circuit 400 due to the signal potential Vin at the drain terminal D as the power supply terminal of the drive transistor 121. A second example of the suppression unit 107 is configured.

  The control signal generation unit 522 is driven for each row or for each pixel circuit P, more preferably for each row and for each pixel circuit P, regardless of the display pattern, based on the signal potential Vin of the input video signal Vsig. This is characterized in that the control signal Vcnt2 is adjusted in a direction in which the voltage V121D at the drain terminal D of the transistor 121 is kept constant. That is, the operation is performed based on the signal potential Vin so that the voltage variation due to the display pattern at the drain terminal D of the driving transistor 121 is suppressed, and more preferably, the voltage V121D at the drain terminal D is constant regardless of the display pattern. Make adjustments.

  By correcting the correction transistor 512 between the power supply line 105DSL and the drain terminal D of the driving transistor 121, the correction transistor 512 functions as a current-voltage conversion unit, and a control signal supplied to the gate terminal G thereof. The voltage drop Vds (= R512 * Ids = R512 * Iel) corresponding to the product of the operating resistance R512 (= f (Vcnt1)) corresponding to Vcnt2 and the drive current Ids (= light emission current Iel) is the drain end of the drive transistor 121. Although it is common to the pixel circuit P of the first embodiment in that it is given to D, it is different in that the control signal Vcnt2 is adjusted so that the operation resistance R512 corresponding to the display pattern is set for each row.

  The correction transistor 512 is disposed between the supply terminal 400H for the first potential Vcc_H and the drain terminal D of the driving transistor 121, and the gate terminal G is optimized by the control signal generator 522 in accordance with the signal potential Vin. By controlling with the control signal Vcnt2, the influence of the voltage drop at the p-type transistor 402 of the output circuit 400 due to the signal potential Vin is suppressed at the drain terminal D as the power supply terminal of the drive transistor 121.

  The operating resistance R512 is dynamically adjusted according to the signal potential Vin so as to cancel the fluctuation of the voltage V105DSL of the power supply line 105DSL according to the display pattern. As a result, the voltage V121D of the drain terminal D of the driving transistor 121 Fluctuation is mitigated. By appropriately setting the magnitude (gain) of the control signal Vcnt2 with respect to the signal potential Vin, in the optimum state, the voltage V121D at the drain terminal D of the drive transistor 121 is maintained constant regardless of the display pattern. .

<Improvement method: operation of the second embodiment>
FIG. 15 is a diagram for explaining the operation and effect of the voltage drop generation unit 522 in the organic EL display device 1 of the second embodiment shown in FIG.

  In adjusting the control signal Vcnt2 so that the operating resistance R512 according to the display pattern is set for each row, the voltage V121D at the drain terminal D of the drive transistor 121 is constant regardless of the display pattern. A control signal Vcnt2 for controlling the operating resistance R512 of the correction transistor 512 is generated using an index value such as an average value or a median value representing the state of the signal potential.

  In the following description, it is assumed that the control signal Vcnt2 is generated according to the average value Vin_Ave of the signal potential Vin for one row. For example, when the control signal Vcnt2_100 for the 100% display row is set lower than the control signal Vcnt2_50 for the 50% display row, the operating resistance R512_100 of the correction transistor 512 in the 100% display row is 50%. The operation resistance R512_50 of the correction transistor 512 in the display row is made smaller (R512_50> R512_100).

  As a result, even if the signal potential Vin is the same, a smaller voltage drop Vds is generated between the drain and source of the correction transistor 512 as the average value Vin_Ave of the signal potential Vin for one row is larger. The voltage V121D at the drain end D is dynamically adjusted so as to suppress variation due to the display pattern (preferably so as to be constant regardless of the display pattern).

  Specifically, in each white display portion of the 50% display row and the 100% display row, the voltage drop Vds_50W due to the correction transistor 512 in the 50% display row is larger, and the voltage at the drain terminal D of the drive transistor 121 is larger. While the decrease in V121D is large, the voltage drop Vds_100W due to the correction transistor 512 in the 100% display row is smaller and the decrease in the voltage V121D at the drain terminal D of the drive transistor 121 is small.

  In the 100% display row and the 50% display row, the actual power supply line 105DSL operates in a direction that cancels out the influence of the fluctuations of the voltages V105DSL_100 and V105DSL_50 with respect to the first potential Vcc_H on the drain terminal D of the drive transistor 121. Thus, the voltage V121D_50W and the voltage V121D_100W act in the direction in which they match. Since the voltage V121D of the drain terminal D of the drive transistor 121 is maintained to be constant, display is performed by optimizing the magnitude (gain) of the control signal Vcnt2 with respect to the average value Vin_Ave of the signal potential Vin for one row. Regardless of the pattern, the potential V105DSL can be maintained constant.

  If the potential V105DSL can be kept constant regardless of the display pattern, a difference in the drive current Ids does not occur, so that horizontal crosstalk can be prevented.

  As mentioned above, although this invention was demonstrated using embodiment, the technical scope of this invention is not limited to the range as described in the said embodiment. Various changes or improvements can be added to the above-described embodiment without departing from the gist of the invention, and embodiments to which such changes or improvements are added are also included in the technical scope of the present invention.

  Further, the above-described embodiments do not limit the invention according to the claims (claims), and all combinations of features described in the embodiments are not necessarily essential to the solution means of the invention. Absent. The embodiments described above include inventions at various stages, and various inventions can be extracted by appropriately combining a plurality of disclosed constituent elements. Even if some constituent requirements are deleted from all the constituent requirements shown in the embodiment, as long as an effect is obtained, a configuration from which these some constituent requirements are deleted can be extracted as an invention.

<Modification of drive timing>
For example, various modifications can be made while setting the timing at which the potential of the power supply line 105DSL transitions from the second potential Vcc_L to the first potential Vcc_H as the period of the reference potential Vo, which is the ineffective period of the video signal Vsig.

  For example, as a first modification, although not shown, the setting method of the sampling period & mobility correction period H can be modified with respect to the drive timing shown in FIG. Specifically, first, the timing t15V at which the video signal Vsig changes from the reference potential Vo to the signal potential Vin is shifted to the latter half of one horizontal period from the driving timing shown in FIG. Narrow the period.

  When the threshold correction operation is completed (when the threshold correction period E is completed), the horizontal drive unit 106 first supplies the video signal line 106HS to the video signal Vsig while the write drive pulse WS remains active H. The period from when the potential Vin is supplied (t16) until the write drive pulse WS is changed to inactive L (t17) is the writing period of the pixel signal Vsig to the storage capacitor 120. This signal potential Vin is held in the form of adding to the threshold voltage Vth of the drive transistor 121. As a result, fluctuations in the threshold voltage Vth of the drive transistor 121 are always canceled, and threshold correction is performed. By this threshold value correction operation, the gate-source voltage Vgs held in the holding capacitor 120 becomes “Vsig + Vth”. At the same time, the mobility correction is executed in the signal writing period t16 to t17. That is, the timings t16 to t17 serve as both a signal writing period and a mobility correction period.

  In the period from t16 to t17 in which the mobility correction is performed, the organic EL element 127 does not emit light because it is actually in the reverse bias state. In the mobility correction period t16 to t17, the drive current Ids flows through the drive transistor 121 while the gate end G of the drive transistor 121 is fixed at the level of the video signal Vsig. The driving timing is the same as that shown in FIG.

  Even at the driving timing of the first modification, the switching operation of the power supply to the drain terminal D of the driving transistor 121 is completely the same as the driving timing shown in FIG. It can be enjoyed in the same manner as the above-described embodiment.

  Each drive unit (104, 105, 106) adjusts the relative phase difference between the video signal Vsig supplied from the horizontal drive unit 106 to the video signal line 106HS and the write drive pulse WS supplied from the write scanning unit 104. Thus, the mobility correction period can be optimized.

  However, the writing & mobility correction preparation period G does not exist, and the timing t16V to t17W becomes the sampling period & mobility correction period H. For this reason, a difference in waveform characteristics due to the influence of the wiring resistance and wiring capacitance of the write scanning line 104WS and the video signal line 106HS may affect the sampling period & mobility correction period H. . Since the sampling potential and the mobility correction time are different between the side closer to the writing scanning unit 104 and the far side (that is, the left and right sides of the screen), a luminance difference occurs between the left and right sides of the screen, and there is a difficulty in being visually recognized as shading. Concerned.

  Further, as a second modification, it is possible to change the off timing (transition timing to the second potential Vcc_L side) of the power supply line 530DSL. Specifically, both the off timing and the on timing of the row can be set to the same horizontal period. For example, in the light emission period of the previous field before the timing t13W when the write drive pulse WS is made active H, the write drive pulse WS is inactive L and the sampling transistor 125 is non-conductive, while the power drive pulse DSL is Since it is at the first potential Vcc_H on the high potential side, regardless of the potential of the video signal line 106HS, the voltage state (the gate-source voltage Vgs of the drive transistor 121) held in the holding capacitor 120 by the operation in the previous field is changed. Accordingly, the drive current Ids is supplied from the drive transistor 121 to the organic EL element 127, and the organic EL element 127 is in a light emitting state.

  Thereafter, a new field of line sequential scanning is entered, and first, the write drive pulse WS is switched from inactive L to active H (t13W). At this time, by setting the potential of the video signal Vsig in the video signal line 106HS to the reference potential Vo (t13V), the gate potential Vg of the drive transistor 121 is initialized. A storage capacitor 120 is connected between the gate terminal G and the source terminal S of the drive transistor 121, and due to the effect of the storage capacitor Cs, the source potential (Vs) is affected by the variation in the gate potential (Vg) of the drive transistor 121. Are linked. This period is referred to as a gate initialization period.

  Next, with the write drive pulse WS kept active H, the power drive pulse DSL applied to the power supply line 105DSL is set to the second potential Vcc_L on the low potential side (t_off: corresponding to the off timing t11). As a result, power supply to the driving transistor 121 is stopped, and the source potential Vs of the driving transistor 121 is initialized to the second potential Vcc_L. This period is referred to as a source initialization period. The gate initialization period and the source initialization period are collectively referred to as a threshold correction preparation period in which the gate potential Vg and the source potential Vs of the driving transistor 121 are initialized.

  In this way, by preparing the gate potential Vg and the source potential Vs of the drive transistor 121, the preparation for the threshold voltage correction operation is completed. Thereafter, the period t13V to t14 of (t14) is the initialization period until the power supply pulse DSL applied to the power supply line 105DSL is set to the first potential Vcc_H.

  Next, the power supply drive pulse DSL applied to the power supply line 105DSL is set to the first potential Vcc_H while the write drive pulse WS remains active H (t14). As a result, the drain current flows into the storage capacitor 120 and enters a threshold correction period in which the threshold voltage Vth of the drive transistor 121 is corrected (cancelled). Hereinafter, the driving timing shown in FIG. 6 and the first modification example corresponding thereto are the same.

  At the drive timing of the second modification, both power supply switching operations are performed during the period of the reference potential Vo of the video signal Vsig. At this time, the sampling transistor 125 is turned on and the gate end G of the drive transistor 121 is set to the reference potential. Since the impedance is fixed to Vo and the impedance is low, the tolerance to the coupling noise caused by the power pulse (the power scan pulse DS on the power scan line 105DS and the power drive pulse DSL on the power supply line 530DSL based thereon) is improved.

<Modification of Pixel Circuit>
In addition, since “dual theory” holds in circuit theory, the pixel circuit P can be modified from this viewpoint. In this case, although illustration is omitted, first, the pixel circuit P shown in FIGS. 11 and 14 is configured using an n-channel transistor, whereas the pixel circuit P using a p-channel transistor is used. Configure. In accordance with this, a change according to the dual reason is made, such as reversing the polarity of the signal potential Vin with respect to the reference potential Vo of the video signal Vsig and the magnitude relation of the power supply voltage.

  For example, in the pixel circuit P having a modification according to the “dual theory”, the storage capacitor 120 is connected between the gate terminal G and the source terminal S of a p-channel type driving transistor (hereinafter referred to as a p-type driving transistor 121p). The source terminal S of the p-type driving transistor 121p is directly connected to the cathode terminal K of the organic EL element 127. The anode end A of the organic EL element 127 has an anode potential Vanode as a reference potential. This anode potential Vanode is connected to a reference power supply (high potential side) common to all pixels for supplying a reference potential.

  The p-type drive transistor 121p passes a drive current Ids that causes the organic EL element 127 to emit light when the first potential Vcc_L on the low voltage side is supplied to the drain terminal D thereof. For this reason, the drain terminal D of the p-type drive transistor 121p is connected to the power supply line 105DSL from the drive scanning unit 105 functioning as a power supply scanner via the p-type voltage drop generation unit 510p. In the voltage drop generation unit 510p, a p-channel correction transistor (hereinafter referred to as a p-type correction transistor 512p) is arranged in cascode connection with respect to the p-type drive transistor 121p. The p-type correction transistor 512p has the same function as the above-described correction transistor 512, and a control signal is supplied to the gate terminal G from the same components as the control signal generation unit 520 and the control signal generation unit 522 described above. To do.

  The power supply line 105DSL itself has a power supply capability for the p-type drive transistor 121p. The drive scanning unit 105 supplies the first potential Vcc_L on the low voltage side corresponding to the power supply voltage and the second potential Vcc_H on the high voltage side by switching to the drain terminal D side of the p-type drive transistor 121p. In other words, the drive scanning unit 105 that drives the pixel circuit P shown in FIGS. 11 and 14 has an active-H power supply capability, while driving the pixel circuit P of a modified example that applies dual reason. The driving scanning unit 105 that has an active L power supply capability. The second potential Vcc_H is set to a potential sufficiently higher than the reference potential Vo of the video signal Vsig on the video signal line 106HS.

  The p-channel type sampling transistor 125 has a gate terminal G connected to the writing scanning line 104WS from the writing scanning unit 104, a source terminal S connected to the video signal line 106HS, and a drain terminal D connected to the p-type driving transistor 121p. Is connected to the gate end G. The gate terminal G is supplied with an active L write drive pulse WS from the write scanning unit 104.

  Even in the organic EL display device of the modified example to which such duality is applied, the influence of the voltage V105DSL of the power supply line 105DSL fluctuating depending on the display pattern is mitigated by the voltage V125D of the drain terminal D of the p-type drive transistor 121p. By doing so, a mechanism for suppressing lateral crosstalk can be applied, and of course, a threshold correction operation, a mobility correction operation, and a bootstrap operation can also be executed.

  In addition, although the modification demonstrated here adds the change according to "the dual reason" with respect to the structure shown in FIG.11 and FIG.14, the method of a circuit change is limited to this is not. In performing the threshold correction operation, the video signal Vsig switched between the reference potential Vo and the signal potential Vin within each horizontal period in accordance with the line sequential scanning in the writing scanning unit 104 is driven so as to be transmitted to the video signal line 106HS. 11 and FIG. 14 described above, the influence that the voltage V105DSL of the power supply line 105DSL fluctuates depending on the display pattern is mitigated by the voltage V125D of the drain terminal D of the p-type driving transistor 121p. By doing so, it is possible to apply the idea of the present embodiment of suppressing lateral crosstalk.

1 is a block diagram showing an outline of a configuration of an active matrix display device which is an embodiment of a display device according to the present invention. It is a figure which shows the 1st comparative example with respect to the pixel circuit of this embodiment. 3 is a timing chart for explaining the operation of the pixel circuit of the first comparative example shown in FIG. 2. It is a figure explaining the influence which the characteristic variation of an organic EL element or a drive transistor has on a drive current. It is a figure explaining the concept of the improvement method of the influence which the characteristic variation of a drive transistor has on a drive current. It is a figure which shows the 2nd comparative example with respect to the pixel circuit of this embodiment. 6 is a timing chart for explaining a basic example of drive timing related to the pixel circuit of the second comparative example shown in FIG. 5 (the same applies to the pixel circuit of the present embodiment). It is an equivalent circuit of light emission period B in the drive timing with respect to the pixel circuit of a 2nd comparative example, and a figure of operation | movement description. It is an equivalent circuit of the discharge period C in the drive timing with respect to the pixel circuit of the 2nd comparative example, and a figure of operation | movement description. It is an equivalent circuit of the initialization period D in the drive timing with respect to the pixel circuit of the 2nd comparative example, and a figure of operation | movement description. It is an equivalent circuit of the threshold value correction period E in the drive timing with respect to the pixel circuit of the 2nd comparative example, and a figure of operation | movement description. It is an equivalent circuit of the period F in the drive timing with respect to the pixel circuit of a 2nd comparative example, and a figure of operation | movement description. It is an equivalent circuit of the writing & mobility correction preparation period G in the drive timing with respect to the pixel circuit of the 2nd comparative example, and a figure of operation | movement description. It is an equivalent circuit of the sampling period & mobility correction period H in the drive timing with respect to the pixel circuit of the 2nd comparative example, and a figure of operation | movement description. It is an equivalent circuit of light emission period I in the drive timing with respect to the pixel circuit of the 2nd comparative example, and a figure of operation explanation. It is a figure explaining the output circuit of a drive scanning part (drive scanner). It is a figure explaining the operation | movement of the output circuit of a drive scanning part, and a pixel circuit in the light emission period of an organic EL element. It is a figure explaining the 1st problem which may arise when it is set as the control system which drives with the power supply voltage which switches the drain terminal of a drive transistor with 1st electric potential and 2nd electric potential. It is a figure explaining the problem given to the drive current of the drive transistor and the light emission current of an organic EL element resulting from the voltage of a power supply line differing with a display pattern. It is a figure which shows 1st Embodiment of the organic electroluminescence display which eliminates the problem demonstrated in FIG. It is FIG. (1) explaining the operation | movement and effect of the voltage drop production | generation part in the organic electroluminescence display of 1st Embodiment shown in FIG. It is FIG. (2) explaining the operation | movement and effect of the voltage drop production | generation part in the organic electroluminescence display of 1st Embodiment shown in FIG. It is a figure which shows 2nd Embodiment of the organic electroluminescence display which eliminates the problem demonstrated in FIG. It is a figure explaining the operation | movement and effect of the voltage drop production | generation part in the organic electroluminescence display of 2nd Embodiment shown in FIG.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Organic EL display device, 100 ... Display panel part, 101 ... Substrate, 102 ... Pixel array part, 103 ... Vertical drive part, 104 ... Write scanning part, 104WS ... Write scanning line, 105 ... Drive scanning part, 105DS ... Drive scanning line, 105DSL ... Power supply line, 106 ... Horizontal drive unit, 106HS ... Video signal line, 107 ... Power supply potential fluctuation suppressing unit, 109 ... Control unit, 120 ... Retention capacitor, 121 ... Drive transistor, 125 ... Sampling transistor DESCRIPTION OF SYMBOLS 127 ... Organic EL element, 400 ... Output circuit, 402 ... P-type transistor, 404 ... N-type transistor, 510 ... Voltage drop generation part, 512 ... Correction transistor, 520, 522 ... Control signal generation part, Cel ... Organic EL element Parasitic capacitance, DSL ... power supply drive pulse, P ... pixel circuit, Vsig ... video signal, Vin ... signal potential, Vo ... reference potential, Vcc_H ... first Place, Vcc_L ... the second potential, WS ... write drive pulse

Claims (15)

A driving transistor for generating a driving current, a holding capacitor connected between a control input terminal and an output terminal of the driving transistor, an electro-optic element connected to an output terminal of the driving transistor, and sampling for writing a signal to the holding capacitor A pixel circuit that includes a transistor and emits a driving current based on a signal held in the holding capacitor by the driving transistor and flows through the electro-optical element is arranged in a matrix. A pixel array section;
A write scanning section for sequentially scanning the pixel circuits by sequentially controlling the sampling transistors in a horizontal period and writing information corresponding to a signal potential of a video signal to each holding capacitor for one row; and the writing The first potential from the supply terminal for the first potential used to flow a driving current to the electro-optic element to each pixel circuit for one row in accordance with the line sequential scanning in the scanning unit, A control unit including a drive scanning unit having an output circuit that switches and outputs the second potential from the supply terminal for the second potential different from the first potential;
The signal potential is provided between a supply terminal for the first potential and the power supply terminal of the drive transistor, and uses the voltage drop caused by the operation current of the drive transistor flowing through its operation resistance. And a power supply potential fluctuation suppressing section that suppresses an influence of a voltage drop in the output circuit due to the power supply terminal of the drive transistor. A driving transistor for generating a driving current, a holding capacitor connected between a control input terminal and an output terminal of the driving transistor, an electro-optic element connected to an output terminal of the driving transistor, and sampling for writing a signal to the holding capacitor A pixel circuit that includes a transistor and emits a driving current based on a signal held in the holding capacitor by the driving transistor and flows through the electro-optical element is arranged in a matrix. A pixel array section;
The first potential from the supply terminal for the first potential used to flow the driving current to the electro-optic element is supplied to the pixel circuit, and the reference potential in the video signal is supplied to the sampling transistor. Prior to performing a threshold correction operation for holding a voltage corresponding to the threshold voltage of the driving transistor in the holding capacitor by conducting the sampling transistor in a time zone, the second potential different from the first potential is used. The preparatory operation for the threshold value correcting operation is performed by turning on the sampling transistor in a time zone in which the second potential from the supply terminal is supplied to the pixel circuit and the reference potential is supplied to the sampling transistor. A control unit for controlling
The signal potential is provided between a supply terminal for the first potential and the power supply terminal of the drive transistor, and uses the voltage drop caused by the operation current of the drive transistor flowing through its operation resistance. And a power supply potential fluctuation suppressing section that suppresses an influence of a voltage drop in the output circuit due to the power supply terminal of the drive transistor. For each row, the output circuit, and a power supply line for connecting the output circuit and the pixel circuit for one row,
The power supply potential fluctuation suppressing unit is represented by the product of its own operating resistance and the current flowing through the drive transistor, which is arranged between the power supply line and the power supply end of the drive transistor for each pixel circuit. The display device according to claim 1, further comprising a current-voltage converter that generates a voltage drop. The display device according to claim 1, wherein the current-voltage conversion unit includes a correction transistor arranged in a cascode connection between the power supply line and a power supply end of the drive transistor. The display device according to claim 4, wherein a constant voltage is supplied to a gate terminal of the correction transistor regardless of a signal potential. The display device according to claim 4, wherein the power supply potential fluctuation suppressing unit further includes a control signal generation unit that generates a control signal for controlling an operating resistance of the current-voltage conversion unit. The power supply potential fluctuation suppressing unit is a control signal for controlling the operating resistance of the current-voltage converter so that voltage fluctuation due to a display pattern at the power supply end of the driving transistor is suppressed based on the signal potential. The display device according to claim 4, further comprising: a control signal generation unit that generates The display device according to claim 7, wherein the control signal generation unit generates a control signal for controlling an operating resistance of the current-voltage conversion unit based on the signal potential for one row. The control unit conducts the sampling transistor during a time period in which a voltage corresponding to the first potential is supplied to the power supply terminal of the driving transistor and a reference potential in the video signal is supplied to the sampling transistor. 2. The display device according to claim 1, wherein the display device is controlled to perform a threshold value correction operation for holding a voltage corresponding to a threshold voltage of the driving transistor in the storage capacitor. 10. The display according to claim 2, wherein the control unit performs control so that the threshold correction operation is repeatedly performed in a plurality of horizontal periods preceding the writing of the signal potential to the storage capacitor. apparatus. Prior to the threshold correction operation, the control unit supplies a voltage corresponding to the second potential to the power supply terminal of the drive transistor and supplies a reference potential of the video signal to the sampling transistor. The display device according to claim 9, wherein the sampling transistor is turned on in a band and controlled to perform a preparatory operation for the threshold correction operation. After the threshold correction operation, the control unit causes the sampling transistor to conduct in a time zone in which the first potential is supplied to the driving transistor and the reference potential is supplied to the sampling transistor, thereby holding the storage capacitor. 10. The display device according to claim 2, wherein when the signal potential is written to the signal, a correction for the mobility of the driving transistor is added to the signal written to the storage capacitor. The display device according to claim 12, wherein the control unit causes the sampling transistor to conduct at a predetermined position within a time zone in which the reference potential is supplied to the sampling transistor for a period shorter than the time zone. . The control unit makes the sampling transistor non-conductive at the time when the signal potential is written to the storage capacitor, stops supply of the video signal to the control input terminal of the drive transistor, and The display device according to claim 1, wherein an operation in which a potential of the control input terminal is interlocked with a potential fluctuation of the output terminal is enabled. A driving transistor for generating a driving current, a holding capacitor connected between a control input terminal and an output terminal of the driving transistor, an electro-optic element connected to an output terminal of the driving transistor, and sampling for writing a signal to the holding capacitor A pixel circuit that includes a transistor and emits a driving current based on a signal held in the holding capacitor by the driving transistor and flows through the electro-optical element is arranged in a matrix. A pixel array unit and a writing scanning unit that sequentially controls the sampling transistors in a horizontal cycle to scan the pixel circuit in a line-sequential manner and writes information corresponding to the signal potential of the video signal to each holding capacitor for one row In order to pass a driving current to the electro-optic element in each pixel circuit for one row in accordance with the line-sequential scanning in the writing scanning unit A first potential to be use, a driving method of the display device and a driving scanning section for outputting by switching between different second potential to the first potential,
A current-voltage converter is provided between the supply terminal for the first potential and the power supply terminal of the drive transistor, and a voltage drop is obtained by multiplying the operating resistance of the current-voltage converter by the current flowing through the drive transistor. By generating, the influence of the voltage drop of the first potential due to the signal potential is suppressed at the power supply end of the drive transistor.

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