JP2008241948A - Display device and its driving method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To prevent an adverse effect of a shading phenomenon in accordance with a threshold correction operation in an organic EL display device having a threshold correction function. <P>SOLUTION: A period of the threshold correction operation is controlled so that a period for making a sampling transistor into a conducted state becomes dominant. When control is performed so that the period for making the sampling transistor into the conducted state, since termination of threshold correction comes to be prescribed by OFF of the sampling transistor, coupling of a scanning drive pulse for turning on/off of light emission control transistor is not generated and the adverse effect of the shading phenomenon in accordance with the threshold correction is not generated. Since the light emission control transistor is operated in a linear area even in a threshold correction period while avoiding generation of the adverse effect of the shading phenomenon in accordance with the threshold correction, there is no risk of increase of power consumption. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a display device having a pixel array portion in which pixel circuits (also referred to as pixels) having electro-optical elements (also referred to as display elements and light-emitting elements) are arranged in a matrix, and a driving method thereof. . More specifically, pixel circuits having electro-optic elements whose luminance changes depending on the magnitude of the drive signal as display elements are arranged in a matrix, each pixel circuit has an active element, and the active element is used for each pixel. The present invention relates to an active matrix display device in which display driving is performed and a driving method thereof.

  As a display element of a pixel, there is a display device using an electro-optical element whose luminance changes depending on an applied voltage or a flowing current. For example, a liquid crystal display element is a typical example of an electro-optical element whose luminance changes depending on an applied voltage, and an organic electroluminescence (Organic Electro Luminescence, Organic EL, Organic) (Light Emitting Diode, OLED; hereinafter referred to as “organic EL”) A typical example is an element. The organic EL display device using the latter organic EL element is a so-called self-luminous display device using an electro-optic element which is a self-luminous element as a pixel display element.

  An organic EL element is an electro-optical element utilizing a phenomenon that light is emitted when an electric field is applied to an organic thin film. Since the organic EL element can be driven with a relatively low applied voltage (for example, 10 V or less), the power consumption is low. Further, since the organic EL element is a self-luminous element that emits light by itself, an auxiliary illumination member such as a backlight that is required in a liquid crystal display device is not required, and the weight and thickness can be easily reduced. Furthermore, since the response speed of the organic EL element is very high (for example, about several μs), no afterimage occurs when displaying a moving image. Because of these advantages, development of flat self-luminous display devices using organic EL elements as electro-optical elements has been actively performed in recent years.

  By the way, in a display device using an electro-optic element such as a liquid crystal display device using a liquid crystal display element and an organic EL display device using an organic EL element, a simple (passive) matrix method and an active device are used as the driving method. A matrix method can be adopted. However, a simple matrix display device has problems such as a simple structure and a difficulty in realizing a large and high-definition display device.

  Therefore, in recent years, a pixel signal supplied to a light emitting element in a pixel has been converted into an active element, for example, an insulated gate field effect transistor (generally a thin film transistor (TFT)) as a switching transistor. Active matrix systems that are used and controlled have been actively developed.

  Here, when the electro-optic element in the pixel circuit is caused to emit light, an input image signal supplied via the video signal line is switched by a switching transistor and a storage capacitor (control input terminal) provided at the gate end (control input terminal) of the drive transistor. The drive signal corresponding to the input image signal is supplied to the electro-optical element.

  In a liquid crystal display device using a liquid crystal display element as an electro-optical element, the liquid crystal display element is a voltage-driven element, and thus the liquid crystal display element is driven with a voltage signal itself corresponding to an input image signal taken into the storage capacitor. On the other hand, in an organic EL display device using an organic EL element as an electro-optical element, the organic EL element is a current-driven element, and therefore, a drive signal (voltage signal) corresponding to an input image signal taken into the storage capacitor. ) Is converted into a current signal by the driving transistor, and the driving current is supplied to the organic EL element.

  In a current-driven electro-optical element, typically an organic EL element, the light emission luminance varies depending on the drive current value. Therefore, in order to emit light with stable luminance, it is important to supply a stable drive current to the electro-optical element. For example, driving methods for supplying a driving current to the organic EL element can be broadly classified into a constant current driving method and a constant voltage driving method (this is a well-known technique, and publicly known literature is not presented here).

  Since the voltage-current characteristic of the organic EL element has a large inclination, when constant voltage driving is performed, a slight voltage variation or a variation in element characteristics causes a large current variation, resulting in a large luminance variation. Therefore, generally, constant current driving using a driving transistor in a saturation region is used. Of course, even with constant current driving, if there is a current variation, luminance variations will be caused, but if the current variation is small, only small luminance variations will occur.

  In other words, even in the constant current driving method, the driving signal written and held in the holding capacitor according to the input image signal may be constant because the light emission luminance of the electro-optic element is unchanged. It becomes important. For example, in order that the light emission luminance of the organic EL element remains unchanged, it is important that the drive current corresponding to the input image signal is constant.

  However, the threshold voltage and mobility of an active element (driving transistor) that drives the electro-optical element vary due to process variations. In addition, characteristics of electro-optical elements such as organic EL elements vary with time. If there is such a variation in characteristics of the active element for driving or a characteristic variation of the electro-optical element, even the constant current driving method affects the light emission luminance.

  Therefore, in order to uniformly control the light emission luminance over the entire screen of the display device, a mechanism for correcting the luminance variation caused by the characteristic variation of the driving active element and the electro-optical element described above in each pixel circuit. Various studies have been made.

JP 2006-215213 A

  For example, in the mechanism described in Patent Document 1, as a pixel circuit for an organic EL element, a threshold correction function for making the drive current constant even when the threshold voltage of the drive transistor varies or changes over time, In order to keep the driving current constant even when the mobility-correction function for making the driving current constant even when the mobility of the organic EL element varies or changes with time, or when the current-voltage characteristic of the organic EL element changes with time A bootstrap function has been proposed.

  However, the mechanism described in Patent Document 1 requires a wiring for supplying a correction potential, a correction switching transistor, and a switching pulse for driving the wiring. A 5TR drive configuration using two transistors is employed, and the configuration of the pixel circuit is complicated. Since there are many components of a pixel circuit, it becomes a hindrance to high definition of a display apparatus. As a result, the 5TR drive configuration makes it difficult to apply to a display device used in a small electronic device such as a portable device (mobile device).

  For this reason, there is a demand for development of a method for suppressing luminance change due to variation in element characteristics while simplifying the pixel circuit. At this time, it should be taken into consideration that a new problem that does not occur in the configuration of the 5TR drive does not occur with the simplification.

  The present invention has been made in view of the above circumstances, and it is a general object of the present invention to provide a display device and a driving method thereof capable of increasing the definition of the display device by simplifying the pixel circuit.

  Further, it is particularly desirable to provide a mechanism that can reduce the influence of the operation of driving the pixel circuit on the image quality (particularly suppressing luminance unevenness) while simplifying the pixel circuit.

  Further, in order to simplify the pixel circuit, it is preferable to provide a mechanism that can suppress a change in luminance due to variation in characteristics of the driving transistor and the light emitting element.

  One embodiment of a display device according to the present invention is a display device that emits electro-optic elements in a pixel circuit based on a video signal. First, in a pixel circuit arranged in a matrix in a pixel array unit, At least a driving transistor that generates a driving current, an electro-optical element connected to the output terminal of the driving transistor, a holding capacitor that holds information (driving potential) corresponding to the signal potential of the video signal, and a signal potential in the video signal in the holding capacitor A sampling transistor for writing information according to the above is provided. In this pixel circuit, the electro-optic element is caused to emit light by generating a drive current based on information held in the holding capacitor by the drive transistor and flowing it through the electro-optic element.

  Since the sampling transistor writes information corresponding to the signal potential to the holding capacitor as the driving potential, the sampling transistor takes in the signal potential at its input end (source end or drain end) and outputs it (source end or drain end) Information corresponding to the signal potential is written in the storage capacitor connected to the other of the above. Of course, the output terminal of the sampling transistor is also connected to the control input terminal of the drive transistor.

  Note that the connection configuration of the pixel circuit shown here is the most basic configuration, and the pixel circuit only needs to include at least each of the above-described components. May be included. Further, the “connection” is not limited to being directly connected, but may be connected via other components.

  Even in a pixel circuit having such a modified mode, as long as the configuration and operation described in this section (means for solving the problem) can be realized, these modified modes are also displayed according to the present invention. 1 is a pixel circuit that implements an embodiment of an apparatus.

  For example, a change such as interposing a switching transistor or a functional unit having a certain function may be added between the connections as necessary. Typically, in order to dynamically control the display period (in other words, the non-light emission time), a switching transistor (light emission control transistor) is driven between the output terminal of the drive transistor and the electro-optical element or driven. A transistor may be disposed between a power supply end (a drain end is a typical example) and a power supply line which is a power supply wiring. Among these, in one embodiment of the display device according to the present invention, at least a light emission control transistor is disposed between a power supply end (a drain end is a typical example) of a driving transistor and a power supply line which is a power supply wiring. The basic features of the structure.

  Further, in the peripheral portion for driving the pixel circuit, for example, the pixel circuit is line-sequentially scanned by sequentially controlling the sampling transistors in the horizontal period, and the signal potential of the video signal is set to each holding capacitor for one row. Write scan unit for writing corresponding information, and output scan drive pulse for controlling power supply applied to power supply end of each drive transistor for one row in accordance with line sequential scanning in write scan unit A control unit including a drive scanning unit is provided. In addition, the control unit is provided with a horizontal driving unit that controls the video signal that is switched between the reference potential and the signal potential within each horizontal period in accordance with the line sequential scanning in the writing scanning unit to be supplied to the sampling transistor.

  The control unit further includes at least a time zone in which a voltage (so-called power supply voltage) corresponding to the first potential used to flow the drive current is supplied to the power supply end of the drive transistor via the light emission control transistor. Control is performed so that a fixed potential for threshold correction operation is supplied to the control input terminal of the drive transistor, and control is performed so as to perform threshold correction operation for holding the voltage corresponding to the threshold voltage of the drive transistor in the storage capacitor. . If necessary, a correction scanning unit for the control is provided. Preferably, a fixed potential for threshold correction operation is output to the video signal during a part of the horizontal scanning period. Thus, the sampling transistor can function as a switch transistor for applying a fixed potential.

  More preferably, the control unit performs control so as to perform a mobility correction operation for adding a correction amount for the mobility of the driving transistor to information written in the storage capacitor. If necessary, a correction scanning unit for the control is provided.

  It is preferable that the correction scanning unit for mobility correction operation and the correction scanning unit for threshold value correction operation are combined. Accordingly, the pixel circuit also causes the light emission control transistor to function as a correction switch transistor that operates in response to a pulse from the correction scanning unit for mobility correction operation or threshold value correction operation.

  The threshold value correcting operation may be repeatedly executed at a plurality of horizontal periods preceding the writing of the signal potential to the storage capacitor as necessary. Here, “as necessary” means a case where a voltage corresponding to the threshold voltage of the driving transistor cannot be sufficiently held in the storage capacitor in the threshold correction period within one horizontal cycle. By executing the threshold correction operation a plurality of times, a voltage corresponding to the threshold voltage of the drive transistor is reliably held in the holding capacitor.

  More preferably, the control unit initializes the potential at the control input terminal and the output terminal of the drive transistor so that the potential difference between both ends is equal to or greater than the threshold voltage prior to the threshold correction operation. Control to execute. More specifically, by setting a storage capacitor between the control input terminal and the output terminal, the potential difference between both ends of the storage capacitor is set to be equal to or higher than the threshold voltage. For this preparatory operation, the pixel circuit is preferably provided with a switch transistor.

  More preferably, after the threshold correction operation, the control unit conducts the sampling transistor during a time period in which the signal potential is supplied to the sampling transistor, thereby writing the signal potential information to the storage capacitor and moving the driving transistor. Control is performed so that the correction for the degree is added to the signal written in the storage capacitor.

  More preferably, when the information corresponding to the signal potential is written to the storage capacitor, the control unit turns off the sampling transistor to stop the supply of the video signal to the control input terminal of the drive transistor, and Control is performed so as to perform a bootstrap operation in which the potential at the control input terminal is linked to the potential fluctuation at the output terminal.

  The control unit preferably executes the bootstrap operation even after the end of the sampling operation, particularly at the beginning of the light emission start. That is, the potential difference between the control input terminal and the output terminal of the drive transistor is maintained constant by turning the sampling transistor non-conductive after the sampling transistor is turned on while the signal potential is supplied to the sampling transistor. Like that.

  In addition, the control unit preferably controls the bootstrap operation so as to realize the temporal variation correction operation of the electro-optic element in the light emission period. For this reason, the control unit continuously keeps the sampling transistor in a non-conductive state during the period in which the drive current based on the information held in the holding capacitor flows to the electro-optic element, so that the control input terminal and the output terminal It is preferable that the voltage can be maintained constant and the electro-optical element correction operation with time is realized.

  Here, as a characteristic matter in the embodiment of the display device according to the present invention, in the first mechanism, the control unit supplies the fixed potential for the threshold correction operation to the control input terminal of the drive transistor. In addition to the control, the threshold correction operation period is controlled so that the period during which the sampling transistor is in the conductive state is dominant.

  Alternatively, in the second mechanism, the control unit controls the fixed potential for the threshold correction operation to be supplied to the control input terminal of the drive transistor, and sets the light emission control transistor in the conductive state during the threshold correction operation period. The period is controlled to be dominant, and the emission control transistor operates in the saturation region during the threshold correction operation period, while the light emission period of the electro-optic element is controlled to operate in the linear region. .

  In both the first and second mechanisms, it is preferable that the voltage across the storage capacitor is set to the threshold voltage of the driving transistor by repeating the threshold correction operation a plurality of times in a time division manner. At this time, it is preferable that the horizontal drive unit outputs a fixed potential for threshold correction operation to the video signal during a part of the horizontal scanning period. Correspondingly, in the case of the first mechanism, the control unit keeps the light emission control transistor in a conducting state during a plurality of threshold correction operations and performs sampling during a fixed potential period in the video signal. Control is performed so that the threshold value correction operation is performed each time by switching the transistor to the conductive state. On the other hand, in the case of the second mechanism, the control unit keeps the sampling transistor conductive during the threshold correction operation for a plurality of times, and sets the light emission control transistor during the fixed potential period in the video signal. Control is performed so that the threshold correction operation is performed each time by switching to the conductive state.

  According to one embodiment of the present invention, as a first mechanism, the threshold correction operation period is controlled so that the period during which the sampling transistor is in a conductive state is dominant. If a control method is adopted in which the conduction period of the light emission control transistor becomes dominant, when the light emission control transistor is used in a linear region to reduce power consumption, it is driven when the light emission control transistor is turned off at the end of threshold correction. Threshold correction due to the coupling of the scan drive pulse that turns on and off the light emission control transistor at the power supply end of the transistor, and the amount of coupling differs depending on whether it is near or far from the scan unit that emits the scan drive pulse As a result, the shading phenomenon is adversely affected.

  On the other hand, if control is performed so that the period during which the sampling transistor is in a conductive state is dominant, the threshold correction end is defined by the sampling transistor being turned off, so the scanning drive for turning on / off the light emission control transistor Pulse coupling does not occur, and there is no adverse effect of shading due to threshold correction. Since the light emission control transistor can be operated in the linear region even during the threshold correction period while preventing the adverse effect of the shading phenomenon associated with the threshold correction, there is no concern about an increase in power consumption.

  In addition, when the threshold correction operation period is controlled so that the period during which the light emission control transistor is in a conductive state is dominant, the shading phenomenon caused by the threshold correction is caused by operating the light emission control transistor in a linear region. Is a factor. Therefore, as in the second mechanism, the adverse effect of the shading phenomenon associated with the threshold correction does not occur by operating the light emission control transistor in the saturation region during the threshold correction period. At the same time, if the light emission control transistor is operated in the linear region during the light emission period, there is no fear of an increase in power consumption during the light emission period.

  Further, in an active matrix display device using a current-driven electro-optic element such as an organic EL element in a pixel circuit, if each pixel circuit has at least a threshold correction function of a drive transistor, the threshold voltage varies. Therefore, a display device with good image quality can be realized. Desirably, a higher quality image can be obtained by providing a mobility correction function of the drive transistor.

  By correcting the threshold fluctuation of the driving transistor with the threshold correction function or correcting the mobility fluctuation of the driving transistor with the mobility correction function, the light emission luminance is kept constant without being affected by these fluctuations and variations. Because it can.

  Here, when the threshold correction function and the threshold correction preparation function (initialization function) that precedes the threshold correction function are realized, if the on / off control for the light emission control transistor is used in combination, the switch transistor for realizing these functions is used. The light emission control transistor 122 can function and is effective. In addition, if a fixed potential for threshold correction operation is output to the video signal during part of the horizontal scanning period, the sampling transistor can effectively function as a switch transistor for applying a fixed potential.

  As a result, based on the 2TR drive configuration, first, threshold correction is performed using a fixed potential supplied from a signal line for video signals while providing a light emission control transistor on the power supply end side of the drive transistor. Therefore, the number of constituent elements and the number of wirings of the pixel circuit can be greatly reduced, the pixel array portion can be reduced, and high-definition of the display device can be easily achieved. While simplifying the pixel circuit, it is possible to realize a function of correcting a luminance change due to a variation in element characteristics. Since the number of elements and the number of wirings are small, it is suitable for high definition, and a small display device that requires high definition display can be easily realized.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

<Overview of display device>
FIG. 1 is a block diagram showing an outline of a configuration of an active matrix display device which is an embodiment of a display device according to the present invention. In the present embodiment, for example, an organic EL element is used as a display element of a pixel, a polysilicon thin film transistor (TFT) is used as an active element, and the organic EL element is formed on a semiconductor substrate on which a thin film transistor is formed. A case where the present invention is applied to a matrix type organic EL display (hereinafter referred to as “organic EL display device”) will be described as an example.

  In the following, an organic EL element will be specifically described as an example of a pixel display element. However, this is merely an example, and the target display element is not limited to an organic EL element. In general, all the embodiments described later can be applied to all light emitting elements that emit light by current drive.

  As shown in FIG. 1, the organic EL display device 1 has an aspect ratio in which a pixel circuit (also referred to as a pixel) 110 having a plurality of organic EL elements (not shown) as display elements has a display aspect ratio. A display panel unit 100 arranged so as to constitute an effective video area of X: Y (for example, 9:16), and a drive that is an example of a panel control unit that generates various pulse signals for driving and controlling the display panel unit 100 A signal generation unit 200 and a video signal processing unit 300 are provided. The drive signal generation unit 200 and the video signal processing unit 300 are built in a one-chip IC (Integrated Circuit).

  As shown in the figure, the product form is provided as an organic EL display device 1 in the form of a module (composite part) including all of the display panel unit 100, the drive signal generation unit 200, and the video signal processing unit 300. For example, the organic EL display device 1 can be provided only by the display panel unit 100. Such an organic EL display device 1 is used in a display unit of a portable music player or other electronic device using a recording medium such as a semiconductor memory, a mini disk (MD), or a cassette tape.

  The display panel unit 100 includes a pixel array unit 102 in which pixel circuits P are arranged in a matrix of n rows × m columns on a substrate 101, a vertical drive unit 103 that scans the pixel circuits P in the vertical direction, and pixels A horizontal driving unit (also referred to as a horizontal selector or a data line driving unit) 106 that scans the circuit P in the horizontal direction, a terminal unit (pad unit) 108 for external connection, and the like are integrated. That is, peripheral drive circuits such as the vertical drive unit 103 and the horizontal drive unit 106 are formed on the same substrate 101 as the pixel array unit 102.

  As the vertical driving unit 103, for example, a writing scanning unit (write scanner WS; Write Scan) 104, a driving scanning unit (drive scanner DS; Drive Scan) 105 (both are shown integrally in the figure), a threshold value, and the like. & Has a mobility correction scanning unit 115.

  For example, the pixel array unit 102 is driven by the writing scanning unit 104, the driving scanning unit 105, and the threshold value & mobility correction scanning unit 115 from one side or both sides in the horizontal direction shown in the figure, and one side in the vertical direction shown in the figure. Or it is driven by the horizontal drive unit 106 from both sides.

  Various pulse signals are supplied to the terminal unit 108 from the drive signal generation unit 200 arranged outside the organic EL display device 1. Similarly, the video signal Vsig is supplied from the video signal processing unit 300.

  As an example, necessary pulse signals such as shift start pulses SPDS, SPWS and vertical scanning clocks CKDS, CKWS, which are examples of vertical write start pulses, are supplied as pulse signals for vertical driving. In addition, necessary pulse signals such as a shift start pulse SPAZ and a vertical scanning clock CKAZ, which are examples of a threshold detection start pulse in the vertical direction, are supplied as pulse signals for correcting the threshold and mobility. In addition, necessary pulse signals such as a horizontal start pulse SPH and a horizontal scanning clock CKH, which are examples of horizontal write start pulses, are supplied as pulse signals for horizontal driving.

  Each terminal of the terminal unit 108 is connected to the vertical driving unit 103 and the horizontal driving unit 106 via a wiring 109. For example, each pulse supplied to the terminal unit 108 is internally adjusted to a voltage level by a level shifter unit (not shown) as necessary, and then supplied to each unit of the vertical driving unit 103 and the horizontal driving unit 106 via a buffer. Supplied.

  Although the pixel array unit 102 is not shown in the drawing (details will be described later), pixel circuits P in which pixel transistors are provided with respect to an organic EL element as a display element are two-dimensionally arranged in a matrix form. On the other hand, scanning lines are wired for each row, and signal lines are wired for each column.

  For example, the pixel array unit 102 includes scanning lines (gate lines) 104WS and 105DS, threshold value & mobility correction scanning lines 115AZ, and signal lines (data lines) 106HS. An organic EL element (not shown) and a thin film transistor (TFT) for driving the organic EL element are formed at the intersection of the two. A pixel circuit P is configured by a combination of an organic EL element and a thin film transistor.

  Specifically, for each pixel circuit P arranged in a matrix, the write scanning lines 104WS_1 to 104WS_n for n rows driven by the write scanning unit 104 with the write drive pulse WS and the drive scanning unit The driving scanning lines 105DS_1 to 105DS_n for n rows driven by the scanning driving pulse DS 105 and the threshold & mobility for n rows driven by the threshold & mobility correction scanning unit 115 by the threshold & mobility correction pulse AZ. Correction scanning lines 115AZ_1 to 115AZ_n are wired for each pixel row.

  The writing scanning unit 104 and the driving scanning unit 105 sequentially select the pixel circuits P via the scanning lines 105DS and 104WS based on the vertical driving system pulse signal supplied from the driving signal generation unit 200. The horizontal drive unit 106 writes an image signal to the selected pixel circuit P via the signal line 106HS based on the horizontal drive pulse signal supplied from the drive signal generation unit 200.

  Each unit of the vertical driving unit 103 scans the pixel array unit 102 line-sequentially, and in synchronization with this, the horizontal driving unit 106 performs line-sequential driving for simultaneously writing one horizontal line of the image signal to the pixel array unit 102. Do. In the case of corresponding to line sequential driving, the horizontal driving unit 106 is configured to include a driver circuit that turns on all the switches provided on the signal lines 106HS of all the columns, and is input from the video signal processing unit 300. In order to simultaneously write the pixel signals to be written to all the pixel circuits P for one line in the row selected by the vertical driving unit 103, all the switches provided on the signal lines 106HS in all the columns are turned on at the same time. Let

  Each unit of the vertical drive unit 103 is configured by a combination of logic gates (including latches), and selects each pixel circuit P of the pixel array unit 102 in units of rows. FIG. 1 shows a configuration in which the vertical drive unit 103 is disposed only on one side of the pixel array unit 102. However, a configuration in which the vertical drive unit 103 is disposed on both the left and right sides with the pixel array unit 102 interposed therebetween is employed. Is also possible. Similarly, FIG. 1 shows a configuration in which the horizontal drive unit 106 is disposed only on one side of the pixel array unit 102, but a configuration in which the horizontal drive unit 106 is disposed on both upper and lower sides with the pixel array unit 102 interposed therebetween is employed. It is also possible.

<Pixel circuit>
FIG. 2 is a diagram showing an example of the pixel circuit P of the present embodiment that constitutes the organic EL display device 1 shown in FIG. Note that a vertical driving unit 103 and a horizontal driving unit 106 provided on the periphery of the pixel circuit P on the substrate 101 of the display panel unit 100 are also shown. FIG. 3 is a diagram for explaining the operating points of the organic EL element and the driving transistor. FIG. 3A is a diagram for explaining the influence of variations in characteristics of organic EL elements and drive transistors on the drive current Ids.

  The pixel circuit P of the present embodiment is characterized in that a drive transistor is basically composed of an n-channel thin film field effect transistor. In addition, a circuit for suppressing fluctuations in the drive current Ids to the organic EL element due to deterioration over time of the organic EL element, that is, driving by correcting a change in current-voltage characteristics of the organic EL element which is an example of an electro-optical element It is characterized in that a drive signal stabilizing circuit (part 1) for realizing a threshold value correcting function and a mobility correcting function for maintaining the current Ids constant is provided. In addition, the present invention is characterized in that it includes a drive signal stabilizing circuit (part 2) that realizes a bootstrap function that keeps the drive current constant even when the current-voltage characteristic of the organic EL element changes with time.

  If all the switch transistors can be driven by n-channel transistors instead of p-channel transistors, a conventional amorphous silicon (a-Si) process can be used in the transistor fabrication. Thereby, the cost of the transistor substrate can be reduced, and the development of the pixel circuit P having such a configuration is expected.

  MOS transistors are used as the transistors including the drive transistor. In this case, for the drive transistor, the gate end is handled as the control input end, and either the source end or the drain end (here, the source end) is handled as the output end, and the other is the power supply end (here, the drain end). ).

  The pixel circuit P of the present embodiment includes a storage capacitor (also referred to as a pixel capacitor) 120, an n-channel driving transistor 121, and an active H driving pulse (scanning driving pulse DS) at a gate terminal G that is a control input terminal. The n-channel type light emission control transistor 122 supplied, the active H drive pulse (write drive pulse WS) is supplied to the gate terminal G which is the control input terminal, and the n-channel type sampling transistor 125 supplied with current flows. And an organic EL element 127 which is an example of an electro-optical element (light emitting element).

  The sampling transistor 125 is a switching transistor provided on the gate end G (control input terminal) side of the driving transistor 121, and the light emission control transistor 122 is also a switching transistor.

  In general, the organic EL element 127 is represented by a diode symbol because of its rectifying property. The organic EL element 127 has a parasitic capacitance (equivalent capacitance) Cel. In the figure, this parasitic capacitance Cel is shown in parallel with the organic EL element 127.

  Here, the pixel circuit P of this embodiment includes a light emission control transistor 122 on the drain terminal D side of the drive transistor 121 and connects the storage capacitor 120 between the gate and source of the drive transistor 121 to thereby form a bootstrap circuit. Is further provided with a switch transistor that constitutes a threshold value & mobility correction circuit.

  Since the organic EL element 127 is a current light emitting element, color gradation is obtained by controlling the amount of current flowing through the organic EL element 127. Therefore, the value of the current flowing through the organic EL element 127 is controlled by changing the voltage applied to the gate terminal G of the drive transistor 121. At this time, a bootstrap circuit and a threshold value & mobility correction circuit are provided so as not to be affected by a change with time of the organic EL element 127 and a variation in characteristics of the drive transistor 121. Therefore, the vertical driving unit 103 that drives the pixel circuit P includes a threshold value & mobility correction scanning unit 115 in addition to the writing scanning unit 104 and the driving scanning unit 105.

  Although only one pixel circuit P is shown in the figure, pixel circuits P having the same configuration are arranged in a matrix as described with reference to FIG. The pixel circuits P arranged in a matrix are scanned by the write scanning lines 104WS_1 to 104WS_n for n rows driven by the write scanning pulse 104 by the write scanning unit 104 and the drive scanning unit 105. In addition to the driving scan lines 105DS_1 to 105DS_n for n rows driven by the driving pulse DS, the threshold & mobility correction for n rows driven by the threshold & mobility correction scanning unit 115 by the threshold & mobility correction scanning unit 115. Scan lines 115AZ_1 to 115AZ_n are wired for each pixel row.

  The bootstrap circuit includes an n-channel detection transistor 124 connected in parallel to the organic EL element 127 and supplied with an active-H threshold value and mobility correction pulse AZ. The storage capacitor 120 is connected between the sources. The storage capacitor 120 functions also as a bootstrap capacitor.

  The threshold & mobility correction circuit includes an n-channel detection transistor 124 to which an active H threshold & mobility correction pulse AZ is supplied between the gate terminal G of the drive transistor 121 and the second power supply potential Vc2. The transistor 124, the drive transistor 121, the light emission control transistor 122, and the storage capacitor 120 connected between the gate and source of the drive transistor 121 are configured. The holding capacitor 120 functions as a threshold voltage holding capacitor that holds the detected threshold voltage Vth.

  In the drive transistor 121, first, the drain end D is connected to the source end S of the light emission control transistor 122. The drain terminal D of the light emission control transistor 122 is connected to the first power supply potential Vc1. The gate end G is supplied with the active H scanning drive pulse DS from the drive scanning unit 105 via the drive scanning line 105DS.

  Here, in the present embodiment, in consideration of low power consumption, when the gate-source voltage is Vgs_122, the threshold voltage is Vth_122, and the drain-source voltage is Vds_122, the light emission control transistor 122 is at least organic EL. The element 127 is operated in a linear region (Vgs_122−Vth_122> Vds_122) during the light emission period. For this reason, the drive scanning unit 105 reduces the amplitude (difference between the L level and the H level) of the scan drive pulse DS so that it does not saturate at least during the light emission period of the organic EL element 127 when the light emission control transistor 122 is turned on. Set.

  The source end S of the driving transistor 121 is directly connected to the anode end A of the organic EL element 127. The connection point is referred to as a node ND121. The cathode terminal K of the organic EL element 127 is connected to a ground wiring Vcath (GND) common to all pixels for supplying a reference potential so that the cathode potential Vcath is supplied.

  The sampling transistor 125 has a gate terminal G connected to the writing scanning line 104WS from the writing scanning unit 104, a drain terminal D connected to the video signal line 106HS, and a source terminal S connected to the gate terminal G of the driving transistor 121. Has been. The connection point is referred to as a node ND122. The gate terminal G of the sampling transistor 125 is supplied with an active H write drive pulse WS from the write scanning unit 104. The sampling transistor 125 may have a connection mode in which the source terminal S and the drain terminal D are reversed. The storage capacitor 120 has one terminal connected to the source terminal S of the driving transistor 121 and the other terminal connected to the gate terminal G of the driving transistor 121.

  The detection transistor 124 is a switching transistor, and the drain terminal D is connected to a node ND121 that is a connection point between the source terminal S of the driving transistor 121 and the anode terminal A of the organic EL element 127. The source terminal S has a reference potential. It is connected to a reference potential Vini (also referred to as a ground potential Vs1) as an example, and a gate terminal G as a control input terminal is connected to a threshold value & mobility correction scanning line 115AZ. The storage capacitor 120 is connected between the gate and source of the drive transistor 121 and the detection transistor 124 is turned on, whereby the potential of the source terminal S of the drive transistor 121 is connected to the reference potential Vini that is a fixed potential via the detection transistor 124. It is configured to do.

  The sampling transistor 125 operates when selected by the write scanning line 104WS, samples the pixel signal Vsig (the signal potential Vin thereof) from the signal line 106HS, and supplies the holding capacitor 120 via the node ND112 with a magnitude corresponding to the signal potential Vin. The voltage is maintained. The potential held in the holding capacitor 120 is ideally the same magnitude as the signal potential Vin, but actually becomes smaller.

  The drive transistor 121 corresponds to the drive potential (the gate-source voltage Vgs of the drive transistor 121 at that time) held in the holding capacitor 120 when the light emission control transistor 122 is turned on under the scan drive pulse DS. The organic EL element 127 is driven by current. The light emission control transistor 122 is turned on when the drive scanning line 105DS is selected, and supplies current to the drive transistor 121 from the first power supply potential Vc1.

  In this way, the drain terminal D side which is the power supply terminal of the drive transistor 121 is connected to the first power supply potential Vc1 via the light emission control transistor 122, and the ON period of the light emission control transistor 122 is controlled to thereby control the organic EL element 127. By adjusting the light emission period and the non-light emission period, it is possible to perform duty driving.

  The detection transistor 124 operates when an active H threshold value & mobility correction pulse AZ is supplied from the threshold value & mobility correction scanning unit 115 to the threshold value & mobility correction scanning line 115AZ, and each is in a selected state. A correction operation (in this case, an operation for correcting variations in threshold threshold voltage Vth and mobility μ) is performed. For example, prior to current driving of the organic EL element 127, the threshold voltage Vth of the drive transistor 121 is detected, and the detected potential is held in the holding capacitor 120 in order to cancel the influence in advance.

  Further, threshold correction is performed using an offset voltage Vofs (also referred to as a reference potential Vo) which is a constant potential (fixed potential) of the video signal Vsig in the video signal line 106HS and a reference potential Vini on the source terminal S side of the detection transistor 124. It is possible to perform a preparatory operation prior to the operation. In this preparatory operation, the potential of the control input terminal (gate terminal G) and the output terminal (source terminal S) of the driving transistor 121 is initially set so that the potential difference (gate-source voltage Vgs) between both terminals becomes equal to or higher than the threshold voltage Vth. It is to become. The offset voltage Vofs is used for an initialization operation prior to the threshold correction operation and also used for precharging the video signal line 106HS in advance.

  As a condition for guaranteeing normal operation of the pixel circuit P, the reference potential Vini is set lower than a level obtained by subtracting the threshold voltage Vth of the drive transistor 121 from the offset voltage Vofs of the video signal Vsig. That is, “Vini <Vofs−Vth”. In other words, “Vofs−Vini> Vth” is satisfied, and the reference potential Vini is set to a potential sufficiently lower than the offset voltage Vofs of the video signal Vsig in the video signal line 106HS.

  The level obtained by adding the threshold voltage VthEL of the organic EL element 127 to the potential Vcath at the cathode end K of the organic EL element 127 is set higher than the reference potential Vini. That is, “Vcath + VthEL> Vini”. This means that the organic EL element 127 is reverse-biased during the preparatory operation prior to the threshold correction operation. The cathode potential Vcath may be considered as 0 V (= ground potential), and may be “VthEL> Vini”.

  Further, the anode potential (source potential Vs of the drive transistor 121) in the threshold correction period is set higher than the level obtained by adding the threshold voltage VthEL of the organic EL element 127 to the potential Vcath of the cathode terminal K of the organic EL element 127. That is, “Vofs−Vth <Vcath + VthEL”. This means that the organic EL element 127 is reverse-biased even during the threshold correction period. The cathode potential Vcath may be considered as 0 V (= ground potential), and may be “Vofs−Vth <VthEL”.

  In the pixel circuit P of the comparative example having such a configuration, the sampling transistor 125 is turned on in response to the write drive pulse WS supplied from the write scanning line 104WS during a predetermined signal writing period (sampling period). The video signal Vsig supplied from the line 106HS is sampled in the storage capacitor 120. The storage capacitor 120 applies an input voltage (gate-source voltage Vgs) between the gate and source of the drive transistor 121 in accordance with the sampled video signal Vsig.

  The drive transistor 121 supplies an output current corresponding to the gate-source voltage Vgs to the organic EL element 127 as a drive current Ids during a predetermined light emission period. When driving the organic EL element 127, the first potential Vcc_H is supplied to the drain terminal D of the driving transistor 121, and the source terminal S is connected to the anode terminal A side of the organic EL element 127, so that the source follower circuit as a whole. Is supposed to form.

  The drive current Ids is dependent on the carrier mobility μ and the threshold voltage Vth in the channel region of the drive transistor 121. The organic EL element 127 emits light with a luminance corresponding to the video signal Vsig (particularly the signal potential Vin) by the drive current Ids supplied from the drive transistor 121. The pixel circuit P according to the present embodiment includes a correction unit including switching transistors (light emission control transistor 122 and detection transistor 124). In order to cancel the dependency of the drive current Ids on the carrier mobility μ, in advance The gate-source voltage Vgs held in the holding capacitor 120 at the beginning of the light emission period is corrected.

  More specifically, the correction means (switching transistors 122 and 124) has a signal writing period corresponding to the write drive pulse WS and the scan drive pulse DS supplied from the write scan line 104WS and the drive scan line 105DS. The drive current Ids is extracted from the drive transistor 121 while the video signal Vsig is sampled, and negatively fed back to the storage capacitor 120 to correct the gate-source voltage Vgs. Further, the correction means (switching transistors 122 and 124) detects and detects the threshold voltage Vth of the driving transistor 121 in advance prior to the signal writing period in order to cancel the dependence of the driving current Ids on the threshold voltage Vth. The threshold voltage Vth is added to the gate-source voltage Vgs.

  In particular, in the pixel circuit P of the present embodiment, the drive transistor 121 is an n-channel transistor and has a drain connected to the positive power supply side and a source connected to the organic EL element 127 side. In this case, the correction means described above extracts the drive current Ids from the drive transistor 121 at the beginning of the light emission period that overlaps the latter part of the signal writing period, and negatively feeds back to the storage capacitor 120 side. At this time, the correcting means causes the drive current Ids extracted from the source end S side of the drive transistor 121 at the beginning of the light emission period to flow into the parasitic capacitance Cel included in the organic EL element 127. Specifically, the organic EL element 127 is a diode-type light emitting element having an anode end A and a cathode end K. The anode end A side is connected to the source end S of the drive transistor 121, while the cathode end K side is grounded. (In this example, the cathode potential Vcath).

  With this configuration, the correction means (switching transistors 122 and 124) sets the anode and cathode of the organic EL element 127 in a reverse bias state in advance, and the drive current Ids extracted from the source terminal S side of the drive transistor 121 is When flowing into the organic EL element 127, the diode-type organic EL element 127 functions as a capacitive element.

  The correcting means can adjust the time width t for extracting the drive current Ids from the drive transistor 121 within the signal writing period, and thereby optimize the negative feedback amount of the drive current Ids with respect to the storage capacitor 120. Here, “optimizing the negative feedback amount” means that the mobility correction can be appropriately performed at any level in the range from the black level to the white level of the video signal potential. . The amount of negative feedback applied to the gate-source voltage Vgs depends on the drive current Ids extraction time. The longer the extraction time, the larger the negative feedback amount.

  For example, the mobility correction period t is automatically set to the video line signal potential by adding a slope to the rise of the voltage of the signal line 106HS which is the video line signal potential or the transition characteristic of the write drive pulse WS of the write scanning line 104WS. To optimize it. That is, the mobility correction period t can be determined by the phase difference between the write scanning line 104WS and the signal line 106HS, and can also be determined by the potential of the signal line 106HS. The mobility correction parameter ΔV is ΔV = Ids · Cel / t. As is clear from this equation, the mobility correction parameter ΔV increases as the drive current Ids, which is the drain-source current of the drive transistor 121, increases. Conversely, when the drive current Ids of the drive transistor 121 is small, the mobility correction parameter ΔV is small. Thus, the mobility correction parameter ΔV is determined according to the drive current Ids. At that time, the mobility correction period t is not necessarily constant, and conversely, it may be preferable to adjust the mobility correction period t according to the drive current Ids. For example, when the drive current Ids is large, the mobility correction period t is preferably set short, and conversely, when the drive current Ids is small, the mobility correction period t is preferably set long.

  Therefore, the rising of the video signal line potential (the potential of the signal line 106HS) or the transition characteristic of the write drive pulse WS of the write scan line 104WS is inclined so that the potential of the signal line 106HS is high (the drive current Ids is reduced). The correction period t is automatically adjusted so that the correction period t is shortened and the correction period t is lengthened when the potential of the signal line 106HS is low (when the drive current Ids is small). In this way, an appropriate correction period can be automatically set following the video signal potential (the signal potential Vin of the video signal Vsig), so that optimum mobility correction can be performed regardless of the brightness of the image and the design. Become.

  The pixel circuit P of the present embodiment shown in FIG. 2 has a display period (in other words, a 2TR driving configuration using one switching transistor (sampling transistor 125) for scanning the video signal Vsig in addition to the driving transistor 121. In this case, the light emission control transistor 122 is provided on the drain terminal D side of the drive transistor 121 in order to dynamically control the non-light emission time, and one switching transistor (sampling transistor) is used for scanning for correcting the threshold value and mobility. 124) is used. In addition, the deterioration of the organic EL element 127 over time and the characteristics of the drive transistor 121 vary depending on the on / off timing settings of the write drive pulse WS, the scan drive pulse DS, and the threshold & mobility correction pulse AZ that control each switching transistor. This is characterized in that the influence on the drive current Ids due to (for example, variations or fluctuations in threshold voltage or mobility) is prevented.

  The pixel circuit P of the present embodiment shown in FIG. 2 is characterized by the connection mode of the storage capacitor 120, and is a drive signal stabilizing circuit (part 2) as a circuit for preventing fluctuations in the drive current due to deterioration of the organic EL element 127 over time. Is configured as a bootstrap circuit. A feature is that it has a drive signal stabilization circuit (part 2) that realizes a bootstrap function that makes the drive current constant even when the current-voltage characteristic of the organic EL element changes with time (to prevent fluctuations in the drive current). It has. Specifically, in the pixel circuit P of this embodiment, the storage capacitor 120 is connected between the gate terminal G (node ND122) of the driving transistor 121 and the source terminal S, and the source terminal S of the driving transistor 121 is directly connected. The organic EL element 127 is connected to the anode end A.

<Basic operation>
First, as a comparative example for explaining the characteristics of the pixel circuit P of the present embodiment shown in FIG. 2, the light emission control transistor 122 and the detection transistor 124 are not provided, and the holding capacitor 120 has one terminal. An operation in the case where the other terminal is connected to the node ND122 and the other terminal is connected to the ground wiring Vcath (GND) common to all the pixels will be described. Hereinafter, such a pixel circuit P is referred to as a pixel circuit P of a comparative example.

  In the pixel circuit P of the comparative example, the potential of the source terminal S (source potential Vs) of the drive transistor 121 is determined by the operating point of the drive transistor 121 and the organic EL element 127, and the voltage value is the gate potential Vg of the drive transistor 121. Will have different values.

  In general, as shown in FIG. 3, the drive transistor 121 is driven in a saturation region. Therefore, the current flowing between the drain end and the source of the transistor operating in the saturation region is Ids, the mobility is μ, the channel width (gate width) is W, the channel length (gate length) is L, and the gate capacitance (per unit area). When the gate oxide film capacitance) is Cox and the threshold voltage of the transistor is Vth, the drive transistor 121 is a constant current source having a value represented by the following equation (1). “^” Indicates a power. As apparent from the equation (1), in the saturation region, the drain current Ids of the transistor is controlled by the gate-source voltage Vgs and operates as a constant current source.

<Iel-Vel characteristics and IV characteristics of light-emitting elements>
In the current-voltage (Iel-Vel) characteristics of a current-driven light-emitting element typified by the organic EL element shown in FIG. 3A (1), the curve indicated by the solid line indicates the characteristic in the initial state, and the curve indicated by the broken line indicates The characteristic after change with time is shown. In general, the IV characteristics of current-driven light-emitting elements such as organic EL elements deteriorate as time passes, as shown in the graph.

  For example, when the light emission current Iel flows through the organic EL element 127 which is an example of the light emitting element, the anode-cathode voltage Vel is uniquely determined. As shown in FIG. 3A (1), during the light emission period, the light emission current Iel determined by the drain-source current Ids (= drive current Ids) of the drive transistor 121 flows through the anode terminal A of the organic EL element 127. As a result, the anode-cathode voltage Vel increases.

  In the pixel circuit P of the comparative example, the anode-cathode voltage Vel with respect to the same light emission current Iel changes from Vel1 to Vel2 due to the change with time of the IV characteristic of the organic EL element 127. Changes, and even if the same gate potential Vg is applied, the source potential Vs of the drive transistor 121 changes, and as a result, the gate-source voltage Vgs of the drive transistor 121 changes. In a simple circuit using an n-channel type as the drive transistor 121, the source end S is connected to the organic EL element 127 side, and therefore, it is affected by the change in the IV characteristics of the organic EL element 127 over time. The amount of current flowing through the organic EL element 127 (light emission current Iel) changes, and as a result, the light emission luminance changes.

  Specifically, in the pixel circuit P of the comparative example, the operating point changes due to the time-dependent change in the IV characteristics of the organic EL element 127, and the source potential Vs of the drive transistor 121 is applied even when the same gate potential Vg is applied. Will change. As a result, the gate-source voltage Vgs of the drive transistor 121 changes. As is apparent from the characteristic equation (1), when the gate-source voltage Vgs varies, the drive current Ids varies even if the gate potential Vg is constant, and the current value flowing through the organic EL element 127 also varies. . Thus, when the IV characteristic of the organic EL element 127 changes, the light emission luminance of the organic EL element 127 changes with time in the pixel circuit P of the comparative example.

  In a simple circuit using an n-channel type as the driving transistor 121, the source terminal S is connected to the organic EL element 127 side, so that the gate-source voltage Vgs changes as the organic EL element 127 changes over time. As a result, the amount of current flowing through the organic EL element 127 changes, and as a result, the light emission luminance changes. A variation in the anode potential of the organic EL element 127 due to a change in characteristics of the organic EL element 127, which is an example of the light emitting element, appears as a variation in the gate-source voltage Vgs of the driving transistor 121, and the drain current (driving current Ids). Cause fluctuations. Variations in the drive current due to this cause appear as variations in light emission luminance for each pixel circuit P, resulting in degradation of image quality.

  In contrast, as will be described in detail later, the sampling transistor 125 is turned off when information corresponding to the signal potential Vin is written in the storage capacitor 120 (and the light emission period of the organic EL element 127 thereafter is continued). In this state, a bootstrap operation is performed with a circuit configuration and a drive timing for realizing a bootstrap function in which the potential Vg of the gate terminal G is interlocked with a change in the source potential Vs of the drive transistor 121. As a result, even if there is an anode potential fluctuation (that is, source potential fluctuation) of the organic EL element 127 due to a change in characteristics of the organic EL element 127 with time, the screen luminance is changed by changing the gate potential Vg so as to cancel the fluctuation. Uniformity (uniformity) can be secured. The bootstrap function can improve the temporal variation correction capability of a current-driven light-emitting element typified by an organic EL element.

  This bootstrap function can be started at the light emission start time when the write drive pulse WS is switched to inactive L and the sampling transistor 125 is turned off, and then the light emission current Iel starts to flow through the organic EL element 127. At the same time, in the process in which the anode-cathode voltage Vel rises until it becomes stable, it also functions when the source potential Vs of the drive transistor 121 varies as the anode-cathode voltage Vel varies.

<Vgs-Ids characteristics of drive transistor>
In addition, due to variations in the manufacturing process of the drive transistor 121, there are variations in characteristics such as threshold voltage and mobility for each pixel circuit P. Even when the driving transistor 121 is driven in the saturation region, even if the same gate potential is applied to the driving transistor 121 due to this characteristic variation, the drain current (driving current Ids) varies for each pixel circuit P, and the emission luminance is reduced. Appears as variations.

  For example, FIG. 3A (2) is a diagram illustrating voltage-current (Vgs-Ids) characteristics focusing on threshold variation of the drive transistor 121. A characteristic curve is given for each of the two drive transistors 121 having different threshold voltages of Vth1 and Vth2.

  As described above, the drain current Ids when the driving transistor 121 operates in the saturation region is expressed by the characteristic formula (1). As apparent from the characteristic equation (1), when the threshold voltage Vth varies, the drain current Ids varies even if the gate-source voltage Vgs is constant. That is, if no countermeasure is taken against the variation of the threshold voltage Vth, the drive current corresponding to Vgs becomes Ids1 when the threshold voltage is Vth1, as shown in FIG. The drive current Ids2 corresponding to the same gate voltage Vgs when is Vth2 is different from Ids1.

  FIG. 3A (3) is a diagram showing voltage-current (Vgs-Ids) characteristics focusing on the mobility variation of the drive transistor 121. Characteristic curves are given for two drive transistors 121 having different mobility in μ1 and μ2.

  As apparent from the characteristic equation (1), when the mobility μ varies, the drain current Ids varies even when the gate-source voltage Vgs is constant. That is, if no countermeasure is taken against the variation in mobility μ, the drive current corresponding to Vgs becomes Ids1 when the mobility is μ1, as shown in FIG. When I is μ2, the drive current corresponding to the same gate voltage Vgs becomes Ids2, which is different from Ids1.

  As shown in FIGS. 3A (2) and 3A (3), if a large difference occurs in Vin-Ids characteristics due to a difference in threshold voltage Vth or mobility μ, even if the same signal potential Vin is applied, the drive current Ids, that is, the light emission luminance differs, and the uniformity of screen luminance cannot be obtained. On the other hand, by setting the drive timing (details will be described later) to realize the threshold value correction function and the mobility correction function, the influence of these fluctuations can be suppressed, and the uniformity of the screen luminance (uniformity) can be secured. .

  Although details will be described later in the threshold value correcting operation and the mobility correcting operation of the present embodiment, the drain-source current can be expressed by expressing the gate-source voltage Vgs at the time of light emission as “Vin + Vth−ΔV”. Ids is not dependent on variations or fluctuations in the threshold voltage Vth, and is not dependent on variations or fluctuations in the mobility μ. As a result, even if the threshold voltage Vth and the mobility μ fluctuate due to the manufacturing process and time, the driving current Ids does not fluctuate and the light emission luminance of the organic EL element 127 does not fluctuate.

<Operation of Pixel Circuit of this Embodiment>
First, the driving timing for the pixel circuit P of the present embodiment will be described from a qualitative viewpoint. As the drive timing in the pixel circuit P of the present embodiment, first, the sampling transistor 125 is turned on in accordance with the write drive pulse WS supplied from the write scan line 104WS, and the video signal supplied from the video signal line 106HS. Vsig is sampled and information corresponding to the signal potential Vin, which is the potential of the video signal Vsig during the effective period, is held in the holding capacitor 120 as a driving potential. This is the same as the case of driving a general pixel circuit.

  The drive transistor 121 is supplied with current from the power supply potential Vc1 and is driven according to the drive potential held in the holding capacitor 120 (potential corresponding to the potential of the video signal Vsig during the effective period: corresponding to the signal potential Vin). Ids is passed through the organic EL element 127.

  The vertical drive unit 103 activates the write drive pulse WS as a control signal for making the sampling transistor 125 conductive in a time zone in which the video signal line 106HS is at the offset voltage Vofs (reference potential Vo) which is the ineffective period of the video signal Vsig. Thus, a voltage corresponding to the threshold voltage Vth of the driving transistor 121 is held in the holding capacitor 120. This operation realizes a threshold correction function. By this threshold value correction function, it is possible to cancel the influence of the threshold voltage Vth of the drive transistor 121 that varies for each pixel circuit P.

  Preferably, the vertical driving unit 103 repeatedly executes the threshold correction operation in a plurality of horizontal periods preceding the sampling of the signal potential Vin in the video signal Vsig, and reliably supplies a voltage corresponding to the threshold voltage Vth of the driving transistor 121. It is held in the holding capacitor 120. In this way, a sufficiently long write time is ensured by executing the threshold correction operation a plurality of times. In this way, a voltage corresponding to the threshold voltage Vth of the drive transistor 121 can be reliably held in advance in the storage capacitor 120.

  The voltage corresponding to the held threshold voltage Vth is used to cancel the threshold voltage Vth of the drive transistor 121. Therefore, even if the threshold voltage Vth of the drive transistor 121 varies for each pixel circuit P, it is completely canceled for each pixel circuit P. Therefore, the uniformity of the image, that is, the uniformity of the light emission luminance over the entire screen of the display device is achieved. Rise. In particular, luminance unevenness that tends to appear when the signal potential is low gradation can be prevented.

  Preferably, the vertical drive unit 103 activates the threshold value & mobility correction pulse AZ (in this example, both in the present example) with the scanning drive pulse DS inactive (L level in this example) prior to the threshold value correction operation. H level) sets (initializes) the source potential Vs of the drive transistor 121 to the reference potential Vini, and activates the write drive pulse WS during the period in which the video signal Vsig is at the offset voltage Vofs (H in this example). Level), the gate potential Vg of the drive transistor 121 is set (initialized) to the offset voltage Vofs, and the voltage across the storage capacitor 120 connected between the gate and the source of the drive transistor 121 is equal to or higher than the threshold voltage Vth. The threshold value correction operation is started after setting to. By such a reset operation (initialization operation) of the gate potential and the source potential, it is possible to reliably execute the subsequent threshold value correction operation.

  In addition, the pixel circuit P of the present embodiment can be provided with a mobility correction function in addition to the threshold value correction function. For example, after the threshold value correcting operation, the vertical driving unit 103 causes the sampling transistor 125 to conduct in a time zone in which the signal potential Vin is supplied to the sampling transistor 125, thereby causing the storage capacitor 120 to receive information corresponding to the signal potential Vin ( After writing the drive potential), the scan drive pulse DS is set to active H while the signal potential Vin is supplied to the gate terminal G of the drive transistor 121, so that the correction for the mobility of the drive transistor 121 is written to the storage capacitor. In addition, the write drive pulse WS is controlled to be inactive L after that.

  The period from when the scanning drive pulse DS is set to active H to when the write drive pulse WS is set to inactive is the mobility correction period. By appropriately setting this period, the correction for the mobility μ of the drive transistor 121 is performed. The amount can be adjusted appropriately.

  In the pixel circuit P of the present embodiment, a bootstrap function is also provided by connecting the storage capacitor 120 between the gate and source of the drive transistor 121. That is, the writing scanning unit 104 cancels the application of the writing driving pulse WS to the writing scanning line 104WS at the stage where the driving potential corresponding to the signal potential Vin of the video signal Vsig is held in the holding capacitor 120 (that is, the IN The sampling transistor 125 is turned off, and the gate terminal G of the driving transistor 121 is electrically disconnected from the video signal line 106HS.

  A storage capacitor 120 is connected between the gate terminal G and the source terminal S of the driving transistor 121, and the gate potential Vg is interlocked with the variation of the source potential Vs of the driving transistor 121 due to the effect of the storage capacitor 120. Thus, the bootstrap function for maintaining the gate-source voltage Vgs constant can be activated.

<Timing chart; comparative example>
FIG. 4 is a timing chart for explaining the operation of the comparative example in the pixel circuit P of the present embodiment. FIG. 4 shows waveforms of the write drive pulse WS, the threshold & mobility correction pulse AZ, and the scan drive pulse DS along the time axis t. As understood from the above description, since the switching transistors 122, 124, and 125 are n-channel type, they are turned on when the pulses DS, WS, and AZ are at the high (H) level, respectively, and when the pulses are at the low (L) level. Turn off. This timing chart also shows the video signal Vsig, the potential change at the gate end G of the drive transistor 121, and the potential change at the source end S, along with the waveforms of the pulses WS, AZ, DS.

  In the explanation and figures, DS (scanning drive pulse DS) and AZ (threshold & mobility correction pulse AZ) are used to distinguish each drive pulse as necessary, such as when different drive pulses exist at the same timing. ), WS (when the write drive pulse is WS), and V (when the video signal is Vsig).

  At the drive timing of the comparative example, the period in which the video signal Vsig is in the ineffective period (signal fixed period) in the offset voltage Vofs (same in all horizontal periods) is the first half of one horizontal period, and the signal potential Vin in the effective period is A period (different for each horizontal period) is defined as the second half of one horizontal period. That is, the video signal Vsig is a pulse that takes a binary value of the offset voltage Vofs and the signal potential Vin in a 1H cycle.

  Then, with the write drive pulse WS set to active H and the sampling transistor 125 turned on, the scan drive pulse DS is set during the offset voltage Vofs in accordance with the video signal Vsig repeated with the offset voltage Vofs and the signal potential Vin. The threshold voltage Vth information is written in the storage capacitor 120 by turning on the light emission control transistor 122 by setting the signal to active H. That is, the threshold correction period is defined by the on period of the light emission control transistor 122 (specifically, the period during which the light emission control transistor 122 is on within the period in which the sampling transistor 125 is on). This is because the period during which the scanning drive pulse DS is active H (the light emission control transistor 122 is on) is dominant (priority).

  In the light emission period B of the previous field before the timing t10, the threshold & mobility correction pulse AZ and the write drive pulse WS are inactive L, and the detection transistor 124 and the sampling transistor 125 are in a non-conductive state, while the scan drive pulse DS Is active H and the light emission control transistor 122 is turned on, whereby the power supply potential Vc1 is supplied to the drain terminal D of the drive transistor 121.

  Therefore, regardless of the potential of the video signal line 106HS, the organic EL element 127 has a voltage state (the gate-source voltage Vgs of the driving transistor 121 = the driving potential) held in the holding capacitor 120 by the operation in the previous field. When the drive current Ids is supplied from the drive transistor 121 and flows into the ground wiring Vcath (GND) common to all the pixels, the organic EL element 127 is in a light emitting state.

  Thereafter, a new field of line-sequential scanning is entered, and the organic EL element is switched by switching the scanning drive pulse DS to inactive L while the threshold & mobility correction pulse AZ and the writing drive pulse WS are in inactive L. The element 127 stops emitting light and enters a non-emission period (t10). At this time, since the write drive pulse WS is inactive L and the sampling transistor 125 is off, the gate terminal G of the drive transistor 121 is high impedance, and the holding capacitor 120 is connected between the gate and the source. Therefore, the source potential Vs and the gate potential Vg are lowered in association with each other so as to maintain the previous gate-source voltage Vgs.

  In the non-light emitting period, a threshold correction preparation operation for initializing the gate potential Vg and the source potential Vs of the drive transistor 121 is performed prior to threshold correction. For example, first, while the scanning drive pulse DS and the write drive pulse WS remain in the inactive L state, the threshold & mobility correction pulse AZ is switched to the active H, the detection transistor 124 is turned on, and the source of the light emission control transistor 122 The potential Vs is initialized to the reference potential Vini (t11 to t12). At this time, since the sampling transistor 125 is turned off, the gate terminal G of the driving transistor 121 has a high impedance, and the source potential Vs is held so that the previous gate-source voltage Vgs is held by the effect of the holding capacitor 120. Following this decrease, the gate potential Vg also decreases.

  Thereafter, the vertical drive unit 103 temporarily inactivates the threshold value & mobility correction pulse AZ by the threshold value & mobility correction scan unit 115 while the scan drive pulse DS and the write drive pulse WS remain in the inactive L state. It returns to L (t12), and is again set to active H (t14). This is because the sampling transistor 125 is turned off immediately before the threshold correction so that the video signal Vsig immediately before the threshold correction is not transmitted to the gate of the driving transistor 121 and is not written in the storage capacitor 120.

  Further, the scanning drive pulse DS is inactive L and the threshold value & mobility correction pulse AZ is in active H state, and written in accordance with the timing (t16V to t18V) when the video signal Vsig is at the offset voltage Vofs. The embedded drive pulse WS is switched to active H, the sampling transistor 125 is turned on, and the gate potential Vg of the drive transistor 121 is initialized to the offset voltage Vofs (t16WS to t18WS).

  As described above, since the offset voltage Vofs and the reference potential Vini are set so as to satisfy “Vofs−Vini> Vth”, the gate-source voltage Vgs of the driving transistor 121, that is, the gate-source voltage of the driving transistor 121. The voltage held in the holding capacitor 120 connected to is set to a voltage exceeding the threshold voltage Vth of the driving transistor 121, and the holding capacitor 120 is reset prior to the threshold correction operation. Further, since “VthEL> Vini” is set, a reverse bias is applied to the organic EL element 127 so that the subsequent threshold value correcting operation is normally performed.

  After completion of the threshold correction preparation operation, the threshold & mobility correction pulse AZ is switched to inactive L to turn off the detection transistor 124 (t20AZ), and the write drive pulse WS is switched to active H to sample the transistor 125. Is turned on (t20AZ). While the write drive pulse WS remains active H, the scan drive pulse DS is switched to active H in accordance with the timing (t22V to t24V) when the video signal Vsig is at the offset voltage Vofs in the first half of one horizontal period. The control transistor 122 is turned on (t22DS). As a result, the drain current is used to charge and discharge the storage capacitor 120 and the organic EL element 127, and information on the threshold voltage Vth of the drive transistor 121 is recorded in the storage capacitor 120.

  Thereafter, in the second half of one horizontal period, before the video signal Vsig becomes the signal potential Vin, the scanning drive pulse DS is switched to inactive L to turn off the light emission control transistor 122, and between the drain and source of the drive transistor 121. The current is prevented from flowing (t24DS), and the potential of the video signal line 106HS is switched from the offset voltage Vofs to the signal potential Vin (t24V). As a result, the gate potential Vg of the drive transistor 121 changes from the offset voltage Vofs to the signal potential Vin. On the other hand, the source potential Vs of the drive transistor 121 maintains the previous potential state because the light emission control transistor 122 is off and the drive current Ids does not flow in principle, but actually the gate potential of the scan drive pulse DS. Under the influence of coupling, current flows between the gate and the source and rises slightly. This point will be described in detail later.

  A voltage corresponding to the threshold voltage Vth is written in the storage capacitor 120 connected between the gate terminal G and the source terminal S of the drive transistor 121. When the threshold correction period is not sufficiently secured, The threshold correction operation ends before the threshold voltage Vth information is sufficiently written, and the correction operation becomes insufficient in the subsequent light emission period.

  In order to avoid this problem, in the comparative example, the threshold voltage correction operation is executed a plurality of times (three times in this example), so that the threshold voltage Vth information is sufficiently written in the storage capacitor 120. In each threshold correction period (t22DS to t24DS), in the drive transistor 121, the source potential Vs rises while the gate potential Vg is suppressed to the offset voltage Vofs, and the gate-source voltage Vgs becomes just the threshold voltage Vth. At that point, the drain current is cut off. When cut off, the source potential Vs of the drive transistor 121 becomes “Vofs−Vth”, and information on the gate-source voltage Vgs (= Vth) is recorded in the storage capacitor 120.

  After the threshold voltage Vth information is written in the storage capacitor 120 and the drive transistor 121 is cut off, the signal potential Vin of the video signal Vsig is supplied to the signal line 106HS (t24V3), so that the storage capacitor 120 corresponds to the signal potential Vin. Write the information you want. Since the signal potential Vin is written to the storage capacitor 120 after the threshold correction operation is completely completed, the signal potential Vin is stored in the storage capacitor 120 in a form that is added to the threshold voltage Vth of the drive transistor 121.

As a result, fluctuations in the threshold voltage Vth of the drive transistor 121 are always canceled, and threshold correction is performed. By this threshold correction, the gate-source voltage Vgs held in the holding capacitor 120 becomes “Vsig + Vth” = “Vin + Vth”.
As described above, when the gate potential Vg of the drive transistor 121 changes from the offset voltage Vofs to the signal potential Vin, it is slightly increased due to the influence of current flowing between the gate and the source.

  Next, in a state where the potential of the video signal line 106HS is at the signal potential Vin, the writing drive pulse WS is switched to inactive L, and the sampling transistor 125 is turned off (t26). Thereafter, the scanning drive pulse DS is switched to active H, and the light emission period starts (t28). Thereafter, the process proceeds to the next frame (or field), and the threshold correction preparation operation, the threshold correction operation, and the light emission operation are repeated again.

  In the light emission period, the drive current Ids flowing through the drive transistor 121 flows through the organic EL element 127, and the anode potential of the organic EL element 127 rises according to the drive current Ids. Let this increase be Vel. Eventually, as the source potential Vs rises, the reverse bias state of the organic EL element 127 is canceled, so that the organic EL element 127 actually starts to emit light by the inflow of the drive current Ids. The rise (Vel) of the anode potential of the organic EL element 127 at this time is nothing but the rise of the source potential Vs of the drive transistor 121, and the source potential Vs of the drive transistor 121 becomes “Vofs−Vth + Vel”.

  A storage capacitor 120 is connected between the gate terminal G and the source terminal S of the drive transistor 121, and a bootstrap operation is performed by the effect of the storage capacitor 120, and the gate-source voltage “Vgs” of the drive transistor 121. = Vin + Vth "is maintained constant, and the gate potential Vg and the source potential Vs of the drive transistor 121 rise. When the source potential Vs of the drive transistor 121 becomes “Vofs−Vth + Vel”, the gate potential Vg becomes “Vin + Vel”.

  The relationship between the drive current Ids and the gate voltage Vgs can be expressed as in Expression (2) by substituting “Vin + Vth” for Vgs in Expression (1) representing the previous transistor characteristics. In formula (2), k = (1/2) (W / L) Cox. From this equation (2), it can be seen that the term of the threshold voltage Vth is canceled and the drive current Ids supplied to the organic EL element 127 does not depend on the threshold voltage Vth of the drive transistor 121. Basically, the drive current Ids is determined by the signal potential Vin of the video signal Vsig. In other words, the organic EL element 127 emits light with a luminance corresponding to the signal potential Vin.

<Adverse effects of threshold correction operation>
5 to 7 are diagrams for explaining the adverse effects of the threshold correction operation at the drive timing of the comparative example shown in FIG. Here, FIG. 5 is a diagram for explaining the state of waveform dullness of the scanning drive pulse DS when the drive timing of the comparative example is used. FIG. 6 is a circuit diagram in the vicinity of the drive transistor 121 and the light emission control transistor 122 for explaining the coupling noise when the light emission control transistor 122 is OFF when the drive timing of the comparative example is used. FIG. 7 is a timing chart showing the relationship between the scanning drive pulse DS and the gate potential Vg (= video signal Vsig), source potential Vs, and drain potential Vd of the drive transistor 121 in FIG.

  In the pixel circuit P of the present embodiment, the number of circuit elements is reduced by adopting the 4TR configuration, thereby reducing the number of transistors required for threshold correction and mobility correction by one than the 5TR configuration.

  In performing threshold correction by adopting the 4TR configuration, a period (signal fixed period) of the offset voltage Vofs of the pulsed video signal Vsig taking the binary values of the offset voltage Vofs and the signal potential Vin within 1H period is used. Threshold correction operation is performed. In particular, at the drive timing of the comparative example, information on the threshold voltage Vth is written in the storage capacitor 120 by turning on the light emission control transistor 122 while the video signal Vsig is in the period of the offset voltage Vofs with the sampling transistor 125 turned on. I have to.

  Consider the horizontal direction of the screen on which the drive scanning unit 105, which is an example of a vertical scanning system scanning unit, is arranged. For example, as schematically shown in FIG. 5, the write scanning unit 104 is arranged on the left and right and the write drive pulse WS is provided only from the left end, and the threshold & mobility correction scanning unit 115 is arranged on the right and only the right end and threshold & move. The degree correction pulse AZ is supplied to the left and right sides of the drive scanning unit 105, and the scanning drive pulse DS is supplied to the pixel circuit P in the pixel array unit 102 from both the left end and the right end.

  When focusing on the scan drive pulse DS, the scan drive pulse DS is commonly supplied from the drive scanning unit 105 to all the pixel circuits P in one row. Due to the influence, the waveform of the pixel circuit P farther from the drive scanning unit 105 (referred to as a far-side pixel) becomes duller than the pixel circuit P closer to the drive scan unit 105 (referred to as a near-side pixel).

  In the example shown in FIG. 5, since the scanning drive pulse DS is supplied from both the left and right sides of the pixel array unit 102, the waveform dullness of the scanning drive pulse DS becomes larger near the center of the panel than at the panel end. At the drive timing of the comparative example in which the information on the threshold voltage Vth is written in the holding capacitor 120 by turning on the light emission control transistor 122 while the video signal Vsig is in the period of the offset voltage Vofs with the sampling transistor 125 turned on, the scanning drive pulse The effect of DS waveform dullness appears in the image as shading.

  With reference to FIG. 6 and FIG. 7, the mechanism of this shading occurrence will be described. As shown in FIG. 6, when attention is paid to the drain terminal D of the driving transistor 121, a parasitic capacitance Cgd1 exists between the gate and the drain, and the drain terminal D of the driving transistor 121 and the gate terminal G of the light emission control transistor 122 It is assumed that a parasitic capacitance Cgd2 exists between them.

  First, in consideration of low power consumption, regarding the light emission control transistor 122, when the gate-source voltage is Vgs_122, the threshold voltage is Vth_122, and the drain-source voltage is Vds_122, in the linear region (Vgs_122-Vth_122> Vds_122). Make it work. In order to operate the light emission control transistor 122 in the linear region, the drive scanning unit 105 sets the amplitude (difference between the L level and the H level) of the scan drive pulse DS to be small so as not to be saturated when the light emission control transistor 122 is turned on. To do.

  Consider the operating points of the gate potential Vg, source potential Vs, and drain potential Vd of the drive transistor 121 at the end of the threshold correction operation, that is, at the timing when the light emission control transistor 122 is turned off. Further, as shown in FIG. 5, the waveform of the scanning drive pulse DS is dull at the center of the panel with respect to the panel edge. In the scanning drive pulse DS in FIG. 7, the solid line shows the waveform at the panel end, and the broken line shows the waveform at the panel center.

  A component in which the scanning drive pulse DS supplied to the gate end of the light emission control transistor 122 is supplied to the drain end D of the drive transistor 121 via the parasitic capacitance Cgd2 is referred to as gate coupling to the drain potential Vd.

  Consider the timing (t24DS) when the light emission control transistor 122 is turned off at the end of the threshold correction operation. The light emission control transistor 122 is cut off from the point in time when the gate potential of the light emission control transistor 122 cuts “power supply potential Vc1 + Vth — 122”.

  In the scan drive pulse DS in FIG. 7, as indicated by a solid line, when the scan drive pulse DS at the gate end of the light emission control transistor 122 is steep, the scan drive pulse DS is inactive L at the output end of the drive scan unit 105. Since the gate potential cuts “power supply potential Vc1 + Vth_122” almost immediately, the gate coupling when the light emission control transistor 122 is turned off becomes the drain potential Vd at the drain potential Vd in FIG. 7 as shown by the solid line. Get in.

  On the other hand, if the waveform of the scan drive pulse DS at the gate end of the light emission control transistor 122 is dull, even if the scan drive pulse DS becomes inactive L at the output end of the drive scan unit 105, Since the timing at which the potential cuts off the power supply potential Vc1 + Vth_122 (cutoff timing) is delayed and gate coupling is applied after the cut-off, the amount of coupling is smaller than that of a pulse with a steep gate potential of the light emission control transistor 122. That is, in the drain potential Vd in FIG. 7, as indicated by a broken line, the drain potential Vd is higher than that in the pixel circuit P in which the scanning drive pulse DS at the gate end of the light emission control transistor 122 is steep.

  As shown by the gate potential Vg in FIG. 7, the signal potential Vin of the video signal Vsig is supplied to the gate terminal G of the drive transistor 121 after the light emission control transistor 122 is completely turned off. Current flows between the drain and source of the transistor. At this time, a larger current flows between the drain and the source in the center of the panel with more coupling than in the panel end with a sharp scan driving pulse DS at the gate end G of the light emission control transistor 122 and less coupling.

  For this reason, as indicated by the source potential Vs in FIG. 7, the gate-source voltage Vgs is reduced at the center of the panel indicated by a broken line with less coupling than at the panel end indicated by a solid line with much coupling. When the source potential Vs is increased (α) due to such gate coupling, the gate-source voltage Vgs written to the storage capacitor 120 during the signal writing period K becomes “Vin−Vth−α”. In the light emission period L, the luminance decreases by the increase α of the source potential Vs. If there is a difference in the increase α of the source potential Vs between the panel edge and the center portion, a luminance difference is generated with respect to the same signal potential Vin.

  The brightness is lower at the center of the panel, resulting in shading. That is, if the coupling is large, a minute current flows between the drain and source of the drive transistor 121 during the threshold correction pause period, the source potential Vs rises, and the gate-source voltage Vgs shrinks. A difference in the gate-source voltage Vgs occurs due to the coupling difference between the scanning drive pulses DS at the panel edge and the center in one threshold value correction pause period, resulting in shading.

  If the threshold value correcting operation is repeated a plurality of times, the operation shown in FIG. 7 is repeated, so that the potential difference between the source potential Vs at the panel edge and the center increases. This is for the following. If there is a difference between the panel edge and the center during the threshold correction pause, the process proceeds to the next threshold correction with the difference. The next threshold correction also changes depending on the difference between the gate-source voltage Vgs and the gate-source voltage Vgs does not match. Thereafter, a difference occurs in the gate-source voltage Vgs by the same mechanism during the pause, and the difference increases as the number of times increases. When the source potential Vs is high, the gate-source voltage Vgs becomes small and the drive current Ids that can be passed through the organic EL element 127 becomes small. Therefore, the brightness at the center of the panel where the scanning drive pulse DS is dull is darker than the panel end. And shading will occur.

  Therefore, in the present embodiment, the driving timing is set such that the information on the threshold voltage Vth is written in the storage capacitor 120 by turning on the light emission control transistor 122 while the video signal Vsig is in the offset voltage Vofs while the sampling transistor 125 is turned on. A mechanism that can improve the shading phenomenon caused by the blunting of the waveform of the scanning drive pulse DS that occurs in some cases. This will be specifically described below.

<Method for suppressing shading phenomenon associated with threshold correction; first example>
FIG. 8 is a timing chart for explaining the operation of the first example of the pixel circuit of the present embodiment. The timing chart of the first example is an application of the first example of the shading phenomenon suppression method associated with threshold correction. Similar to the comparative example, the waveforms of the write drive pulse WS, the threshold & mobility correction pulse AZ, and the scan drive pulse DS are represented along the time axis t. As understood from the above description, since the switching transistors 122, 124, and 125 are n-channel type, they are turned on when the pulses DS, WS, and AZ are at the high (H) level, respectively, and when the pulses are at the low (L) level. Turn off. This timing chart also shows the video signal Vsig, the potential change at the gate end G of the drive transistor 121, and the potential change at the source end S, along with the waveforms of the pulses WS, AZ, DS.

  In the explanation and figures, DS (scanning drive pulse DS) and AZ (threshold & mobility correction pulse AZ) are used to distinguish each drive pulse as necessary, such as when different drive pulses exist at the same timing. ), WS (when the write drive pulse is WS), and V (when the video signal is Vsig).

  In the suppression method of the first example, similarly to the driving timing of the comparative example, the light emission control is performed by setting the scanning drive pulse DS to the active level during the period in which the video signal Vsig is the offset voltage Vofs while the sampling transistor 125 is turned on. By turning on the transistor 122, information on the threshold voltage Vth is written in the storage capacitor 120, and the active level of the scanning drive pulse DS is set to a magnitude that allows the light emission control transistor 122 to operate in the saturation region during the threshold correction period. The light emission period is characterized in that the light emission control transistor 122 is set to a size for operating in a linear region. This will be specifically described below.

  The shading phenomenon caused by the waveform dullness of the scanning drive pulse DS accompanying the threshold correction is different in the amount of gate coupling when the light emission control transistor 122 that stops the threshold correction operation is turned off in the in-plane position of the pixel array unit 102. It is caused by that. The main cause is that the light emission control transistor 122 is operated in a linear region. That is, in the pixel in which the scanning drive pulse DS is dull at the gate end G of the light emission control transistor 122, the light emission control transistor 122 is operated in the linear region, so that the potential of the scan drive pulse DS is “power supply potential Vc1 + Vth — 122”. Since the cut-off occurs at the cut point, the gate coupling amount differs between the pixel having a sharp change in the scan driving pulse DS at the gate end of the light emission control transistor 122 and the pixel having a dull change. It appears as a shading phenomenon.

  Therefore, in the suppression method of the first example, the light emission control transistor 122 is used not in the linear region but in the saturation region. The operation region of the light emission control transistor 122 at the time of threshold correction is set as a saturation region. If the light emission control transistor 122 is used in the saturation region, the light emission control transistor 122 starts to be cut off from the moment when the light emission control transistor 122 is turned off. Even for pixels whose change is dull, the gate coupling amounts match. Thereby, the shading phenomenon does not occur. However, in order to operate the light emission control transistor 122 in the saturation region, the amplitude of the scanning drive pulse DS generated by the drive scanning unit 105 must be increased, and there is a concern that the power consumption may be adversely affected.

  On the other hand, in order to suppress the shading phenomenon caused by the waveform dullness of the scanning drive pulse DS accompanying the threshold correction, it is necessary to operate the light emission control transistor 122 in the saturation region, and at least the light emission of the organic EL element 127. It is considered that no inconvenience occurs even if the operation is performed in the linear region (Vgs_122−Vth_122> Vds_122) during the period.

  Therefore, in the suppression method of this first example, as shown in the timing chart of FIG. 8, the drive scanning unit 105 uses the scan drive pulse DS and the active level to operate the light emission control transistor 122 in the saturation region during the threshold correction period. (H level in this example) is set to be relatively large (referred to as Hh level for distinction), while in the light emission period of the organic EL element 127, the active level (in this example, the light emission control transistor 122 is operated in a linear region). (H level) is set relatively small (referred to as Hm level for distinction). In other words, the active level of the scanning drive pulse DS is set to the binary value of the Hh level and the Hm level, and as a whole, ternary driving is performed.

  In this case, the specification of the drive scanning unit 105 must be changed to be compatible with ternary drive instead of binary drive, which complicates the circuit specification. Even if it remains as it is, the advantage that the shading phenomenon accompanying the threshold correction can be suppressed without increasing the power consumption can be obtained.

<Method for suppressing shading phenomenon associated with threshold correction; second example>
FIG. 9 is a timing chart for explaining the operation of the second example of the pixel circuit of the present embodiment. The timing chart of the second example is an application of the second example of the shading phenomenon suppression method associated with threshold correction.

  FIG. 9 shows waveforms of the write drive pulse WS, the threshold & mobility correction pulse AZ, and the scan drive pulse DS along the time axis t. As understood from the above description, since the switching transistors 122, 124, and 125 are n-channel type, they are turned on when the pulses DS, WS, and AZ are at the high (H) level, respectively, and when the pulses are at the low (L) level. Turn off. This timing chart also shows the video signal Vsig, the potential change at the gate end G of the drive transistor 121, and the potential change at the source end S, along with the waveforms of the pulses WS, AZ, DS.

  Basically, the same driving is performed with a delay of one horizontal scanning period for each row of the write scanning line 104WS and the threshold & mobility correction scanning line 115AZ. Each timing and signal in the figure are indicated by the same timing and signal as the timing and signal of the first row regardless of the processing target row. Then, when it is necessary to distinguish between rows in the description, the processing target row is indicated by a reference with “_” in the timing and signal. Also, in the explanation and figures, when different drive pulses exist at the same timing, DS (scanning drive pulse DS), AZ (threshold & mobility correction pulse) to distinguish each drive pulse as necessary AZ), WS (when the write drive pulse is WS), and V (when the video signal is Vsig).

  As in the first example (also the comparative example), the drive timing to which the suppression method of the second example is applied is a period in which the video signal Vsig is in the ineffective period offset voltage Vofs (same in all horizontal periods). Let the first half of one horizontal period be the first half of one horizontal period, and let the period in the signal potential Vin (which varies from one horizontal period to another) be the effective period. That is, the video signal Vsig is a pulse that takes a binary value of the offset voltage Vofs and the signal potential Vin in a 1H cycle.

  Then, the suppression method of the second example is such that, with the scanning drive pulse DS set to active H and the light emission control transistor 122 turned on, the offset voltage is adjusted in accordance with the video signal Vsig repeated with the offset voltage Vofs and the signal potential Vin. It is characterized in that information on the threshold voltage Vth is written in the storage capacitor 120 by turning on the sampling transistor 125 by setting the write drive pulse WS to active H during the period of Vofs. That is, the threshold correction period is defined by the on period of the sampling transistor 125 (specifically, the period in which the sampling transistor 125 is on within the period in which the light emission control transistor 122 is on). Although similar to the first example (also the comparative example), the threshold correction period is different in that the period of active H (the sampling transistor 125 is on) of the write drive pulse WS is dominant (priority).

  For a certain row (here, the first row), in the light emission period B of the previous field before timing t10, the threshold & mobility correction pulse AZ and the write drive pulse WS are inactive L, and the detection transistor 124 and sampling While the transistor 125 is non-conductive, the scanning drive pulse DS is active H and the light emission control transistor 122 is turned on, so that the power supply potential Vc1 is supplied to the drain terminal D of the drive transistor 121.

  Therefore, regardless of the potential of the video signal line 106HS, the organic EL element 127 has a voltage state (the gate-source voltage Vgs of the driving transistor 121 = the driving potential) held in the holding capacitor 120 by the operation in the previous field. When the drive current Ids is supplied from the drive transistor 121 and flows into the ground wiring Vcath (GND) common to all the pixels, the organic EL element 127 is in a light emitting state.

  Thereafter, a new field of line sequential scanning is entered. First, the drive scanning unit 105 drives the drive scan line in the first row while the threshold value & mobility correction pulse AZ and the write drive pulse WS are inactive L. The scanning drive pulse DS applied to 105DS is switched from active H to inactive L (t30). As a result, the light emission control transistor 122 is turned off and the drive transistor 121 is disconnected from the power supply potential Vc1, so that the organic EL element 127 stops emitting light and enters a non-emission period. At the timing t30, the control transistors 122, 124, and 125 are turned off. At this time, since the write drive pulse WS is inactive L and the sampling transistor 125 is off, the gate terminal G of the drive transistor 121 is high impedance, and the holding capacitor 120 is connected between the gate and the source. Therefore, the source potential Vs and the gate potential Vg are lowered in association with each other so as to maintain the previous gate-source voltage Vgs.

  Next, the vertical drive unit 103 switches the threshold value & mobility correction pulse AZ to active H by the threshold value & mobility correction scan unit 115 while the scan drive pulse DS and the write drive pulse WS remain in the inactive L state. Then, the detection transistor 124 is turned on (t31). As a result, the reference potential Vini is set to the voltage of the node ND121, that is, the other end of the storage capacitor 120 and the source end S of the driving transistor 121, and the source potential Vs is initialized. A period (t31 to t42WS) until the threshold correction period starts is an initialization period C of the source potential Vs. At this time, since the write drive pulse WS is inactive L and the sampling transistor 125 is off, the gate terminal G of the drive transistor 121 is high impedance, and the holding capacitor 120 is connected between the gate and the source. Therefore, the gate potential Vg is lowered following the drop of the source potential Vs so as to maintain the previous gate-source voltage Vgs.

  Thereafter, the vertical drive unit 103 maintains the scan drive pulse DS and the write drive pulse WS in the inactive L state, and the threshold & mobility correction scan unit 115 outputs the threshold & mobility correction pulse AZ to the inactive L level. (T32). Further, the vertical drive unit 103 activates the write drive pulse WS by the write scan unit 104 in accordance with the timing (t34V to t38V) when the scan drive pulse DS is inactive L and the video signal Vsig is at the offset voltage Vofs. The sampling transistor 125 is turned on by switching to H (t34WS), and the threshold & mobility correction scanning unit 115 switches the threshold & mobility correction pulse AZ to active H and turns on the detection transistor 124 (t34AZ).

  Regarding the write drive pulse WS, the active H period (t34WS to t38WS) is completely included in the period (t34V to t38V) in which the video signal Vsig is at the offset voltage Vofs. On the other hand, the timing t34AZ at which the threshold & mobility correction pulse AZ becomes active H may be within the period (t34V to t38V) in which the video signal Vsig is at the offset voltage Vofs, and preferably in the vicinity immediately after t34V, It should be roughly the same. On the other hand, the timing at which the threshold value & mobility correction pulse AZ becomes inactive L may be any time point before the threshold value correction operation is started. For example, as shown in the figure, the video signal Vsig is a signal potential Vin. The timing t39AZ within a certain period (t38V to t42V) may be used.

  As a result, the offset voltage Vofs is set to the voltage of the node ND122, that is, the gate terminal G of the drive transistor 121, and the gate potential Vg is initialized. At this time, since the threshold & mobility correction pulse AZ is active H and the detection transistor 124 is turned on, the fluctuation when the gate potential Vg transitions to the offset voltage Vofs does not affect the source potential Vs. The period (t34WS to t42WS) until the threshold correction period is started is the initialization period D of the gate potential Vg.

  In this way, by initializing the gate potential Vg and the source potential Vs of the drive transistor 121, the preparation for the threshold correction operation is completed. The initialization period C of the source potential Vs and the initialization period D of the gate potential Vg are collectively referred to as a threshold correction preparation period for initializing the gate potential Vg and the source potential Vs of the drive transistor 121.

  As described above, since the offset voltage Vofs and the reference potential Vini are set so as to satisfy “Vofs−Vini> Vth”, the gate-source voltage Vgs of the driving transistor 121, that is, the gate-source voltage of the driving transistor 121. The voltage held in the holding capacitor 120 connected to is set to a voltage exceeding the threshold voltage Vth of the driving transistor 121, and the holding capacitor 120 is reset prior to the threshold correction operation. Further, since “VthEL> Vini” is set, a reverse bias is applied to the organic EL element 127 so that the subsequent threshold value correcting operation is normally performed.

  After the threshold correction preparation operation is completed, the vertical drive unit 103 sets the scan drive pulse DS to active H by the drive scanning unit 105 and turns on the light emission control transistor 122 (t42DS). Further, the write scanning unit 104 switches the write drive pulse WS to active H in accordance with the timing (t42V to t44V) when the video signal Vsig is at the offset voltage Vofs, and the sampling transistor 125 is turned on (t42WS).

  Note that the timing t42WS and the timing t42DS may be substantially the same or may be in the same order. This is because the threshold correction period is defined in the period in which the write drive pulse WS is active H while the scan drive pulse DS is in active H. Of course, in practice, the threshold correction period is defined by the period in which the light emission control transistor 122 and the sampling transistor 125 to which the pulses DS and WS are supplied are actually turned on.

  As a result, the drain current is used to charge and discharge the storage capacitor 120 and the organic EL element 127, and the threshold correction period E in which information for correcting (cancelling) the threshold voltage Vth of the drive transistor 121 is recorded in the storage capacitor 120. to go into. This threshold value correction period E continues until the timing (t44DS) when the scanning drive pulse DS is set to inactive L.

  Preferably, the threshold correction period E (t42WS, t42DS to t44DS: “t42WS, t42DS” means whichever is later, and so on) is within a time zone (t42V to t44V) in which the video signal Vsig is at the offset voltage Vofs. Is completely included. This is because when the light emission control transistor 122 is on and the video signal Vsig is at the signal potential Vin and the write drive pulse WS is active H, the sampling operation (signal potential of the signal potential Vin to the holding capacitor 120 is performed. This is because the gate potential Vg becomes equal to or higher than the offset voltage Vofs, resulting in inconvenience as the threshold correction operation. During the threshold correction operation, the gate potential Vg of the driving transistor 121 is fixed to the offset voltage Vofs, and a current corresponding to the gate-source voltage Vgs (initially Vofs−Vini after initialization) is applied to the organic EL element. 127 (parasitic capacitance Cel) increases the potential of the node ND121 (source potential Vs), and the gate potential Vg at the time of cut-off is held in the holding capacitor 120 is the threshold correction of the present embodiment. This is because it is a basic operation.

  In the threshold correction period E, the gate terminal G of the driving transistor 121 is held at the offset voltage Vofs of the video signal Vsig, and a drain current flows until the source potential Vs of the driving transistor 121 rises and the driving transistor 121 is cut off. Try to. When cut off, the source potential Vs of the drive transistor 121 becomes “Vofs−Vth”. That is, since the equivalent circuit of the organic EL element 127 is represented by a parallel circuit of a diode and a parasitic capacitance Cel, as long as “Vel ≦ Vcath + VthEL”, that is, the leakage current of the organic EL element 127 is greater than the current flowing through the drive transistor 121. Is considerably small, the current of the driving transistor 121 is used to charge and discharge the storage capacitor 120 and the parasitic capacitor Cel.

  As a result, when a drain current flows through the drive transistor 121, the voltage Vel at the anode end A of the organic EL element 127, that is, the potential of the node ND121 increases with time. Then, when the potential difference between the potential of the node ND121 (source potential Vs) and the voltage of the node ND122 (gate potential Vg) is just the threshold voltage Vth, the driving transistor 121 is turned off from the on state, and the drain current does not flow. The threshold correction period ends. That is, after a lapse of a certain time, the gate-source voltage Vgs of the drive transistor 121 takes the value of the threshold voltage Vth, and this information is held in the holding capacitor 120 connected between the gate and source.

  After the threshold voltage Vth information is written in the storage capacitor 120 and the driving transistor 121 is cut off, the writing scanning unit 104 is in a state where the scanning driving pulse DS is inactive L and the writing driving pulse WS is active H. The signal potential Vin of the video signal Vsig is supplied to the signal line 106HS (t44V to t47V). In this process, the write scanning unit 104 switches the write drive pulse WS to inactive L before the video signal Vsig switches to the offset voltage Vofs (t46).

  As a result, information corresponding to the signal potential Vin is written in the storage capacitor 120. Since the signal potential Vin is written to the storage capacitor 120 after the threshold correction operation is completely completed, the signal potential Vin is stored in the storage capacitor 120 in a form that is added to the threshold voltage Vth of the drive transistor 121. As a result, fluctuations in the threshold voltage Vth of the drive transistor 121 are always canceled, and threshold correction is performed. By this threshold correction, the gate-source voltage Vgs held in the holding capacitor 120 becomes “Vsig + Vth” = “Vin + Vth”.

  Next, the drive scanning unit 105 switches the scan drive pulse DS to active H (t48). As a result, the light emission control transistor 122 is turned on, so that the drive current Ids corresponding to the gate-source voltage Vgs (= Vin + Vth) at that time flows to the drive transistor 121 and proceeds to the light emission period L. Thereafter, the process proceeds to the next frame (or field), and the threshold correction preparation operation, the threshold correction operation, and the light emission operation are repeated again.

  Here, when the write drive pulse WS is set to inactive L, the gate terminal G of the drive transistor 121 is disconnected from the video signal line 106HS. Since the application of the video signal Vsig to the gate terminal G of the drive transistor 121 is released, the gate potential Vg of the drive transistor 121 can be changed in conjunction with the source potential Vs. That is, a bootstrap operation is possible.

  In the light emission period L, the drive current Ids flowing through the drive transistor 121 flows through the organic EL element 127, and the anode potential of the organic EL element 127 rises according to the drive current Ids. Let this increase be Vel. Eventually, as the source potential Vs rises, the reverse bias state of the organic EL element 127 is canceled, so that the organic EL element 127 actually starts to emit light by the inflow of the drive current Ids. The rise (Vel) of the anode potential of the organic EL element 127 at this time is nothing but the rise of the source potential Vs of the drive transistor 121, and the source potential Vs of the drive transistor 121 becomes “Vofs−Vth + Vel”.

  A storage capacitor 120 is connected between the gate terminal G and the source terminal S of the drive transistor 121, and a bootstrap operation is performed by the effect of the storage capacitor 120, and the gate-source voltage “Vgs” of the drive transistor 121. = Vin + Vth "is maintained constant, and the gate potential Vg and the source potential Vs of the drive transistor 121 rise. When the source potential Vs of the drive transistor 121 becomes “Vofs−Vth + Vel”, the gate potential Vg becomes “Vin + Vel”.

  The relationship between the drive current Ids and the gate voltage Vgs can be expressed as shown in Equation (2), the term of the threshold voltage Vth is canceled, and the drive current Ids supplied to the organic EL element 127 is the threshold voltage of the drive transistor 121. Since the drive current Ids is determined by the signal potential Vin of the video signal Vsig without depending on Vth, the organic EL element 127 emits light with a luminance corresponding to the signal potential Vin.

  Here, the organic EL element 127 has its IV characteristic changed as the light emission time becomes longer. Therefore, the potential of the node ND121 also changes with time. However, even if the anode potential fluctuates due to such deterioration of the organic EL element 127 with time, the gate-source voltage Vgs held in the holding capacitor 120 is always kept constant at “Vin + Vth”.

  Since the drive transistor 121 operates as a constant current source, the IV characteristic of the organic EL element 127 changes with time, and even if the source potential Vs of the drive transistor 121 changes accordingly, the drive transistor 121 drives the drive transistor 121. Since the gate-source potential Vgs 121 is kept constant (≈Vin + Vth), the current flowing through the organic EL element 127 does not change, and thus the light emission luminance of the organic EL element 127 is also kept constant.

  Regardless of the characteristic variation of the organic EL element 127, an operation for correction (operation based on the effect of the storage capacitor 120) for maintaining the gate-source voltage of the driving transistor 121 constant and maintaining the luminance constant is performed. This is called a bootstrap operation. By this bootstrap operation, even if the IV characteristic of the organic EL element 127 changes with time, it is possible to display an image without luminance deterioration associated therewith. Therefore, even if the IV characteristic of the organic EL element 127 deteriorates, the constant current Ids always flows, so that the organic EL element 127 continues to emit light with the luminance according to the pixel signal Vsig, and the luminance changes. Absent.

  In addition, a threshold correction circuit, which is an example of a drive signal stabilization circuit that corrects the threshold voltage Vth of the drive transistor 121 and maintains the drive current constant, is configured so that the threshold correction operation functions. As the gate-source potential Vgs reflecting the threshold voltage Vth of the drive transistor 121, a constant current Ids that is not affected by variations in the threshold voltage Vth can be passed.

  Since not only the bootstrap operation but also the threshold correction operation is performed, the gate-source voltage Vgs maintained in the bootstrap operation is adjusted by a voltage corresponding to the threshold voltage Vth. Is not affected by variations in the threshold voltage Vth of the drive transistor 121, and is not affected by deterioration over time of the organic EL element 127. A stable gradation corresponding to the input signal potential Vin can be displayed, and a high-quality image can be obtained.

  In addition, the pixel circuit P can be configured using only n-channel transistors including the driving transistor 121 and the sampling transistor 125 in the periphery thereof, and an amorphous silicon (a-Si) process is also used in TFT fabrication. Therefore, the cost of the TFT substrate can be reduced.

<Method of suppressing shading phenomenon associated with threshold correction; third example>
FIG. 10 is a timing chart for explaining the operation of the third example of the pixel circuit of this embodiment. FIG. 10 shows waveforms of the write drive pulse WS, the threshold & mobility correction pulse AZ, and the scan drive pulse DS along the time axis t. As understood from the above description, since the switching transistors 122, 124, and 125 are n-channel type, they are turned on when the pulses DS, WS, and AZ are at the high (H) level, respectively, and when the pulses are at the low (L) level. Turn off. This timing chart also shows the video signal Vsig, the potential change at the gate end G of the drive transistor 121, and the potential change at the source end S, along with the waveforms of the pulses WS, AZ, DS.

  Basically, the same driving is performed with a delay of one horizontal scanning period for each row of the write scanning line 104WS and the threshold & mobility correction scanning line 115AZ. Each timing and signal in the figure are indicated by the same timing and signal as the timing and signal of the first row regardless of the processing target row. In the description, when it is necessary to distinguish between rows, the processing target row is indicated by a reference with “_” in the timing and signal. Also, in the explanation and figures, when different drive pulses exist at the same timing, DS (scanning drive pulse DS), AZ (threshold & mobility correction pulse) to distinguish each drive pulse as necessary AZ), WS (when the write drive pulse is WS), and V (when the video signal is Vsig).

  The third example suppression method is characterized in that the threshold correction operation is repeated a plurality of times while applying the second example of the shading phenomenon suppression method associated with threshold correction. That is, at the drive timing shown in FIG. 10, the threshold value correction operation is repeated a plurality of times (for example, three times) every horizontal period including the effective period and the ineffective period of the video signal Vsig. Regarding the switching timing (t62V, t64V) of the effective period and the ineffective period of the video signal Vsig and the switching timing (t62DS, t64DS) of the scanning drive pulse DS active and inactive, Distinguish by indicating with a reference without _.

  In the drive timing shown in FIG. 10, the threshold value correcting operation is repeated a plurality of times with one horizontal period as a processing cycle. One horizontal period is a processing cycle of the threshold correction operation because the gate potential Vg of the drive transistor 121 is set prior to the threshold correction operation before the sampling transistor 125 samples the signal potential Vin in the storage capacitor 120 for each row. Is set to the offset voltage Vofs, and after the initialization operation to set the source potential Vs to the reference potential Vini, the time period in which the video signal line 106HS is at the offset voltage Vofs while the sampling transistor 125 remains conductive. This is because the threshold correction operation is performed to turn on the light emission control transistor 122 and hold the voltage corresponding to the threshold voltage Vth of the drive transistor 121 in the holding capacitor 120.

  Since the time zone in which the video signal line 106HS is at the offset voltage Vofs appears every horizontal period and exists in the first half of the video signal Vsig as described above and is narrower than one horizontal period, inevitably the threshold correction period is It becomes shorter than one horizontal period. Therefore, due to the difference between the capacitance Cs of the storage capacitor 120, the reference potential Vini and the offset voltage Vofs, and other factors, the storage capacitor 120 holds an accurate voltage corresponding to the threshold voltage Vth in this short threshold correction period. Cases that cannot be partitioned can also occur. The reason why the threshold correction operation is executed a plurality of times is to cope with this. That is, the voltage corresponding to the threshold voltage Vth of the drive transistor 121 is reliably held by repeatedly executing the threshold correction operation in a plurality of horizontal cycles preceding the sampling (signal writing) of the signal potential Vin to the holding capacitor 120. The capacity 120 is held.

  As a basic mechanism of drive timing, threshold correction and signal writing are performed within one horizontal scanning period. However, the number of pixels on the panel has increased, and the field frequency has been increased for higher image quality. In this case, since one horizontal scanning period is shortened, there is a possibility that threshold correction cannot be performed sufficiently. On the contrary, if the threshold correction period is secured to some extent, the signal writing time is compressed, so that the video signal Vsig (signal potential Vin) may not be sufficiently written to the storage capacitor 120. As an improvement, the threshold correction operation is performed a plurality of times to cope with higher definition and higher image quality of the panel.

  Then, the suppression method of the third example is such that the offset voltage Vofs and the signal potential Vin are set in a state in which the light emission control transistor 122 is turned on by continuously setting the scanning drive pulse DS to active H during the threshold correction operation for a plurality of times. In accordance with the video signal Vsig repeated in step S1, the threshold voltage Vth is written in the storage capacitor 120 by turning on the sampling transistor 125 by setting the write drive pulse WS to active H during the offset voltage Vofs. . That is, the remaining threshold correction period excluding the first and last threshold correction periods is an on period of the sampling transistor 125 (specifically, a period in which the sampling transistor 125 is on in a period in which the light emission control transistor 122 is on). It has features in the points to be defined. Although it is similar to the first example (also the comparative example), it differs in that the period of active H (the sampling transistor 125 is on) of the write drive pulse WS becomes dominant (priority).

  The reason for excluding the first threshold correction period is that the start point of the threshold correction period is defined when both the write drive pulse WS and the scan drive pulse DS become active H. In addition, when the signal writing is performed in the period of the first signal potential Vin after the last threshold correction period except for the last threshold correction period, the start point of the last threshold correction period is This is because the end point of the final threshold correction period is defined when the scanning drive pulse DS becomes inactive L while the write drive pulse WS is defined as active H. When signal writing is performed without performing signal writing in the first signal potential Vin period after the final threshold correction period, the end point of the final threshold correction period is the write drive pulse WS. Is defined at the time when becomes inactive L, and the final threshold correction period is also the on period of the sampling transistor 125 (specifically, the period in which the sampling transistor 125 is on in the period in which the light emission control transistor 122 is on). ).

  In the new field of line sequential scanning, first, the drive scanning unit 105 supplies the drive scan line 105DS of the first row with the threshold value & mobility correction pulse AZ and the write drive pulse WS being inactive L. The scanning drive pulse DS is switched from active H to inactive L (t50).

  As a result, the light emission control transistor 122 is turned off and the drive transistor 121 is disconnected from the power supply potential Vc1, so that the organic EL element 127 stops emitting light and enters a non-emission period. At the timing t50, the control transistors 122, 124, and 125 are turned off. At this time, since the write drive pulse WS is inactive L and the sampling transistor 125 is off, the gate terminal G of the drive transistor 121 is high impedance, and the holding capacitor 120 is connected between the gate and the source. Therefore, the source potential Vs and the gate potential Vg are lowered in association with each other so as to maintain the previous gate-source voltage Vgs.

  Next, the vertical drive unit 103 switches the threshold value & mobility correction pulse AZ to active H by the threshold value & mobility correction scan unit 115 while the scan drive pulse DS and the write drive pulse WS remain in the inactive L state. Then, the detection transistor 124 is turned on (t51 to t56). As a result, the reference potential Vini is set to the voltage of the node ND121, that is, the other end of the storage capacitor 120 and the source end S of the driving transistor 121, and the source potential Vs is initialized. The period (t51 to t62DS, t62WS) until the threshold correction operation is started is the initialization period C of the source potential Vs.

  At this time, since the write drive pulse WS is inactive L and the sampling transistor 125 is off, the gate terminal G of the drive transistor 121 is high impedance, and the holding capacitor 120 is connected between the gate and the source. Therefore, the gate potential Vg is lowered following the drop of the source potential Vs so as to maintain the previous gate-source voltage Vgs.

  Thereafter, the vertical drive unit 103 causes the write scan unit 104 to write the write drive pulse WS while the scan drive pulse DS is in the inactive L state and the threshold & mobility correction pulse AZ is in the active H state. Switching to active H, the sampling transistor 125 is turned on (t54WS), and after the threshold value & mobility correction pulse AZ becomes inactive L, the writing drive pulse WS is switched to inactive L (t58WS). As a result, the offset voltage Vofs is set to the voltage of the node ND122, that is, the gate terminal G of the drive transistor 121, and the gate potential Vg is initialized. The period (t54WS to t62DS, t62WS) until the threshold correction operation is started is the initialization period D of the gate potential Vg. Since the source potential Vs is not affected by the coupling at the timing of the gate potential Vg = Vofs of the drive transistor 121, the detection transistor 124 driven by the threshold & mobility correction pulse AZ is turned on and the source is set to Vini.

  The period (t54WS to t55WS) of the offset voltage Vofs of the video signal Vsig is included in the period (t54WS to t55WS) in which the write drive pulse WS is active H. Preferably, it is included a plurality of times (in this example, it is set twice).

  In this example, since the threshold value & mobility correction pulse AZ is in the inactive L state in the second half of the period (t54WS to t55WS) in which the write drive pulse WS is active H, the gate potential Vg is set to the offset voltage Vofs. The fluctuation at the time of transition to affects the source potential Vs.

  Also in this example, since the offset voltage Vofs and the reference potential Vini are set so as to satisfy “Vofs−Vini> Vth”, the gate-source voltage Vgs of the driving transistor 121, that is, the voltage held in the holding capacitor 120 is The voltage is set to exceed the threshold voltage Vth of the drive transistor 121, and the storage capacitor 120 is reset prior to the threshold correction operation. Further, since “VthEL> Vini” is set, a reverse bias is applied to the organic EL element 127 so that the subsequent threshold value correcting operation is normally performed.

  After completing the threshold correction preparation operation, the vertical drive unit 103 sets the scan drive pulse DS to active H by the drive scanning unit 105 to turn on the light emission control transistor 122 (t62DS1). Also, the write scanning unit 104 switches the write drive pulse WS to active H in accordance with the timing (t62V1 to t64V1) when the video signal Vsig is at the offset voltage Vofs, and the sampling transistor 125 is turned on (t62WS1).

  As a result, the drain current is used to charge and discharge the storage capacitor 120 and the organic EL element 127, and the first threshold correction is performed to record information for correcting (cancelling) the threshold voltage Vth of the drive transistor 121 in the storage capacitor 120. Enter period E. This first threshold value correction period E continues until the timing (t64WS1) when the write drive pulse WS is made inactive L.

  Preferably, the period (t62WS to t64WS) in which the write drive pulse WS is set to active H is completely included in the time zone (t62V to t64V) in which the video signal Vsig is at the offset voltage Vofs. Note that the timing t62WS and the timing t62DS may be substantially the same or may be in the same order. This is because the threshold correction period is defined in the period in which the write drive pulse WS is active H while the scan drive pulse DS is in active H. Of course, in practice, the threshold correction period is defined by the period in which the light emission control transistor 122 and the sampling transistor 125 to which the pulses DS and WS are supplied are actually turned on.

  In this example, the write drive pulse WS is first switched to active H (t62WS1) so that the video signal Vsig is completely included in the timing (t62V1 to t64V1) at the offset voltage Vofs, and then the write drive pulse. The scanning drive pulse DS is switched to active H (t62DS1) within a period in which WS is active H (t62WS1 to t64WS1).

  In the first threshold correction period E, the gate terminal G of the drive transistor 121 is held at the offset voltage Vofs of the video signal Vsig, and the drain current is increased until the source potential Vs of the drive transistor 121 rises and the drive transistor 121 is cut off. Tries to flow. When cut off, the source potential Vs of the drive transistor 121 becomes “Vofs−Vth”. That is, since the equivalent circuit of the organic EL element 127 is represented by a parallel circuit of a diode and a parasitic capacitance Cel, as long as “Vel ≦ Vcath + VthEL”, that is, the leakage current of the organic EL element 127 is greater than the current flowing through the drive transistor 121. Is considerably small, the current of the driving transistor 121 is used to charge and discharge the storage capacitor 120 and the parasitic capacitor Cel.

  As a result, when a drain current flows through the drive transistor 121, the voltage Vel at the anode end A of the organic EL element 127, that is, the potential of the node ND121 increases with time. Then, when the potential difference between the potential of the node ND121 (source potential Vs) and the voltage of the node ND122 (gate potential Vg) is just the threshold voltage Vth, the driving transistor 121 is turned off from the on state, and the drain current does not flow. The threshold correction period ends. That is, after a lapse of a certain time, the gate-source voltage Vgs of the drive transistor 121 takes the value of the threshold voltage Vth, and this information is held in the holding capacitor 120 connected between the gate and source.

  Here, a voltage corresponding to the threshold voltage Vth is written into the storage capacitor 120 connected between the gate terminal G and the source terminal S of the drive transistor 121. In practice, however, the first threshold correction period E is from the timing (t62WS1) at which the write drive pulse WS is set to active H to the timing (t64WS1) at which the write driving pulse WS is returned to inactive L. If this period is not sufficiently ensured, it ends before that. It becomes.

  Specifically, when the gate-source voltage Vgs becomes Vx1 (> Vth), that is, when the source potential Vs of the drive transistor 121 becomes “Vofs−Vx1” from the reference potential Vini on the low potential side. It will end. For this reason, Vx1 is written to the storage capacitor 120 at the time when the first threshold value correction period E is completed (t64WS1).

  Next, the writing scanning unit 104 inputs the writing driving pulse WS before the video signal Vsig becomes the signal potential Vin in the second half of one horizontal period while the scanning driving pulse DS remains in the active H state. The light emission control transistor 122 is turned off by switching to the active L (t64WS1), and the horizontal driver 106 samples the signal potential for the pixels in the other rows, so the potential of the video signal line 106HS is changed from the offset voltage Vofs to the signal potential. Switch to Vin (t64V1). As a result, the potential of the write scanning line 104WS (write drive pulse WS) becomes low level, while the video signal line 106HS changes to the signal potential Vin.

  As described above, the period t62WS to t64WS in which the write drive pulse WS is active H (that is, the period in which the sampling transistor 125 is turned on) is completely included in the period t62V to t64V in which the video signal Vsig is at the offset voltage Vofs. In other words, in other words, the period t64V to t62V in which the video signal Vsig is at the signal potential Vin is completely included in the period in which the sampling transistor 125 is reliably turned off.

  Here, in the period t64WS to t62WS in which the sampling transistor 125 is turned off, the light emission control transistor 122 is in a conductive (on) state, and a voltage corresponding to the threshold voltage Vth is sufficient for the storage capacitor 120 in the first threshold correction period E. Therefore, the gate-source voltage Vgs of the drive transistor 121 is larger than the threshold voltage Vth (Vgs> Vth). When the light emission control transistor 122 is turned on in such a state, a drain current flows through the drive transistor 121, the source potential Vs rises, and the gate potential Vg rises, so-called bootstrap operation (denoted as BST in the figure). ) Is performed.

  In the first half of the next one horizontal period (1H), the horizontal drive unit 106 switches the potential of the video signal line 106HS from the signal potential Vin to the offset voltage Vofs (t62V2), and then the write scanning unit 104 writes a write drive pulse. WS is switched to active H (t62WS2). As a result, the drain current flows into the storage capacitor 120 with the gate potential Vg of the drive transistor 121 set to the offset voltage Vofs, and information for correcting (cancelling) the threshold voltage Vth of the drive transistor 121 is recorded in the storage capacitor 120. A second threshold correction period (referred to as a second threshold correction period G) is entered. This second threshold value correction period G continues until the timing (t64WS2) when the write drive pulse WS is set to inactive L.

  In the second threshold correction period G, the same operation as the first threshold correction period E is performed. Specifically, the gate terminal G of the drive transistor 121 is held at the offset voltage Vofs of the video signal Vsig, and the gate potential Vg is instantaneously switched from the immediately preceding potential to the offset voltage Vofs. Thereafter, the drain current tends to flow until the source potential Vs of the driving transistor 121 rises and the driving transistor 121 is cut off. When cut off, the source potential Vs of the drive transistor 121 becomes “Vofs−Vth”.

  However, the second threshold correction period G is from the timing when the write drive pulse WS is set to active H (t62WS2) to the timing when it returns to inactive L (t64WS2), and when this period is not sufficiently secured, It will end before. This is the same as in the first threshold correction period E, and when the gate-source voltage Vgs becomes Vx2 (<Vx1 and> Vth), that is, the source potential Vs of the drive transistor 121 is “Vo−Vx1”. It ends when “Vo-Vx2” is reached. For this reason, Vx2 is written to the storage capacitor 120 at the time (t64WS2) when the second threshold correction period G is completed.

  Similarly, after the scanning drive pulse DS is once changed to inactive L (t64WS2), the third threshold correction period (the third threshold correction period I and the first half of the next horizontal period (1H)) is set. (T62WS3). This third threshold value correction period I continues until the timing (t64WS3) when the write drive pulse WS is made inactive L.

  In the third threshold correction period I, the same operation as the first threshold correction period E and the second threshold correction period G is performed. Specifically, the gate terminal G of the drive transistor 121 is held at the offset voltage Vofs of the video signal Vsig, and the gate potential is instantaneously switched from the immediately preceding potential to the offset voltage Vofs. Thereafter, the drain current tends to flow until the source potential Vs of the driving transistor 121 rises and the driving transistor 121 is cut off. The drain current is cut off when the gate-source voltage Vgs is just equal to the threshold voltage Vth. When cut off, the source potential Vs of the drive transistor 121 becomes “Vofs−Vth”.

  That is, the gate-source voltage Vgs of the drive transistor 121 takes the value of the threshold voltage Vth by the processing in the threshold correction period that is performed a plurality of times (three times in this example). Here, actually, a voltage corresponding to the threshold voltage Vth is written in the storage capacitor 120 connected between the gate terminal G and the source terminal S of the driving transistor 121.

  In addition, in the threshold correction periods E, G, and I for three times, in order to prevent the drain current from flowing exclusively to the storage capacitor 120 side (when Cs << Cel) and not to the organic EL element 127 side, As described above, “Vofs−Vth” so that the source potential Vs in the threshold correction periods E, G, and I does not exceed the threshold voltage VthEL of the organic EL element 127 so that the organic EL element 127 is cut off. By setting <VthEL + Vcath ”, the organic EL element 127 is maintained in the reverse bias state.

  When the organic EL element 127 is placed in a reverse bias state during the threshold correction periods E, G, and I, since it is in a cutoff state (high impedance state), it does not emit light and is not a diode characteristic but a simple capacitance characteristic. Will come to show. Therefore, the drain current (drive current Ids) flowing through the drive transistor 121 is a capacitance “C = Cs + Cel” obtained by combining both the capacitance value Cs of the storage capacitor 120 and the capacitance value Cel of the parasitic capacitance (equivalent capacitance) Cel of the organic EL element 127. It will be written. As a result, the drain current of the driving transistor 121 flows into the parasitic capacitance Cel of the organic EL element 127 and starts charging. As a result, the source potential Vs of the drive transistor 121 increases.

  After the threshold voltage Vth information is written in the storage capacitor 120 and the driving transistor 121 is cut off, the driving scanning unit 105 switches the scanning driving pulse DS to inactive L (t65). Thereafter, the horizontal drive unit 106 supplies the signal potential Vin of the video signal Vsig to the signal line 106HS (t66V to t67V) while the scanning drive pulse DS remains in the inactive L state, and the video signal Vsig is at the signal potential Vin. Within the period (t66V to t67V), the write scanning unit 104 sets the write drive pulse WS to active H to turn on the sampling transistor 125 (t66WS to t67WS).

  In this example, signal writing is not performed in the period of the signal potential Vin immediately after the third threshold value correction period I, and the writing drive pulse WS is set to active H in accordance with the timing of the signal potential Vin appearing after one horizontal period. I try to make it. Even in such driving, signal writing is performed immediately after threshold correction.

  Accordingly, since the signal potential Vin is supplied to the gate terminal of the driving transistor 121, the gate potential Vg of the driving transistor 121 changes from the offset voltage Vofs to the signal potential Vin, and information corresponding to the signal potential Vin is stored in the storage capacitor 120. Written. A period (t66WS to t67WS) in which the write drive pulse WS is set to active H after the threshold correction operation is completely completed is a signal writing period K (sampling period) in which the signal potential Vin is written to the storage capacitor 120. The signal potential Vin is held in the holding capacitor 120 in a form that is added to the threshold voltage Vth of the driving transistor 121.

  As a result, fluctuations in the threshold voltage Vth of the drive transistor 121 are always canceled, and threshold correction is performed. By this threshold correction, the gate-source voltage Vgs held in the holding capacitor 120 becomes “Vsig + Vth” = “Vin + Vth”.

  Next, the drive scanning unit 105 switches the scan drive pulse DS to active H (t68). As a result, the light emission control transistor 122 is turned on, so that the drive current Ids corresponding to the gate-source voltage Vgs (= Vin + Vth) at that time flows to the drive transistor 121 and proceeds to the light emission period L. In the light emission period L, as in the second example, the gate potential Vg of the drive transistor 121 can be changed in conjunction with the source potential Vs, and a bootstrap operation is possible. Thereafter, the process proceeds to the next frame (or field), and the threshold correction preparation operation, the threshold correction operation, and the light emission operation are repeated again.

  As described above, at the drive timing of the third example, as a threshold correction mechanism, the operation is performed within a plurality of horizontal scanning periods assigned to a plurality of rows, and the storage capacitor 120 is charged to the threshold voltage Vth in a time division manner. The sampling transistor 125 is supplied from the signal line 106HS during the signal supply period in which the signal line 106HS (that is, the video signal Vsig) becomes the signal potential Vin within the horizontal scanning period assigned to the write scanning line 104WS as the signal writing target. The video signal Vsig (signal potential Vin) is sampled in the holding capacitor 120.

  On the other hand, the correction means realized by controlling the on / off timing of the light emission control transistor 122, the detection transistor 124, and the sampling transistor 125 is a signal within each horizontal scanning period allocated to the write scanning lines 104WS of a plurality of rows. During the signal fixing period in which the line 106HS becomes the offset voltage Vofs having a constant potential, the threshold voltage Vth of the drive transistor 121 is detected, and the storage capacitor 120 is charged to the threshold voltage Vth in a time division manner. The signal fixed period in which the video signal Vsig is at the offset voltage Vofs is a period that separates the horizontal scanning periods sequentially assigned to the signal lines 106HS. As an example, the horizontal blanking period can be allocated so as to include the horizontal blanking period.

  The correction unit charges the storage capacitor 120 to the threshold voltage Vth in a time-sharing manner in each signal fixing period (offset voltage Vofs period). After the correction unit charges the holding capacitor 120 in each signal fixing period, the sampling transistor 125 is turned off before the signal line 106HS is switched from the offset voltage Vofs, which is a constant potential, to the signal potential Vin, so that the holding capacitor 120 is connected to the signal line. It is preferable to be electrically disconnected from 106HS. This is because by releasing the application of the video signal Vsig, the Vg of the driving transistor 121 can be increased, and a bootstrap operation that increases with the source potential Vs is enabled. Needless to say, the sampling transistor 125 is turned on in the signal writing period K.

  In the driving timing of the third example, the threshold correction operation (the operation of holding the information on the threshold voltage Vth in the storage capacitor 120) is executed a plurality of times as in the comparative example. Unlike the comparative example, except for the first time, the threshold correction period is defined by the on period of the sampling transistor 125 (specifically, the period in which the sampling transistor 125 is on within the period in which the light emission control transistor 122 is on). Thus, threshold correction is performed by turning on / off the sampling transistor 125.

  Even if the scanning drive pulse DS is active H in a plurality of threshold correction periods, the write drive pulse WS is inactive L during the period of the signal potential Vin, and the signal potential Vin is at the gate end G of the drive transistor 121. Will not be transmitted. Therefore, when the threshold correction operation is performed a plurality of times as in the comparative example, the source potential Vs rises due to the gate coupling phenomenon of the scanning drive pulse DS to the drain end D of the drive transistor 121. Therefore, it is possible to prevent the shading phenomenon accompanying the threshold correction. Further, since shading due to gate coupling of the scanning drive pulse DS can be avoided, the light emission control transistor 122 can be operated in the linear region even during the threshold correction period, and the specification of the drive scanning unit 105 is not complicated. That's it.

  In the drive timing shown in FIG. 10, the signal writing period K is provided separately from a plurality of threshold correction periods, but this is not essential. For example, after the final threshold correction period (the third threshold correction period I in the previous example), the timing may be changed to the signal writing period K by setting the same timing as in the second example.

  That is, after the threshold voltage Vth information is written in the storage capacitor 120 and the driving transistor 121 is cut off, the first half of one horizontal scanning period (period of the offset voltage Vofs) elapses, and the video signal Vsig becomes the signal potential Vin. Change. When the video signal Vsig is at the signal potential Vin, information on the signal potential Vin is written in the storage capacitor 120.

  For this reason, the write drive pulse WS is input before the video signal Vsig is switched to the signal potential Vin at each time (the first and second times in this example) except the threshold correction operation of the last time (the third time in this example). In the last threshold correction operation, the scanning drive pulse DS is set to inactive L before the video signal Vsig is switched to the signal potential Vin in preparation for the writing of the signal potential Vin. Even when Vsig is switched to the signal potential Vin, the write drive pulse WS is kept active H. Thus, since the signal potential Vin is supplied to the gate terminal of the drive transistor 121 with the light emission control transistor 122 turned off, the gate potential Vg of the drive transistor 121 changes from the offset voltage Vofs to the signal potential Vin, and the storage capacitor Information corresponding to the signal potential Vin is written in 120.

<About mobility correction>
If the timing t68 at which the scanning drive pulse DS that defines the start of the light emission period L is set to active H is set within the signal writing period K (t68μ: refer to the dotted line in the figure), the signal potential Vin is applied to the storage capacitor 120. Or the information on the signal potential Vin is written to the storage capacitor 120 and the light emission control transistor 122 is turned on while the sampling transistor 125 is turned on. Accordingly, the drain current can be passed through the driving transistor 121 while writing the information of the signal potential Vin to the holding capacitor 120, and the mobility for adding the correction amount for the mobility of the driving transistor 121 to the driving signal written to the holding capacitor 120. Correction can be performed.

  That is, the scanning drive pulse DS is set to active H before the timing t67WS at which the signal writing period K ends, and the light emission control transistor 122 is turned on. As a result, the drain terminal D of the drive transistor 121 is connected to the first power supply potential Vc1 via the light emission control transistor 122, so that the pixel circuit P proceeds from the non-light emission period to the light emission period.

  As described above, the mobility of the driving transistor 121 is corrected during the period t68 μ to t67WS in which the sampling transistor 125 is still in the on state and the light emission control transistor 122 is in the on state. Adjustment of the mobility of the driving transistor 121 of each pixel is optimized by adjusting the overlapping period (referred to as mobility correction period) of the active period of the writing drive pulse WS and the scanning drive pulse DS. That is, the mobility correction is appropriately executed in the period t68μ to t67WS where the rear part of the signal writing period and the head part of the light emission period overlap.

  Note that, at the beginning of the light emission period in which the mobility correction is performed, the organic EL element 127 does not emit light because it is actually in a reverse bias state. In the mobility correction period t68μ to t67WS, the drive current Ids flows through the drive transistor 121 while the gate terminal G of the drive transistor 121 is fixed at a potential corresponding to the video signal Vsig (specifically, the signal potential Vin).

  Here, by setting “Vofs−Vth <VthEL”, the organic EL element 127 is placed in a reverse bias state, so that it exhibits simple capacitance characteristics instead of diode characteristics. Therefore, the drive current Ids flowing through the drive transistor 121 is written to the capacitance “C = Cs + Cel” obtained by combining both the capacitance value Cs of the storage capacitor 120 and the capacitance value Cel of the parasitic capacitance (equivalent capacitance) Cel of the organic EL element 127. . As a result, the source potential Vs of the drive transistor 121 increases. Let this increase be ΔV.

  This increase ΔV, that is, the negative feedback amount ΔV, which is a mobility correction parameter, is eventually subtracted from the gate-source voltage Vgs held in the holding capacitor 120, so that negative feedback is applied. As described above, the mobility μ can be corrected by negatively feeding back the drive current Ids of the drive transistor 121 to the gate-source voltage Vgs of the drive transistor 121. Note that the negative feedback amount ΔV can be optimized by adjusting the time width of the mobility correction period t68μ to t67WS.

  The higher the video signal Vsig, the greater the drive current Ids and the greater the absolute value of ΔV. Therefore, mobility correction according to the light emission luminance level can be performed. Further, when considering the driving transistor 121 having a high mobility and the driving transistor 121 having a low mobility, if the video signal Vsig is constant, the absolute value of ΔV increases as the mobility μ of the driving transistor 121 increases.

  In other words, the source potential of the driving transistor 121 having high mobility during the mobility correction period is significantly increased compared to the driving transistor 121 having low mobility. Further, negative feedback is applied so that the potential difference between the gate and the source becomes smaller and the current hardly flows as the source potential increases greatly. Since the negative feedback amount ΔV increases as the mobility μ increases, it is possible to remove variations in the mobility μ for each pixel. Even in the drive transistor 121 having different mobility, the same drive current Ids can be passed through the organic EL element 127. By adjusting the mobility correction period, the magnitude of the negative feedback amount ΔV can be set to an optimum state.

  In the light emission period L after the mobility correction, the gate end G of the drive transistor 121 is disconnected from the video signal line 106HS, so that the application of the signal potential Vin to the gate end G of the drive transistor 121 is released, and the gate of the drive transistor 121 The potential Vg can be increased. At this time, the drive current Ids flowing through the drive transistor 121 flows through the organic EL element 127, and the anode potential of the organic EL element 127 rises according to the drive current Ids. Let this increase be Vel. At this time, since the gate-source voltage Vgs of the drive transistor 121 is constant due to the effect of the storage capacitor 120, the drive transistor 121 passes a constant current (drive current Ids) to the organic EL element 127. As a result, a voltage drop occurs, and the potential Vel at the anode end A of the organic EL element 127 (= potential at the node ND121) rises to a voltage at which the current called the drive current Ids can flow through the organic EL element 127. Meanwhile, the gate-source voltage Vgs held in the holding capacitor 120 maintains the value of “Vsig + Vth−ΔV”.

  Eventually, as the source potential Vs rises, the reverse bias state of the organic EL element 127 is canceled, so that the organic EL element 127 actually starts to emit light by the inflow of the drive current Ids. The rise (Vel) of the anode potential of the organic EL element 127 at this time is nothing but the rise of the source potential Vs of the drive transistor 121, and the source potential Vs of the drive transistor 121 becomes “−Vth + ΔV + Vel”.

  The relationship between the drive current Ids at the time of light emission and the gate voltage Vgs can be expressed as in Expression (3) by substituting “Vsig + Vth−ΔV” into Vgs in Expression (1) representing the previous transistor characteristics. it can.

  In Equation (3), k = (1/2) (W / L) Cox. From this equation (3), it can be seen that the term of the threshold voltage Vth is canceled and the drive current Ids supplied to the organic EL element 127 does not depend on the threshold voltage Vth of the drive transistor 121. Basically, the drive current Ids is determined by the signal voltage Vsig of the video signal. In other words, the organic EL element 127 emits light with a luminance corresponding to the video signal Vsig.

  At that time, the signal potential Vin is corrected by the feedback amount ΔV. This correction amount ΔV works so as to cancel the effect of the mobility μ located in the coefficient part of the equation (3). Therefore, the drive current Ids substantially depends only on the signal potential Vin. Since the drive current Ids does not depend on the threshold voltage Vth, even if the threshold voltage Vth varies depending on the manufacturing process, the drain-source drive current Ids does not vary, and the light emission luminance of the organic EL element 127 does not vary.

  By configuring the mobility correction circuit, the light emission control transistor interlocked with the writing operation of the video signal Vsig by the sampling transistor 125 within the period of the signal potential Vin of one horizontal period composed of the offset voltage Vofs and the signal potential Vin. Due to the operation in the mobility correction period 122, a constant current Ids that is not affected by the variation in the carrier mobility μ can flow as the gate-source potential Vgs reflecting the carrier mobility μ of the driving transistor 121. Therefore, the image can be displayed with stable gradation corresponding to the input pixel signal, and a high-quality image can be obtained.

  As mentioned above, although this invention was demonstrated using embodiment, the technical scope of this invention is not limited to the range as described in the said embodiment. Various changes or improvements can be added to the above-described embodiment without departing from the gist of the invention, and embodiments to which such changes or improvements are added are also included in the technical scope of the present invention.

  Further, the above embodiments do not limit the invention according to the claims (claims), and all combinations of features described in the embodiments are not necessarily essential to the solution means of the invention. Absent. The embodiments described above include inventions at various stages, and various inventions can be extracted by appropriately combining a plurality of disclosed constituent elements. Even if some constituent requirements are deleted from all the constituent requirements shown in the embodiment, as long as an effect is obtained, a configuration from which these some constituent requirements are deleted can be extracted as an invention.

<Modification of Pixel Circuit and Drive Timing>
For example, since “dual theory” holds in circuit theory, the pixel circuit P can be modified from this point of view. In this case, although not shown in the figure, the pixel circuit P having the 4TR configuration shown in FIG. 2 is configured using the n-channel driving transistor 121, whereas the p-channel driving transistor (hereinafter referred to as p) is used. The pixel circuit P is configured using a type driving transistor 121p. In accordance with this, the other transistors 122, 124, and 125 are also p-channel type to which an active L drive pulse is supplied, and the polarity of the signal potential Vin of the video signal Vsig and the magnitude relation of the power supply voltage are reversed. Make changes according to the reason.

  In the organic EL display device of the modified example in which the transistor is made p-type by applying such dual reason, the on-period of the sampling transistor 125 is changed in the same manner as the organic EL display device of the basic example of n-type described above. By controlling so as to define the threshold correction period, it is possible to prevent a shading phenomenon associated with threshold correction. Of course, since shading due to gate coupling of the scanning drive pulse DS can be avoided, the light emission control transistor can be operated in the linear region even during the threshold correction period, and the specification of the drive scanning unit does not have to be complicated.

  In addition, although the modification demonstrated here adds the change according to "the dual reason" with respect to 4TR structure shown in FIG. 2, the method of a circuit change is not limited to this. . For example, with respect to the 4TR configuration shown in FIG. 2, only the light emission control transistor 122 can be a p-channel type, or only the sampling transistor 125 can be a p-channel type. The same thing can be applied to the 4TR configuration shown in FIG. 2 in which a change according to “dual theory” is added, and only the light emission control transistor 122 can be made an n-channel type, or Only the sampling transistor 125 may be an n-channel type. In any case, the drive transistor 121 only needs to be controlled so that the threshold correction period is defined by the ON period of the sampling transistor during the threshold correction operation.

FIG. 1 is a block diagram showing an outline of a configuration of an active matrix display device which is an embodiment of a display device according to the present invention. FIG. 2 is a diagram illustrating an example of the pixel circuit of the present embodiment. It is a figure explaining the operating point of an organic EL element and a drive transistor. It is a figure explaining the influence which the characteristic variation of an organic EL element or a drive transistor has on drive current Ids. 6 is a timing chart for explaining the operation of a comparative example in the pixel circuit of the present embodiment. It is a figure explaining the condition of the waveform dullness of the scanning drive pulse when it is set as the drive timing of a comparative example. It is a figure explaining the coupling noise at the time of the light emission control transistor at the time of setting it as the drive timing of a comparative example. 7 is a timing chart showing the relationship between the gate potential (= video signal), source potential, and drain potential of the drive transistor in FIG. 6 and the scan drive pulse. 6 is a timing chart for explaining the operation of the first example of the pixel circuit of the embodiment. 6 is a timing chart illustrating an operation of a second example of the pixel circuit of the embodiment. 6 is a timing chart for explaining the operation of the third example of the pixel circuit according to the embodiment.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Organic EL display device, 101 ... Substrate, 102 ... Pixel array part, 103 ... Vertical drive part, 104 ... Write scanning part, 104WS ... Write scanning line, 105 ... Drive scanning part, 106 ... Horizontal drive part, 106HS DESCRIPTION OF SYMBOLS ... Video signal line, 109 ... Control part, 115 ... Threshold value & mobility correction scanning part, 115AZ ... Threshold value & mobility correction scanning line, 120 ... Retention capacity, 121 ... Drive transistor, 122 ... Light emission control transistor, 124 ... Detection transistor , 125 ... sampling transistor, 127 ... organic EL element, AZ ... threshold & mobility correction pulse, Cel ... parasitic capacitance of the organic EL element, DS ... scanning drive pulse, P ... pixel circuit, Vsig ... video signal, WS ... writing Drive pulse

Claims (12)

A driving transistor that generates a driving current, an electro-optical element connected to the output terminal of the driving transistor, a holding capacitor that holds information according to the signal potential of a video signal, and information that corresponds to the signal potential is written to the holding capacitor A sampling transistor, and a light emission control transistor arranged between a power supply end of the drive transistor and a power supply line to adjust a light emission period of the electro-optic element, and driving current based on information held in the holding capacitor A pixel array unit in which pixel circuits that emit light from the electro-optical element by being generated by a driving transistor and flowing through the electro-optical element are arranged in a matrix;
By sequentially controlling the sampling transistor in a horizontal cycle, the pixel circuit is line-sequentially scanned, and a write scan pulse for writing information corresponding to the signal potential of the video signal to each holding capacitor for one row is output from the sampling transistor. A writing scanning unit that outputs to the video signal, and a control unit that includes a horizontal driving unit that supplies a video signal for one row to the video signal line in accordance with a signal potential writing operation by the sampling transistor.
The control unit performs control so that a fixed potential for threshold correction operation for holding a voltage corresponding to the threshold voltage of the drive transistor in the storage capacitor is supplied to a control input terminal of the drive transistor, and A display device characterized in that a threshold correction operation period is controlled so that a period during which the sampling transistor is in a conductive state is dominant. The display device according to claim 1, wherein the control unit repeats the threshold correction operation a plurality of times in a time-sharing manner to set the voltage across the storage capacitor to the threshold voltage of the driving transistor. The horizontal driving unit outputs a fixed potential for the threshold correction operation to the video signal in a part of a horizontal scanning period,
The control unit maintains the light emission control transistor in a conducting state during a plurality of threshold correction operations, and switches the sampling transistor to a conducting state during the fixed potential period in the video signal. The display device according to claim 2, wherein the display device is controlled to perform the threshold correction operation each time. A driving transistor that generates a driving current, an electro-optical element connected to the output terminal of the driving transistor, a holding capacitor that holds information according to the signal potential of a video signal, and information that corresponds to the signal potential is written to the holding capacitor A sampling transistor, and a light emission control transistor arranged between a power supply end of the drive transistor and a power supply line to adjust a light emission period of the electro-optic element, and driving current based on information held in the holding capacitor A pixel array unit in which pixel circuits that emit light from the electro-optical element by being generated by a driving transistor and flowing through the electro-optical element are arranged in a matrix;
By sequentially controlling the sampling transistor in a horizontal cycle, the pixel circuit is line-sequentially scanned, and a write scan pulse for writing information corresponding to the signal potential of the video signal to each holding capacitor for one row is output from the sampling transistor. A writing scanning unit that outputs to the video signal, and a control unit that includes a horizontal driving unit that supplies a video signal for one row to the video signal line in accordance with a signal potential writing operation by the sampling transistor.
The control unit performs control so that a fixed potential for threshold correction operation for holding a voltage corresponding to the threshold voltage of the drive transistor in the storage capacitor is supplied to a control input terminal of the drive transistor, and The threshold correction operation period is controlled so that the period during which the light emission control transistor is in a conductive state is dominant, and the light emission control transistor operates in a saturation region during the threshold correction operation period, while the electro-optics A display device, wherein a light emission period of the element is controlled so that the light emission control transistor operates in a linear region. The display device according to claim 4, wherein the control unit repeats the threshold correction operation a plurality of times in a time-sharing manner to set the voltage across the storage capacitor to the threshold voltage of the driving transistor. The horizontal driving unit outputs a fixed potential for the threshold correction operation to the video signal in a part of a horizontal scanning period,
The control unit maintains the sampling transistor in a conductive state during the threshold correction operation over a plurality of times, and switches the light emission control transistor to a conductive state during the fixed potential period in the video signal. The display device according to claim 3, wherein control is performed so that the threshold correction operation is performed each time. Prior to the threshold correction operation, the control unit performs control so as to perform a preparatory operation for the threshold correction operation in which the voltage across the storage capacitor is set to be equal to or higher than the threshold voltage of the drive transistor. The display device according to claim 1, wherein the display device is a display device. In the pixel circuit, the storage capacitor is disposed between a control input terminal and the output terminal of the drive transistor, and in addition to the drive transistor, the sampling transistor, and the light emission control transistor, a voltage across the storage capacitor Has a switch transistor disposed between a reference potential for setting so as to be equal to or higher than a threshold voltage of the driving transistor and the output terminal of the driving transistor,
The display device according to claim 7, wherein the control unit brings the switch transistor into a conductive state during the preparatory operation for the threshold correction operation. The control unit performs control so as to perform a mobility correction operation for adding, to the information written in the storage capacitor, a correction for the mobility of the drive transistor after the threshold correction operation. 5. The display device according to 1 or 4. The control unit makes the sampling transistor non-conductive when information corresponding to the signal potential is written to the storage capacitor and stops the supply of the video signal to the control input terminal of the drive transistor, The display device according to claim 1, wherein an operation in which a potential of the control input terminal is interlocked with a potential fluctuation of the output terminal of the driving transistor is enabled. A driving transistor that generates a driving current, an electro-optical element connected to the output terminal of the driving transistor, a holding capacitor that holds information according to the signal potential of a video signal, and information that corresponds to the signal potential is written to the holding capacitor A sampling transistor, and a light emission control transistor arranged between a power supply end of the drive transistor and a power supply line to adjust a light emission period of the electro-optic element, and driving current based on information held in the holding capacitor A driving method of a pixel circuit that emits light from the electro-optical element by being generated by a driving transistor and flowing through the electro-optical element,
Control is performed such that a fixed potential for threshold correction operation for holding a voltage corresponding to the threshold voltage of the drive transistor in the storage capacitor is supplied to the control input terminal of the drive transistor, and the period of the threshold correction operation Is controlled so that a period during which the sampling transistor is in a conductive state is dominant. A driving transistor that generates a driving current, an electro-optical element connected to the output terminal of the driving transistor, a holding capacitor that holds information according to the signal potential of a video signal, and information that corresponds to the signal potential is written to the holding capacitor A sampling transistor, and a light emission control transistor arranged between a power supply end of the drive transistor and a power supply line to adjust a light emission period of the electro-optic element, and driving current based on information held in the holding capacitor A driving method of a pixel circuit that emits light from the electro-optical element by being generated by a driving transistor and flowing through the electro-optical element,
Control is performed such that a fixed potential for threshold correction operation for holding a voltage corresponding to the threshold voltage of the drive transistor in the storage capacitor is supplied to the control input terminal of the drive transistor, and the period of the threshold correction operation Is controlled so that the period during which the light emission control transistor is in a conductive state is dominant, and during the threshold correction operation period, the light emission control transistor operates in a saturation region, while the light emission period of the electro-optical element is A driving method, wherein the light emission control transistor is controlled to operate in a linear region.