JP2008092611A - Inverter device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an inverter device which reduces losses of reactors and switching elements, and suppresses the deterioration of the efficiency of a motor. <P>SOLUTION: In the inverter device 1 having an inverter circuit which is connected to a load composed of a plurality of phases, the inverter device comprises, at each phase, a boosting circuit 6U constituted of: the first reactor connected to the positive electrode end side of a DC power supply 2; the second reactor connected to the negative electrode side of the DC power supply; a first capacitor connected between the input end of the first reactor and the output end of the second reactor; and a second capacitor connected between the output end of the first reactor and the input end of the second reactor. At the output end side of each boosting circuit, the switching element at the positive side and the switching element at the negative side are connected in series, a series circuit formed by connecting each switching element and a diode in parallel is connected to the output end side, and the load is connected to the connecting point of the switching elements of the series circuit. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to an inverter device having a booster circuit that boosts a DC power supply.

  In hybrid vehicles, fuel cell vehicles, electric vehicles, and the like, a driving force is generated by an electric motor (hereinafter referred to as a motor) and transmitted to an axle. In order to obtain an optimum driving force according to the running state of the vehicle, the power supply voltage of the DC power supply is boosted to a desired voltage by a booster circuit, and the driving force of the motor is obtained based on the boosted voltage.

  As a booster circuit that realizes high output and high efficiency, an impedance (Z) source booster circuit described in Patent Document 1 has been proposed. The Z source booster circuit includes a first reactor connected to the positive electrode end side of the DC power source 2, a second reactor connected to the negative electrode end side of the DC power source, an input end of the first reactor, and an output end of the second reactor. And a second capacitor connected between the output terminal of the first reactor and the input terminal of the second reactor. An inverter circuit is connected to the output side of the booster circuit.

  In the inverter circuit, for the U, V, and W phases, an IGBT module in which an IGBT element (Insulated Gate Bipolar mode Transistor) (switching element) and a free wheel diode are connected in reverse parallel constitutes each arm of the three-phase inverter circuit. The IGBT module constituting the upper arm (high side) and the IGBT module constituting the lower arm (low side) are connected in series to constitute a three-phase inverter circuit. Conventionally, the booster circuit is provided in common for the U, V, and W phases of the inverter circuit. The inverter circuit is controlled by a pulse width modulation (PWM) system so that each phase current matches the target current for the U phase, the V phase, and the W phase for each carrier cycle.

The Z source booster circuit accumulates magnetic energy by charging the first and second reactors and discharges the first and second capacitors during a short period in which the upper and lower arms of any of U, W, and W are short-circuited. In the zero vector period in which all the upper and lower arms of the U, V, and W phases are short-circuited, and in the energization period by PWM control, the first and second reactors are discharged and the first and second capacitors are charged. To increase the pressure. At this time, current flows through the first and second reactors in the short period, the zero vector period, and the energization period. When increasing the boost voltage, the short period is lengthened. Conventionally, the target boost voltage Vo ^* of the booster circuit has been set to be common to the U, V, and W phases, for example, Vo ^* = {(Vd ^* ) ² + (Vq ^* ) ² )} ^1/2 . However, Vd ^* is a d-axis target voltage, and Vq ^* is a q-axis target voltage.
US Patent Application Publication No. 2003/0231518

However, the conventional booster circuit has the following problems because it is boosted based on the boost command value Vo ^* common to the U, V, and W phases.
(1) To increase the boost command value Vo ^* , it is necessary to lengthen the short period. In the short period, the charging current flows through the first and second reactors. There is a problem that the charging current of the second reactor is increased, and the loss due to the resistance of the first and second reactors is increased. In addition, during the energization period and the zero vector period, a discharge current flows through the first and second reactors. However, when the boosted voltage Vo is large, the discharge currents of the first and second reactors increase, and the first and second reactors There is a problem that the loss due to the resistance increases. Conventionally, since the boost command voltage Vo ^* is a high voltage common to U, V, and W, there is a problem that loss due to resistance of the first and second reactors becomes large. In particular, since the loss is proportional to the square of the current, the loss increases as the current increases. There is a problem that the first and second reactors generate heat due to loss.
(2) The switching element of the inverter circuit is ON / OFF controlled by a pulse width modulation method. When the switching element is turned ON / OFF, both the terminal voltage and current of the switching element pass through the non-saturation region and take an intermediate value, and thus switching loss of the element occurs between them. This switching loss increases as the boosted voltage Vo increases. However, as described above, since the boost command value Vo ^* increases, there is a problem that the boosted voltage Vo increases and the switching loss increases. is there. Normally, the short period is provided by short-circuiting the upper and lower arms of the phase to be changed at the timing when the U, V, and W phase PWM control patterns are changed. A large short circuit current flows and the switching loss becomes large.
(3) Although a motor is connected to the inverter circuit as a load, the following ripple current Irip shown by the equation (1) is included in the phase current of the motor.

Irip = (Vo−Vemf) / L (1)
However, Vo is a boosted voltage, Vemf is a counter electromotive voltage generated in the motor coil in proportion to the rotation speed of the motor, and L is an inductance of the motor coil. Conventionally, since the boosted voltage Vo is a high voltage common to the U, V, and W phases, the boosted voltage Vo increases, and the ripple current Irip increases accordingly. When the ripple current Irip increases, there is a problem that eddy current flows in the iron core of the motor, the motor iron loss increases, and the efficiency of the motor decreases.

  The present invention has been made in view of the above-described problems, and an object of the present invention is to provide an inverter device that reduces the loss of the reactor and the switching element and suppresses the decrease in the efficiency of the motor.

  According to invention of Claim 1, it is an inverter apparatus which has an inverter circuit connected to the load which consists of several phases, Comprising: The 1st reactor connected to the positive electrode end side of DC power supply, The negative electrode terminal of the said DC power supply A second reactor connected to the side, a first capacitor connected between an input end of the first reactor and an output end of the second reactor, an output end of the first reactor and an input of the second reactor A booster circuit configured to include a second capacitor connected to each end of the booster circuit. The booster circuit is provided for each phase, and a positive-side switching element and a negative-side switching element are connected in series on the output end side of each booster circuit. A series circuit in which a diode is connected in parallel with each of the switching elements is connected, and the load is connected to a connection point between the switching elements of the series circuit. Inverter apparatus is provided, wherein.

  According to a second aspect of the present invention, in the first aspect of the present invention, the load is a three-phase motor having a stator and a rotor, and a rotation angle of the rotor relative to the stator with respect to a phase current of each phase flowing through the load. Provided is an inverter device characterized in that when the phase current of the phase in a predetermined range is less than or equal to a predetermined value, the boosted voltage by the phase booster circuit is controlled to be smaller than in the case of other phase currents Is done.

  According to a third aspect of the present invention, in the first aspect of the invention, the phase voltage command value calculating means for calculating the phase voltage command value of each phase based on the command torque of the load, and the step-up circuit for each phase There is provided an inverter device comprising boost command value calculation means for calculating a boost command value based on the phase voltage command value of the phase.

  According to a fourth aspect of the present invention, in the third aspect of the present invention, the boost command value calculation means is a boost circuit corresponding to the phase voltage command value based on the phase voltage command value and the voltage of the DC power supply. The inverter device is characterized in that it is determined whether or not boosting is necessary, and when the boosting is not necessary, the switching elements on the positive side and the negative side of the phase are not short-circuited.

  According to a fifth aspect of the present invention, in the fourth aspect of the present invention, when it is determined that the boost command value calculation means needs to be boosted, the step-up command phase voltage command value of the boost circuit that needs to be boosted is determined. An inverter device is provided that increases the step-up command value for the phase as the absolute value increases.

  According to a sixth aspect of the present invention, in the fifth aspect of the present invention, when the boost command value calculating means determines that the boost is necessary, the absolute value of the boost command value of the boost circuit requiring the boost is determined. An inverter device is provided in which the value is equal to the absolute value of the phase voltage command value of the phase.

  According to the first aspect of the present invention, since the booster circuit is provided for each phase, the optimum boosted voltage can be obtained for each phase, so that the switching loss of the switching element is reduced, and the loss flowing through the first and second reactors is reduced. In addition, the load efficiency can be improved.

  According to the invention of claim 2, when the load is a three-phase motor having a stator and a rotor, the phase current of each phase indicated by the rotation angle of the rotor with respect to the stator is a low-current phase current in a predetermined range. Since the contribution to the command torque is small, the boosted voltage by the booster circuit of the phase is controlled to be smaller than that of the other phase currents, so that the boosted voltage can be reduced and the switching loss of the switching element is reduced. Loss flowing through the first and second reactors can be reduced and load efficiency can be improved.

  According to the third aspect of the present invention, the boost command value for each phase booster circuit is calculated based on the phase voltage command value, so that the optimum boost command value can be obtained, and the switching loss of the switching element is reduced. It is possible to reduce the loss flowing through the two reactors and improve the load efficiency.

  According to the fourth aspect of the present invention, when there is no need for boosting, the switching element on the positive electrode side and the negative electrode side of the phase can be prevented from being short-circuited, so that the short period can be eliminated. The loss flowing through the first and second reactors can be reduced, and the load efficiency can be improved.

  According to the fifth aspect of the present invention, when it is determined that boosting is necessary, the phase boost command value is increased as the absolute value of the phase voltage command value of the booster circuit that requires boosting increases. It becomes an optimal boost command value, and it is possible to reduce the switching loss of the switching element, reduce the loss flowing through the first and second reactors, and improve the load efficiency.

  According to the sixth aspect of the present invention, the boost command value can be minimized, the switching loss of the switching element is reduced, the loss flowing through the first and second reactors is reduced, and the load efficiency is improved. Can be optimized.

  FIG. 1 is a configuration diagram of an inverter device 1 shown according to an embodiment of the present invention. As shown in FIG. 1, the inverter device 1 includes a DC power source 2, a smoothing capacitor 4, diodes DU, DV, DW, transistors (IGBT elements) QiU, QiV, QiW, booster circuits 6U, 6V, 6W, and a bridge circuit 7U, A booster / bridge circuit 8U, 8V, 8W comprising 7V, 7W, a motor 9, a battery voltage sensor 10, phase current sensors 12U, 12V, 12W, a position detection sensor 14 and an ECU 16 are provided.

  The DC power supply 2 is a power storage device for supplying electric power to the motor 9 through the booster circuits 6U, 6V, 6W and the bridge circuits 7U, 7V, 7W, and is a lithium ion battery, a nickel metal hydride battery, or the like. A plurality of battery blocks in which batteries are modularized are connected in series. The DC power supply 2 may be a capacitor.

  The diodes DU, DV, DW have anodes connected to the DC power supply 2 (DC power supply) and cathodes connected to the first reactors L1U, L1V, L1W of the booster circuits 6U, 6V, 6W. IGBT elements QiU, QiV, and QiW are connected in parallel to diodes DU, DV, and DW, and form switching elements.

  The diodes DU, DV, DW are for turning on in the energization period and zero vector period, and turning off in the short period. Transistors QiU, QiV, and QiW are provided as switching elements that switch on and off between the DC power supply 2 and the booster circuits 6U, 6V, and 6W according to a gate signal input from the ECU 16.

  The collectors of the IGBT elements QiU, QiV, QiW are connected to the first reactors L1U, L1V, L1W, respectively, and the emitters are connected to the positive terminal of the DC power supply 2. The IGBT elements QiU, QiV, and QiW are appropriately turned on by the ECU 16 when the load current of the motor 9 is low, or turned on when the DC power source 2 is charged with the generated power generated by driving the motor 9 by an external motor (not shown). Or turned on when the DC power source 2 is charged with regenerative power generated by regenerative braking of the motor 9.

  The smoothing capacitor 4 is connected in parallel to the positive electrode end and the negative electrode end of the DC power supply 2 and removes noise due to switching of the IGBT elements UH, UL, VH, VL, WH, WL.

  The step-up circuits 6U, 6V, 6W are provided for each phase, and are connected to the cathodes of the diodes DU, DV, DW (the positive end side of the DC power supply 2), respectively, and the DC power supply 2 The second reactors L2U, L2V, and L2W respectively connected to the negative electrode end sides of the first reactors L2U, L1V, and L1W, and the second reactors L2U, L2V, and L2W, respectively. First capacitors C1U, C1V, C1W, and second capacitors C2U, C2V, C2W connected between the output terminals of the first reactors L1U, L1V, L1W and the input terminals of the second reactors L2U, L2V, L2W, respectively The Z source booster circuit is configured to include:

  In a short period in which the IGBT elements UH and UL of the bridge circuit 7U are short-circuited by the control of the ECU 16, the booster circuit 6U includes the first reactor L1 → UH → UL → first capacitor C1U → first reactor L1U and second reactor L2U → A current flows through the second capacitor C2U → UH → UL → second reactor L2U, the magnetic energy is accumulated by charging the first and second reactors L1U and L2U, and the first and second capacitors C1U and C2U are discharged. The voltage is boosted to the boosted voltage Vou determined by the time length of the period. The boosting operation in the short period of the boosting circuits 6V and 6W is the same as that of the above-described boosting circuit 6U. As described above, the booster circuits 6U, 6V, and 6W independently boost the desired boosted voltages Vou, Vov, and Vow, respectively.

  In the step-up circuit 6U, the DC power source 2 → the diode DU → the first reactor L1U → the second capacitor C2U → the DC power source 2 and the DC power source 2 → the diode DU → the first capacitor C1U → the second reactor during the energization period and the zero vector period. L2U → DC power source 2, first and second reactors L1U and L2U are discharged, and first and second capacitors C1U and C2U are discharged. The discharge of the first and second reactors L1V, L1W, L2V, L2W and the discharge operation of the first and second capacitors C1V, C1W, C2V, C2W during the energization period and zero vector period of the booster circuits 6V, 6W The operation is the same as that of the circuits 6V and 6W.

  The bridge circuits 7U, 7V, and 7W have positive output terminals on the positive side of each phase of the Z source booster circuits 6U, 6V, and 6W, positive power supply lines on each phase of the U, V, and W phases, and output terminals on the negative side on the U, It is a bridge circuit of each phase connected to each phase negative power supply line of V, W phase, and a bridge circuit 7U, 7V, 7W constitutes a three-phase inverter circuit.

  In the bridge circuits 7U, 7V, and 7W, an IGBT module in which an IGBT element (switching element) and a free wheel diode are connected in reverse parallel constitutes each arm of the three-phase inverter circuit. The IGBT modules constituting the upper and lower arms of the U phase, V phase, and W phase are connected in series to form a bridge circuit.

  IGBT element UH and flywheel diode DUH constitute the upper arm (high side) of the U phase. Further, IGBT element VH and flywheel diode DVH constitute an upper arm of V phase, and IGBT element WH and flywheel diode DWH constitute an upper arm of W phase.

  IGBT element UL and flywheel diode DUL constitute the lower arm (low side) of the U phase. The IGBT element VL and the flywheel diode DVL constitute a lower arm of the V phase, and the IGBT element WL and the flywheel diode DWL constitute a lower arm of the W phase.

  The collectors of the IGBT elements UH, VH, and WH are connected to the output end sides of the Z source boost circuits 6U, 6V, and 6W of the first reactors L1U, L1V, and L1W, respectively. The emitters of the IGBT elements UL, VL, WL are connected to the output end sides of the Z source boost circuits 6U, 6V, 6W of the second reactors L2U, L2V, L2W, respectively. Flywheel diodes DUH, DVH, DWH, DUL, DVL, DWL are connected between the collectors of the IGBT elements UH, VH, WH, UL, VL, WL so that the direction from the emitter to the collector is the forward direction. Has been.

  A pulse signal (gate signal) for turning ON / OFF the IGBT elements UH, UL, VH, VL, WH, WL by pulse width modulation is input from the ECU 16 to the gates of the IGBT elements UH, UL, VH, VL, WH, WL. . The emitters of the IGBT elements UH, VH, and WH and the collectors of the IGBT elements UL, VL, and WL are connected to the U, V, and W phase coil terminals of the motor 9, respectively.

  A PWM control pattern by pulse width modulation is represented by (***) in the order of U, V, and W phases. * Is 1/0, is 1 when UH, VH, and WH are in the ON state, and is 0 when UH, VH, and WH are in the OFF state. When UH, VH, and WH are in the ON state, UL, VL, and WL are in the OFF state.

  In a period other than the zero vector, for example, when the PWM control pattern is (100), UH is ON, VH and WH are OFF, UL is OFF, VL and WL are ON, and DUL, DVH, DWH is turned off. At this time, the current flowing in the motor 9 is: DC power source 2 → DU → first reactor L1U → UH → U phase coil of the motor 9 → V and W phase coils of the motor 9 → VL, WL → second reactors L2V and L2W → It flows with DC power supply 2. The U-phase current iu is a current corresponding to the sum of the U-phase and V-phase interphase voltage Vuv = Vou + Vov and the U-phase and W-phase interphase voltage Vuw = Vou + Vow. That is, the U-phase current iu is a current corresponding to the sum of the U-phase and V-phase voltages and the U-phase and W-phase voltages.

  For example, when the PWM control pattern is (110), UH and VH are in the ON state, WH is in the OFF state, UL and VL are in the OFF state, WL is in the ON state, and DUL, DVL, and DWH are in the OFF state. Become. At this time, the current flowing through the motor 9 is: DC power source 2 → diodes DU, DV → first reactors L1U, L1V → UH, VH → U and V phase coils of the motor 9 → W phase coils of the motor 9 → WL → second Reactor L2W → DC power supply 2 flows. Further, the U-phase and V-phase boosted voltages Vou and Vov are set to Vou> Vov. Since Vou> Vov, DVH is turned ON. A current iuv corresponding to the interphase voltage (Vou−Vov) between the U phase and the V phase flows from the first reactor L1U to the first reactor L1V through UH and DVH. That is, the U-phase current iu is a current corresponding to the sum of the U-phase and W-phase voltages and the U-phase and V-phase voltages. In the case of other PWM control patterns other than the zero vector, the current flowing through the motor 9 is the same as described above.

  In the (111) zero vector, UH, VH, and WH are on, UL, VL, and WL are off, and DUL, DVL, and DWL are off. The U-phase, V-phase, and W-phase boosted voltages Vou, Vov, and Vow are set to Vou> Vov and Vow. Since Vou> Vov and Vow, DVH and DWH are turned ON. A current iuv corresponding to the interphase voltage (Vou−Vov) between the U phase and the V phase flows from the first reactor L1U to the first reactor L1V through UH and DVH. Further, a phase current iuw corresponding to the interphase voltage (Vou-Vow) between the U phase and the W phase flows from the first reactor L1U to the first reactor L1W side through UH and DWH. Therefore, the U-phase current iu is (iuv + ivw). That is, the U-phase current iu is a current corresponding to the sum of the U-phase and V-phase interphase voltages and the U-phase and W-phase interphase voltages.

  In the (000) zero vector, UH, VH, and WH are in an OFF state, UL, VL, and WL are in an ON state, and DUH, DVH, and DWH are in an OFF state. The U-phase, V-phase, and W-phase boosted voltages Vou, Vov, and Vow are set to Vou> Vov and Vow. Since −Vou <−Vov and −Vow, DVL and DWL are turned ON. A current -ivu corresponding to the phase voltage (-Vov + Vou) between the U-phase and the V-phase flows from the second reactor L2V to the second reactor L2U through DVL and UL. In addition, a phase current -iwu corresponding to the phase voltage (-Vow + Vou) between the U phase and the W phase flows from the second reactor L2W to the second reactor L2U side through DWL and UL. Therefore, the U-phase current iu is (−ivu−iwv). That is, the U-phase current iu is a current corresponding to the voltage between the U-phase and the V-phase and between the U-phase and the W-phase.

  Similarly, the V phase current iv is a current corresponding to the sum of the interphase voltage of the V phase and the U phase and the interphase voltage of the V phase and the W phase. Similarly, the W-phase current iw is a current corresponding to the sum of the W-phase and U-phase voltages and the W-phase and V-phase voltages.

Therefore, in the pulse width modulation method, the average phase voltage in the carrier period is equal to the phase voltage command values Vu ^* , Vv ^* , Vw ^*, and therefore corresponds to the interphase voltage of the phase voltages Vu ^* , Vv ^* , Vw ^*. Phase currents iu, iv, iv flow, and command torque can be supplied to the motor 9.

  The motor 9 is a three-phase power device, for example, a DC brushless motor mounted as a drive source on a vehicle such as a hybrid vehicle, a fuel cell vehicle, or an electric vehicle. The battery voltage sensor 10 detects the power supply voltage Vbatt of the DC power supply 2. The phase current sensors 12U, 12V, and 12W are sensors that detect the phase currents iu, iv, and iw of the U, V, and W phases flowing through the motor 9. The position detection sensor 14 is a sensor that detects a relative rotation angle θm between the stator and the rotor of the motor 9. The output signals of the sensors 10, 12U, 12V, 12W, 14 are input to the ECU 16, converted from analog signals to digital signals by the analog / digital converter, and processed by the ECU 16.

The ECU 16, which is an electronic control unit, functions as motor control means for controlling the drive and regenerative operation of the motor 9, and as shown in FIG. 2, the target Vd ^* , Vq ^* calculation means 52, the phase voltage command value Vu. ^* , Vv ^* , Vw ^* calculation means 54, boost command values Vou ^* , Vov ^* , Vow ^* calculation means 56, short time calculation means 58, pulse width determination means 60 and gate signal output means 62 are realized by executing a program or the like. It has the function to do.

The target Vd ^* , Vq ^* calculating means 52 performs feedback control of current on the dq coordinates forming the rotation orthogonal coordinates, and includes an accelerator opening sensor (not shown) that detects the accelerator opening related to the accelerator operation of the driver, A target d-axis current id ^* and a target are calculated from a torque command value for the motor 9 corresponding to a driving state of the vehicle calculated from detection signals by sensors such as on / off of a brake switch (not shown) related to a driver's brake operation. The q-axis current iq ^* is calculated. The d-axis current id obtained by converting the target d-axis current id ^* , the target q-axis current iq ^* , the rotation angle θm, and the detected values of the U-phase current iu, the V-phase current iv and the W-phase current iw onto the dq coordinate. And the q-axis current iq, the deviations between the d-axis current id and the q-axis current iq and the target d-axis current id ^* and the target q-axis current iq ^*, and the deviation between the rotational speed of the motor 9 and the rotational speed command value are as follows. The target d-axis voltage Vd ^* and the target q-axis voltage Vq ^* are calculated so as to be zero.

The phase voltage command values Vu ^* , Vv ^* , Vw ^* calculation means 54, for example, perform coordinate conversion of the target d-axis voltage Vd ^* and the target q-axis voltage Vq ^*, and phase U, V, W phases to be applied to the motor 9. The voltage command values Vu ^* , Vv ^* , Vw ^* are calculated.

The step-up command values Vou ^* , Vov ^* , Vow ^* calculation means 56 are provided for the boost circuits 6U, 6V, 6W based on the voltage Vbatt and the phase voltage command values Vu ^* , Vv ^* , Vw ^* output from the battery voltage sensor 10. This is means for calculating the boost command values Vou ^* , Vov ^* , Vow ^* and is calculated as follows, for example.
(A) It is determined whether or not the booster circuit 6U needs to be boosted. That is, when the voltage Vbatt detected by the battery voltage sensor 10 and the phase voltage command value Vu ^* are compared, and the phase voltage command value Vu ^* is within the range of the power supply voltage of the DC power supply 2, the boosting circuit 6U performs boosting. It is determined that it is not necessary. Otherwise, it is determined that boosting by the booster circuit 6U is necessary. The booster circuits 6V and 6W are the same as the booster circuit 6U.
(B) if there is a need for boosting by the boosting circuit 6U, the phase when the voltage command value Vu ^* is increased, the boost command value Vou ^* is calculates the boost command value Vou ^* to be larger. For example, boost command value Vou ^* = phase voltage command value Vu ^* . The booster circuits 6V and 6W are the same as the booster circuit 6U. As described above, the boost command values Vou ^* , Vov ^* , and Vow ^* are separately calculated for the U, V, and W phases. Note that the boost command values Vou ^* , Vov ^* , Vow ^* may be larger than the phase voltage command values Vu ^* , Vv ^* , Vw ^* .

The short time calculation means 58 sets the short period = 0 for the U phase when boosting by the boosting circuit 6U is not necessary, and from the boost command value Vou ^* and the voltage of the DC power supply 2 when boosting is necessary. The short time Ts is calculated so that the boost voltage Vou of the booster circuit 6U matches the boost command value Vou ^* . For the V phase and the W phase, the short time Ts is calculated as in the U phase.

The pulse width determining means 60 uses the phase voltage command value Vu ^* and the carrier cycle Tc as a cycle, and corresponds to the boost command value Vou ^* (the boost command value is the voltage Vbatt of the DC power supply 2 when boosting is not required). A PWM control signal to be applied to the gates of the U-phase IGBT elements UH and UL is obtained by a pulse width modulation method based on a triangular wave carrier signal having an amplitude. Similarly, for the V and W phases, PWM control signals to be applied to the gates of VH, VL, WH, and WL are obtained.
For example, when the boost command value Vou ^* = phase voltage command value Vu ^* , UH is in the ON state throughout the carrier cycle. The pulse width modulation method may be, for example, a space vector modulation method other than the triangular wave carrier modulation method.

  The gate signal output means 62 is synchronized with the carrier cycle Tc and the short period continues for the short time Ts of the U, V, and W phases calculated by the short time calculation means 58. For example, UH, VH, and WH are turned on. A gate signal is output so as to short-circuit at the timing. After the short period, the gate signal is output so that UH, VH, and WH are turned on for the pulse width determined by the pulse width determining means 60, and then the gate signal is output so that UH, VH, and WH are turned off. If pulse width + short period> carrier period, pulse width = carrier period-short period.

  Hereinafter, the control method for the inverter according to the present embodiment will be described with reference to FIGS. Since the control for the U, V, and W phases is substantially the same, FIGS. 3 and 5 show the U phase. In step S2, the command torque and the command rotation speed are input. In step S4, the rotation angle θm is detected from the position detection sensor 14, the U-phase current iu, the V-phase current iv and the W-phase current iw are detected from the phase current sensors 12U, 12V and 12W, and the rotation speed of the motor 9 is detected from a rotation speed sensor (not shown). Is done. In step S6, the following processing is performed.

A target d-axis current id ^* and a target q-axis current iq ^* are calculated from the torque command value. The d-axis current id obtained by converting the target d-axis current id ^* , the target q-axis current iq ^* , the rotation angle θm, and the detected values of the U-phase current iu, the V-phase current iv and the W-phase current iw onto the dq coordinate. And the q-axis current iq, the deviations between the d-axis current id and the q-axis current iq and the target d-axis current id ^* and the target q-axis current iq ^*, and the deviation between the rotational speed of the motor 9 and the rotational speed command value The target d-axis voltage Vd ^* and the target q-axis voltage Vq ^* are calculated so that becomes zero. Then, the target d-axis voltage Vd ^* and the target q-axis voltage Vq ^* are coordinate-converted, and U, V, and W phase voltage command values Vu ^* , Vv ^* , and Vw ^* to be applied to the motor 9 are calculated. The following steps S8 to S16 are executed separately for the U, V, and W phases. Here, the U phase will be described.

As shown in FIG. 4, the phase voltage command value Vu ^* becomes a sine wave with respect to time. It is assumed that the sensor output value of the battery voltage sensor 10 has a voltage range of the DC power supply 2 in the range of −Vbatt to Vbatt, and the battery voltage sensor 10 detects the battery voltage Vbatt. In step S8, the absolute value of the phase voltage command value Vu ^* is compared with the battery voltage Vbatt detected by the battery voltage sensor 10, and it is determined whether or not | Vu ^* | is larger than Vbatt.

  If the determination is affirmative, it is determined that boosting by the booster circuit 7U is necessary, and the process proceeds to step S12. If a negative determination is made, it is determined that boosting by the booster circuit 7U is not necessary, and the process proceeds to step S14. For example, at time t1 to t2 and time t3 to t4, it is determined that boosting by the booster circuit 6U is necessary, and at time t2 to t3, it is determined that boosting by the booster circuit 6U is not necessary.

In step S10, the boost command value Vou ^* is calculated from the phase voltage command value Vu ^* . For example, the boost command value Vou ^* = the absolute value of the phase voltage command value Vu ^* . In step S12, a short time Ts for boosting is calculated from the boost command value Vou ^* and the battery voltage Vbatt. At this time, as shown in FIG. 4, when the absolute value of the boost command value Vou ^* increases, the time length of the short period Ts increases. In step S14, the pulse is based on the phase voltage command value Vu ^* and the triangular wave carrier signal having an amplitude corresponding to the boost command value Vou ^* (or Vou ^* = Vbatt if no boost is required) with the carrier cycle Tc as the cycle. The high-level pulse width of the gate signal applied to the UH gate is obtained by the width modulation method. The pulse width modulation method may be, for example, a space vector modulation method other than the triangular wave carrier modulation method.

For example, as shown in FIG. 5, the boost command value Vou ^*, for example, as compared with the triangular wave carrier signal CS and the phase voltage command values Vu ^* having amplitude equal to the battery voltage Vbatt of the DC power source 2, applied to the gate of the UH A high level pulse width W1 of the gate signal to be calculated is calculated. At this time, the average emitter voltage of UH (the voltage of the U-phase coil terminal) in the carrier period is equal to the phase voltage command value Vu ^* , and when the UH is in the ON state, the U-phase coil terminal voltage is set to the boost command value Vou ^* . Will be equal.

Meanwhile, conventionally, the step-up command value Vo ^* is U, V, is a common pressure in W phase, and the triangular wave carrier signal CS 'having an amplitude value equal to the boost command value Vo ^* phase voltage command value Vu ^* comparison Then, the high-level pulse width W2 of the gate signal applied to the UH gate is calculated. At this time, when UH is in the ON state, the U-phase coil terminal voltage is equal to the boost command value Vo ^* and is higher than the boost command value Vou ^* . In FIG. 5, ViUH indicates the level of the UH gate signal according to the present embodiment, and V′iUH indicates the level of the UH gate signal according to the prior art.

In step S16, UH and UL are both ON for the short time Ts at the timing when the UH is turned from the OFF state to the ON state, for example, after the short time Ts has elapsed from the timing when the UH is turned from the OFF state to the ON state. By outputting a gate signal so as to turn OFF, a short period of time Ts is provided, and after the UL is turned OFF, the ON state of UH is continued by the pulse width calculated in step S14. For example, in FIG. 5, the boost command value Vou ^* = Vbatt and there is no need for boosting by the boosting circuit 6U, so no short period is provided. On the other hand, a short period Ts ′ is conventionally provided for boosting to the boost command value Vo ^* .

As a result, the boost voltage Vou of the boost circuit 6U becomes equal to the boost command value Vou ^* . Further, the average voltage of the U-phase coil of the motor 9 in the carrier cycle Tc is equal to the phase voltage command value Vu ^* . For the V and W phases, processing similar to that for the U phase is executed, and the average voltages of the U and W phase coils of the motor 9 in the carrier cycle Tc become equal to the phase voltage command values Vv ^* and Vw ^* . Therefore, the phase currents iu, iv, iw of the motor 9 become current values based on the interphase voltages of the phase voltage command values Vu ^* , Vv ^* , Vw ^* , and the command torque can be supplied to the motor 9.

As shown in FIG. 6, the boosted voltage Vou in the region B where the phase voltage command value Vu ^* becomes equal to or lower than the battery voltage Vbatt of the DC power supply 2 and the phase current iu decreases becomes equal to Vbatt. Further, the boosted voltage Vou in A region raised by the phase current iu is greater than the phase voltage command value Vu ^* of the DC power supply 2 battery voltage Vbatt is between the phase voltage command values Vu ^*. The relationship between the phase currents iv and iw and the boosted voltages Vov and Vow is the same as the relationship between the phase current iu and the boosted voltage Vou.

The first embodiment has the following effects.
(1) Since the boosted voltages Vou, Vov, Vow of the booster circuits 6U, 6V, 6W are set to voltages according to the phase voltage command values Vu ^* , Vv ^* , Vw ^* , the first reactors L1U, L1V in the energization period , L1W and the second reactors L2U, L2V, L2W, the current flowing in the conventional reactor is smaller than that of the conventional one, and the loss due to the resistance of the first and second reactors can be suppressed. Further, when boosting is not required for each booster circuit 6U, 6V, 6W, boosting is not performed, and boosted voltages Vou, Vov, Vow of each booster circuit 6U, 6V, 6W are set to phase voltage command value Vu ^*. , Vv ^* , and Vw ^* , the short-circuit period is eliminated, and the short-circuit period is shorter than in the prior art. Therefore, the current flowing during the short-circuit period is reduced, and the resistances of the first and second reactors are reduced. The loss due to can be suppressed.
(2) Since the boosted voltages Vou, Vov, Vow of the booster circuits 6U, 6V, 6W are voltages corresponding to the phase voltage command values Vu ^* , Vv ^* , Vw ^* , switching loss can be suppressed.
(3) Since the boosted voltages Vou, Vov, Vow of the booster circuits 6U, 6V, 6W are the voltages indicated by a1 in FIG. 7 corresponding to the phase voltage command values Vu ^* , Vv ^* , Vw ^* , the formula (1 ) Is smaller than the conventional voltage b1, and the ripple current Irip1 in a2 can be made smaller than the conventional ripple current Irip2. Therefore, motor iron loss can be suppressed, and reduction in motor efficiency can be suppressed.

FIG. 8 is a functional block diagram of the ECU 100 according to the second embodiment applied to the configuration of the inverter device 1 similar to FIG. 1, and the same reference numerals are given to substantially the same components as those in FIG. It is attached. In the present embodiment, the calculation methods of the boost command values Vou ^* , Vov ^* , and Vow ^* are different. The boost command values Vou ^* , Vov ^* , Vow ^* calculation means 102 compares the phase voltage command values Vu ^* , Vv ^* , Vw ^* with the battery voltage Vbatt of the DC power supply 2 as shown in FIG. In the B region where boosting by 6U, 6V, 6W is not necessary, the boosted voltages Vou, Vov, Vow are made equal to the battery voltage of the DC power supply 2, and in the A region where boosting by the boosting circuits 6U, 6V, 6W is necessary. The command values Vou ^* , Vov ^* , Vow ^* are set to a constant boost command voltage Vo ^* , for example, Vo ^* = {(Vd ^* ) ² + (Vq ^* ) ² } ^1/2 . The effect in the B region of the second embodiment is the same as that of the first embodiment.

In the present embodiment, the boost command values Vou ^* , Vov ^* , Vow ^* of the boost circuits 6U, 6V, 6W are calculated based on the phase voltage command values Vu ^* , Vv ^* , Vw ^* and the battery voltage Vbatt of the DC power supply 2. However, the boost command values Vou ^* , Vov ^* , Vow ^* can be calculated from the phase currents iu, iv, iw detected by the phase current sensors 12U, 12V, 12W.

For example, the phase of the phase currents iu, iv, iw indicated by the rotation angle θm detected by the position detection sensor 14 has a predetermined range in which the phase current for each phase has a small contribution for realizing the command torque, For example, when the phase currents iu, iv, iw are near zero and the absolute values of the phase currents iu, iv, iw are below a certain value, the boost command values Vou ^* , Vov ^* , Vow ^* are set to a certain value, The battery voltage Vbatt of the power supply 2 is set, and the other boost command values Vou ^* , Vov ^* , Vow ^* are set to, for example, V0 ^* = {(Vd ^* ) ² + (Vq ^* ) ² } ^1/2 .

The boost command values Vou ^* , Vov ^* , and Vow ^* may be in two stages as shown in FIG. 9, or in three or more stages. The calculated boost command values Vou ^* , Vov ^* , Vow ^* are used, for example, in the next carrier cycle at the time of calculation, and are subjected to pulse width modulation.

It is a figure which shows the inverter apparatus by embodiment of this invention. It is a block diagram of 1st Embodiment which concerns on the motor control of ECU in FIG. It is a flowchart which shows the motor control method by 1st Embodiment of this invention. It is a time chart which shows the motor control method by 1st Embodiment of this invention. It is a time chart which shows the motor control method by 1st Embodiment of this invention. It is a wave form diagram which shows the pressure | voltage rise command value by 1st Embodiment of this invention. It is a figure for demonstrating the effect of this invention. It is a block diagram concerning motor control by a 2nd embodiment of the present invention. It is a wave form diagram which shows the pressure | voltage rise command value by 2nd Embodiment of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Inverter apparatus 2 DC power supply 6U, 6V, 6W Boost circuit 7U, 7V, 7W Bridge circuit 9 Motor 10 Battery voltage sensor 12U, 12V, 12W Phase current sensor 14 Position detection sensor 16 ECU
56 Boost command values Vou ^* , Vov ^* , Vow ^* calculation means 58 Short period calculation means 60 Pulse width determination means

Claims (6)

An inverter device having an inverter circuit connected to a load composed of a plurality of phases,
A first reactor connected to the positive electrode end side of the DC power source, a second reactor connected to the negative electrode end side of the DC power source, and an input end of the first reactor and an output end of the second reactor A booster circuit configured to include a first capacitor connected and a second capacitor connected between the output terminal of the first reactor and the input terminal of the second reactor is provided for each phase.
On the output terminal side of each booster circuit, a positive-side switching element and a negative-side switching element are connected in series, and a series circuit in which a diode is connected in parallel with each of the switching elements is connected. The inverter is characterized in that the load is connected to a connection point between the switching elements.   When the load is a three-phase motor having a stator and a rotor, and the phase current of each phase flowing through the load is within a predetermined range of the rotation angle of the rotor with respect to the stator, the phase current of the phase is a predetermined value or less. 2. The inverter device according to claim 1, wherein the boosted voltage by the phase booster circuit is controlled to be smaller than that of other phase currents.   Phase voltage command value calculation means for calculating a phase voltage command value of each phase based on the command torque of the load, and a booster that calculates a boost command value for the boost circuit of each phase based on the phase voltage command value of the phase The inverter device according to claim 1, further comprising command value calculation means.   The boost command value calculating means determines whether or not boosting is required by a boosting circuit corresponding to the phase voltage command value based on the phase voltage command value and the voltage of the DC power supply. 4. The inverter device according to claim 3, wherein the switching elements on the positive electrode side and the negative electrode side of the phase are not short-circuited.   When it is determined that boosting is necessary, the boost command value calculating means calculates the absolute value of the boost command value of the phase as the absolute value of the phase voltage command value of the boosting circuit that requires boosting increases. The inverter device according to claim 4, wherein the inverter device is enlarged.   When it is determined that boosting is required, the boost command value calculation means is configured so that the absolute value of the boost command value of the booster circuit that requires boosting is equal to the absolute value of the phase voltage command value of the phase. The inverter device according to claim 5.