JP2008054466A - Motor controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor controller which improves current controllability by lessening the dead time, irrespective of the magnitude of the carrier period. <P>SOLUTION: When the time required for the time for one current control calculation from reading of a phase current value and a rotation angle up to the calculation of a voltage command value is the current-control calculation time, then it is constituted that a point of time which is prior only by the current control calculation time from the next voltage command value output timing is set as start timing of the current control calculation. In other words, it is so constituted that reading is performed at the time that is as close as possible to the voltage command value output point of time (the time prior to the voltage command value output point of time by only the current control calculation time). Since the time that is faster only by the current control calculation time from the voltage command value output timing is set as the start timing of the current control calculation, irrespective of the PWM carrier period, the dead time is shortened, even if the PWM carrier period becomes larger with respect to the current control calculation time, and current controllability is improved. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a motor control device, and more particularly to a technique for setting calculation start timing in a control calculation.

As described in Patent Document 1 below, the phase current value and rotation angle of the motor are captured at the timing of a predetermined phase of the PWM carrier signal (for example, peak or valley: the peak of a triangular wave), and the difference between the phase current value and the current command value A control device is disclosed that calculates a voltage command value based on the rotation angle and outputs the calculated voltage command value at the time of the predetermined phase in the next carrier cycle.
JP-A-2005-57901

As described above, conventionally, the reading of the phase current and the rotation angle and the output timing of the voltage command value are determined according to the carrier period. If this carrier cycle is changed in accordance with the rotation speed of the electric motor, there is an advantage that the switching loss of the inverter can be reduced and the temperature rise of the switching element can be suppressed by extending the carrier cycle during low-speed rotation. However, in this configuration, the dead time from the reading of the phase current and the rotation angle to the actual output of the voltage command value is determined according to the carrier cycle. There is a problem that the control is delayed due to the increase, and the current controllability is deteriorated.
The present invention solves the above-described problem, and an object thereof is to provide an electric motor control device capable of improving current controllability by reducing dead time regardless of the carrier period.

  In order to achieve the above object, in the present invention, when the time required for one current control calculation from the reading of the phase current value and the rotation angle to the calculation of the voltage command value is defined as the current control calculation time, A time point before the current control calculation time from the voltage command value output timing is set as a start timing of the current control calculation. That is, in the present invention, the reading is performed at a time point as close as possible to the voltage command value output time point (a time point before the voltage command value output time point by the current control calculation time).

  In the present invention, since the time point before the current control calculation time from the voltage command value output timing is the current control calculation start timing regardless of the PWM carrier cycle, the PWM carrier cycle becomes longer than the current control calculation time. However, the wasted time can be shortened. Therefore, current controllability can be improved.

FIG. 1 is a circuit block diagram showing an example of an electric motor control device to which the present invention is applied. In this example, the case of a three-phase motor is illustrated, but the present invention is not limited to this.
In FIG. 1, 1 is a DC power source, 2 is an inverter, 3 is an electric motor, 4 is a control unit (details will be described later), 5 is a voltage sensor for detecting the voltage Vdc of the DC power source 1, 6 is each phase current i _{u of} the electric motor, A current sensor 7 detects i _v , i _w , and 7 is an angle sensor (resolver, encoder, etc.) that detects the rotor rotation angle (rotation phase) θ (electrical angle) of the motor. Further, Q1 to Q6 in the inverter 2 are switching elements (IGBT or the like), and D1 to D6 are drive signals for driving the switching elements Q1 to Q6, respectively. T ^* is a torque command value given from the outside, for example, a signal set according to an accelerator signal or a vehicle speed signal.

The control unit 4 includes, for example, a CPU and an electronic circuit, and inputs a DC power supply voltage Vdc, motor phase currents i _u , i _v , i _w , a motor rotation angle θ, and a torque command value T ^* , and performs vector control. The current control calculation and the PWM calculation are performed, and the drive signals D1 to D6 are output, whereby the switching elements Q1 to Q6 of the inverter 2 are controlled to open and close to supply current to the motor 3, and the motor 3 is set to the torque command value T ^*. Drive according to.

FIG. 2 is a block diagram showing an internal configuration of the control unit 4 in FIG.
In FIG. 2, the inverter 2 and the electric motor 3 are the same as those in FIG. 1, and a portion surrounded by a broken line corresponds to the control unit 4 in FIG. 1.
In the control unit 4, the current map 11 is based on the torque command value T ^* [N · m] given from the outside and the rotor angular velocity ω [rad / s] of the motor, and the current command values (i and d-axis) _d ^* [A], i _q ^* [A]) is obtained by map drawing. The rotor angular velocity ω is obtained by differentiating the rotor rotation angle θ by the differentiating unit 15.

Deviations between the current command values i _d ^* and i _q ^* and the actual d-axis current i _d and q-axis current i _q are obtained, and the results are sent to the dq-axis voltage command value calculation unit 12. The actual d-axis and q-axis currents i _d ^* and i _q are obtained by the dq-axis conversion unit 16 using the read phase currents i _u , i _v and i _w of the motor and the rotation angle θ of the motor. Obtained by performing three-phase to two-phase conversion.
In dq-axis voltage command value computing unit 12, * current command value _{i ^d, i} ^{q *} and the actual d-axis current _i d, the deviation between the q-axis current _{i q ⁽ⁱ} d * -i _d and _i ^{q *} - The proportional-integral control is performed from i _q ) using the following equation (Equation 1) to calculate the d-axis voltage command value v _d ^* and the q-axis voltage command value v _q ^* .

但し、（数１）式において、
ｖ_ｄ ^＊…ｄ軸電圧指令値［Ｖ］ ｖ_ｑ ^＊…ｑ軸電圧指令値［Ｖ］
ｉ_ｄ ^＊…ｄ軸電流指令値［Ａ］ ｉ_ｑ ^＊…ｑ軸電流指令値［Ａ］
ｉ_ｄ…ｄ軸電流［Ａ］
ｉ_ｑ…ｑ軸電流［Ａ］
Ｋ_ｐｄ…ｄ軸比例ゲイン Ｋ_ｐｑ…ｑ軸比例ゲイン
Ｋ_ｉｄ…ｄ軸積分ゲイン Ｋ_ｉｑ…ｑ軸積分ゲイン
ｓ…ラプラス演算子
なお、ここで非干渉制御を用いても良い。

However, in Equation (1),
v _d ^* ... d-axis voltage command value [V] v _q ^* ... q-axis voltage command value [V]
i _d ^* ... d-axis current command value [A] i _q ^* ... q-axis current command value [A]
i _d ... d-axis current [A]
i _q q-axis current [A]
K _pd ... d-axis proportional gain K _pq ... q-axis proportional gain K _id ... d-axis integral gain K _iq ... q-axis integral gain s ... Laplace operator Here, non-interference control may be used.

Next, in the two-phase / three-phase conversion unit 13, the d-axis voltage command value v _d ^* and the q-axis voltage command value v _q ^* are converted into the three-phase voltage command value v _u ^* , using the following equation (2): Convert to v _v ^* , v _w ^* .

但し、（数２）式において、
ｖ_ｕ ^＊…ｕ相電圧指令値［Ｖ］ ｖ_ｖ ^＊…ｖ相電圧指令値［Ｖ］
ｖ_ｗ ^＊…ｗ相電圧指令値［Ｖ］ ｖ_ｄ ^＊…ｄ軸電圧指令値［Ｖ］
ｖ_ｑ ^＊…ｑ軸電圧指令値［Ｖ］ θ…回転子回転角（電気角）［ｒａｄ］
なお、回転角θは遅れ補償をした値を用いても良い。

However, in the formula (2),
v _u ^* ... u-phase voltage command value [V] v _v ^* ... v-phase voltage command value [V]
v _w ^* ... w-phase voltage command value [V] v _d ^* ... d-axis voltage command value [V]
v _q ^* ... q-axis voltage command value [V] θ ... rotor rotation angle (electrical angle) [rad]
The rotation angle θ may be a value with delay compensation.

Then, the PWM conversion unit 14, the three-phase voltage command value of the ^{_{v u *, v v *,}} v w * and the power supply voltage Vdc and the three-phase voltage command value based on the PWM carrier signal period _{T 0 v} ^{u *} , V _v ^* , v _w ^* are converted into PWM signals. In other words, to calculate the phase pulse width _t u of the PWM _signal, t v, a _{t w} by using the following equation (3). For example, a triangular wave or a sawtooth wave is used as the PWM carrier signal.

但し、（数３）式において、
ｔ_ｕ…ｕ相パルス幅［ｓ］ ｔ_ｖ…ｖ相パルス幅［ｓ］ ｔ_ｗ…ｗ相パルス幅［ｓ］
Ｔ_０…ＰＷＭキャリア信号周期［ｓ］ Ｖdc…電源電圧［Ｖ］
次に、デッドタイム補償部１７では、三相電流値（ｉ_ｕ、ｉ_ｖ、ｉ_ｗ）によって決まる補償値をマップ引きにより求め、それぞれ上記各相のパルス幅に加える。
なお、この場合の三相電流値（ｉ_ｕ、ｉ_ｖ、ｉ_ｗ）は、ｄｑ軸電流指令値を三相変換した三相電流指令値でも良いし、ｄｑ軸電流指令値に電流応答相当のフィルタをかけたｄｑ軸電流推定値を三相変換した三相電流推定値でも良い。
また、前記ｄｑ軸変換部１６における三相電流値（ｉ_ｕ、ｉ_ｖ、ｉ_ｗ）からｄｑ軸電流への変換は、下記（数４）式で行う。

However, in the formula (3),
t _u ... u-phase pulse width [s] t _v ... v-phase pulse width [s] t _w ... w-phase pulse width [s]
T ₀ ... PWM carrier signal cycle [s] Vdc ... Power supply voltage [V]
Next, the dead time compensation unit 17 obtains a compensation value determined by the three-phase current values (i _u , i _v , i _w ) by map drawing, and adds each to the pulse width of each phase.
In this case, the three-phase current values (i _u , i _v , i _w ) may be three-phase current command values obtained by three-phase conversion of the dq-axis current command values. A three-phase current estimated value obtained by three-phase conversion of the filtered dq-axis current estimated value may be used.
Further, the conversion from the three-phase current values (i _u , i _v , i _w ) to the dq-axis current in the dq axis conversion unit 16 is performed by the following equation (Equation 4).

但し、（数４）式において、
ｉ_ｕ：ｕ相電流値［Ａ］ ｉ_ｖ：ｖ相電流値［Ａ］ ｉ_ｗ：ｗ相電流値［Ａ］
θ…回転子回転角［ｒａｄ］
この後、デッドタイム補償部１７から出力されたＰＷＭ信号（パルス幅）に従ってインバータ２の各スイッチング素子を開閉駆動することにより、電動機３に電圧を印加して駆動する。

However, in Equation (4),
i _u : u-phase current value [A] i _v : v-phase current value [A] i _w : w-phase current value [A]
θ: Rotor rotation angle [rad]
Thereafter, each switching element of the inverter 2 is driven to open and close in accordance with the PWM signal (pulse width) output from the dead time compensation unit 17 to drive the motor 3 by applying a voltage.

The above configuration is the same as the current feedback control by the conventional vector control. In the present invention, the three-phase voltage command values v _u ^* , v _v ^* , v in the two-phase three-phase conversion unit 13 are read from the respective phase currents i _u , i _v , i _w and the rotor rotation angle θ shown in FIG. _This relates to the start timing of the current control calculation up to the calculation of _w ^* .

Hereinafter, the start timing of the current control calculation of the conventional example and the present invention will be compared and described.
FIG. 7 is a diagram showing the relationship between the PWM carrier signal and the current control calculation timing in the conventional example. In FIG. 7, a triangular wave indicates a carrier signal, and shows a case where the carrier period is large in the order of (a) <(b) <(c). 3 to 7, the black thick one-dot chain line indicates the output time point of the three-phase voltage command value, and the black square indicates the current control calculation time.

  The current calculation start timing is determined by the valley of the PWM carrier signal, that is, the point A in FIG. 3C. At this time, the current of each phase and the rotation angle are read, and then the voltage command value is calculated. The calculation result is output at the next trough of the PWM carrier signal. Therefore, a dead time corresponding to one cycle is generated in the PWM carrier signal from the reading of each phase current and the rotation angle to the output of the voltage command value as the calculation result. In the case of FIG. 7A in which the rotation speed of the electric motor is large and the cycle of the PWM carrier signal is short, the dead time is also short, but when the cycle of the PWM carrier signal is long as shown in FIG. As a result, the control is delayed, and the current control characteristics are degraded.

(First embodiment)
FIG. 3 is a diagram showing a PWM carrier signal and calculation start timing in the first embodiment of the present invention. FIG. 3 shows a case where the calculation start timing is determined using a timer other than the PWM carrier signal. Also in this case, the output timing of the three-phase voltage command value is at the time of the valley (predetermined phase) of the PWM carrier signal, but the current control calculation start timing is set by the current control calculation start timing setting timer.

As shown in FIG. 3, the current control calculation start timing setting timer is reset at the same cycle as the cycle of the PWM carrier signal, and is shifted and counted by the current control calculation time before the PWM carrier cycle.
Here, the current control calculation time means a time required for one current control calculation from the reading of the three-phase current value and the rotation angle to the calculation of the three-phase voltage command value.
Therefore, if the time point when the timer is reset is set as the calculation start timing, the time point that is the current control calculation time before the next voltage command value output timing can be set as the current control calculation start timing. With this configuration, the dead time from the reading of each phase current and rotation angle to the output of the calculated voltage command value is always substantially equal to the current control calculation time regardless of the PWM carrier cycle. It becomes equivalent and can be made as short as possible.

  Note that the actual current control calculation time varies somewhat in the practical calculation process, and the longest time is defined as the current control calculation time. That is, the current control calculation time is the longest time required from the reading of each phase current value and rotation angle to the calculation of the three-phase voltage command value, and before the next voltage command value output timing. Thus, the timing after the one cycle before the carrier cycle is set as the start timing of the current control calculation.

In practice, it is not necessary to accurately match the current control calculation time, and it may be set to a good time close to the current control calculation time. The same applies to the following embodiments.
Also, the current control calculation time is set according to the current control calculation load determined based on the CPU performing the motor control, and can be shortened by using a high-speed CPU, but the cost is high. Become.

(Second embodiment)
FIG. 4 is a diagram showing a PWM carrier signal and current calculation start timing in the second embodiment.
In this embodiment, a reset value variable timer is used as the current control calculation start timing setting timer, the next current control calculation time is calculated during the current control calculation, and the current control calculation time is calculated according to the current control calculation time obtained by the calculation. The timer reset value is set, and the current control calculation start timing is set to the time when the timer that has started counting from the predetermined phase (peak or valley) of the PWM carrier signal is reset.

If the current control calculation time changes while performing current control with the PWM carrier signal, the next current control calculation time is calculated during the current current control calculation as described above, and the current control calculation start timing is set. By changing the reset value of the timer, the dead time can always be minimized.
The reset timer does not count until the valley of the PWM carrier signal comes, and starts counting from the time of the valley of the PWM carrier signal after the next current control calculation.
If the current control calculation time changes in the middle, a flag is set according to the change in the calculation contents, and the current control calculation time setting value, that is, the current control calculation start timing setting timer is reset according to the flag. Set to change the value.

(Third embodiment)
FIG. 5 is a diagram showing a PWM carrier signal and current calculation start timing in the third embodiment.
In FIG. 5, the cycle of the current control calculation start timing setting timer is set to substantially the same value as the longest time of the current control calculation time, the current control calculation is repeated regardless of the voltage command value output timing, and the final calculation result Are sequentially stored, and the stored value at the time of the next voltage command value output timing (carrier signal valley) is output as the voltage command value.
With this configuration, as shown in FIG. 5, even when the carrier cycle and the current control calculation time change every time, the dead time can be shortened without measuring the PWM carrier signal and the current control time. .

(Fourth embodiment)
FIG. 6 is a diagram showing a PWM carrier signal and current calculation start timing in the fourth embodiment.
In this embodiment, the current calculation start timing is set using a PWM carrier signal without using another timer. As shown in FIG. 6, when the PWM carrier signal can be measured, a threshold value for the current control calculation time is provided for the PWM carrier signal, and when the threshold value falls below this threshold value, the current control calculation is started. The current control calculation can be started at a timing obtained by subtracting the current control calculation time from the timing at which the voltage command value is output. That is, the value of the PWM carrier signal is read, and the time when the PWM carrier signal reaches the threshold set according to the current control calculation time is set as the calculation start timing. In this embodiment, since no other timer is required, the cost can be reduced.

  The above embodiments may be combined. In the above description, the configuration in which the three-phase voltage command value is output at the time of the valley of the PWM carrier signal has been described as an example. However, the peak of the PWM carrier signal indicated by point B in FIG. The same applies to the configuration output at, and not only valleys and peaks, but also other predetermined phases of the PWM carrier signal may be used.

The circuit block diagram which shows an example of the electric motor control apparatus to which this invention is applied. The block diagram which shows the internal structure of the control part 4 in FIG. The figure which shows the PWM carrier signal and calculation start timing in 1st Example of this invention. The figure which shows the PWM carrier signal and calculation start timing in 2nd Example of this invention. The figure which shows the PWM carrier signal and calculation start timing in 3rd Example of this invention. The figure which shows the PWM carrier signal and calculation start timing in 4th Example of this invention. The figure which shows the relationship between the PWM carrier signal and current control calculation timing in a prior art example.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... DC power supply 2 ... Inverter 3 ... Electric motor 4 ... Control part 5 ... Voltage sensor 6 ... Current sensor 7 ... Angle sensor 11 ... Current map 12 ... dq axis voltage command value calculating part 13 ... Two-phase three-phase conversion part 14 ... PWM Conversion unit 15 ... Differentiation unit 16 ... dq axis conversion unit 17 ... Dead time compensation unit

Claims (7)

Read the phase current value and rotation angle of the motor at the timing of the predetermined phase of the PWM carrier signal, calculate the voltage command value for controlling the motor based on the difference between the phase current value and the current command value and the rotation angle, In the motor controller that outputs the calculated voltage command value at the time of the predetermined phase in the next carrier cycle of the PWM carrier signal,
When the time required for one current control calculation from the reading of the phase current value and the rotation angle to the calculation of the voltage command value is defined as a current control calculation time, the current control calculation time from the next voltage command value output timing An electric motor control device characterized in that a time point just before is set as the start timing of the current control calculation.   The current control calculation time is the longest time required to calculate the voltage command value from the reading of the phase current value and rotation angle, and is before the longest time before the next voltage command value output timing, The motor control device according to claim 1, wherein a timing later than one cycle before the carrier cycle is set as a start timing of the current control calculation.   A timer that is reset in the same period as the PWM carrier cycle is shifted and counted by the current control calculation time before the PWM carrier period, and the time point when the timer is reset is set as the calculation start timing. The electric motor control device according to claim 1 or 2.   A reset value variable timer is provided, the next current control calculation time is calculated during the current current control calculation, the reset value of the timer is set according to the current control calculation time obtained by the calculation, and the PWM carrier signal The motor control device according to claim 1, wherein the calculation start timing is a time when the timer that has started counting from a predetermined phase is reset.   Regardless of the voltage command value output timing, the current control calculation is repeatedly performed and the final calculation result is sequentially stored, and the stored value at the time of the next voltage command value output timing is output as the voltage command value. The electric motor control device according to claim 1 or 2.   The value of the PWM carrier signal is read, and a point in time when the PWM carrier signal reaches a threshold set according to a current control calculation time is set as the calculation start timing. Electric motor control device.   The motor control according to claim 1, wherein the current control calculation time is set according to a current control calculation load determined based on a CPU performing motor control. apparatus.

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