JP2007324018A - Power control device for high-frequency dielectric heating, and its control method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a power control device for high-frequency dielectric heating without being influenced by the type of a magnetron, dispersion of characteristics or power voltage fluctuation. <P>SOLUTION: An input current and an input voltage of an inverter circuit converting a voltage of an A.C. power to high-frequency power by rectifying and switching it at a high frequency are detected; input current waveform information 90 and input voltage waveform information 94 are obtained by rectifying the detected current and the detected voltage; and a switching frequency control signal 92 obtained by combining the input current waveform information 90 and power control information 91 by further adding the input voltage waveform information 94 in non-oscillation of the magnetron is converted to drive signals of semiconductor switching elements 3 and 4 of the inverter circuit. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to power control for high-frequency dielectric heating using a magnetron, such as a microwave oven, and more particularly to high-frequency dielectric heating that is not affected by differences in the characteristics and types of magnetron characteristics and differences in magnetron anode temperature, etc. Is.

  Since the conventional power source mounted on the high-frequency heating device is heavy and large, it has been desired to reduce its size and weight. For this reason, downsizing, light weight, and cost reduction by switching power supply have been actively promoted in various fields. In particular, in a high-frequency heating apparatus that cooks food using microwaves generated by a magnetron, the power source for driving the magnetron is required to be smaller and lighter, and is realized by a switched inverter circuit.

  Among these, in particular, the high-frequency inverter circuit targeted by the present invention is of a bridge resonance type circuit system using one or two bridge arms composed of two switching elements connected in series (for example, , See Patent Document 1).

  The above switching is coupled with the fact that the magnetron is a non-linear load indicated by the VAK (anode-cathode voltage) -Ib characteristic in FIG. 21, and the current waveform of the commercial power supply supplied to the magnetron driving power supply has many harmonic components. The problem of becoming a waveform including is left.

  On the other hand, the absolute value of the harmonic component becomes higher as the power consumption of the magnetron driving power source for satisfying the demand for shortening the cooking time of the microwave oven is increased, making it difficult to suppress the harmonic current of the power source.

  Various control methods for suppressing the harmonic current have been proposed (see, for example, Patent Document 2).

  FIG. 20 shows an example of a magnetron driving power source (inverter power source) of the high-frequency heating device. DC power supply 1, leakage transformer 2, first semiconductor switching element 3, second semiconductor switching element 4, first capacitor 5 (snubber capacitor), second capacitor 6 (resonance capacitor), third capacitor 7 ( Smoothing capacitor), drive unit 13, full-wave voltage doubler rectifier circuit 11, and magnetron 12.

  The DC power supply 1 applies full-wave rectification to the commercial power supply and applies the DC voltage VDC to the series circuit of the second capacitor 6 and the primary winding 8 of the leakage transformer 2. The first semiconductor switching element 3 and the second semiconductor switching element 4 are connected in series, and the series circuit (resonance circuit) of the primary winding 8 and the second capacitor 6 of the leakage transformer 2 is the second semiconductor switching. The element 4 is connected in parallel.

  The first capacitor 5 is connected in parallel to the second semiconductor switching element 4 and has a snubber role for suppressing inrush current (voltage) generated during switching. The AC high voltage output generated in the secondary winding 9 of the leakage transformer 2 is converted into a DC high voltage by the voltage doubler rectifier circuit 11 and applied between the anode and cathode of the magnetron 12. The tertiary winding 10 of the leakage transformer 2 supplies current to the cathode of the magnetron 12.

The first semiconductor switching element 3 and the second semiconductor switching element 4 are composed of an IGBT and a flywheel diode connected in parallel thereto. Needless to say, the first and second semiconductor switching elements 3 and 4 are not limited to this type, and thyristors, GTO switching elements, and the like may be used.

  The drive unit 13 has an oscillating unit for generating drive signals for the first semiconductor switching element 3 and the second semiconductor switching element 4 therein, and a rectangular wave having a predetermined frequency is generated by the oscillating unit. The DRIVE signal is given to the semiconductor switching element 3 and the second semiconductor switching element 4. Immediately after one of the first semiconductor switching element 3 or the second semiconductor switching element 4 is turned off, the voltage across the other semiconductor switching element is high. If turned on at this point, a spike-like excessive current flows, which is unnecessary. Loss and noise occur. However, by providing a dead time, the turn-off is delayed until the voltage at both ends is reduced to about 0 V, so that the unnecessary loss and noise generation can be prevented. Of course, it works in the same way when switching in reverse.

  Since the DRIVE signal given from the driving unit 13 and the detailed operation of each semiconductor switching element in each operation mode are described in Patent Document 1, they are omitted.

  The circuit configuration of FIG. 20 is characterized in that the applied voltage to the first semiconductor switching element 3 and the second semiconductor switching element 4 is equivalent to the DC power supply voltage VDC even in Europe 240V, which is the highest voltage of a general household power supply. That is, 240√2 = 339V. Therefore, even when an abnormal time such as an induced lightning surge or a return from an instantaneous voltage stop is assumed, the first semiconductor switching element 3 and the second semiconductor switching element 4 can use inexpensive elements with a breakdown voltage of about 600V.

  Next, FIG. 23 shows resonance characteristics in this type of inverter power supply circuit (a series resonance circuit is constituted by an inductance L and a capacitance C).

  FIG. 23 is a diagram showing frequency-current characteristics when a constant voltage is applied. The horizontal axis corresponds to the switching frequency, and the vertical axis corresponds to the current flowing through the primary side of the leakage transformer.

  Since the impedance of the series resonance circuit is minimum at the resonance frequency f0 and increases as the distance from the resonance frequency increases, the current I1 becomes maximum at the resonance frequency f0 and the frequency range is increased to f1 to f3 as shown in the figure. As the current decreases, the current I1 decreases.

  In actual inverter operation, the frequency range (solid line part I1) of f1 to f3 higher than the resonance frequency f0 is used. Further, when the input power source is an alternating current such as a commercial power source, a nonlinear load of the magnetron as will be described later The switching frequency is changed according to the phase of the commercial power supply in accordance with the characteristics.

  Using the resonance characteristics of FIG. 23, the phase around 90 degrees and 270 degrees at which the instantaneous voltage of the commercial power supply, which is relatively not required for the boost ratio of the magnetron applied voltage to the commercial power supply voltage, is the highest at each high frequency output, The switching frequency is set to the highest.

  For example, when the microwave oven is used at 200 W, the frequency is near f3, lower than 500 W, and lower than 1000 W.

  Naturally, since the input power or the input current is controlled, this frequency changes due to changes in the commercial power supply voltage, the magnetron temperature, and the like.

Also, at the phase near 0 ° and 180 ° at which the instantaneous voltage of the commercial power supply is lowest, the switching frequency is lowered to the vicinity of the resonance frequency f0 in accordance with the characteristics of the magnetron that does not oscillate at high frequency unless a high voltage is applied. The boost ratio of the magnetron applied voltage to the voltage is increased, and the phase width of the commercial power source that emits radio waves from the magnetron is set to be wide.

  In this way, by changing the inverter operating frequency for each power supply phase, it is possible to realize a current waveform having a large number of fundamental wave components (commercial power supply frequency) and a small number of harmonic components.

  However, the non-linear characteristics of the magnetron vary depending on the type of magnetron, and also vary depending on the magnetron temperature and the object to be heated (load) in the microwave oven.

  FIG. 21 is a graph showing the anode-cathode applied voltage-anode current characteristics of the magnetron, where (a) shows the difference depending on the type of magnetron, (b) shows the difference due to the matching with the power supply of the magnetron, and (c) shows the temperature of the magnetron. The vertical axis represents the anode-cathode voltage and the horizontal axis represents the anode current in common to (a) to (c).

  Therefore, looking at (a), A, B, and C are characteristic diagrams of three types of magnetrons. In the case of magnetron A, the current flows only as little as IA1 until VAK becomes VAK1 (= ebm). . However, when VAK exceeds VAK1, the current IA starts to increase rapidly. In this region, IA changes greatly with a slight difference in VAK. Next, in the case of magnetron B, VAK2 (= ebm) is lower than VAK1, and in the case of magnetron C, VAK3 (= ebm) is lower than VAK2. As described above, the non-linear characteristics of the magnetron differ depending on the types A, B, and C of the magnetron. Therefore, in the case of a modulation waveform adapted to a magnetron having a low ebm, the input current waveform is distorted when a magnetron having a high ebm is used. . Conventional devices could not cope with these problems. Therefore, it has been a challenge to make a high-frequency dielectric heating circuit that is not affected by these types.

  Similarly, regarding (b), the characteristic diagrams of the three types of magnetrons show good and bad impedance matching of the heating chamber as seen from the magnetron. When the impedance matching is good, VAK1 (= ebm) is the maximum and becomes smaller as it gets worse. Thus, since this nonlinear characteristic of the magnetron is greatly different depending on whether the impedance matching is good or bad, it is a problem to produce a high-frequency dielectric heating circuit that is not affected by these types.

  Similarly, looking at (c), the characteristics of the three types of magnetrons show the temperature of the magnetron. When the temperature is low, VAK1 (= ebm) is maximum, and ebm gradually decreases as the temperature gradually increases. Therefore, when the temperature of the magnetron is adjusted to the lower side, the input current waveform may be distorted when the temperature of the magnetron is increased.

  As described above, the non-linear characteristics of the magnetron are greatly different depending on the temperature difference of the magnetron. Therefore, it is an object to produce a high-frequency dielectric heating circuit which is not affected by the kind of the magnetron.

In order to solve these problems, as shown in FIG. 22, by using a modulation waveform obtained by processing and shaping the commercial power supply voltage waveform detected by the power supply voltage detection means 27 by the modulation unit 21, There has been proposed a method of driving the semiconductor switching elements 3 and 4 by modulating the drive pulse frequency and performing waveform shaping by the “expected control method” so that the input current waveform approaches a sine wave.
JP 2000-58252 A JP 2004-6384 A

  However, even with such a “prospect control method”, variations in the types and characteristics of the magnetron described above, ebm (anode-cathode voltage) fluctuation due to magnetron anode temperature and load in the microwave oven, and power supply voltage fluctuation On the other hand, it was found that waveform shaping was not able to follow up.

  Furthermore, since the output voltage waveform of the smoothing circuit immediately before the start of the first on-operation of the first semiconductor switching element 3 becomes a direct current regardless of the phase of the commercial power supply, the modulation waveform obtained by processing and shaping the commercial power supply voltage waveform is adopted. Along with this, the commercial power supply phase at the start of the ON operation is controlled to the phase where the ON time width (1 / frequency) determined from this modulation waveform is the smallest, that is, around 90 degrees and 270 degrees, and an excessive voltage is applied to the magnetron. There is a problem that control adjustment for this is complicated.

  In addition, in order to realize the power supply current waveform shaping that follows the characteristic fluctuation of the magnetron described above, a waveform reference signal is created, and the drive pulse frequency of the semiconductor switching element is modulated and controlled so that the input current waveform follows this waveform. However, there is a problem that the circuit configuration is complicated and large.

  As is well known, since a magnetron is a kind of vacuum tube, there is a delay time (hereinafter abbreviated as start-up time) from when an electric current is supplied to the heater until the electromagnetic wave is oscillated and output. Although the startup time is shortened by increasing the heater current, the impedance between the anode and cathode of the magnetron is infinite within the startup time, so the voltage applied to both ends may become excessively high. There is a problem that measures to prevent this harmful effect are necessary.

  Therefore, the present invention is configured so that the applied voltage does not become excessive with respect to the withstand voltage of each part during non-oscillation within the magnetron start-up time, and the start-up time is shortened, while the magnetron type and its characteristics vary. Even if there is a change, ebm (anode-cathode voltage) fluctuations due to magnetron anode temperature and load in the microwave oven, and even power supply voltage fluctuations are not affected by them, improving operating efficiency An object of the present invention is to provide a high-frequency dielectric heating power control apparatus and a control method thereof.

  The present invention is a high-frequency dielectric heating power control apparatus that controls an inverter circuit that rectifies the voltage of an AC power supply, modulates the switching frequency of high-frequency switching of a semiconductor switching element, and converts it to high-frequency power. An input current detector that detects input current to the inverter circuit and outputs input current waveform information; and an input voltage that detects input voltage to the inverter circuit from the AC power source and outputs input voltage waveform information A detection switch; an oscillation detection unit that detects oscillation of the magnetron; and a changeover switch that causes the input voltage detection unit to output the input voltage waveform information during a period until the oscillation detection unit detects oscillation of the magnetron. And the input current waveform information and the input output during a period until the oscillation of the magnetron is detected. And a voltage waveform information, and a converter for converting the drive signal of the semiconductor switching elements of the inverter circuit.

Connected between the input current detection unit and the conversion unit, the input current waveform information, the input voltage waveform information output in a period until the oscillation of the magnetron is detected, and any location of the inverter circuit And a control circuit that mixes power control information for controlling the current or voltage at a predetermined value to generate a switching frequency control signal, and the converter suppresses the peak of the voltage applied to the magnetron. As described above, the switching frequency control signal may be converted into the drive signal.

  Preferably, the conversion unit includes a frequency limiting unit, and an upper limit or a lower limit is provided for the high-frequency switching frequency.

  The conversion unit may further include a duty control unit, and at least a range in which the high frequency switching frequency is limited to an upper limit of the frequency limiting unit may be configured to be supplemented by duty control that controls an on-duty of the high frequency switching.

  The mixing circuit mixes the input current waveform information, the input voltage waveform information, and power control information for controlling the output of the input current detection unit to a predetermined value, and generates a switching frequency control signal. You may comprise as follows. Further, the input current waveform information and the input voltage waveform information are directly input to the mix circuit, and the mix circuit adds and inverts the directly input input current waveform information and input voltage waveform information, and the power control You may make it the structure which mixes with information.

  The input current detection unit may include a current transformer that detects the input current and a rectifier circuit that rectifies and outputs the detected input current. Further, a comparison circuit that outputs the power control information in comparison with the input current and the output setting signal can be further provided.

  On the other hand, the input current detector may be configured to detect and output a unidirectional current after rectifying the input current of the inverter circuit. Here, the input current detection unit includes a shunt resistor that detects a unidirectional current after rectifying the input current of the inverter circuit, and an amplifier circuit that amplifies a voltage generated at both ends of the shunt resistor. Can be configured. Further, the output obtained by the amplifier circuit is directly input to the mix circuit as the input current waveform information, and the power control information is output by comparing the output obtained by the amplifier circuit with an output setting signal. May further be provided.

  The mix circuit may include a configuration that cuts a high frequency component of the power control information, and a configuration in which a circuit configuration is switched between when the input current is increased and when the input current is decreased. The mix circuit may be configured such that a time constant increases when the input current is increased, and a time constant decreases when the input current is decreased.

  Further, the mix circuit may be configured such that resonance circuit voltage control information for controlling the resonance circuit voltage of the resonance circuit to a predetermined value is input, and the circuit configuration is switched according to the magnitude of the resonance circuit voltage. . Here, the mix circuit may be configured such that the time constant increases when the resonance voltage is low and the time constant decreases when the resonance voltage is high.

  Furthermore, the input current detection unit can be provided with a filter circuit that attenuates high-frequency parts such as a high-order frequency part and a high-frequency switching frequency of a commercial power supply, and phase lead compensation can be added to the filter circuit. .

  The input voltage detection unit includes a pair of diodes that detect an input voltage from the AC power supply to the inverter circuit, and a shaping circuit that shapes and outputs the input voltage detected by the diode. Can be configured. The shaping circuit may have a configuration that attenuates a higher-order frequency portion of the input voltage, and may further include phase lead compensation.

  Further, the oscillation detection unit is configured by an oscillation detection circuit connected between the input current detection unit and the input voltage detection unit, and the changeover switch is provided between the oscillation detection circuit and the input voltage detection unit. It can also be provided at the connection point.

  In addition, the conversion unit may be configured by a frequency modulation circuit that generates a drive signal for the semiconductor switching element by superimposing a carrier wave whose frequency is set by the switching frequency control signal and a slice control signal.

Furthermore, the present invention provides a power control method for high-frequency dielectric heating that controls an inverter circuit that rectifies the voltage of an AC power supply, modulates a high-frequency switching frequency of a semiconductor switching element, and converts it into high-frequency power. Detecting an input current from the AC power source to the inverter circuit, obtaining input current waveform information corresponding to the input current, and detecting an input voltage from the AC power source to the inverter circuit; Obtaining the input voltage waveform information corresponding to the input voltage; detecting the magnetron oscillation; outputting the input voltage waveform information in a period until the magnetron oscillation is detected; , The input current waveform information and output in the period until the magnetron oscillation is detected And said input voltage waveform information, and converting the drive signal of the semiconductor switching elements of the inverter circuit,
Is provided.

  According to the present invention, the input current waveform information of the inverter circuit that rectifies the AC power supply voltage and converts it into alternating current of a predetermined frequency is converted into the drive signal of the semiconductor switching element of the inverter circuit so that the instantaneous fluctuation is suppressed. The For example, input current waveform information is converted into a frequency of an on / off drive signal of a semiconductor switching element of an inverter circuit by a frequency modulation method and used. Therefore, a control loop for correcting the input current is configured such that the switching frequency is increased in the portion where the input current is large and the switching frequency is decreased in the portion where the input current is small. Therefore, even if there are variations in the type of magnetron and its characteristics, ebm (anode-cathode voltage) fluctuations due to magnetron anode temperature and load in the microwave oven, and even power supply voltage fluctuations, these effects The input current waveform shaping that is not subject to this can be obtained with a simpler configuration, and the stable output of the magnetron can be achieved with a simple configuration.

  Further, since the input voltage waveform information is also input to the correction loop when the magnetron is not oscillating, there is an effect that the startup time of the magnetron is shortened.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.

(Embodiments 1 and 2)
FIG. 1 is a block diagram illustrating a high frequency heating apparatus according to Embodiment 1 of the present invention. In FIG. 1, the high-frequency heating device includes an inverter circuit 40, a control circuit 45 that controls the first and second semiconductor switching elements 3 and 4 of the inverter, and a magnetron 12. The inverter circuit 40 includes an AC power supply 50, a diode bridge rectifier circuit 60, a smoothing circuit 61, a resonance circuit 36, first and second semiconductor switching elements 3 and 4, and a voltage doubler rectifier circuit 11.

The AC voltage of the AC power supply 50 is rectified by a diode bridge type rectifier circuit 60 including four diodes 63, and converted to the DC power supply 1 through a smoothing circuit 61 including an inductor 64 and a third capacitor 7. Thereafter, the transformer 41 is converted into high-frequency alternating current by the resonance circuit 36 including the first capacitor 5, the second capacitor 6 and the primary winding 8 of the transformer 41, and the first and second semiconductor switching elements 3 and 4. A high-frequency and high-voltage is induced in the secondary winding 9.

  The high frequency and high voltage induced in the secondary winding 9 is applied between the anode 69 and the cathode 70 of the magnetron 12 through the voltage doubler rectifier circuit 11 including the capacitor 65, the diode 66, the capacitor 67, and the diode 68. Is done. Further, the transformer 41 has a tertiary winding 10, which heats the heater (cathode) 70 of the magnetron 12. The above is the inverter circuit 40.

  Next, the control circuit 45 that controls the first and second semiconductor switching elements 3 and 4 of the inverter circuit 40 will be described. First, a current detection unit including a CT (Current Transformer) 71 provided between the AC power supply 50 and the diode bridge type rectifier circuit 60 is connected to the rectifier circuit 72, and the CT 71 and the rectifier circuit. 72 constitutes an input current detector for detecting the input current to the inverter circuit. The input current to the inverter circuit is insulated and detected by CT 71, and its output is rectified by rectifier circuit 72, and input current waveform information 90 is generated.

  The current signal obtained by the rectifier circuit 72 is smoothed by the smoothing circuit 73, and the signal from the output setting unit 75 that outputs the output setting signal corresponding to the other heating output setting is compared by the comparison circuit 74. . Note that the comparison circuit 74 compares the input current signal smoothed by the smoothing circuit 73 with the setting signal from the output setting unit 75 in order to control the magnitude of power. Accordingly, the anode current signal of the magnetron 12 or the collector current signals of the first and second semiconductor switching elements 3 and 4 can be used as the input signal instead of the input current signal smoothed by the smoothing circuit 73. . That is, the comparison circuit 74 outputs the power control information 91 for controlling the output of the input current detection unit to be a predetermined value, but the comparison circuit 74 and the power control information 91 are not essential for the present invention as will be described later. .

  Similarly, the input current detection is performed by the current detection unit composed of the shunt resistor 86 provided between the diode bridge type rectifier circuit 60 and the smoothing circuit 61 of the second embodiment shown in FIG. The output may be used as the input current waveform information 90. The shunt resistor 86 detects the input current after being rectified in a single direction by the diode bridge type rectifier circuit 60.

  On the other hand, in the present embodiment, the control circuit 45 detects a voltage of the AC power supply 50 and a pair of diodes 46 that rectifies, and a shaping circuit 47 that shapes the rectified voltage and generates input voltage waveform information 49. The input voltage detection part which consists of these is also provided. Further, the control circuit 45 includes an oscillation detection circuit 48 that constitutes an oscillation detection unit that detects whether the current signal obtained by the rectifier circuit 72 is at a predetermined level and whether the magnetron is oscillating. The oscillation detection circuit 48 detects that the magnetron has started to oscillate according to the level of the current signal, and divides the state before detection into a non-oscillation state and the state after detection as an oscillation state. If it is determined that there is no oscillation, the oscillation detection circuit 48 turns on the changeover switch SW3 disposed between the shaping circuit 47 and the mix circuit 81. In other words, the changeover switch SW3 outputs the input voltage waveform information 49 to the input voltage detector during a period until the oscillation detection circuit 48 detects the oscillation of the magnetron 12. It should be noted that the magnetron repeats oscillation / non-oscillation in accordance with the cycle of the commercial power supply even after the magnetron oscillation starts. However, the non-oscillation, that is, non-oscillation after the oscillation starts here causes the changeover switch SW3 to Turning on has nothing to do with the present invention.

In this embodiment, the input current waveform information 90, the power control information 91 from the comparison circuit 74, and the input voltage waveform information 49 (when SW3 is ON) are also added and mixed and filtered by the mix circuit 81 to control the switching frequency. A signal 92 is output. The sawtooth wave 84 output from the sawtooth wave generating circuit 83 is frequency-modulated by the switching frequency control signal 92 as described later.

  The comparator 82 compares the sawtooth wave 84 and a slice control signal 87, which will be described later, to convert it into a rectangular wave, and the resulting rectangular wave is applied to the gates of the first and second semiconductor switching elements 3 and 4 via a driver. It is done. The input current waveform information detection system is configured such that the sawtooth wave from the sawtooth wave generation circuit 83 frequency-modulated by the switching frequency control signal 92 is compared with the comparator 82 and the semiconductor switching element of the inverter circuit is controlled to be turned on / off. It is simplified. In particular, the present embodiment employs a simplified configuration in which the input current waveform information 90 is directly input to the mix circuit 81.

  The portion that generates the drive signals for the first and second semiconductor switching elements 3 and 4 from the switching frequency control signal 92 described above has a high switching frequency in the portion where the input current from the AC power supply 50 is large, and the input current is high. The small portion may be configured as a conversion unit that converts the switching frequency control signal 92 into a drive signal for the semiconductor switching element of the inverter circuit so that the switching frequency is low, and the present invention is not limited to this configuration.

  In particular, in the present invention, the conversion unit drives the input current waveform information 90 and the input voltage waveform information 49 output during the period until the oscillation of the magnetron 12 is detected to drive the semiconductor switching elements 3 and 4 of the inverter circuit. Convert to signal.

  The on / off control of the semiconductor switching elements 3 and 4 with respect to the input current waveform information 90 is converted with a polarity such that the switching frequency is high when the input current is large and the switching frequency is low when the input current is small. Therefore, in order to obtain such a waveform, the input current waveform information is used after being inverted in the mix circuit 81 described later.

  FIG. 3 is a detailed circuit diagram of the sawtooth wave (carrier wave) generation circuit 83. The outputs of the comparators 164 and 165 are input to the S terminal and the R terminal of the SR flip-flop 166, respectively. Charging / discharging of the capacitor 163 is switched according to the output polarity of the non-Q terminal of the SR flip-flop 166, and is charged with the current I10 when Hi and discharged with the current I11 when Lo. The non-Q terminal of the SR flip-flop 166 is set by receiving the output Hi of the comparator 164 when the potential of the capacitor 163 exceeds V1, and is reset by receiving the output Hi of the comparator 165 when the potential falls below V2. Switch to Hi.

  With this configuration, the potential of the capacitor 163 becomes a sawtooth waveform (triangular wave), and this signal is conveyed to the comparator 82.

  The charging / discharging currents I10 and I11 to the capacitor 163 are determined by reflecting a current I12 obtained by dividing the potential difference between the voltage of the switching frequency control signal 92 and Vcc by the resistance value. The slope changes. Accordingly, the switching frequency is determined by the magnitudes of I10 and I11 reflecting the switching frequency control signal.

  FIG. 5A shows an example of the mix circuit 81. The mix circuit 81 has three input terminals. The power control information 91 is added to one of them, and the input voltage waveform information 49 is added to the other via the input current waveform information 90 and SW3. The input current waveform information 90 is input to the mix circuit 81 and inverted by the inverting circuit to become a correction signal.

Further, as shown in FIG. 5B, a high-frequency cut filter is configured between the power control information 91 and the output as shown by the AC equivalent circuit in the mix circuit 81. As a result, the high-frequency component included in the power control that has interfered with the input current waveform information 90 for shaping the input current waveform is cut by this filter.

  Further, as shown in FIG. 5C, a low-frequency cut filter is configured between the input current waveform information 90 and the input voltage waveform information 49 and the output as shown by an AC equivalent circuit in the mix circuit 81. . Therefore, the power control information 91 is converted into a DC component of the output of the mix circuit 81, and the input current waveform information 90 and the input voltage waveform information 49 are converted into an AC component.

  In the first embodiment, as described above, the input current waveform information 90 or a signal obtained by adding the input voltage waveform information 49 to the input current waveform information 90 when the magnetron is not oscillating is transmitted to the semiconductor switching elements 3 and 4 of the inverter circuit. It is used by converting to a switching frequency. In general, inverters used in microwave ovens are well known. 50 to 60 cycles of commercial AC power is rectified and converted to DC, and the converted DC power is converted to high frequency of, for example, about 20 to 50 KHz by the inverter. The converted high frequency is boosted by a step-up transformer, and a high voltage rectified by a voltage doubler rectifier circuit is applied to the magnetron.

  As a circuit system of the inverter, for example, switching is performed by using one semiconductor switching element that is often used in a commercial power supply such as a 100V area, and the output is changed by changing the on-time of the switching pulse. An on-time modulation method using a voltage resonance type circuit and two semiconductor switching elements 3 and 4 connected in series as shown in FIG. 1 of the present invention, FIG. There are two types, a (half) bridge type voltage resonance type circuit system in which the switching frequency is controlled to change the output. The bridge-type voltage resonance circuit system is a system that can be simply configured and controlled such that the output decreases when the switching frequency is increased, and the output increases when the switching frequency is decreased.

  FIG. 6 is a diagram for explaining a waveform obtained by the first embodiment of the present invention. This example is a situation when the magnetron is oscillating normally, that is, during normal operation. At this time, the oscillation detection circuit 48 determines that the magnetron is under normal operation from the current value obtained from the rectifier circuit 72, and turns off SW3. Therefore, during the actual operation, the diode 46 and the shaping circuit 47 do not act, and the input voltage waveform information 49 is not generated.

  6A shows a case where the input current is large, and FIG. 6B shows a case where the input current is small. As will be described later, the solid line represents the signal shape after correction by the power control apparatus of the present invention, which is mainly used in the following description, and the broken line represents the signal shape of the instantaneously changing output from the AC power supply 50 before correction. Represent.

  6A, the waveform of the input current waveform information from (a) from the top is the input current waveform information 90 which is the output of the rectifier circuit 72 in FIG. 1 and the output of the amplifier circuit 85 in FIG. The dotted line shows the waveform before correction resulting from the non-linear load characteristic of the magnetron. 6A is a switching frequency control signal 92 which is a correction output of the mix circuit 81. The switching frequency control signal 92 follows the input current waveform information 90 and the power control information 91 and has a magnitude. In addition, in order to complement and correct the distortion of the input current, it is output as an inverted waveform of (A).

  (C) in FIG. 6 (a) shows a sawtooth wave (carrier wave) and a slice control signal frequency-modulated by the switching frequency control signal shown in (b), and the first and second semiconductors shown in (2). A drive signal that is an on / off signal of the switching elements 3 and 4 is generated.

  These two drive signals have a complementary relationship of on / off.

  The comparator 82 inputs a frequency-modulated sawtooth wave (carrier wave) 84 and the slice control signal 87 and compares them, and the drive signals for the first and second semiconductor switching elements obtained are a sawtooth wave as shown in FIG. Similar frequency modulation is applied.

  That is, as shown in the figure, in the portion where the amplitude value of the switching frequency control signal is large (0 degree, near 180 degrees, where the input current is small), the frequency of the sawtooth wave is low. It is corrected. In addition, the portion where the amplitude value of the switching frequency control signal is small (90 °, 270 °, the portion where the input current is large) is corrected to a polarity that reduces the input current from the resonance characteristics described above because the frequency of the sawtooth wave is high. A pulse train having a frequency such as (d) is output as a drive signal for the semiconductor switching element. That is, since the switching frequency control signal (b) is inverted as the correction waveform with respect to the input current waveform information (b), the input current waveform information (b) has a large input (near 90 degrees and 270 degrees). In the part where the frequency is increased as in the pulse train signal of (d) and the input of the input current waveform information (a) is small (near zero crossing of 0 degree and 180 degrees), the frequency is lowered. This is converted to an inverted output. As a result, an effect of correcting the input waveform can be obtained, but this effect is particularly great in the vicinity of the zero cross.

  The lower waveform (e) indicates the switching frequency of the first and second semiconductor switching elements 3 and 4, and is a correction frequency switching frequency control signal (b) obtained by inverting the input current waveform information shown in (a). By frequency-modulating a high-frequency sawtooth wave and comparing the frequency-modulated sawtooth wave with the slice control signal, inverter conversion is performed to a high frequency such as 20 KHz to 50 KHz, and the drive signal (d) is generated. In response to the drive signal (e), the semiconductor switching elements 3 and 4 are turned on / off, and high-frequency power is input to the primary side of the step-up transformer to generate a high voltage boosted to the secondary side of the step-up transformer. (E) plots the frequency information on the Y-axis to visualize how the frequency of each pulse of this on / off signal (d) changes within the period of the commercial power supply. Each point is connected.

  The above description shows the same signal as that when the input current from the AC power supply 50 is obtained in an ideal state (for example, a sine wave). However, in general, the input current from the AC power supply 50 deviates from an ideal sine wave and fluctuates when viewed instantaneously. A broken line signal indicates such a realistic state. As indicated by the broken line, even in an instantaneous period of a half cycle (0 to 180 degrees) of the commercial power supply, the actual signal generally deviates from the ideal signal state, and instantaneous fluctuations are generally generated. Such a signal shape is generated by a transformer, a boosting action by a voltage doubler circuit, a smoothing characteristic of the voltage doubler circuit, and a magnetron characteristic in which an anode current flows only when the voltage is only ebm or higher. That is, it can be said that it is an unavoidable variation in an inverter circuit for a magnetron.

  In the power control apparatus of the present invention, the input current waveform information (see (A)) indicated by the broken line reflecting the fluctuation state of the input current is obtained from the input current detection unit, and the subsequent input current waveform information is based on this input current waveform information. Control is made. This control is performed so that the instantaneous fluctuation of the input current waveform information that appears in a period such as a half cycle is suppressed so as to approach an ideal signal as indicated by an arrow. This suppression is achieved by adjusting the drive signals of the first and second semiconductor switching elements 3 and 4. Specifically, when the input current waveform information 90 is smaller than the ideal signal, the above-described frequency becomes lower and correction for increasing the input current is performed. When the input current waveform information is larger than the ideal signal, the above-described frequency becomes higher and correction for reducing the input current is performed. Even in the case of instantaneous fluctuations for a shorter period, the changed waveform is reflected in the frequency information, and the same correction as described above is performed.

Due to the instantaneous fluctuation suppressing action of the first and second semiconductor switching elements 3 and 4 to which the drive signal is given, the input current waveform information 90 is corrected as indicated by an arrow, and an input close to the ideal wave is always a magnetron. Will be given. Illustrations of (B) and (C) after correction are omitted. Here, the ideal signal described above is a virtual signal, but this signal is a sine wave.

  That is, in a short period such as a half cycle of the commercial power supply, the instantaneous error or the sum of the correction amount of the ideal signal waveform and the input current waveform information is controlled (power control) by the other means. It is almost zero. In addition, since the portion where the input current does not flow due to the non-linear load is corrected in the flowing direction, the portion where the input current is large is decreased to establish the above substantially zero. This is to correct even the nonlinear load so that the current waveform can be regarded as a linear load.Since the commercial power supply voltage waveform is a sine wave, the ideal waveform is a sine wave. become.

  In this way, the input current is corrected with the reverse polarity of the waveform so as to cancel the change in the input current waveform and the excess or deficiency with respect to the ideal waveform. Therefore, a sudden current change in the commercial power supply cycle caused by the non-linear load of the magnetron, that is, distortion is canceled out by this control loop, and the input current waveform shaping is performed.

  Furthermore, since the control loop operates with the input current waveform information that follows the instantaneous value of the input current in this way, even if there are variations in the type and characteristics of the magnetron, the temperature of the anode of the magnetron and the microwave oven Even if there is an ebm (anode-cathode voltage) fluctuation due to the load and a power supply voltage fluctuation, the input current waveform can be shaped without being affected by them.

  In particular, in the present invention, the semiconductor switching element is controlled based on instantaneously varying input current waveform information. Instantaneous fluctuation of the input current is directly input to the mix circuit 81 in the form of input current waveform information and reflected in the switching frequency control signal 92. Therefore, the distortion of the input current waveform is suppressed and the followability to the instantaneous fluctuation is excellent. A driving signal for the semiconductor switching element can be obtained.

  The subject of the present invention is to convert input current waveform information having this information into a drive signal for a semiconductor switching element of an inverter circuit so that distortion and instantaneous fluctuation of the input current waveform are suppressed. The power control information 91 is not essential for achieving the purpose. The power control information 91 is information for controlling power fluctuations in a long period, that is, a period longer than the commercial power supply period, and corrects instantaneous fluctuations for a short period such as a half cycle of AC targeted by the present invention. It is not information. Therefore, the use of the mix circuit 81, the comparator 82, and the sawtooth wave generation circuit 83 is merely an example of the embodiment, and the one corresponding to the conversion unit that performs the conversion described above exists between the input current detection unit and the semiconductor switching element. do it.

  Even when the power control information is used, it is not essential to input the power control information 91 for controlling the output of the input current detection unit to a predetermined value as in the above-described embodiment. That is, in the above-described embodiment, the power control information 91 originates from the current detector 71 and the rectifier circuit 72 (FIG. 1) or the shunt resistor 86 and the amplifier circuit 85 (FIG. 2) that detect the input current. Information for controlling the current or voltage at an arbitrary position of the inverter circuit 40 to be a predetermined value can be input to the mix circuit 81 as power control information. For example, the resonance circuit voltage information 42 of the resonance circuit 62 as shown in FIGS. 1 and 2 is used as it is, or the information after the smoothing by the smoothing circuit 73 and the comparison with the output setting signal in the comparison circuit 74 is power controlled. Can be used as information.

Next, FIG. 6B shows a comparison with FIG. 6A when the input current is small, and FIG. 6G shows the input current waveform information when the input is small. ) Corresponds to (a), (H) is a switching frequency control signal, (Li) is a switching frequency of the semiconductor switching element, and corresponds to (B) and (E) in FIG. . Although not shown in the drawing, the sawtooth wave comparison process shown in FIGS. 6A and 6D is of course the same process.

  The above description based on FIG. 6 is related to the normal operation of the magnetron. Next, the operation when the magnetron is activated will be described. The time of start-up refers to a state of a preparatory stage before the oscillation is started although a voltage is applied to the magnetron but no oscillation is generated.

  When the magnetron is started (corresponding to non-oscillation), the impedance between the anode and the cathode becomes equal to infinity, unlike during steady operation. Then, since the difference between the steady operation and the start-up affects the state of the input current via the transformer 41, the oscillation detection circuit 48 starts the magnetron from the current value obtained from the rectifier circuit 72. It can be determined whether or not it is sometimes. If it is determined that it is at startup, the oscillation detection circuit 48 turns on SW3. Therefore, at the time of start-up, the diode 46 and the shaping circuit 47 act and the input voltage waveform information 49 is generated.

  By the way, in the present invention, the voltage from the commercial AC power supply 50 is multiplied by the power control by the switching frequency control system, that is, the commercial power supply voltage is amplitude-modulated by the power control by the switching frequency modulation system, Applied to the side. The peak value of the applied voltage on the primary side is related to the applied voltage to the magnetron 12, and the area defined by the applied voltage and the elapsed time is related to the power supplied to the heater.

  In the present invention, when the input current waveform information 90 is small, the input voltage waveform information 49 is input to the mix circuit 81 via the changeover switch SW3. In other words, a format is adopted in which the input voltage compensates for the shortage of the input current as a reference signal particularly at the time of startup.

  FIG. 7 is a diagram for comparing and explaining the operation when the input voltage waveform information is added and when the input voltage waveform information is not added. FIG. 7A shows the switching frequency when the input voltage waveform information is not added in order from the top. The waveforms of the control signal, switching frequency, transformer primary applied voltage, magnetron applied voltage, and heater input power are shown.

  FIG. 7B illustrates the operation when the input voltage waveform information is added (when activated). FIGS. 7A and 7B both show a case where the peak value of the applied voltage on the primary side of the transformer is limited by the configuration of the seventh embodiment described later. Further, in FIG. 7B, the peak of the transformer primary application voltage and the magnetron application voltage is suppressed by the action of the added input voltage waveform information, and the waveform shows a trapezoid. 7B also shows the waveforms of the switching frequency control signal, the switching frequency, the transformer primary application voltage, the magnetron application voltage, and the heater input power from the top as in FIG. 6A.

  As shown in FIG. 7, since the switching frequency of the semiconductor switching element is low in the vicinity of the phase of 0 degree and 180 degrees, the transformer primary side applied voltage and the magnetron applied voltage have a relatively large amplification width. On the other hand, in the vicinity of the phase of 90 degrees and 270 degrees, the switching frequency of the semiconductor switching element is high, so that the amplification width is relatively suppressed. From the relative relationship with the amplification width at the phase of 0 degree and 180 degrees, the entire waveform diagram Becomes a trapezoid and shows a shape in which the peak is suppressed.

Comparing the magnetron applied voltages in FIG. 7A and FIG. 7B, when the magnetron applied voltage is the same, the waveform area indicating the heater input power is larger in FIG. 7B. That is, since the heater input power is increased in FIG. 7B, the heater is heated in a short time, and the startup time can be shortened.

  The oscillation detection circuit in this case uses, for example, the characteristic that the input current increases when the magnetron starts oscillating, and the output of the input current detection unit is compared with the oscillation detection threshold level by a comparator or the like, and the output is latched. There is. The detected value is output to SW3.

  FIG. 8 is a diagram showing an example of an adding / inverting circuit for adding the input current waveform information and the input voltage waveform information used in the first embodiment of the present invention. This adding / inverting circuit is provided in the mix circuit 81 as shown in FIGS.

  The input current waveform information 90 and the input voltage waveform information 49 are each input to a buffer transistor, and the output is input to two transistors having a common collector resistance. The buffer transistor is provided to prevent interference between the input current waveform information 90 and the input voltage waveform information 49. A current (emitter current) corresponding to the magnitude of the input signal flows through the emitter resistance of each of the two transistors, and a voltage drop occurs in the common collector resistance according to the value obtained by adding the respective emitter currents.

  Here, as the emitter voltage increases, the current increases and the voltage drop increases, that is, the collector voltage decreases, so that the collector voltage is inverted in polarity with respect to the input signal. Also, the signal conversion coefficient varies depending on the resistance value ratio between the collector resistance and the emitter resistance. In terms of interference with the power control signal, it is more effective to convert the impedance of the signal at the common collector connection point through a buffer and connect it to the subsequent capacitor. Thus, this circuit adds the two signals, inverts them, and outputs them.

(Embodiment 2)
In the second embodiment of the present invention, the input current waveform information and a signal to which the input voltage waveform information is further added when the magnetron is not oscillating are mixed with the power control information from the comparison circuit 74 and filtered to perform an inverter circuit. This is a configuration of a control circuit (conversion unit) in which the semiconductor switching elements 3 and 4 are used after being converted into on / off drive signals.

  According to this configuration, it is not necessary to process commercial power supply voltage waveform information in accordance with the non-linear load characteristic of the magnetron, the frequency modulation signal generation circuit is simplified, and simplification and miniaturization can be achieved. Further, the input voltage waveform information 49 is added to the input current waveform information 90 with a simple configuration to increase the heater power at the time of start-up and shorten the start-up time, so that an excessive voltage between the anode 69 and the cathode 70 of the magnetron is obtained. Since the safety measures to prevent the application are added, the reliability of the product is improved.

  Further, by configuring as described above, the control loop using the input current waveform information 90 is specialized for waveform shaping of the input current, and the control loop using the power control information 91 is specialized for power control. The mutual control itself maintains the conversion efficiency as a configuration that does not interfere with each other.

(Embodiment 3)
The third embodiment relates to an input current detection unit. As shown in FIG. 1, the input current detection unit detects an input current of an inverter circuit with a CT 71 or the like, and rectifies and outputs it from a rectification circuit 72. It is comprised as follows. In this configuration, since the input current is detected using CT or the like, a large signal can be extracted while maintaining insulation, so that the effect of shaping the input current waveform is great and the quality of the input current is improved.

In the example shown in FIG. 2, the input current detection unit detects a unidirectional current rectified by the rectifier circuit 60 of the inverter circuit via a shunt resistor 86 disposed between the rectifier circuit 60 and the smoothing circuit 61. The voltage generated at both ends is amplified by an amplifier circuit (amplifier) 85 and output. This configuration has an advantage that the input current detection unit can be configured at low cost because the detection unit does not need to be insulated from the electronic circuit and does not need to be rectified.

  Further, the amplifier circuit 85 of the input current detection unit shown in FIG. 2 is configured to attenuate high-frequency portions such as high-order frequency portions of commercial power supplies and high-frequency switching frequencies, thereby preventing unnecessary resonance. Specifically, as shown in the detailed diagram of the input current detection unit in FIG. 4, as shown in FIG. 4A, the amplifier circuit 85 uses a high-frequency cut capacitor, and a high-order frequency portion of the commercial power supply. In addition, high-frequency portions such as a high-frequency switching frequency are attenuated.

  Further, by inserting a high-frequency cut capacitor of the amplifier circuit 85, a phase delay compensation is added by inserting a resistor in series with the capacitor to generate a phase delay as shown in the phase characteristic diagram of FIG. 4B. This prevents transient time delays and ensures the stability of the control loop. The rectifier circuit 72 of FIG. 1 can also use a configuration that attenuates the high-frequency portion or a configuration that adds phase lead compensation to prevent a transient time delay. As shown in FIG. 9, the same configuration can be used in the shaping circuit 47 of the input voltage waveform information 49 creation unit.

(Embodiment 4)
The fourth embodiment relates to the mix circuit 81 shown in FIGS. 1 and 2, and receives input current waveform information 90, input voltage waveform information 49, and power control information 91 as shown in FIG. 5A. Three terminals are provided. With this configuration, the heater input power is compensated, and the startup time can be shortened.

  The input current waveform information 90 and the input voltage waveform information 49 (when SW3 is on) are input to an addition / inversion circuit as shown in FIG. 8, and an addition inversion process is performed. The processed signal and the power control information 91 are input to a filter circuit composed of C, R1, and R2, respectively, and after filtering, are output as a switching frequency control signal 92 to the sawtooth wave generation circuit. The filter circuit is configured to cut the high frequency component of the power control output 91 as shown in the equivalent circuit diagram of FIG. With this configuration, the high frequency component that hinders the shaping of the input current waveform is cut, so that the quality of the input current waveform is improved. On the other hand, as shown in the equivalent circuit diagram of FIG. 5C, a low-frequency cut filter is configured for the input current waveform information 90 and the input voltage waveform information 49 to perform waveform maintenance.

(Embodiment 5)
As shown in the configuration diagram of the mix circuit according to the fifth embodiment of FIG. 10, the fifth embodiment of the present invention includes the input current waveform information of the input current detector, the input voltage waveform information of the input voltage detector, and the The characteristic of the mix circuit that synthesizes the power control information for controlling the output of the input current detection unit to be a predetermined value is controlled with a difference between the input current increase control and the decrease control.

  In the configuration diagram of FIG. 10 (a), SW1 is turned on / off by the power control information 91, and the switching frequency control signal 92 is lowered / increased. At the time of input current increase control, SW1 is turned off and the switching frequency control signal is gradually increased with the time constant of C * R2 as shown in the equivalent circuit of FIG. It is controlled to become.

At the time of input current reduction control, SW1 is turned on and the switching frequency control signal is rapidly lowered with a time constant of C * {R1 * R2 / (R1 + R2)} as shown in the equivalent circuit of FIG. Thus, the switching frequency of the semiconductor switching element is controlled to be high. That is, the circuit configuration of the mix circuit 81 is switched between the input current increase control and the input current decrease control. In particular, the time constant is set large during input current increase control, and the time constant is set small during input current decrease control.

  By providing a difference in this way, a control characteristic that normally responds gently and a control characteristic that reduces the input current with a quick response when the input current rises transiently for some reason to prevent component destruction, etc. it can. In addition, the stability of the control characteristics against the non-linear load of the magnetron can be ensured.

(Embodiment 6)
In the sixth embodiment of the present invention, as shown in the configuration diagram of the mix circuit according to the sixth embodiment in FIG. 11, the resonance voltage control information 93 for controlling the resonance circuit voltage information 26 of the resonance circuit to a predetermined value, Input to the mix circuit 81.

  As shown in FIG. 11, the SW2 is on / off controlled by the resonance voltage control information 93 that compares the resonance voltage of the resonance circuit with the reference value. When the resonant voltage is low, SW2 is turned off and the switching frequency control signal is gradually increased with a time constant of C * R2 to control the switching frequency of the semiconductor switching element to be low. If the resonant voltage is high, turn on SW2 and control the C * {R2 * R3 / (R2 + R3)} time constant to rapidly lower the switching frequency control signal to increase the switching frequency of the semiconductor switching element. Yes. That is, the circuit configuration of the mix circuit 81 is switched according to the magnitude of the resonance voltage of the resonance circuit. In particular, the time constant increases when the resonance voltage is low, and decreases when the resonance voltage is high.

  FIG. 12 shows a time-series chart relating to magnetron oscillation detection, and also shows changes in the anode current and the resonance voltage of the resonance circuit accompanying changes in the input current. Before the oscillation of the magnetron 12, the secondary side impedance of the transformer 41 is very large, that is, the impedance between the anode and the cathode of the magnetron is infinite. Therefore, almost no power is consumed by the secondary load of the transformer, and the resonance voltage control information 93 reflecting the resonance voltage of the resonance circuit is controlled (restricted) to a predetermined value. Therefore, the input current to the oscillation detection circuit 48 is Small (Iin1 in FIG. 12).

  On the other hand, after the start of oscillation of the magnetron 12, the impedance between the anode and the cathode of the magnetron becomes small, and the secondary side impedance of the transformer also becomes small. Accordingly, since such a heavy load (magnetron) is driven while the resonance voltage of the resonance circuit is controlled (limited) to a predetermined value, the input current to the oscillation detection circuit 48 becomes larger than before the oscillation starts. (Iin2 in FIG. 12).

  The oscillation detection threshold level of the oscillation detection circuit 48 described above is preset between Iin1 and Iin2 as shown in FIG. That is, it is determined that the input current clearly differs between before and after the start of oscillation while the resonance voltage of the resonance circuit is maintained at a constant level. In the illustrated example, after the input current to the oscillation detection circuit 48 starts increasing with the increase of the anode current, the time required to reach the threshold level is t1, and then the oscillation detection circuit 48 determines the start of oscillation. The time required for is t2. At this time, even if the oscillation starts for the time t3 = t1 + t2, the resonance voltage control of the resonance circuit functions until it is determined that the oscillation starts.

This control is effective in preventing application of an excessive voltage to the magnetron when the magnetron is not oscillating, that is, when the power control is not functioning. In addition, after the magnetron oscillation starts, it is preferable to set the reference value to be compared with the resonance voltage of the resonance circuit larger than before the magnetron oscillation starts in order to disable this control and not affect the power control. .

(Embodiment 7)
In the seventh embodiment of the present invention, the switching frequency is limited as shown in the block diagram of the switching frequency limiting circuit according to the seventh embodiment in FIG.

  The frequency modulation signal 94 input to the sawtooth wave generation circuit 83 is transmitted through the first limit circuit 95 that depends on the fixed voltage V1 to the switching frequency control signal 92 and the first limit circuit 96 that depends on the fixed voltage V2. Created subject to potential and maximum potential limitations.

  This potential limitation is based on the relationship between the switching frequency control signal 92 and the switching frequency, and the former is limited to the maximum switching frequency and the latter is limited to the minimum switching frequency.

  The first limiting circuit 95 limits the maximum frequency to prevent an increase in switching loss of the semiconductor switching elements 3 and 4 when the switching frequency is increased.

  The resonance circuit 62 causes abnormal resonance when the switching frequency approaches the resonance frequency and causes destruction of the semiconductor switching element. The second limiting circuit 96 has a function of limiting the minimum frequency to prevent this phenomenon. .

(Embodiment 8)
In the eighth embodiment of the present invention, as shown in the block diagram of the slice control signal generation circuit according to the eighth embodiment in FIG. Complement with power control by duty control.

  FIG. 14 is a diagram showing the relationship between the on-duty of the first transistor 3 and the high-frequency power in the bridge resonance type inverter. The peak occurs when the on-duty is 50%, and the high-frequency power decreases as the duty decreases or increases below 50%.

  Further, since the on-duty of the second transistor is in a complementary relationship with the on-duty of the first transistor, the X-axis numerical values 0 and 100 in FIG.

  In order to reduce the high-frequency output, that is, to reduce the input current, the switching frequency control signal 92 is changed in the direction of increasing the switching frequency as described above, but the frequency modulation signal 94 limits the frequency by the first limiting circuit 95. This power control will not function during the period of reception. The slice control signal generation circuit 97 is configured to receive the same fixed voltage V1 and the switching frequency control signal 92 as the first limit circuit 95, and to pass the current I20 and change the slice control signal 87 during the period.

  In FIG. 15, the potential of the switching frequency control signal 92 is taken on the X axis, and various signals affected by this signal are taken on the Y axis. (A) shows the switching frequency and the frequency modulation signal 94, and the maximum frequency is limited below the voltage V1, and the minimum frequency is limited above V2. (B) shows that the slice control signal 87 changes within the range of the voltage V1 or less. (C) and (D) show the on-duty of the first and second semiconductor switching elements 3 and 4 that change in response to the slice control signal 87 as will be described later.

FIG. 16 is a visualization of the duty changes of (c) and (d) of FIG. 15. The first and second values derived from this signal and the sawtooth wave 84 through the comparator 82 follow the change of the slice control signal 87. The on-duty of the second semiconductor switching elements 3 and 4 changes.

  Since the slice control signal 87 does not change during the period when the frequency is not limited by the first limiting circuit 95 described above, the on-duty is kept in the vicinity of 50%. The range where the control does not function is complemented by reducing the on-duty to lower the high-frequency power.

  In order for the complementation to be established, the change start point of the slice control signal 87 with respect to the potential of the switching frequency control signal 92 only needs to include V1 at which power control by frequency modulation does not function, and is limited to V1. It is not a thing.

  Although a new reference potential is required, if it is configured to change from a potential higher than V1, the ratio of the high switching frequency decreases, so that the switching loss of the semiconductor switching element can be reduced.

(Embodiment 9)
The ninth embodiment of the present invention relates to a resonance circuit. As shown in the configuration diagram of FIG. 17, the resonance circuit 98 includes a first capacitor 5, a second capacitor 6, and a primary winding 8 of a transformer 41. In the resonant circuit 36, the first capacitor 5 is omitted.

  In this embodiment as well, the power supply harmonic current can be suppressed by converting the input current waveform information into the switching frequency control signal and modulating the switching frequency of the semiconductor switching element of the inverter circuit, as in the previous embodiment. become.

(Embodiment 10)
Embodiment 10 of the present invention relates to the configuration of an inverter circuit. As shown in FIG. 18, first and second series circuits comprising two semiconductor switching elements in a DC power source obtained by rectifying a commercial power source. 99, 100 are connected in parallel, and one end of the resonance circuit 98 to which the primary winding 8 of the transformer 41 and the second capacitor 6 are connected is at the midpoint of one series circuit, and the other end is at the other series circuit. It is configured to connect to the midpoint.

  Also in this embodiment, the power supply harmonic current can be suppressed by converting the input current waveform information into the switching frequency control signal and modulating the switching frequency of the semiconductor switching element of the inverter circuit, as in the previous embodiment. become.

(Embodiment 11)
The eleventh embodiment of the present invention relates to the configuration of an inverter circuit. As shown in FIG. 19, a first series circuit 99 composed of two semiconductor switching elements is connected in parallel to a DC power source obtained by rectifying a commercial power source. In the AC equivalent circuit, the resonance circuit 98 in which the primary winding 8 of the transformer 41 and the second capacitor 6 are connected is connected to the middle point of the first series circuit 99 and the other end is connected to the DC power source 101. It is configured to connect to one end.

  Also in this embodiment, as in the first embodiment, the input current waveform information is converted into a switching frequency control signal, and the switching frequency of the semiconductor switching element of the inverter circuit is modulated, so that Suppression becomes possible.

  Although various embodiments of the present invention have been described above, the present invention is not limited to the matters shown in the above-described embodiments, and those skilled in the art can make modifications and applications based on the description and well-known techniques. This is also the scope of the present invention, and is included in the scope for which protection is sought.

  According to the power control for high frequency dielectric heating of the present invention, a control loop for correcting the input current by inverting so that the portion where the input current is large is small and the small portion is large is configured. Therefore, even if there are variations due to magnetron types and characteristics, anode-cathode voltage fluctuations, power supply voltage fluctuations, etc., input current waveform shaping that is not affected by these effects can be obtained with a simpler configuration. Can be achieved with a simple configuration. In addition, since the input voltage waveform information is also input to the correction loop when the magnetron is not oscillating, the magnetron start-up time is shortened.

Configuration diagram of power control apparatus for high frequency dielectric heating according to Embodiment 1 of the present invention Configuration diagram of power control apparatus for high frequency dielectric heating having input current detection unit according to embodiment 3 of the present invention Circuit diagram showing details of sawtooth wave generating circuit shown in FIG. Detailed view of the input current detector shown in FIG. Circuit diagram of a mix circuit according to Embodiment 4 of the present invention The figure which shows the basic waveform of each part of the power control apparatus for high frequency dielectric heating shown in FIG. Waveform diagram of each part at the time of input voltage waveform information addition of the power control device for high frequency dielectric heating shown in FIG. The figure which shows an example of the addition / inversion circuit shown in FIG. Detailed circuit diagram of the shaping circuit shown in FIG. Configuration diagram of a mix circuit according to a fifth embodiment of the present invention Configuration diagram of a mix circuit according to Embodiment 6 of the present invention Time-series chart of magnetron oscillation detection Configuration diagram of switching frequency limiting circuit and slice control signal generating circuit according to Embodiments 7 and 8 of the present invention Power control special drawing related to the eighth embodiment of the present invention The figure which shows the relationship of the various signals in connection with Embodiment 7, 8 of this invention A diagram of the change of some of the signals shown in FIG. Configuration diagram of power control apparatus for high frequency dielectric heating according to Embodiment 9 of the present invention Configuration diagram of power control apparatus for high frequency dielectric heating according to Embodiment 10 of the present invention Configuration diagram of power control apparatus for high frequency dielectric heating according to Embodiment 11 of the present invention Configuration diagram of a conventional high-frequency heating device Magnetron anode-cathode applied voltage vs. anode current characteristics Configuration diagram of a conventional power control device for high frequency dielectric heating Resonant circuit characteristic diagram of power control device for high frequency dielectric heating

Explanation of symbols

DESCRIPTION OF SYMBOLS 3 1st semiconductor switching element 4 2nd semiconductor switching element 5 1st capacitor | condenser 6 2nd capacitor | condenser 7 3rd capacitor | condenser 8 Primary winding 9 Secondary winding 10 Tertiary winding 11 Voltage doubler rectifier circuit 12 Magnetron 40 Inverter circuit 41 Transformer 42 Resonance circuit voltage information 45 Control circuit 46 Diode (input voltage detector)
47 Shaping circuit (input voltage detector)
48 Oscillation Detection Circuit 49 Input Voltage Waveform Information 50 AC Power Supply 51 DC Power Supply 60 Diode Bridge Type Rectifier Circuit 61 Smoothing Circuit 62 Resonance Circuit 63 Diode 64 Inductor 65 Capacitor 66 Diode 67 Capacitor 68 Diode 69 Anode 70 Cathode 71 Current Detection Unit (Input Current) Detection unit)
72 Rectifier circuit (input current detector)
73 Smoothing circuit 74 Comparison circuit 75 Output setting unit 81 Mix circuit (conversion unit)
82 Comparator (conversion unit)
83 Sawtooth wave generation circuit (conversion unit)
84 Sawtooth wave 85 Amplifier circuit 86 Shunt resistor 87 Slice control signal 90 Input current waveform information 91 Power control information 92 Switching frequency control signal 93 Resonance voltage control information 94 Frequency modulation signal 95 First limit circuit 96 Second limit circuit 97 Slice Control signal generation circuit 98 Resonance circuit 99 First series circuit 100 Second series circuit 163 Capacitor 164 Comparator 165 Comparator 166 SR flip-flop

Claims (25)

A DC circuit obtained by rectifying the voltage of the AC power supply is connected to a series circuit consisting of two semiconductor switching elements, a resonant circuit, and a leakage transformer, and the switching frequency of the semiconductor switching element is modulated and converted to high-frequency power. A high-frequency dielectric heating power control device for controlling an inverter circuit for driving a magnetron for applying an output generated on a secondary side of a leakage transformer to a magnetron and energizing the magnetron;
An input current detection unit that detects an input current from the AC power source to the inverter circuit and outputs input current waveform information;
An input voltage detector that detects an input voltage to the inverter circuit and outputs input voltage waveform information;
An oscillation detector for detecting the oscillation of the magnetron;
In a period until the oscillation detection unit detects the oscillation of the magnetron, the changeover switch for outputting the input voltage waveform information to the input voltage detection unit,
A converter that converts the input current waveform information and the input voltage waveform information output in a period until the oscillation of the magnetron is detected into a drive signal for a semiconductor switching element of the inverter circuit;
A power control apparatus for high frequency dielectric heating comprising: A series circuit composed of two sets of two semiconductor switching elements is connected in parallel to a DC power source obtained by rectifying the voltage of the AC power source, and a primary winding and a capacitor of a leakage transformer between the midpoints of the series circuit Of the rectifying means connected to the secondary winding of the leakage transformer. A power control device for high-frequency dielectric heating that controls an inverter circuit for driving a magnetron that applies an output to a magnetron to energize it,
An input current detection unit that detects an input current from the AC power source to the inverter circuit and outputs input current waveform information;
An input voltage detector that detects an input voltage to the inverter circuit and outputs input voltage waveform information;
An oscillation detector for detecting the oscillation of the magnetron;
In a period until the oscillation detection unit detects the oscillation of the magnetron, the changeover switch for outputting the input voltage waveform information to the input voltage detection unit,
A converter that converts the input current waveform information and the input voltage waveform information output in a period until the oscillation of the magnetron is detected into a drive signal for a semiconductor switching element of the inverter circuit;
A power control apparatus for high frequency dielectric heating comprising: A series circuit composed of two semiconductor switching elements is connected in parallel to a DC power source obtained by rectifying the voltage of the AC power source, and one end of a resonant circuit in which a primary winding of a leakage transformer and a capacitor are connected to the semiconductor switching circuit. The other end of the element is connected to the DC power source in an AC equivalent circuit, the high frequency switching frequency of the semiconductor switching element is modulated and converted into high frequency power, and connected to the secondary winding of the leakage transformer. A power control device for high-frequency dielectric heating that controls an inverter circuit for driving a magnetron that applies an output of a rectifying means to a magnetron to energize the magnetron,
An input current detection unit that detects an input current from the AC power source to the inverter circuit and outputs input current waveform information;
An input voltage detector that detects an input voltage to the inverter circuit and outputs input voltage waveform information;
An oscillation detector for detecting the oscillation of the magnetron;
In a period until the oscillation detection unit detects the oscillation of the magnetron, the changeover switch for outputting the input voltage waveform information to the input voltage detection unit,
A converter that converts the input current waveform information and the input voltage waveform information output in a period until the oscillation of the magnetron is detected into a drive signal for a semiconductor switching element of the inverter circuit;
A power control apparatus for high frequency dielectric heating comprising: 4. The power control device for high frequency dielectric heating according to claim 1, wherein the power control device is connected between the input current detection unit and the conversion unit, and is a period until the input current waveform information and oscillation of the magnetron are detected. A mix circuit that mixes the input voltage waveform information output in step 1 and power control information for controlling current or voltage at an arbitrary position of the inverter circuit to a predetermined value, and generates a switching frequency control signal; And a high-frequency dielectric heating power control device that converts the switching frequency control signal into the drive signal so that a peak of a voltage applied to the magnetron is suppressed. 5. The high frequency dielectric heating power control apparatus according to claim 4, wherein the conversion unit includes a frequency limiting unit that sets an upper limit or a lower limit to the high frequency switching frequency. 6. The power control apparatus for high frequency dielectric heating according to claim 5, wherein the conversion unit further includes a duty control unit that controls an on-duty of the high frequency switching, and at least the high frequency switching frequency is limited to an upper limit of the frequency limiting unit. The power control device for high-frequency dielectric heating, in which the range to be performed is set with the operation range of the duty control unit so as to be supplemented by duty control. 5. The power control apparatus for high frequency dielectric heating according to claim 4, wherein the mix circuit controls the input current waveform information, the input voltage waveform information, and an output of the input current detection unit to a predetermined value. A power control device for high frequency dielectric heating that mixes power control information to generate a switching frequency control signal. 5. The power control apparatus for high-frequency dielectric heating according to claim 4, wherein the input current waveform information and the input voltage waveform information are directly input to the mix circuit, and the mix circuit directly receives input current waveform information and A power control device for high frequency dielectric heating that adds and inverts input voltage waveform information and mixes it with the power control information. 4. The power control apparatus for high-frequency dielectric heating according to claim 1, wherein the input current detector includes a current transformer that detects the input current, and a rectifier circuit that rectifies and outputs the detected input current. A power control device for high frequency dielectric heating. 8. The power control apparatus for high frequency induction heating according to claim 7, further comprising a comparison circuit that outputs the power control information in comparison with the input current and an output setting signal. 8. The high frequency dielectric heating power control apparatus according to claim 7, wherein the input current detection unit is configured to detect and output a unidirectional current after rectifying the input current of the inverter circuit. Power control device for heating. 12. The power control device for high frequency dielectric heating according to claim 11, wherein the input current detection unit detects a unidirectional current after rectifying the input current of the inverter circuit, and both ends of the shunt resistor. The output obtained by the amplifier circuit is directly input to the mix circuit as the input current waveform information, and the output obtained by the amplifier circuit is compared with the output setting signal. A high frequency induction heating power control apparatus further comprising a comparison circuit that outputs the power control information. 5. The power control apparatus for high frequency dielectric heating according to claim 4, wherein the mix circuit has a configuration for cutting a high frequency component of the power control information. 5. The high frequency dielectric heating power control device according to claim 4, wherein the mix circuit switches a circuit configuration between the input current increase control and the input current decrease control. 6. apparatus. 15. The power control apparatus for high frequency dielectric heating according to claim 14, wherein the mix circuit has a time constant that increases during the increase control of the input current and decreases during a decrease control of the input current. Power control device. 5. The power control apparatus for high-frequency dielectric heating according to claim 4, wherein resonance circuit voltage control information for controlling the resonance circuit voltage to a predetermined value is input to the mix circuit, and depending on the magnitude of the resonance circuit voltage. High-frequency dielectric heating power control device that switches the circuit configuration. 17. The power control apparatus for high-frequency dielectric heating according to claim 16, wherein the mix circuit increases a time constant when the resonant circuit voltage is low and decreases a time constant when the resonant circuit voltage is high. Power control device for high frequency dielectric heating. 4. The high frequency dielectric heating power control apparatus according to claim 1, wherein the input current detection unit includes a filter circuit that attenuates a high frequency part such as a high order frequency part and a high frequency switching frequency of an AC power source. Power control device. 19. The power control apparatus for high frequency dielectric heating according to claim 18, wherein the input current detection unit is configured by adding phase lead compensation to the filter circuit. 4. The power control apparatus for high frequency dielectric heating according to claim 1, wherein the input voltage detection unit detects a set of diodes that detect an input voltage from the AC power source to the inverter circuit, and the diodes. A power control device for high-frequency dielectric heating, comprising: a shaping circuit that shapes and outputs a waveform of the input voltage. 20. The power control apparatus for high frequency dielectric heating according to claim 19, wherein the shaping circuit has a configuration for attenuating a higher order frequency portion of the input voltage. 21. The power control apparatus for high frequency dielectric heating according to claim 20, wherein the shaping circuit further includes phase lead compensation. 4. The power control apparatus for high frequency dielectric heating according to claim 1, wherein the oscillation detection unit includes an oscillation detection circuit connected between the input current detection unit and the input voltage detection unit, A high frequency dielectric heating power control apparatus in which a changeover switch is provided at a connection point between the oscillation detection circuit and the input voltage detection unit. 5. The power control device for high-frequency dielectric heating according to claim 4, wherein the conversion unit superimposes a carrier wave, whose frequency is set by the switching frequency control signal, and a slice control signal, thereby driving the semiconductor switching element. A power control device for high frequency dielectric heating composed of a frequency modulation circuit for generating a power. A power control method for high-frequency dielectric heating that controls an inverter circuit that rectifies the voltage of an AC power supply, modulates the switching frequency of high-frequency switching of a semiconductor switching element, and converts it to high-frequency power,
Detecting an input current from the AC power source to the inverter circuit;
Obtaining input current waveform information corresponding to the input current;
Detecting an input voltage from the AC power source to the inverter circuit;
Obtaining input voltage waveform information corresponding to the input voltage;
Detecting oscillation of the magnetron;
Outputting the input voltage waveform information in a period until oscillation of the magnetron is detected;
Converting the input current waveform information and the input voltage waveform information output in a period until the oscillation of the magnetron is detected into a drive signal for a semiconductor switching element of the inverter circuit;
A power control method for high-frequency dielectric heating comprising: