JP2007116750A - Amplifier including variable inductor and radio terminal comprising amplifier - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a variable gain amplifier capable of reducing a change of input impedance when switching an amplifier stage. <P>SOLUTION: In an amplifier, a variable gain amplifier state is constituted by connecting in parallel a plurality of amplifier stages (A1, A2, ...An) and to this variable gain amplifier stage, an input impedance control circuit is connected. The control circuit includes a variable resistor (Rx) in its input section, and a change of input impedance caused by switching a gain is compensated by controlling a resistance value of this variable resistor (Rx). <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

    The present invention relates to a variable inductor, and more particularly, to a variable inductor that changes inductance using a plurality of inductors coupled to a circuit composed of active elements.

  The present invention also relates to an oscillator using an inductor and a wireless terminal, and also relates to improvement of circuit design technology for an amplifier and a wireless terminal and a gain variable method for the amplifier.

  Generally, in order to make the characteristics of an electronic circuit variable, the characteristics of active elements or values of passive elements included in the circuit are changed. With respect to an active element, the characteristics of the active element can be changed by changing the bias voltage applied to the active element. As for the passive element, a variable resistance element can be easily realized by using a passive element, for example, an on-resistance of a MOSFET, and a variable capacitance element can be easily realized by using a PN junction.

  With respect to an inductor as a passive element, it is generally difficult to make the inductance variable while maintaining good characteristics. Non-Patent Document 1 (ELECTRONICS LETTERS 2nd January, Vol 28, No. 1, pp 78-80, 1992) discloses a method in which an inductor is formed using an active element to make the inductance variable. However, since an active element is used for the inductor, there is a problem that noise and distortion characteristics are poor.

  Therefore, various techniques for making the inductance variable without forming the inductor by an active element have been proposed. However, each proposal has a problem in practical use. For example, Patent Document 1 (Japanese Patent Application Laid-Open No. Hei 8-162331) discloses a technique in which a switch is inserted in the middle of an inductor and the inductance is changed by turning the switch on and off. This method has a problem that the performance of the variable inductor is deteriorated by the on-resistance of the switch.

  Patent Document 2 (Japanese Patent Laid-Open No. 2000-223317) discloses a method for physically changing the shape of an inductor with a laser beam. Since this method requires physical adjustment after the inductor is manufactured, there is a problem that the manufacturing cost is high and it is difficult to change the inductance in a state where the circuit is operating.

  Patent Document 3 (Japanese Patent Laid-Open No. 7-320942) proposes a technique for configuring a variable inductor by utilizing mutual coupling of a plurality of inductors. In this method, the shape of the inductor is physically changed in order to change the mutual coupling coefficient. Therefore, there is a problem in that the circuit constituting the variable inductor is reduced in size and cost.

  DRPehlke et al., In US Pat. No. 5,994,985, separates an input signal into two using a directional coupler and flows to two inductors coupled to each other. A technique for changing the inductance by controlling the amplitude and phase of a signal is proposed. However, since the directional coupler is generally not suitable for integration, there is a problem that it is difficult to realize a variable inductor by an integrated circuit.

  By realizing the variable inductor, the oscillation frequency (f) of a voltage controlled oscillator (VCO) including an LC resonance circuit (inductor: inductance L, capacitor: capacitance C) can also be controlled. The oscillation frequency (f) of the voltage controlled oscillator including this LC resonance circuit is generally expressed by f = 1 / [2π (LC) 1/2], and if the inductance L or the capacitance C is controlled, the oscillation will occur. The frequency (f) is controlled. However, in the conventional LC resonance circuit, since it is difficult to realize a variable inductor as described above, generally, the capacitance C is made variable, for example, the reverse bias applied voltage to the PN junction diode is changed. Thus, the capacitance C is variable, and the oscillation frequency is changed by changing the capacitance.

  When such a voltage controlled oscillator is built on a semiconductor substrate as an integrated circuit, that is, when it is integrated into an IC, parasitic capacitance, for example, parasitic capacitance of an inductor, drain parasitic capacitance of a MOS transistor, and gate parasitic of a MOS transistor Such a parasitic capacitance is unavoidably generated due to capacitance and the like, and there is a problem that these parasitic capacitances reduce the fluctuation range of the variable capacitor C of the LC resonance circuit. For example, if the capacitance C variation is ΔC, the design variation rate is assumed to be ΔC / C, but the parasitic capacitance is added to the denominator as a non-variable portion. Therefore, in practice, there is a problem that the variation rate becomes small as ΔC / (C + parasitic capacitance). Although depending on the circuit design and the like, if the capacitance C and the parasitic capacitance are approximately the same, the rate of change is reduced to approximately ½.

  Due to the presence of such parasitic capacitance, the ratio of the variable capacitance C to the capacitance value of the LC resonance circuit has to be reduced. Therefore, compared with the rate of change of the variable capacitor C, the change of the capacitance value in the LC resonance circuit is reduced, and as a result, the variable range of the oscillation frequency is narrowed.

  However, today, the frequency band used in mobile phones, wireless LAN devices, etc. is expanded, and there are cases where a plurality of frequency bands are supported by a single device, and the demand for expansion of the variation range of the oscillation frequency has increased. Yes. From this point of view, realization of a variable inductor is desired.

  The variable inductor can also be applied to an amplifier including an inductor. For example, in an amplifier having a degeneration inductor, if the inductance is reduced, the gain and noise characteristics of the amplifier are improved, but the distortion characteristics are degraded. Conversely, when the inductance is increased, the distortion characteristics are improved, but the gain and noise characteristics are deteriorated. Because of the trade-off relationship, the inductance value is determined so as to obtain a desired characteristic when designing the amplifier.

  In order to achieve low distortion characteristics while maintaining high gain and low noise characteristics, it is common to increase the amount of current. In the amplifier used in the receiver of the wireless terminal, the characteristics required for the amplifier vary depending on the magnitude of the received signal. In general, when the received signal is small, it is important to amplify with low noise. Therefore, the amplifier is required to have good gain and noise characteristics. On the other hand, when the received signal is large, the distortion characteristics are required to be good.

  In an amplifier provided with a conventional degeneration inductor, the inductance is fixed, so that the amount of supply current is controlled in order to change the characteristics of the amplifier. That is, when it is desired to improve the distortion characteristics, the current is controlled so that the amount of supplied current is increased. However, increasing the amount of current to change the characteristics of the amplifier has a problem of increasing power consumption.

  Further, Non-Patent Document 4 discloses a circuit example of a variable gain amplifier. In this variable gain amplifier, the first-stage grounded-emitter circuit composed of the first transistor Q1 is always operated, and the gain switching is performed by switching the second to fourth transistors Q2 to Q4 constituting the next-stage base grounded circuit. It is realized by. Since the first transistor Q1 is operating, the input impedance does not change greatly even when the gain is switched, but a constant current is always consumed, and the distortion characteristics are also substantially constant.

However, the conventional circuit disclosed in Non-Patent Document 4 has a problem that a large current is consumed even when the gain is low, and the distortion characteristic is comparable to that when the gain is high. Basically, to achieve an amplification stage with high gain and good distortion characteristics, a certain amount of current consumption is required, but if the gain is not so high or if the signal is attenuated, that much current will be consumed. Thus, it is possible to realize an amplification stage with good distortion characteristics. However, when a plurality of different amplification stages are used by switching, there is a problem that the input impedance changes.
JP-A-8-162331 JP 2000-223317 A JP 7-320942 A US Pat. No. 5,994,985 ELECTRONICS LETTERS 2nd January, Vol 28, No.1, pp 78-80, 1992 MTMurphy, "Applying the Series Feedback Technique to LNA Design," MICROWAVE JOURNAL, Nov., pp.143-152, 1989 JCRudell, et al., "A 1.9GHz Wide-Band IF Double Conversion CMOS Integrated Receiver for Cordless Telephone Applications," ISSCC97, pp.304-305, 1997 Dual-Band High-Linearity Variable-Gain Low-Noise Amplifier for Wireless Applications KLFong, “Dual-Band High-Linearity Variable-Gain Low-Noise Amplifier for Wireless Applications,” IEEE ISSCC99, pp224-225, 1999

    As described above, the variable inductor according to the conventional technique has a problem in electrical characteristics, and there is a problem that it is difficult to reduce the size, reduce the cost, and make an integrated circuit.

In wireless terminals such as mobile phones, there is a strong demand for adaptive characteristic changes in amplifier characteristics according to the received signal level, while on the other hand, low power consumption is also strongly demanded. In an amplifier using a fixed inductor for degeneration, in order to improve the distortion characteristics, the current amount must be increased, resulting in an increase in power consumption.

  The conventional circuit disclosed in Non-Patent Document 4 has a problem in that a large current is consumed even when the gain is low, and the distortion characteristic is comparable to that when the gain is high. Basically, to achieve an amplification stage with high gain and good distortion characteristics, a certain amount of current consumption is required, but if the gain is not so high or if the signal is attenuated, that much current will be consumed. Thus, it is possible to realize an amplification stage with good distortion characteristics. However, when a plurality of different amplification stages are used by switching, there is a problem that the input impedance changes.

  The present invention has been made in view of the above-described circumstances, and its object is to provide an amplification stage having a high gain and good distortion characteristics by passing a certain amount of current consumption when a desired gain is high. Provided is a variable gain amplifier capable of operating an amplification stage having good distortion characteristics with low current consumption when the desired gain is low, and reducing the change in input impedance when the amplification stage is switched There is.

According to this invention,
A variable gain amplification stage in which a plurality of amplification stages are connected in parallel;
An input impedance adjustment circuit that includes a variable resistor connected to the input unit and compensates for a change in input impedance that occurs when the gain is switched by adjusting the resistance value of the variable resistor;
An amplifier is provided.

  According to the present invention, an amplifier capable of controlling amplification characteristics by changing an inductance value can be obtained.

    DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, a variable inductor and an oscillator incorporating a variable inductor according to an embodiment of the invention with reference to the drawings, a radio terminal equipped with the oscillator, an amplifier incorporating a variable inductor in the circuit, and the amplifier A wireless terminal equipped with the above will be described in detail.

  In the following description, an example in which an FET is used as a transistor which is an active element used in a distributor of a variable inductor circuit will be described. However, a variable inductor circuit can also be realized by using a bipolar transistor instead of an FET. .

<Variable inductor>
(First embodiment)
First, a variable inductor according to a basic embodiment of the present invention will be described. 1 and 2 show a circuit configuration of a variable inductor according to an embodiment of the present invention. An input signal (Input) input to the signal input terminal 11 is distributed to a plurality of signal paths 13a, 13b,..., 13n by a distributor 12 configured using active elements. Inductors 14a, 14b,... 14n are inserted in the signal paths 13a, 13b,. The inductors 14a, 14b,... 14n are made of, for example, spiral conductive wires and are arranged close to each other so as to be mutually coupled.

  The inductors 14a, 14b,... 14n generate magnetic fluxes depending on the magnitude of the input signals, and the magnetic fluxes also act on other adjacent inductors to mutually couple with other inductors. Arise. Therefore, each of the inductors 14a, 14b,... 14n has a self-inductance caused by a magnetic flux generated by itself and a mutual inductance determined by a magnetic flux generated by other inductors mutually coupled. For example, the inductance of the inductor 14a, that is, the inductance between the terminals 15 and 16 (referred to as variable inductor terminals) at both ends of the inductor 14a is the self-inductance Lsa of the inductor 14a, the inductor 14a and the other inductors 14b,. Are determined by mutual inductances Mab,. Here, depending on the direction of the mutually coupled inductors (direction that takes into account both the current and winding direction), the mutually coupled inductors are also in a relationship that reinforces the magnetic flux generated by the inductors. , Can also be set to weaken each other. Therefore, the inductance La between the variable inductor terminals 15 and 16 can be increased or decreased with respect to the self-inductance Lsa.

  Signals distributed from the distributor 12 to the signal paths 13a, 13b,..., 13n are supplied to the inductors 14a, 14b,. Here, the mutual inductance Mab,..., Man, that is, the mutual coupling of the inductors 14a, 14b,... 14n is adjusted by adjusting the distribution ratio of the signal levels distributed from the distributor 12 to the inductors 14a, 14b,. The amount of magnetic flux can be controlled. Therefore, the inductance between the variable inductor terminals 15 and 16 can be set to a desired value.

  The distributor 12 is configured by an active element such as a transistor as will be described later. Therefore, unlike a conventional variable inductor circuit using a directional coupler, the distributor 12 shown in FIGS. 1 and 2 can be easily integrated. In the circuit shown in FIG. 1, the distributor 12 is connected between the signal input terminal 11 and one end of the inductors 14a, 14b,. In the circuit shown in FIG. 2, one end and the other end of inductors 14a, 14b,.

  Next, some more specific embodiments of the variable inductor shown in FIGS. 1 and 2 will be described.

(Second Embodiment)
3 and 4 show a variable inductor according to an embodiment in which the distributor 12 includes a transistor circuit whose source is grounded. The circuit according to this embodiment corresponds to a specific circuit example of the basic circuit configuration shown in FIG. An input signal from the signal input terminal 11 is amplified by a FET whose source is grounded, for example, MOSFETs (hereinafter simply referred to as transistors) 21a and 21b, distributed to two signal paths, and inserted into this signal path. It is supplied to the inductors 14a and 14b. The gate terminals of the transistors 21a and 21b are connected to the signal input terminal 11, the source terminal is connected to one end of the inductors 14a and 14b, and the drain terminal is connected to the output terminals 22a and 22b of the variable inductor. In this variable inductor, the signal current amplified by the MOSFETs 21a and 21b is output from the output terminals 22a and 22b of the variable inductor.

  The other ends of the inductors 14a and 14b are connected to one end of the current sources 23a and 23b and one end of the capacitors 24a and 24b, and are grounded with respect to the AC component (high frequency component). The other ends of the current sources 23a and 23b and the other ends of the capacitors 24a and 24b are connected to the ground. Of the current flowing through the inductors 14a and 14b, the direct current component flows through the current sources 23a and 23b, and the alternating current component (high frequency component) is bypassed by the capacitors 24a and 24b.

  In the variable inductors shown in FIG. 3 and FIG. 4, the black dots attached near the symbols representing the inductors 14a and 14b represent the winding start positions of the inductors. It is assumed that the magnetic flux generated with the same line direction is in phase.

  As shown in FIG. 3, when the inductors 14a and 14b are in the same direction, the inductors 14a and 14b are generated in directions in which magnetic fluxes reinforce each other. Therefore, if the self-inductance of the inductor 14a alone is Lsa, the self-inductance of the inductor 14b alone is Lsb, and the mutual coupling coefficient between the inductors 14a and 14b is kab, the effective inductance of the inductor 14a considering the mutual coupling, that is, An inductance La between the variable inductor terminals 15 and 16 is expressed by the following formula (1).

La = Lsa + kab · Lsb (1)
When the mutual inductance between the inductors 14a and 14b is Mab, this equation (1) is expressed by equation (2).

La = Lsa + Mab (2)
On the other hand, as shown in FIG. 4, when the inductors 14a and 14b are reversed, the inductors 14a and 14b are generated so as to weaken the magnetic flux. This inductance La is expressed by equation (3) or (4).

La = Lsa−kab · Lsb (3)
Or La = Lsa-Mab (4)
Here, the mutual coupling coefficient kab, that is, the mutual inductance Mab is determined by the physical arrangement of the inductors 14a and 14b, the sizes of the transistors 21a and 21b, the current values Ia and Ib of the current sources 23a and 23b, and the like. Therefore, the value of the inductance La can be made variable by adjusting the current values Ia and Ib and the sizes of the transistors 21a and 21b without changing the physical arrangement of the inductors 14a and 14b.

  For example, if the current source 23b is a variable current source that can control the current value Ib by an external control signal, and the current value Ib is continuously changed, the mutual inductance Mab is changed accordingly. Thereby, the inductance La between the variable inductor terminals 15 and 16 is continuously changed. When the current source 23b is turned on / off, the inductance La can be switched in a binary manner between (Lsa + Mab) or (Lsa−Mab) and Lsa.

  In the embodiment shown in FIG. 3 and FIG. 4, the grounded source circuit using FET is used for the distributor 12, but a grounded emitter circuit using bipolar transistors can also be used. In the case of a grounded emitter circuit, the gate terminal, drain terminal and source terminal of the FET may be replaced with the base terminal, collector terminal and emitter terminal of the bipolar transistor, respectively. In the circuits shown in FIGS. 3 and 4, two inductors are used. However, the same configuration as that of the present embodiment can be applied to a circuit using three or more inductors.

  Furthermore, as a modified example of this embodiment, the distributor 12 is configured by a plurality of transistors in which the gate terminals or base terminals of each distribution are commonly connected to the signal input terminal 11 via at least one inductor of the plurality of inductors. May be. As another modification of the present embodiment, each gate terminal or base terminal is commonly connected to the signal input terminal 11, and the drain terminal or collector terminal is distributed by a plurality of transistors each connected to one end of a plurality of inductors. A vessel 12 may be configured.

(Third embodiment)
FIG. 5 shows a variable inductor according to a third embodiment of the present invention in which a transistor circuit having a gate grounded to the distributor 12 is used. The signal input terminal 11 is connected to the source terminals of at least one first transistor 31a, 31b,..., 31n and the source terminals of a plurality of second transistors 32a, 32b,. The drain terminals of the first transistors 31a, 31b,..., 31n are connected to one end of the first inductor 14a, and the gate terminals are connected to the control signal input terminals 33a, 33b,. The drain terminals of the second transistors 32a, 32b,..., 32c are commonly connected to one end of the second inductor 14b, and the gate terminals are respectively connected to the control signal input terminals 34a, 34b,.

  Control signals φ33a, φ33b,..., Φ33n are input to the control signal input terminals 33a, 33b,. Control signals φ34a, φ34b,..., Φ34n are input to the control signal input terminals 34a, 34b,. When the control signals φ33a, φ33b,..., Φ33n and φ34a, φ34b,..., Φ34n are changed in binary, the signal level distribution ratio to the inductors 14a and 14b, that is, the ratio of the current flowing through the inductors 14a and 14b. Changed. As a result, the mutual inductance Mab between the inductors 14a and 14b is changed, and as a result, the effective inductance La (inductance between the terminals 15 and 16) of the inductor 14a can be changed.

  The signal level distribution ratio to the inductor 14a is determined by the number of transistors that are turned on by the control signals φ33a, φ33b,..., Φ33n among the transistors 31a, 31b,. Similarly, the distribution ratio of the signal level to the inductor 14b is determined by the number of transistors that are turned on by the control signals φ34a, φ34b,..., Φ34n among the transistors 32a, 32b,.

  The control signals φ33a, φ33b,..., Φ33n and φ34a, φ34b,..., Φ34n are analog signals, and the currents flowing through 31a, 31b, ..., 31n and the transistors 32a, 32b,. Thus, the inductance La between the terminals 15 and 16 may be continuously changed.

  In FIG. 4, a plurality of first transistors 31a, 31b,..., 31n are connected to the inductor 14a. However, if the variable range of inductance may be small, the number of first transistors may be one. Similarly, a plurality of second transistors 32a, 32b,..., 32n are connected to the inductor 14b. However, if the variable range of inductance may be small, a single second transistor may be used.

  In the variable inductor according to this embodiment, a gate grounded circuit using an FET is used for the distributor 12, but a base grounded circuit using a bipolar transistor can also be used. In the case of the grounded base circuit, the gate terminal, drain terminal and source terminal of the FET may be replaced with the base terminal, collector terminal and emitter terminal of the bipolar transistor, respectively.

  Further, as a modification of the present embodiment, the distributor 12 has a source terminal or an emitter terminal connected to the signal input terminal 11 via at least one first inductor, and a gate terminal or a base terminal connected to the control signal input terminal. And at least one second transistor whose source terminal or emitter terminal is connected to the 11 signal input terminal via the first inductor, and whose drain terminal or collector terminal is mutually coupled to the first inductor. It may be configured by at least one second transistor that is connected to one end of the inductor and whose gate terminal or base terminal is connected to the control signal input terminal.

(Fourth embodiment)
FIG. 6 shows a variable inductor according to a fourth embodiment of the present invention that uses a source follower circuit for the distributor 12. The signal input terminal 11 is connected to gate terminals of a plurality (two in the example of FIG. 4) of transistors 41a and 41b. The drain terminals of the transistors 41a and 41b are connected to a power supply Vdd that is a constant potential point, and the source terminals are connected to one ends of the inductors 14a and 14b, respectively. Current sources 43a and 43b are further connected to the source terminals of the transistors 41a and 41b. Therefore, the transistors 41a and 41b operate as a source follower circuit.

  Here, as in the second embodiment, the current source 43b is a variable current source capable of controlling the current value Ib by an external control signal. If this current value Ib is continuously changed, the inductor is accompanied accordingly. The mutual inductance Mab between 14a and 14b is changed, and the inductance La between the terminals 15 and 16 changes continuously. When the current source 43b is turned on / off, the inductance La is binary-switched between (Lsa + Mab) or (Lsa-Mab) and Lsa, where Lsa is the self-inductance of the inductor 14a.

  In the present embodiment, an FET FET emitter follower circuit is used for the distributor 12. However, it is obvious that an emitter follower circuit using a bipolar transistor can also be used.

  Further, as a modification of the present embodiment, each gate terminal or base terminal is connected to the signal input terminal 11 via an inductor, and each drain terminal or collector terminal is distributed by a plurality of transistors connected to a constant potential point. A vessel 12 may be configured.

(Fifth embodiment)
FIG. 7 shows a variable inductor according to a fifth embodiment of the present invention in which a cascode connection circuit is used for the distributor 12. The signal input terminal 11 is connected to the gate terminals of the first and second transistors 51 and 52. The source terminals of the transistors 51 and 52 are connected to current sources 53 and 54, respectively. The drain terminal of the first transistor 51 is connected to the source terminal of the third transistor 55, and the drain terminal of the second transistor 52 is a plurality (two in the example of FIG. 5) of the fourth transistors 56a and 56b. Commonly connected to the source terminals. That is, the transistor 51 and the transistor 55 are cascode-connected, and the transistor 52 and the transistors 56a and 56b are cascode-connected.

  The drain terminals of the third transistor 55 and the fourth transistors 56a, 56b are connected to one ends of the inductors 14a, 14b, 14c, respectively. The gate terminal of the third transistor 55 is connected to the control signal input terminal 58, and the gate terminals of the fourth transistors 56a and 56b are connected to the control signal input terminals 57a and 57b, respectively. As indicated by the black circles in the vicinity of the symbols representing the inductors 14a, 14b, and 14c, the inductor 14b is in a direction in which the magnetic flux is strengthened with respect to the inductor 14a, and the inductor 14c is in a direction in which the magnetic flux is weakened with respect to each other. Each is arranged.

  Now, with the control signal φ input to the control signal input terminal 58 and the transistor 55 turned on, the control signal φ + is input to the control signal input terminal 57a, and the control signal φ− is input to the control signal input terminal 57b. Thus, when the transistor 56a connected to the inductor 14b is turned on and the transistor 56b connected to the inductor 14c is turned off, the inductance La between both ends 15 and 16 of the inductor 14a is larger than the self-inductance Lsa of the inductor 14a. Become. Conversely, when the control signal φ− is input to the control signal input terminal 57a and the control signal φ + is input to the control signal input terminal 57b, the transistor 56a is turned off and the transistor 56b is turned on, both ends of the inductor 14a. The inductance La between 15 and 16 is smaller than the self-inductance Lsa.

  When the cascode connection circuit is used for the distributor 12 as described above, a signal is selectively transmitted to the inductors 14b and 14c by turning on and off the fourth transistors 56a and 56b that are cascode-connected to the second transistor 52. As a result, the inductance La between the terminals 15 and 16 can be increased or decreased.

  In this embodiment, an FET cascode connection circuit is used for the distributor 12, but it is apparent that a cascode connection circuit using a bipolar transistor can also be used.

(Sixth embodiment)
FIG. 8 shows a variable inductor according to a fifth embodiment of the present invention. The signal input terminal 11 is connected to the gate terminal of the transistor 61 constituting the amplifier 60. A current source 63 and an AC (high frequency) bypass capacitor 64 are connected to the source terminal of the transistor 61. One end of the inductor 14 a and the input terminal of the buffer circuit 62 are connected to the drain terminal of the transistor 61, and one end of the other inductor 14 b is connected to the output terminal of the buffer circuit 62.

  The buffer circuit 62 is configured by a circuit that can be formed on an integrated circuit, such as a source ground circuit or a source follower circuit, and its gain is variable. When the gain of the buffer circuit 62 is changed, the signal level distribution ratio to the inductors 14a and 14b is changed as in the previous embodiments, so that the mutual inductance is changed and the inductance between the variable inductor terminals 15 and 16 is changed. be able to.

  In this embodiment, an FET is used for the transistor 61 in the distributor 12, but a bipolar transistor may be used as in the previous embodiments.

(Seventh embodiment)
In the embodiments described so far, the case where all the distributors 12 have a single-phase circuit configuration has been described, but the distributor 12 may be configured by a differential circuit. As an example of a variable inductor according to the seventh embodiment of the present invention, FIG. 9 shows an example in which the distributor 12 has a differential circuit configuration. In the circuit shown in FIG. 9, a differential amplifier cascode-connected to the distributor 12 is used.

  The differential input terminals 11a and 11b are connected to the gate terminals of the differential pair transistors 71a and 71b and the gate terminals of the differential pair transistors 72a and 72b, respectively. The common source terminals of the differential pair transistors 71a and 71b are connected to the current source 78, and the drain terminals of the differential pair transistors 72a and 72b are connected to the current source 79, respectively. The drain terminals of the differential pair transistors 71a and 71b are connected to the source terminals of the transistors 73a and 73b, respectively. The drain terminals of the differential pair transistors 72a and 72b are the source terminals of the transistors 74a and 74b and the sources of the transistors 75a and 75b. Connected to each terminal.

  The gate terminals of the transistors 73a, 73b, 74a, 74b, 75a, and 75b are connected to the control signal input terminals, respectively. The drain terminals of the transistors 73a and 73b are connected to the differential output terminals 22a and 22b, respectively, and to one end of the inductors 76a and 76b. The drain terminals of the transistors 74a and 75b are commonly connected to one end of the inductor 77a, and the drain terminals of the transistors 74b and 75a are commonly connected to one end of the inductor 77b. The other ends of the inductors 76a, 76b, 77a and 77b are connected to a power source Vdd which is a constant potential point.

  Signals Input + and Input− having opposite phases are input to the differential input terminals 11a and 11b. Thus, it is equivalent that the opposite-phase signals Input + and Input− are input and the signs of the mutual inductance M12 between the inductors 76a and 77a and the mutual conductance M13 between the inductors 76b and 77b are inverted. Accordingly, the mutual coupling related to the mutual conductances M12 and M13 can be controlled by the control signals φ + and φ− so as to cancel the magnetic flux and to strengthen the magnetic flux, thereby changing the inductor. Is possible.

  Further, FIG. 10 shows an example in which a distributor is configured using a differential circuit using bipolar transistors. Although there are differences in the transistors used, the principle is the same as the circuit shown in FIG.

  As described above, according to the variable inductor according to the embodiment of the present invention, the inductor can be electromagnetically changed using the distributor that can be formed on the integrated circuit. Moreover, such a variable inductor has good electrical characteristics, can be easily reduced in size and cost, and is suitable for integration into an integrated circuit.

<Oscillator with variable inductor and radio terminal with this oscillator>
Next, an oscillator incorporating the above-described variable inductor according to the present invention and a radio terminal equipped with this oscillator will be described.

  FIG. 11 is a block diagram schematically showing an oscillator according to an embodiment of the present invention.

  The oscillator shown in FIG. 11 includes a core circuit unit (VCO core) 102 including an LC resonance circuit having an inductor L0, and an inductor L11 that is electromagnetically coupled to the inductor L0 with a coupling coefficient k1, and is supplied to the inductor L11. The oscillation frequency control unit 104-1 (Frequency Controller) capable of controlling the current to be supplied is provided.

  A control signal Control_1 is input to the oscillation frequency control unit 104-1, and at least one or both of the amplitude and phase of the current flowing through the inductor L11 is changed according to the control signal Control_1. As a result, the inductance of the inductor L0 of the core circuit unit 102 coupled to the inductor L11 is changed to change the oscillation frequency. For example, when a current in a direction that generates a magnetic field in a direction that strengthens the magnetic field generated by the inductor L0 flows through the inductor L11, the inductance value of the inductor L0 increases, and is proportional to the -1/2 power of the inductance value. The oscillation frequency becomes smaller. Further, when a current in a direction that weakens the magnetic field generated by the inductor L0 flows through the inductor L11, the inductance value of the inductor L0 decreases, and as a result, the oscillation frequency increases. By controlling the current amplitude with the same direction of the current flowing through the inductor L0 and the inductor L11, it is possible to control the increase or decrease of the inductor L0.

  In general, the frequency control range obtained by changing the capacitance C is about Cmax / Cmin = 2. On the other hand, in the method of changing the inductance, for example, assuming that the currents flowing through the inductor L and the inductor L11 are the same value and k = about 0.7, Lmin = (1−k) L and Lmax = (1 + k) L. , Lmax / Lmin = about 6 and a large fluctuation range can be realized.

  Normally, when the oscillation frequency is changed with a variable capacitor, only a change in the range of 5 to 10% of the oscillation frequency can be obtained. However, in an oscillator having a variable inductor as shown in FIG. The oscillation frequency can be changed to a range of 100%.

  The number of inductors L11 is not limited to one, and a plurality of inductors L11 to L1n such as an inductor L11 may be provided as shown in FIG. In the circuit shown in FIG. 11, n oscillation frequency control units 104-n are provided, and the inductors L11 to L1n of the respective oscillation frequency control units are electromagnetically coupled with the inductance L of the VCO circuit 102 and the coupling coefficients (k1 to kn). Is bound to.

  When only one inductor L11 is electromagnetically coupled to one oscillation frequency control unit 104-1, that is, the inductor L of the core circuit unit 102, the current amplitude or phase flowing through the inductor L11 can be changed. As a result, the inductance value of the inductor L0 can be changed.

  In a circuit including a plurality of inductors L11 to L1n, the VCO circuit 102 can be controlled in various ways. For example, even if a current having the same coupling coefficient as that of the inductor L11 and the same current as the current flowing through the inductor L11 is passed through the inductor L1n, the number of inductors L1n through which this current is turned on and off is changed. The activated L1n that is electromagnetically coupled to the inductor L0 may be varied.

  Further, inductors L11 to L1n having different coupling coefficients k1 to kn are prepared, and these inductors L11 to L1n can be switched and used. Furthermore, the current flowing through each of the inductors L11 to L1n is changed, so that the core circuit unit 102 can be controlled more finely.

  In FIG. 12, an oscillation signal (voltage signal) is input to the oscillation frequency control circuit 104 from the differential LC resonance type voltage control oscillation circuit 106 corresponding to the VCO core circuit 102, and this oscillation signal is subjected to voltage-current conversion. A circuit example supplied to inductors L1 and L2 coupled to inductors L01 and L02 of the core circuit unit 102 is shown.

  In the circuit shown in FIG. 12, one end of an inductor L01 is connected to a current source 105 having a power supply voltage Vdd, and the other end is connected to a variable capacitor VC1 including a drain of a MOS transistor T1 and a diode. The source of the MOS transistor T1 is grounded.

  Similarly, one end of an inductor L02 is connected to a current source 105 having a power supply voltage Vdd, and the other end is connected to a variable capacitor VC2 formed of a drain and a diode of a MOS transistor T2. The source of this MOS transistor T2 is grounded.

  The drain of the MOS transistor T1 and the gate of the MOS transistor T2 are connected, and similarly, the drain of the MOS transistor T2 and the gate of the MOS transistor T1 are connected.

  The capacitance control voltage Vctrl is supplied to the variable capacitors VC1 and VC2, and the capacitances of the variable capacitors VC1 and VC2 are determined. The resonance frequency is determined by the parallel connection (L1-VC1) and (L2-VC2) of the variable capacitors VC1, VC2 and the inductors L1, L2.

  In FIG. 12, variable capacitors VC1 and VC2 are used, but fixed capacitors may be used. When a fixed-capacitance capacitor is used, there is no change in frequency control depending on the variable capacitors VC1 and VC2, and the operation is not changed.

  The oscillator 106 outputs an output 1 (Output_1) from the drain of the MOS transistor T1 and an output 2 (Output_2) from the drain of the MOS transistor T2.

  In order to change the oscillation frequency of this output signal, output 1 and output 2 (Output_1, Output_2) are input to the current-voltage converter circuit (VI Converter) 108 of the oscillation frequency control circuit 104 and are electromagnetically coupled to the inductor L01. The current flowing through the inductor L1 coupled with the coefficient k1 and the inductor L2 electromagnetically coupled with the inductor L02 with the coupling coefficient k2 is controlled.

  Here, k1 = 0.7, k2 = 0.7, and the inductance values are the same for each inductor (L01, L02, L1, L2), and the same current flows through each inductor. Assume.

  When the direction of the current between the electromagnetically coupled inductors is a direction in which the magnetic field is strengthened, the inductance value of the inductor L01 increases from L0 to 1.7L0.

  Further, when a current flows in a direction in which the magnetic field is weakened, the inductance value of the inductor L01 decreases to 0.3L0.

  Therefore, if the lower oscillation frequency is near fLo = 2 GHz, the higher oscillation frequency is fHi = (0.3 / 1.7) −1 / 2 · fLo, and an oscillation frequency of 4 GHz or higher is obtained. You can see that

  FIG. 13 is a specific example in which the voltage-current conversion circuit 108 shown in FIG. 12 is configured by cascode-connected transistors. The differential LC resonance type voltage-controlled oscillation circuit 16 has the same circuit configuration as that shown in FIG. 12, and the same reference numerals are given in FIG. The frequency controller 104 will be described below.

  The inductor L1 is connected to the drain of the transistor T12, and the source of the transistor T12 is connected to the drain of the transistor T11 whose output 1 (Output_1) is input to the gate. The source of the transistor T11 is grounded through the current source 110. Further, the source of the transistor T13 having the drain connected to the inductor L2 is connected to the drain of the transistor T11. A control signal φ + is input to the gate of the transistor T12, and a control signal φ− is input to the gate of the transistor T13. That is, the inductor L1 is connected to the transistor T12 that is cascode-connected to the transistor T11 that is grounded through the current source and the output 1 (Output_1) is supplied to the gate, and the control signal φ + is applied to the gate of the transistor T12. Supplied. Further, the inductor L2 is connected to the transistor T13 that is cascode-connected to the transistor T11, and the control signal φ− is supplied to the gate of the transistor T13.

  Similarly, the drain of the transistor T15 is connected to the inductor L2, and the source of the transistor T15 is connected to the drain of the transistor T14 whose output 2 (Output_2) is input to the gate. The source of the transistor T14 is grounded through a current source. Further, the source of the transistor T16 having the drain connected to the inductor L1 is connected to the drain of the transistor T14. A control signal φ + is input to the gate of the transistor T15, and a control signal φ− is input to the transistor T16. That is, the inductor L2 is connected to the transistor T14 that is cascode-connected to the transistor T14 that is grounded via the current source and the output 2 (Output_2) is supplied to the gate, and the control signal φ + is supplied to the gate of the transistor T15. The The inductor L1 is connected to the transistor T16 that is cascode-connected to the transistor T14, and the control signal φ− is supplied to the gate of the transistor T16.

  By changing the control signals φ + and φ−, for example, the direction of the current flowing through the inductors L1 and L2 can be reversed. Further, the amplitude of the current flowing through the inductors L1 and L2 can be changed by appropriately setting the potentials of the control signals φ + and φ−.

  Thus, by controlling the control signals φ + and φ−, the current flowing through the inductors L1 and L2 can be controlled, and as a result, the inductance values of the inductors L01 and L02 coupled to the inductors L1 and L2 can be controlled. Become. For example, when the current flowing through the inductor L01 (L02) and the current flowing through the inductor L1 (L2) are in phase, the value of the inductance of the inductor L01 (L02) increases, and as a result, the oscillation frequency decreases. Further, when the current flowing through the inductor L01 (L02) and the current flowing through the inductor L1 (L2) are in opposite phases, the inductance value of the inductor L01 (L02) is reduced, and as a result, the oscillation frequency is increased.

  FIG. 14 shows an oscillator according to an embodiment in which a plurality of differential pairs are arranged.

  One end of the inductor L01 is connected to the current source having the power supply voltage Vdd, and the other end is connected to the drain of the MOS transistor T1 and the capacitor C1. The source of the MOS transistor T1 is grounded via a current source I0.

  Similarly, one end of the inductor L02 is connected to the current source having the power supply voltage Vdd, and the other end is connected to the capacitor C2 including the drain of the MOS transistor T2 and the diode. The source of the transistor T2 is grounded via the current source I0.

  The drain of the transistor T1 and the gate of the transistor T2 are connected. Similarly, the drain of the transistor T2 and the gate of the transistor T1 are connected.

  In the circuit shown in FIG. 14, the capacitances of the capacitors C1 and C2 are fixed, but a variable capacitor may be used as in FIG.

  In the circuit shown in FIG. 14, the oscillation frequency control unit is composed of a plurality of differential pairs. An inductor L1n having one end connected to the power supply voltage Vdd, for example, the inductor L11, is electromagnetically coupled to the inductor L01 of the VCO circuit 106 by a coupling coefficient k1n, for example, a coupling coefficient k11, and the other end is a MOS transistor T1n, for example, The drain of the transistor T11 is connected. A capacitor C1 (n), for example, a capacitor C11 is branched and connected between the drain of the transistor T1n and the inductor L1n. An output 1 (Output_1) of the VCO circuit is supplied to the gate of the transistor T1n, and the source of the transistor T1n is grounded via a variable current source In, for example, a variable current source I1.

  Similarly, an inductor L2n having one end connected to the power supply voltage Vdd, for example, an inductor L21, is electromagnetically coupled to an inductor L02 of the VCO circuit with a coupling coefficient k2n, for example, a coupling coefficient k21, and the other end is connected to a MOS transistor T2n. For example, it is connected to the drain of the MOS transistor T21. A capacitor C2n, for example, a capacitor C21 is branched and connected between the drain of the transistor T2n and the inductor L1n.

  The output 2 (Output_2) of the VCO circuit 106 is supplied to the gate of the transistor T2n, and the source of the transistor T2n is grounded via the variable current source In.

  The capacitor C1n and the capacitor C2n, for example, the capacitor C11 and the capacitor C21, are connected to the other end connected to the inductor.

  A plurality of such differential pairs are prepared, and the oscillation values are controlled by changing the inductance values of the inductors L01 and L02 of the VCO circuit 106 by changing or turning ON / OFF the current values of the individual variable current sources I0 to In. can do.

  FIG. 15 shows an oscillator according to an embodiment using a variable phase shifter and a variable gain amplifier.

  The VCO circuit 106 has the same circuit configuration as that shown in FIG. Note that buffer circuits 112 and 114 having a large impedance as viewed from the outside are connected to the output terminal.

  Output signals (Output_1, Output_2) from the oscillator are input to the variable phase circuits 116 and 118 of the oscillation frequency control circuit 104. The signal of the oscillation circuit is appropriately controlled by the variable phase shifters 120 and 122, the phase shift is shifted, and then the current is controlled via the variable gain amplifiers 124 and 126, and electromagnetically generated by the inductor L01 and the coupling coefficient k1. A current having a controlled phase / current amplitude value is supplied to the inductor L1 coupled to.

  Similarly, the current flowing through inductor L2 coupled with inductor L02 with coupling coefficient k2 is similarly controlled.

  By appropriately controlling the phase shift and the current value, the inductances of the inductors L01 and L02 of the VCO circuit 106 can be made variable, so that the oscillation frequency can be changed and controlled.

  In the above-described circuit example, the differential type has been described, but the same can be applied to a single phase. A circuit example in which a variable inductor is applied to a Colpitts oscillation circuit will be described with reference to FIG.

  The inductor L, one end of which is connected to the power supply voltage Vdd, is connected to the drain of the MOS transistor T, and the source of the transistor T is grounded via the resistor R. The gate of the transistor T is grounded. The inductor L0 and the transistor T are branched from the connection point, and the capacitors C1 and C2 are connected in series and grounded. The source of the transistor T1 is connected to the connection point between the capacitor C1 and the capacitor C2.

  The output from the connection point between the inductor L0 and the capacitor C1 is input to the frequency control unit 128 (Frequency Control), connected to the power supply voltage Vdd, and coupled to the inductor L1 electromagnetically coupled to the inductor L0 with the coupling coefficient k. It is possible to control the current flowing and change the oscillation frequency.

  In the embodiment described above, the MOS transistor is used, but it goes without saying that a circuit having the same function can be realized by using a bipolar transistor or the like.

  Further, the inductor constituting the inductor variable type oscillator described above, that is, the inductor L0 on the LC resonance circuit side and the inductor L1 electromagnetically coupled with the coupling coefficient k can be realized with various configurations. For example, as shown in FIG. 17, the spiral conductor constituting the inductor L can be configured to be symmetrical with the spiral conductor constituting the inductor L1. In this case, the respective conductors intersect at the central portion of FIG. 17, but this intersection can be realized with one layer of wiring except for the intersecting region by configuring the conductors to overlap with each other through an insulating film. it can.

  As shown in FIG. 18, when realizing a plurality of inductors L11... L1n electromagnetically coupled to the inductor L01, one turn L11... L1n is also arranged on the inner circumference side of one turn L01. The structure to do can be taken.

  The conductor patterns need not be arranged on the same plane. For example, L01, L11... L1n can be stacked using multilayer wiring as shown in FIG. In FIG. 19, an insulating layer exists between the coil conductors constituting the inductors L01, L11... L1n, but in FIG. 19, the insulating layer is omitted for the sake of simplicity of the drawing. Yes.

  By using an oscillator with a wide variable frequency range in this way, a wireless terminal suitable for a plurality of communication systems having different frequency bands can be reduced in size. That is, in a conventional wireless terminal, it is necessary to mount an oscillator corresponding to each oscillation frequency in order to supply an oscillation frequency corresponding to each communication system. However, since the oscillator according to the embodiment of the present invention has a wide variable frequency range, it can be applied to a communication system in which various oscillation frequencies are used by one oscillator. FIG. 20 shows a block of wireless terminals that support various communication systems that use different communication frequencies, for example, communication frequencies of 2 GHz, 2.4 GHz, and 5 GHz. The signals of the individual communication systems are antennas 130-1, 130-2, 130-3, low noise amplifiers 132-1, 132-2, 132-3, mixers (MIX) 134-1, 134. The signals are supplied to intermediate frequency processing units (IF) 136-1, 136-2, and 136-3 via -2 and 134-3. Outputs from the intermediate frequency processing units (IF) 136-1, 136-2, and 136-3 are supplied to the baseband processing unit. Each mixer (MIX) 134-1, 134-2, 134-3 is supplied with a signal having an oscillation frequency corresponding to the communication system. In this circuit example, a local signal having a suitable frequency is supplied from the oscillator 138 to each wireless system. The oscillator according to the embodiment of the present invention described above is used as the oscillator 138 that supplies the oscillation frequency. A signal having a desired frequency obtained by changing the inductance is switched by a switch and supplied to the mixer MIX of each communication system.

  As described above, according to the radio terminal according to the embodiment of the present invention, a plurality of communication systems having different frequency bands, such as mobile phones (PDC, W-CDMA, etc.), Bluetooth, etc., without having a plurality of oscillators. (Registered trademark), wireless LAN (2.4 GHz, 5 GHz) and the like can be supported by one wireless terminal. According to the oscillator according to the embodiment of the present invention, a wide variable frequency range can be realized, and as a result, an oscillation frequency can be supplied to a plurality of communication systems with one oscillator.

<Amplifier with variable inductor and radio terminal having this amplifier>
Next, an amplifier including a variable inductor according to an embodiment of the present invention will be described.

  FIG. 21 is a block diagram schematically showing an amplifier according to an embodiment of the present invention.

  The amplifier shown in FIG. 21 includes an amplifier circuit 140 including an inductor La, and a control unit 142 including an inductor Lc electromagnetically coupled to the inductor L with a coupling coefficient k.

  The control unit 142 receives an input signal Input_1 to the amplifier circuit 140 and supplies a signal current to the inductor Lc. The characteristic of the output Output_1 of the amplifier is changed according to the value of the inductance of the inductor La that changes according to the signal current supplied to the inductor Lc.

  In the circuit shown in FIG. 21, only one inductor Lc is shown, but the number is not limited to one, and a plurality of inductors Lc-1 to Lc-n may be provided. Further, when a plurality of inductors Lc-1 to Lc-n are provided, various controls can be realized. For example, an inductor Lc-n having the same coupling coefficient and the same flowing current is prepared, and the number of Lc-1 to Lc-n through which the current flows is changed by turning on / off the current, and the inductor L It is possible to cope by changing the activated Lc-n that is electromagnetically coupled.

  It is also possible to prepare inductors Lc-1 to Lc-n having different coupling coefficients k1 to kn and use them by switching. Furthermore, finer control can be performed by changing the current flowing through each of the inductors Lc-1 to Lc-n.

  The inductance of the inductor La can be apparently changed also by changing the phase in the same manner as the current amount control including the ON / OFF control.

  Further, the simplest control may be a system in which one inductor Lc is prepared and the current flowing through the inductor Lc is binary controlled. Also in this control system, the effect of varying the amplification characteristics can be sufficiently exhibited.

  A specific example of the differential amplifier will be described with reference to FIG.

  As shown in FIG. 22, the source of the MOS transistor M1 whose input signal Input_1 is input to the gate is connected to the current source I1 via the degeneration inductor L1. The other end of the current source I1 is grounded. The inductor L3 is connected to the drain of the MOS transistor M1, and the other end of the inductor L3 is connected to the power supply potential Vdd.

  The input signal Input_2 is inputted to the gate of the MOS transistor M2 constituting the differential pair, and the degeneration inductor L2 is connected to the source of the MOS transistor M2, similarly to the connection of the inductor L1, the MOS transistor M1, and the inductor L3. Is connected to the drain, the inductor L2 is connected to the current source I1, and the inductor L4 is connected to the power supply potential Vdd.

  Output signals Output_1 and Output_2 are output from the drains of the MOS transistors M1 and M2, respectively. The control unit includes MOS transistors M3 and M4 that input the input signals Input_1 and Input_2 of the differential amplifier to the gates, respectively. The source of the MOS transistor M3 is connected to an inductor L5 coupled to the inductor L1 with a coupling coefficient k1, and is grounded via a variable current source I2. Similarly, the source of the MOS transistor M4 is connected to the variable current source I2 via an inductor L6 coupled to the inductor L2 with a coupling coefficient k2. The drains of the MOS transistors M3 and M4 are connected to the power supply potential Vdd.

  In order to change the characteristics of the output signal from the differential amplifier, such as the distortion characteristics, the current value of the variable current source I2 is changed. The current change of the variable current source I2 is assumed to be a binary value of current ON / OFF. When the variable current source I2 is ON, it is assumed that the pair of inductors L1 and L5 and the pair of inductors L2 and L6 that are electromagnetically coupled to each other are configured to weaken the magnetic field. In such a circuit, when the variable current source I2 is ON, the inductance of the degeneration inductor looks small, so that high gain and low noise can be realized. That is, a high gain / low noise mode can be achieved. Further, when the variable current source I2 is OFF, a magnetic field in a direction that cancels the magnetic field generated by the inductors L1 and L2 is not generated, so that the degeneration inductance appears larger than when the variable current source I2 is ON. It will be. In this case, good distortion characteristics can be obtained. That is, a mode in which distortion characteristics are emphasized can be set.

  Therefore, when configuring the differential amplifier, the degeneration inductor may be designed with a relatively large inductance.

  In the above description, the current value of the variable current source I2 is controlled by the binary value of On / OFF. However, desired amplifier characteristics can be obtained by changing the current value stepwise or continuously.

  FIG. 23 is a block diagram schematically showing another embodiment of the present invention.

  The amplifier shown in FIG. 23 includes a first amplifier circuit 150 including an inductor La and a second amplifier 152 including an inductor Lc electromagnetically coupled to the inductor La with a coupling coefficient k.

  The second amplifier 152 receives the same input signal Input1 as the first amplifier 150, functions as a controller for controlling the amplifier characteristics of the first amplifier 150, such as distortion characteristics, and the like. The output of the amplifier 152 is added to the output of the first amplifier 150 and output as an output signal Output_1.

Similar to the above-described embodiment, the second amplifier 152 corresponding to the control unit receives the input signal Input_1 to the amplifier circuit 150 and supplies a signal current to the inductor Lc. The characteristic of the output Output_1 of the first amplifier 150 changes according to the inductance value of the inductor L that changes depending on the signal current supplied to the inductor LC. At this time, the gain of the input signal Input_1 is increased by returning the output of the second amplifier 152 to the output Output_1.
FIG. 24 shows a specific circuit example of the differential amplifier shown in FIG.

  The source of the MOS transistor M1 to which the input signal Input_1 is input to the gate is connected to the current source I1 through the degeneration inductor L1. The other end of the current source I1 is grounded. The inductor L3 is connected to the drain of the MOS transistor M1, and the other end of the inductor L3 is connected to the power supply potential Vdd.

  The input signal Input_2 is input to the gate of the MOS transistor M2 constituting the differential pair. Similarly to the connection of the inductor L1, the transistor M1, and the inductor L3, the degeneration inductor L2 is connected to the source of the MOS transistor M2. The inductor L4 is connected to the drain, the inductor L2 is connected to the current source I1, and the inductor L4 is connected to the power supply potential Vdd.

  Output signals Output_1 and Output_2 are output from the drains of the MOS transistors M1 and M2, respectively. The second amplifier 152 corresponding to the control unit includes a MOS transistor M3 and a MOS transistor M4 that input the input signals Input_1 and Input_2 of the differential amplifier to the gates, respectively.

  The source of the MOS transistor M3 is connected to an inductor L5 coupled to the inductor L1 with a coupling coefficient k1, and is grounded via a variable current source I2. Similarly, the source of the MOS transistor M4 is connected to the variable current source I2 via an inductor L6 coupled to the inductor L2 with a coupling coefficient k2.

  The drain of the MOS transistor M3 is connected to the drain of the MOS transistor M1, and a current from the variable current source I2 flowing through the MOS transistor M3 is added to the output signal Output_1. Similarly, the drain of the MOS transistor M4 is connected to the drain of the MOS transistor M2, and a current from the variable current source I2 flowing through the MOS transistor M4 is added to the output signal Output_2.

  By adopting such a circuit configuration, the current flowing through the control unit can be used as the output of the amplifier circuit, and the current utilization efficiency of the entire amplifier is increased.

  Depending on the design, for example, if the current flowing through a normal amplifier is about 5 mA, it may be necessary to pass the current up to about 10 mA in order to achieve low distortion characteristics. On the other hand, in the circuit shown in FIG. 24, when high gain and low noise are realized, if it is set so that I1 = I2 = 2.5 mA, a characteristic equivalent to that when a current of 5 mA is flowed in a normal amplifier is obtained. You can expect to get. On the other hand, when the low distortion characteristic is realized, the low distortion characteristic can be realized by the effect of the large degeneration inductor even if the current source I2 is turned off and only 2.5 mA is supplied to the current source I1.

  In the above circuit, the characteristics of the amplifier can be made variable, but the input impedance of the amplifier also changes accordingly. In general, it is not desirable for the input impedance of the amplifier to change. Therefore, it is preferable to provide an input impedance control section at the input section of the amplifier.

  With reference to FIG. 25, a circuit including a variable resistor as an input impedance control unit will be described.

  In the circuit shown in FIG. 25, a variable resistor Rv is inserted between the input terminal Input_1 and the input terminal Input_2. The circuit shown in FIG. 25 has the same configuration as that of FIG. 24 except for the variable resistor Rv.

  The variable resistor Rv may be configured using an FET as shown in FIG. 26A, or may be a combination of a fixed resistor and a switch as shown in FIG.

  In the circuit shown in FIG. 25, as shown in FIG. 27, the input impedance of the amplification stage is expressed by the following equation (5).

Zin = L gm / C + j (ωL−1 / ωC). . . (5)
In this equation (5), when the bias current is flowing, there is a real part defined by the first term of equation (5), but when the bias current is turned off, gm = 0. The input impedance has no real part. For this reason, in order to make the input impedance when the bias current is OFF close to the input impedance when the bias current is supplied, it is necessary to compensate the real part of the impedance. For this purpose, the aforementioned variable resistor Rv is used to compensate the real part of the impedance.

  In order to simplify the description, the resistance value of the variable resistor Rv is binary, that is, open / ON. In the circuit shown in FIG. 25, it is assumed that the inductor L1 and the inductor L5, and the inductor L2 and the inductor L6 are arranged in the direction in which the magnetic fluxes cancel each other. Under this assumption, when both the current source I1 and the current source I2 are ON, the total degeneration inductance is reduced and the device operates with high gain and low noise. Here, it is assumed that the variable resistor Rv is in an open state. On the other hand, when only the current source I1 is turned on and the current source I2 is turned off, the total degeneration inductance increases and operates with low distortion, but no bias current flows through the transistors M3 and M4. The real part of the impedance disappears. In order to compensate the real part of this impedance, if the resistance value of the variable resistor Rv is turned ON and an appropriate resistance value is set for the variable resistor Rv, the input impedance close to that when operating in the high gain / low noise mode can be obtained. .

  When the current control by the variable current source I2 is controlled by two or more values such as stepwise or continuous, the resistance value of the variable resistor Rv may be controlled by stepwise or continuous change. In such a case, the voltage φ applied to the gate may be changed in a type using an FET as shown in FIG.

  In the circuit shown in FIG. 25, FIG. 28 shows a circuit configuration in which the gain can be made variable even with a gate grounding circuit in the subsequent stage.

  The MOS transistors M11a and M11b whose gates are grounded are interposed between the MOS transistor M1 and the inductor L3 in FIG. 24. The transistor M11a is turned on / off by a gate signal φ2, and the gate of the transistor M11b is connected to a power source, for example. Always on.

  The MOS transistor M12 is arranged so that the inductor L3 and the MOS transistor M11a are bypassed and the drain of the MOS transistor M1 is connected to the power supply potential Vdd. This MOS transistor M12 is driven by a gate signal φ1.

  Similarly, MOS transistors M21a and M21b are interposed between the MOS transistor M2 and the inductor L4, the transistor M21 is turned on / off by the gate signal φ2, and the transistor M21b is always turned on. The MOS transistor M22 is arranged so as to bypass the inductor L4 and the MOS transistor M21 and to connect the drain of the MOS transistor M2 to the power supply potential Vdd. This MOS transistor M22 is driven by a gate signal φ1.

  It is assumed that the transistors M11a and M12 and the transistors M21 and M22 are configured by MOS transistors having the same size, and the transistors M11b and M21b are configured by MOS transistors having a small size. When the gate signal φ1 is turned off and the gate signal φ2 is turned on, a signal supplied via the transistor M11a, the transistor M11b, the transistor M21a, and the transistor M21b is output. On the other hand, when the gate signal φ1 is turned on and the gate signal φ2 is turned off, only the signals supplied via the transistors M11b and M21b are output, and the other signals are passed through the transistors M12 and M22. The gain is low. By using this gain switching operation in combination with the above-described gain switching for changing the degeneration inductor, a wider gain switching width can be realized.

  FIG. 29 shows a circuit example in which an inductance variable circuit is applied for input impedance adjustment.

Instead of the variable resistor Rv shown in the circuit shown in FIG. 25, an impedance adjustment circuit 160 having a circuit configuration similar to that of the transistor M1, the inductor L1, the transistor M2, the inductor L2, the transistor M3, the inductor L5, the transistor M4, and the inductor L6. Is provided. That is, the drain of the MOS transistor M5 is connected to the power supply potential Vdd, the source is connected to the current source I3 via the inductor L7, and the input signal Input_1 is supplied to the gate.
The drain of the MOS transistor M6 that forms a differential pair with the MOS transistor M5 is connected to the power supply potential Vdd, the source is connected to the current source I3 via the inductor L8, and the input signal Input_2 is supplied to the gate. .
As circuits for controlling the inductances of the inductors L7 and L8, MOS transistors M7 and M8 for inputting the input signals Input_1 and Input2 of the differential amplifier 162 to the gates, respectively, are provided. The source of the MOS transistor M7 is connected to an inductor L7 and an inductor L9 having a coupling coefficient k3, and is grounded via a variable current source I4. Similarly, the source of the MOS transistor M8 is connected to the variable current source I4 via an inductor L10 having an inductor L8 and a coupling coefficient k4. The drains of the MOS transistors M7 and M8 are connected to the power supply potential Vdd.

  In order to simplify the description, the inductances of the inductors L1 to L10 are set to the same value, and the coupling coefficient k1 = k3 = k2 = k4. Variable current sources I2, I4. It is assumed that inductors that change and couple with the current ON / OFF values are connected so as to weaken the magnetic field.

  When the variable current source I2 is turned on, the inductance values of the degeneration inductors L1 and L2 appear small and operate in the high gain / low noise mode. If the variable current source I4 is turned off at this time, the impedance adjustment circuit operates in a pseudo low distortion mode, and the total input impedance is that of the high gain / low noise mode circuit and the low distortion mode circuit. It becomes the value connected in parallel.

  On the other hand, when the variable current source I2 is turned OFF, the inductance values of the degeneration inductors L1 and L2 appear large and operate in the low distortion mode. If the variable current source I4 is turned on at this time, the impedance adjustment circuit operates in a pseudo high gain / low noise mode, and the total input impedance is. The high gain / low noise mode circuit and the low distortion mode circuit are connected in parallel and do not change even when the operation mode is switched.

  Furthermore, in the above-described embodiment, the circuit example of the differential pair has been described, but the same effect can be obtained even with a single-phase circuit. FIG. 30 is a circuit diagram showing an embodiment applied to a single-phase amplifier circuit.

As shown in FIG. 30, the source of the MOS transistor M1 to which the input signal Input_1 is input to the gate is connected to the current source I1 via the degeneration inductor L. The other end of the current source I1 is grounded. The inductor L3 is connected to the drain of the MOS transistor M1, and the other end of the inductor L3 is connected to the power supply potential Vdd. An output signal Output_1 is output from the drain of the MOS transistor M1.
The control unit includes a MOS transistor M2 that inputs an input signal Input_1 to the gate. An inductor LC having an inductor L and a coupling coefficient k is connected to the source of the MOS transistor M2, and grounded via the variable current source I2. Has been. The drain of the MOS transistor M3 is connected to the drain of the MOS transistor M1, and the current from the variable current source I2 flowing through the MOS transistor M2 is added to the output signal Output_1.

  The value of the inductance of the degeneration inductor L can be controlled by ON / OFF control of the power source of the variable current source I2, for example.

  The degeneration inductor described above and the control inductor electromagnetically coupled with the coupling coefficient k, such as the inductors L1 and L2 shown in FIG. 21, can be realized with various configurations. For example, as shown in FIG. 17, the spiral conductor constituting the degeneration inductor L and the spiral conductor constituting the control inductor LC can be configured to be symmetrical. In this case, the respective conductors intersect at the center of the drawing, and by forming this portion via an insulating film, it can be realized by a single layer wiring except for the intersecting region. The inductor is not limited to the configuration shown in FIG. 17, and may have any configuration as long as it can be electromagnetically coupled, and may be configured with a conductor pattern as shown in FIG. 18 or FIG. 19.

  Although the MOS transistor is used in the circuit according to the embodiment of the present invention described above, it goes without saying that an active element such as another transistor may be used.

  The amplifier according to the embodiment of the present invention can be used for a wireless communication terminal such as a mobile phone. An example is shown in FIG. FIG. 31 shows a block diagram of the wireless communication terminal.

  An RF input signal from the antenna (ANT) is supplied to the RF signal processing unit (RF). That is, in the RF signal processing unit (RF), the signal is supplied to the RF bandpass filter 1 (RF-BPF1), the low noise amplifier (LNA), and the RF bandpass filter 2 (RF-BPF2) via the switch (T / R). The multiplier (DC) multiplies the local signal (RF-VCO) by the multiplier (DC) and frequency-converts it to an intermediate frequency signal. This intermediate frequency signal is supplied to an intermediate frequency processing unit (IF-Stage) and a baseband signal processing unit (BB-Stage).

  The low noise amplifier (LNA) is supplied with a gain control signal (Gain Control) from a received electric field strength determination unit (RSSI) in the baseband signal processing unit (BB-Stage).

  The signal to be transmitted is processed in the reverse manner. That is, the signals supplied from the baseband signal processing unit (BB-Stage) and the intermediate frequency processing unit (IF-Stage) are processed by the RF signal processing unit (RF). In the RF signal processing unit, the intermediate frequency signal is multiplied by a local signal (RF-VCO) by a multiplier (UC) and frequency-converted, and the converted signal is passed through an RF bandpass filter (BP-BPF). The power is supplied to the power amplifier (PA) and is supplied to the antenna (ANT) through the switch (T / R).

  The amplifier described above is used for such a low noise amplifier (LNA) of a wireless terminal.

  There are various standards and standards for wireless terminals. When the level of an input RF signal is small, an amplifier characteristic for amplifying a signal with low noise may be required for a low noise amplifier (LNA). is there. In such a case, the LNA degeneration inductance value is controlled to be small. On the other hand, when the RF signal is sufficiently large, it is important not to distort the RF signal. Therefore, the inductance value for LNA degeneration is controlled to be large.

  Such characteristic control of the low noise amplifier (LNA) can be performed using, for example, a gain control signal (Gain Control). The amplifier characteristic control may be performed based on a standard other than RSSI.

  As described above, by using the amplification characteristic variable amplifier of the present invention for the LNA of the RF processing unit of the wireless terminal, it is possible to adaptively realize the amplification characteristic of the desired LNA without increasing the current consumption.

In the amplifier described with reference to FIGS. 21 to 31, at least a pair of inductances La, Lc, L1, L5, L2, L6, L7, L9, L8, and L10 are:
It is assumed that they are mutually coupled with coupling coefficients k1, k2, k3, k4 and have mutual inductance. However, an amplifier can be realized without a pair of inductances being coupled to each other and having no mutual inductance. The amplifier using the mutual inductance has an advantage that a large degeneration can be realized and a good distortion characteristic can be realized with a smaller current. On the other hand, an amplifier having an inductance having no mutual inductance has an advantage that the design is easy because it is not necessary to use an inductor having a complicated shape.

  Hereinafter, an amplifier including an inductor having no mutual inductance according to another embodiment of the present invention will be described in detail with reference to FIGS.

  In the following description, all examples using bipolar transistors will be described, but other active elements such as FETs may be used.

  FIG. 32 shows a basic circuit configuration of an amplifier according to another embodiment of the present invention. As shown in FIG. 32, a plurality of amplification stages A1 to An are connected in parallel to the input side Input, and the amplification stages A1 to An are selectively operated, thereby realizing a gain switching function. . Here, the amplification stages A1 to An may have the same amplification characteristics, may have different amplification characteristics, or may have the same and different amplification characteristics. When the gain is switched, the input impedance is changed, but the change of the input impedance Zin is compensated by the variable resistor Rx. That is, even when any one of the amplification stages A1 to An is operating, the value of the variable resistor Rx is set to an appropriate value so that the input impedance Zin of the entire amplifier does not change greatly. Here, for example, the amplification stage A1 is an amplification stage having high gain and good distortion characteristics with a certain amount of current consumption, and the amplification stage A2 is an amplification stage having low gain and good distortion characteristics with low current consumption. Then, when a high gain is required, the amplification stage A1 is operated, and when a low gain is required, the amplification stage A2 is operated, thereby realizing desired gain and distortion characteristics and requiring current consumption. It can be minimized. By setting the variable resistor Rx to an appropriate value, it is possible to suppress the change in the input impedance Zin when the gain is changed by switching the operation of the amplification stage A1 and the amplification stage A2.

  FIG. 33 shows a circuit according to the first embodiment for realizing the variable resistor Rx of the circuit of FIG. The variable resistor Rx for adjusting the input impedance is realized by using fixed resistors R1 to Rm and switches SW1 to SWm. By appropriately switching the switches SW1 to SWm, the fixed resistors R1 to Rm are selected, and a desired resistance value is given to the input side resistor Rx.

  FIG. 34 shows a circuit according to the second embodiment for realizing the variable resistor Rx of the circuit of FIG. The variable resistor Rx for adjusting the input impedance is composed of an FET 172, and a desired resistance value can be realized by applying an appropriate control voltage to the control gate Vctrl of the FET 172.

  FIG. 35 shows a circuit according to a third embodiment for realizing the variable resistor Rx of the circuit of FIG. In the circuit shown in FIG. 35, two emitter-grounded amplification stages A1 and A2 are connected in parallel, and a MOSFET is connected to the input stage to adjust the input impedance. The degeneration inductor L1 of the amplification stage A1 has a small inductance value so that a high gain can be realized. Further, the degeneration inductor L2 of the amplification stage A2 has a large inductance value so as to realize a low gain and good distortion characteristics. In the circuit shown in FIG. 35, the result of simulating the input impedance (admittance) of the part excluding the MOSFET 172 in the input part is shown in FIG. FIG. 36 shows a first case in which the circuit constant and the operating point are appropriately set, the amplification stage A1 is turned on, and the amplification stage A2 is turned off to realize high gain, and the amplification stage A1 is turned off, and The reflection coefficient in the second case where the amplification stage A2 is turned on to realize a low gain is shown on the admittance. As is apparent from FIG. 36, when the appropriate resistance is connected in parallel, that is, the MOSFET 172 connected to the input unit is turned ON, the impedance in the first and second cases shown in FIG. It is possible to be the same. The current consumption when realizing the high gain is about 3 mA, the current consumption when realizing the low gain is about 1.5 mA, and the current consumption is also reduced when the gain is low. When the gain is low, the current consumption is small, but since the value of the degeneration inductor L2 is large, good distortion characteristics can be realized.

  FIG. 37 shows a circuit according to a fourth embodiment for realizing the variable resistor Rx of the circuit of FIG. In the circuit shown in FIG. 37, two differential amplifier stages A1 and A2 are connected in parallel, and a MOSFET 172 is connected to the input stage to adjust the input impedance. Here, when realizing a high gain, both amplification stages A1 and A2 are turned ON. When the two amplifiers A1 and A2 are operated, the degeneration inductor also looks small, so that a high gain can be realized. Also, when realizing a low gain, only the amplification stage A1 is turned on and the amplification stage A2 is turned off, so that only the inductor L1 can be seen in the degeneration inductor, which is a large degeneration and good. Distortion characteristics can be realized.

  FIG. 38 shows a circuit according to a fifth embodiment for realizing the variable resistor Rx of the circuit of FIG. In the circuit of FIG. 38, a series circuit of fixed resistors R1 to R3 and switches SW1 to SWm is connected in parallel between inputs Input_1 and Input_2. In this circuit, the input side resistor Rx can be appropriately set by selectively turning on and off the switches SW1 to SWm. That is, by appropriately switching the switches SW1 to SWm, the fixed resistors R1 to Rm are selected, and a desired resistance value is given to the input side resistor Rx.

  FIG. 39 shows a circuit example in which the amplifier circuit shown in FIG. 32 is applied to a low noise amplifier of a radio. In the circuit of the wireless terminal shown in FIG. 39, the RF input signal from the antenna (ANT) is input to the RF signal processing unit (RF), and this RF signal processing unit (RF) will be described with reference to FIG. To a low noise amplifier (LNA). An output signal from the low noise amplifier (LNA) is multiplied by a local signal (RF-VCO) by a multiplier (DC) and frequency-converted to an intermediate frequency signal. The intermediate frequency signal is supplied to the intermediate frequency amplifier (IF-AMP) of the intermediate frequency processing unit (IF-Stage) via the band pass filter (BPF). The output from the intermediate frequency amplifier (IF-AMP) is supplied to the baseband signal processing unit (BB-Stage) via the quadrature demodulator (QDEM) and processed.

  In the circuit of this wireless terminal, the amplifier circuit shown in FIG. 32 is applied to a low noise amplifier, realizing gain switching and making the input impedance constant. The gain switching function expands the dynamic range of the radio and does not change the input impedance of the low-noise amplifier. Therefore, it is possible to easily match a desired impedance such as 50Ω with a single input matching circuit. In addition, although a circuit used for a radio device is required to have low power consumption, a low noise amplifier to which the present invention is applied can be operated with a minimum necessary current. It also leads to power consumption.

  In addition, this invention is not limited to the said embodiment as it is, In an implementation stage, a component can be deform | transformed and embodied in the range which does not deviate from the summary.

  In addition, various inventions can be formed by appropriately combining a plurality of components disclosed in the embodiment. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, you may combine the component covering different embodiment suitably.

1 is a circuit diagram schematically showing a variable inductor according to a first embodiment of the present invention. FIG. 6 is a circuit diagram schematically showing a modification of the variable inductor shown in FIG. 1. FIG. 6 is a circuit diagram schematically showing a variable inductor according to a second embodiment of the present invention. FIG. 4 is a circuit diagram schematically showing a modification of the variable inductor shown in FIG. 3. FIG. 5 is a circuit diagram schematically showing a variable inductor according to a third embodiment of the present invention. It is a circuit diagram which shows roughly the variable inductor which concerns on 4th Embodiment of this invention. FIG. 9 is a circuit diagram schematically showing a variable inductor according to a fifth embodiment of the present invention. It is a circuit diagram which shows roughly the variable inductor which concerns on 6th Embodiment of this invention. It is a circuit diagram which shows roughly the variable inductor which concerns on 7th Embodiment of this invention. It is a circuit diagram which shows roughly the variable inductor which concerns on 7th Embodiment of this invention. 1 is a block diagram schematically showing an oscillator according to an embodiment of the present invention. FIG. 5 is a block diagram schematically showing an oscillator according to another embodiment of the present invention. FIG. 13 is a block diagram schematically showing a circuit in which the voltage-current conversion circuit shown in FIG. 12 is configured by cascode-connected transistors. FIG. 10 is a block diagram schematically showing a transmitter according to still another embodiment of the present invention in which a plurality of differential pairs are arranged. FIG. 6 is a block diagram schematically showing an oscillator according to still another embodiment of the present invention using a variable phase shifter and a variable gain amplifier. FIG. 10 is a block diagram schematically showing a Colpitts oscillation circuit according to still another embodiment of the present invention. FIG. 17 is a plan view schematically showing a circuit pattern of the inductor shown in FIGS. 1 to 16. FIG. 17 is a plan view schematically showing another circuit pattern of the inductor shown in FIGS. 1 to 16. FIG. 17 is a plan view schematically showing another circuit pattern of the inductor shown in FIGS. 1 to 16. FIG. 17 is a block diagram schematically illustrating a wireless terminal including the oscillator illustrated in FIGS. 1 to 16. 1 is a block diagram schematically showing an amplifier according to an embodiment of the present invention. FIG. 22 is a circuit diagram showing a specific circuit of the differential amplifier shown in FIG. 21. It is a block diagram which shows the differential amplifier which concerns on other embodiment of this invention. FIG. 24 is a circuit diagram showing a specific circuit of the differential amplifier shown in FIG. 23. FIG. 24 is a circuit diagram showing another specific circuit of the differential amplifier shown in FIG. 23. (A) And (b) shows the circuit example of the variable resistance shown by FIG. FIG. 26 is an equivalent circuit diagram showing the input impedance of the amplification stage shown in FIGS. 24 and 25. FIG. 26 is a circuit diagram showing a circuit according to a modification of the circuit shown in FIG. 25. FIG. 24 is a circuit diagram showing still another specific circuit of the differential amplifier shown in FIG. 23. FIG. 24 is a circuit diagram showing a circuit according to a modification of the differential amplifier shown in FIG. 23. FIG. 31 is a block diagram schematically showing a wireless terminal incorporating the amplifier described with reference to FIGS. 23 to 30. It is a block diagram which shows the basic circuit structure of the amplifier which concerns on other embodiment of this invention. It is a block diagram which shows the circuit which implement | achieves the variable resistance shown by FIG. FIG. 33 is a block diagram showing another circuit that realizes the variable resistance shown in FIG. 32. FIG. 33 is a block diagram showing still another circuit that realizes the variable resistor shown in FIG. 32. FIG. 36 is a graph showing the result of simulating the input impedance of a part excluding the MOSFET of the input part in the circuit shown in FIG. FIG. 33 is a block diagram showing still another circuit for realizing the variable resistance shown in FIG. 32. FIG. 34 is a block diagram showing another circuit that realizes the variable resistor shown in FIG. 33. FIG. 33 is a block diagram illustrating a circuit example of a wireless terminal in which the amplifier circuit illustrated in FIG. 32 is applied to a low noise amplifier of a wireless device.

Explanation of symbols

DESCRIPTION OF SYMBOLS 11 ... Signal input terminal 12 ... Divider 13a-13n ... Signal path 14a-14c, 76a, 76b, 77a, 77b ... Inductor 15, 16 ... Variable inductor terminal 21a, 21b ... Common source transistor 22 ... Load circuit 23a, 23b ... Current sources 24a, 24b ... high frequency bypass capacitors 31a to 31n ... first transistor (gate grounded transistor)
32a to 32n ... second transistor (grounded gate transistor)
34a to 34n: control signal input terminals 41a, 41b: transistors of the source follower circuit 43a, 43b ... current source 51 ... first transistor 52 ... second transistor 53, 54 ... current source 55 ... third transistor (cascode transistor) )
56a, 56b ... fourth transistor (cascode transistor)
57a, 57b ... control signal input terminal 61 ... transistor 62 ... buffer circuit 63 ... current source 64 ... high frequency bypass capacitor 71a, 71b ... first differential pair transistor 72a, 72b ... second differential pair transistor 73a, 73b 74a, 74b, 75a, 75b ... cascode transistors L0, L11, L1n, L01, L02, L1, L2, L2n ... inductors k, k1, kn, k1n, k2n ... coupling coefficients VC1, VC2 ... variable capacitors C1, C2, C11, C21, C1n, C2n: Capacitor L, L1, L2: Degeneration inductor (first inductor)
L ', L5, L6 ... second inductor 102 ... core circuit (VCO core)
104, 104-1 to 104-n ... Oscillation frequency control unit 105 ... Current source 106 ... Voltage controlled oscillation circuit VC1, VC2 ... Variable capacitance T1, T2 ... MOS transistors T11-T16 ... Transistors T11-T1n ... Transistors T21-T2n ... Transistors I0 to In ... variable current sources 116, 118 ... variable phase circuits 120, 122 ... variable phase shifters 130-1, 130-2, 130-3 ... antennas 132-1, 132-2, 132-3 ... low noise Amplifier (Low Noise Amplifier)
134-1, 134-2, 134-3 ... Mixer (MIX)
136-1, 136-2, 136-3, intermediate frequency processing section (IF)
M1, M2, M11a, M11b, M21a, M21b, M22 ... MOS transistors

Claims (5)

A variable gain amplification stage in which a plurality of amplification stages are connected in parallel;
An input impedance adjustment circuit that includes a variable resistor connected to the input unit and compensates for a change in input impedance that occurs when the gain is switched by adjusting the resistance value of the variable resistor;
An amplifier comprising:     2. The amplifier according to claim 1, wherein the variable resistor connected to the input includes a plurality of fixed resistors and a switch for selecting the fixed resistors.     2. The amplifier according to claim 1, wherein the amplification stage has different amplification characteristics, and the gain of the variable gain amplification stage is changed by selectively operating an operating amplification stage.     2. The amplifier according to claim 1, wherein the amplification stages have the same amplification characteristic, and the gain of the variable gain amplification stage is changed by selectively operating an operating amplification stage.     A wireless communication apparatus comprising the amplifier according to claim 1 as a low noise amplifier of an RF unit of the wireless communication apparatus.

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