JP2007103668A - Light emitting device driving circuit and optical transmitter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a light emitting device driving circuit capable of adjusting the conversion gain when converting the photo-current from a monitoring photoreceptor into a voltage signal, and reducing the effect of the adjustment of the conversion gain on the frequency characteristic; and to provide an optical transmitter. <P>SOLUTION: The driving circuit 1a includes a current-voltage converter 5a for converting the photo-current Imon from a monitoring photodiode 22 into a light quantity signal Smon; a reference signal generator 13 for generating a reference signal Vref of a voltage corresponding to a transmission signal Sin; and a driving current generator 19 for generating a driving current Id modulated according to the transmission signal Sin, for adjusting the current amount of the driving current Id according to the difference between the light quantity signal Smon and the reference signal Vref, and for supplying the driving current Id to a laser diode 21. The current-voltage converter 5a includes a common-base TIA circuit, and the transimpedance of the common-base TIA circuit is variable. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a light emitting element driving circuit and an optical transmitter used for optical communication.

  In order to stabilize the signal light intensity and the extinction ratio of a light emitting element such as a laser diode, a light emitting element driving circuit that controls the amount of driving current based on the photocurrent from a light receiving element for monitoring is known. For example, a semiconductor laser driving circuit disclosed in Patent Document 1 detects an average level of the magnitude of photocurrent from a monitor photodiode in an AC-coupled laser driving circuit, and detects the bias current supplied to the laser diode. The size is controlled according to the average level. The drive circuit controls the modulation degree of the bias current according to the average level and high level (peak value) of the magnitude of the photocurrent from the monitor photodiode.

JP-A-4-13979

  The light emission characteristics of a light emitting element such as a laser diode and the coupling efficiency of an optical signal to a monitoring light receiving element are slightly different for each individual. Therefore, in the light emitting element driving circuit as described above, when controlling the amount of drive current based on the photocurrent from the light receiving element for monitoring, the conversion gain when converting the photocurrent to current voltage is set to the light emission characteristic of the light emitting element. It is desirable that the optical signal can be adjusted in accordance with the coupling efficiency of the optical signal to the monitoring light receiving element.

  However, in a negative feedback type transimpedance amplifier that has been widely used as a current-voltage conversion circuit, a low-pass filter is configured by a resistance element for realizing the transimpedance and the internal capacitance of the light receiving element. Therefore, if the conversion gain (transimpedance) of the current-voltage converter circuit is changed, the high-frequency cutoff frequency of the current-voltage converter circuit will fluctuate, and detection errors will occur when detecting the peak and bottom levels of the current-voltage converter circuit output. Therefore, it is not preferable.

  The present invention has been made in view of the above problems, and can adjust the conversion gain when converting the photocurrent from the light receiving element for monitoring into a voltage signal, and reduce the influence on the frequency characteristics by adjusting the conversion gain. An object of the present invention is to provide a light emitting element driving circuit and an optical transmitter.

  In order to solve the above-described problems, the light-emitting element driving circuit according to the present invention causes the light-emitting element to generate signal light corresponding to the transmission signal and controls the signal light based on the photocurrent from the light-receiving element that detects the signal light. A light-emitting element driving circuit, a first current-voltage conversion unit that converts a photocurrent from a light-receiving element into a light amount signal that is a voltage signal, and a reference signal generation that generates a reference signal that is a voltage signal corresponding to a transmission signal And a drive current that is modulated according to the transmission signal, adjusts the amount of the drive current according to the difference between the light amount signal and the reference signal, and supplies the drive current to the light emitting element And the first current-voltage converter includes a common base type transimpedance amplifier circuit, and the transimpedance of the common base type transimpedance amplifier circuit is variable. Characterized in that there.

  In the above-described light emitting element driving circuit, the first current-voltage converter includes a common base type transimpedance amplifier (TIA) circuit. Originally, the common base type TIA circuit easily picks up noise as compared with the negative feedback type TIA circuit, and thus is not suitable for the light emitting element driving circuit. However, the present inventors do not configure a low-pass filter by the transimpedance resistance element of the common base type TIA circuit and the internal capacitance of the light receiving element, and even if the transimpedance is changed, the frequency characteristic of the closed-loop control system changes almost. I found it not. That is, according to the above-described light emitting element driving circuit, the first current-voltage conversion unit includes the common base type TIA circuit and the transimpedance is variable, so that the photocurrent from the light receiving element (for monitoring) is changed to the voltage. The conversion gain at the time of conversion into a signal (light quantity signal) can be adjusted, and the influence on the frequency characteristics due to the adjustment of the conversion gain can be reduced.

  The light emitting element driving circuit may further include a gain control unit that provides a gain control signal for changing the transimpedance of the common base type transimpedance amplifier circuit to the first current-voltage conversion unit. Thereby, the conversion gain of the first current-voltage conversion unit can be easily changed in accordance with variations in characteristics of the light emitting elements.

  In the light emitting element driving circuit, the reference signal generation unit generates a reference current corresponding to the transmission signal, a second current-voltage conversion unit that converts the reference current into a reference signal, and a reference current generation And a variable capacitance portion connected between the signal line connecting the second current-voltage conversion portion and the constant potential line, and the second current-voltage conversion portion includes a common base type transimpedance amplifier circuit. The transimpedance of the common base type transimpedance amplifier circuit may be variable.

  As described above, the reference signal generation unit includes the reference current generation unit and the second current-voltage conversion unit, and the second current-voltage conversion unit is a transimpedance variable common base type like the first current-voltage conversion unit. By having the TIA circuit, when the conversion characteristics such as the gain of the first current-voltage converter change due to a temperature change or the like, the conversion characteristics of the second current-voltage converter also change in the same manner. Can be offset. Further, by having a variable capacitance portion between the signal line connecting the reference current generation portion and the second current-voltage conversion portion and the constant potential line, it results from the internal capacitance of the light receiving element and the parallel capacitor capacitance for noise removal. A frequency characteristic similar to the frequency characteristic of the light quantity signal can be given to the reference signal, and the internal capacitance of each light receiving element can be easily handled. As a result, the drive current amount can be controlled with higher accuracy.

  In addition, the light emitting element driving circuit may include a capacitance control unit that provides a capacitance control signal for changing the capacitance of the variable capacitance unit to the variable capacitance unit. Thereby, the capacitance value of the variable capacitance portion can be easily changed according to the variation in the internal capacitance of the light receiving element.

  The optical transmitter of the present invention is an optical transmitter that outputs signal light according to a transmission signal, a light emitting element that generates signal light, a light receiving element that detects signal light from the light emitting element, and a transmission A light-emitting element drive circuit that supplies a drive current modulated in accordance with a signal to the light-emitting element, and the light-emitting element drive circuit converts a photocurrent from the light-receiving element into a light amount signal that is a voltage signal; A reference signal generation unit that generates a reference signal that is a voltage signal corresponding to the transmission signal, and generates a drive current, adjusts the amount of drive current according to the difference between the light amount signal and the reference signal, and drives the drive A drive current generation unit for supplying current to the light emitting element, and the current-voltage conversion unit includes a common base type transimpedance amplifier circuit. Wherein the dance is variable. According to this optical transmitter, the current-voltage conversion unit includes a common base type TIA circuit, and the transimpedance is variable, so that the conversion of the photocurrent from the light receiving element (for monitoring) into a voltage signal is performed. The gain can be adjusted, and the influence on the frequency characteristics due to the adjustment of the conversion gain can be reduced.

  According to the light emitting element driving circuit or the optical transmitter of the present invention, it is possible to adjust the conversion gain when converting the photocurrent from the monitoring light receiving element into a voltage signal, and to reduce the influence on the frequency characteristics by adjusting the conversion gain. it can.

  Hereinafter, preferred embodiments of a light emitting element driving circuit and an optical transmitter according to the present invention will be described in detail with reference to the drawings. In the description of the drawings, the same or corresponding parts are denoted by the same reference numerals, and redundant description is omitted.

(First embodiment)
FIG. 1 is a block diagram showing a configuration of a light emitting element driving circuit 1a according to a preferred embodiment of the present invention. A light-emitting element driving circuit 1a shown in the figure is a circuit for driving the optical module 2, and constitutes the optical transmitter according to the present embodiment together with the optical module 2. The optical module 2 has a small package configuration such as TOSA (Transmitter Optical Sub-Assembly). The optical module 2 detects (monitors) a light emitting element such as a laser diode 21 that generates the signal light P and the signal light P from the laser diode 21 (more precisely, light from the light reflection end face of the laser diode 21). And a light receiving element for monitoring such as a diode 22. The laser diode 21 receives a drive current Id including a modulation current Imod and a bias current Ibias corresponding to the transmission signals Sp and Sn from the outside from the light emitting element drive circuit 1a, and the signal light P which is a laser light corresponding to the drive current Id. Is generated.

Here, FIG. 2A is a diagram illustrating an example of output characteristics of the laser diode 21. 2A, the horizontal axis indicates the forward current (LD drive current) flowing through the laser diode 21, and the vertical axis indicates the light output intensity of the laser diode 21. FIG. Further, the graph G1 in the figure shows the output characteristics when the temperature of the laser diode 21 is a certain value T ₁ [° C.], and the graph G2 shows the temperature of the laser diode 21 at T ₂ (> T ₁ ) [° C. ] Shows the output characteristics. Ith ₁ in the graph G1 and Ith ₂ in the graph G2 indicate the threshold current values of the laser diode 21 at the temperatures T ₁ and T ₂ , respectively.

FIG. 2B is a graph showing an example of a time waveform of the signal light P output from the laser diode 21. The signal light intensity PL at the low level and the signal light intensity PH at the high level are alternately displayed. The waveform that repeats is shown. FIG. 2C is a graph showing an example of a time waveform of the drive current Id necessary for outputting the signal light P shown in FIG. In FIG. 2C, a graph G3 is an example of a time waveform of the drive current Id required at the temperature T ₁ [° C.], and a graph G4 shows the drive current Id required at the temperature T ₂ [° C.]. It is an example of a time waveform. Note that Ibias ₁ in FIG. 2C indicates a bias current value when the laser driving circuit is AC-coupled to the laser diode. In addition, Ibias ₂ in FIG. 2C indicates a bias current value when the laser drive circuit is DC coupled to the laser diode. The drive current Id is formed by convolution of the modulation current Imod with the bias current Ibias ₁ (or Ibias ₂ ).

As shown in FIG. 2A, the output characteristics of the laser diode 21 vary greatly with changes in temperature. That is, as the element temperature of the laser diode 21 increases, the laser oscillation threshold current value increases, and a larger drive current is required to obtain a predetermined light output intensity. Therefore, in order to keep the signal light intensity PH and the extinction ratio (PH / PL) constant regardless of the element temperature of the laser diode 21, as shown in graphs G3 and G4 in FIG. 2C, the bias current Ibias ₁ It is necessary to control the magnitudes of (Ibias ₂ ) and modulation current Imod according to the light output intensity. Therefore, the light emitting element driving circuit 1 a controls the signal light intensity PH and the extinction ratio (PH / PL) of the signal light P based on the photocurrent Imon from the photodiode 22.

  Refer to FIG. 1 again. The light emitting element driving circuit 1a of the present embodiment is a so-called AC coupling type laser driving circuit. The light emitting element drive circuit 1a includes a modulation current generator 3, a bias current source 4, current-voltage converters 5a and 5b, level hold units 6a and 6b, error amplifiers 17a and 7b, a reference current generator 8, a variable capacitor 9, The switch elements 10a to 10f and the control unit 16 are included.

The modulation current generator 3 constitutes a drive current generator 19 in the present embodiment together with a bias current source 4 to be described later. The drive current generator 19 is a circuit part for supplying the laser diode 21 with the drive current Id modulated according to the transmission signals Sn and Sp. The modulation current generator 3 generates modulation currents Imod ₁ and Imod ₂ corresponding to the transmission signals Sn and Sp among the drive current Id. Specifically, the modulation current generator 3 includes two input terminals for inputting transmission signals Sp (positive phase) and Sn (negative phase), which are differential signals including transmission data, from the outside, and the transmission signal Sn. and two output terminals for outputting to the laser diode 21 modulated current Imod ₂ in accordance with the modulation current Imod ₁ and the transmission signal Sp corresponding to. The output terminal for outputting the modulation current Imod ₁ is electrically connected to the anode of the laser diode 21 via a capacitor (capacitance element) 11a for cutting low frequency components. The output terminal for outputting the modulation current Imod ₂ is electrically connected to the cathode of the laser diode 21 via a capacitor (capacitance element) 11b for cutting low frequency components.

The modulation current generator 3 further includes an input terminal for inputting a difference signal Sd_bt from an error amplifier 17b described later. The modulation current generator 3 adjusts the magnitudes of the modulation currents Imod ₁ and Imod ₂ so that the magnitude of the difference signal Sd_bt becomes small.

  The bias current source 4 generates a forward bias current Ibias out of the drive current Id. Specifically, one end of the bias current source 4 is electrically connected to the cathode of the laser diode 21 via an inductor 12b for cutting high frequency components. The other end of the bias current source 4 is electrically connected to a reference potential line 15 such as a GND line. The anode of the laser diode 21 is electrically connected to the power supply potential line 14 via an inductor 12a for cutting high frequency components. A power supply voltage Vcc is supplied to the power supply potential line 14. The bias current source 4 further has an input terminal for inputting a difference signal Sd_pk from an error amplifier 17a described later. The bias current source 4 adjusts the magnitude of the bias current Ibias so that the magnitude of the difference signal Sd_pk becomes small.

  In the present embodiment, the bias current source 4 is connected between the cathode of the laser diode 21 and the reference potential line 15, but the bias current source is connected between the anode of the laser diode and the power supply potential line 14. It may be connected.

  The current-voltage converter 5a is a first current-voltage converter in the present embodiment, and is a circuit for converting the photocurrent Imon from the photodiode 22 into a light amount signal Smon that is a voltage signal. As will be described later, the current-voltage conversion unit 5a includes a common base type TIA circuit, and the transimpedance of the common base type TIA circuit is variable. The transimpedance of the common base type TIA circuit is changed according to gain control signals ZTSEL1 and ZTSEL2 from the control unit 16 described later.

  Here, FIG. 3 is a graph showing an example of a time waveform of the photocurrent Imon from the photodiode 22 and an example of a time waveform of the light amount signal Smon corresponding to the photocurrent Imon in this embodiment. is there. As shown in FIGS. 3A and 3B, the current-voltage converter 5a of the present embodiment outputs a peak level as the light amount signal Smon corresponding to the low level of the photocurrent Imon, and the high level of the photocurrent Imon. The bottom level is output as the light quantity signal Smon corresponding to Therefore, the photocurrent Imon and the light amount signal Smon are reversed from each other and have complementary time waveforms.

  The level hold unit 6a includes a peak detection unit 61a and a bottom detection unit 62a. The peak detector 61a is a circuit for detecting the peak level of the light amount signal Smon. The peak detector 61a generates a peak level signal Spk indicating the peak level of the light amount signal Smon, and provides it to the error amplifier 17a via the switch element 10a. The bottom detection unit 62a is a circuit for detecting the bottom level of the light amount signal Smon. The bottom detector 62a generates a bottom level signal Sbt indicating the bottom level of the light amount signal Smon, and provides the bottom level signal Sbt to the error amplifier 17b via the switch element 10b. The peak level of the light amount signal Smon corresponds to the low level of the transmission signals Sp and Sn (that is, the low level of the signal light P), and the bottom level of the light amount signal Smon is the high level of the transmission signals Sp and Sn. (That is, the high level of the signal light P).

  The error amplifier 17a is a circuit for amplifying a difference between a peak level signal Spk from the peak detector 61a and a peak level reference voltage Vpk from a peak detector 61b described later, and providing the amplified signal to the bias current source 4. Specifically, the error amplifier 17a includes an operational amplifier, the non-inverting input terminal of the error amplifier 17a is electrically connected to the peak detector 61a, and the inverting input terminal of the error amplifier 17a is the peak detector 61b. And are electrically connected. The output terminal of the error amplifier 17a is electrically connected to the bias current source 4 via the switch element 10e. The error amplifier 17a generates a difference signal Sd_pk indicating the difference between the peak level signal Spk and the peak level reference voltage Vpk, and provides the difference signal Sd_pk to the bias current source 4.

  The error amplifier 17b is a circuit for amplifying a difference between a bottom level signal Sbt from the bottom detection unit 62a and a bottom level reference voltage Vbt from a bottom detection unit 62b described later, and providing the amplified current to the modulation current generation unit 3 It is. Specifically, the error amplifier 17b includes an operational amplifier, the non-inverting input terminal of the error amplifier 17b is electrically connected to the bottom detection unit 62a, and the inverting input terminal of the error amplifier 17b is the bottom detection unit 62b. And are electrically connected. The output terminal of the error amplifier 17b is electrically connected to the modulation current generator 3 through the switch element 10f. The error amplifier 17 b generates a difference signal Sd_bt that indicates the difference between the bottom level signal Sbt and the bottom level reference voltage Vbt, and provides the difference signal Sd_bt to the modulation current generator 3.

  The reference current generation unit 8 constitutes the reference signal generation unit 13 in the present embodiment together with a variable capacitance unit 9 and a current-voltage conversion unit 5b described later. The reference signal generation unit 13 is a circuit for generating a reference signal Vref that is a voltage signal corresponding to the transmission signals Sp and Sn from the outside. The reference current generator 8 generates a reference current Iref corresponding to the transmission signals Sp and Sn. This reference current Iref is a pulse current corresponding to the transmission signals Sp and Sn. That is, the reference current Iref includes the same data component as the transmission signals Sp and Sn, and has a rise and a fall corresponding to the vibration of the transmission signals Sp and Sn. For example, the reference current generator 8 can generate a modulation component by performing voltage-current conversion on the transmission signals Sp and Sn, and can generate the reference current Iref by convolution of the modulation component with a predetermined bias component.

  The current-voltage converter 5b is a second current-voltage converter in the present embodiment, and is a circuit for converting the reference current Iref from the reference current generator 8 into a reference signal Vref that is a voltage signal. The function and internal configuration of the current-voltage converter 5b are the same as those of the current-voltage converter 5a described above. That is, the current-voltage converter 5b of the present embodiment includes a common base type TIA circuit, and the transimpedance of the common base type TIA circuit is variable. The transimpedance of the common base type TIA circuit is changed according to gain control signals ZTSEL1 and ZTSEL2 from the control unit 16 described later. The current-voltage converter 5b outputs a peak level as the reference signal Vref corresponding to the low level of the reference current Iref, and outputs a bottom level as the reference signal Vref corresponding to the high level of the reference current Iref. Therefore, the reference current Iref and the reference signal Vref are reversed from each other and have a complementary time waveform.

  The variable capacitance unit 9 is connected between a signal line connecting the reference current generation unit 8 and the current-voltage conversion unit 5 b and a constant potential line such as a reference potential line 15. The variable capacitance unit 9 is configured so that the capacitance value can be changed according to the capacitance control signals BW_AD1 to BW_AD3 from the control unit 16.

  The variable capacitance unit 9 is a circuit for similarly giving the reference signal Vref the influence on the light quantity signal Smon by the internal capacitance of the photodiode 22 and the parallel capacitor capacitance for removing crosstalk noise. That is, in the light emitting element driving circuit 1a, a low-pass filter is configured by the parallel capacitance of the photodiode 22 and the input impedance of the current-voltage conversion unit 5a. Therefore, the light amount signal Smon is subjected to band limitation by the low-pass filter. This low-pass filter has an effect of removing noise convoluted with the photocurrent Imon. On the other hand, also in the reference signal generation unit 13, the variable capacitance unit 9 is connected between the signal line connecting the reference current generation unit 8 and the current-voltage conversion unit 5 b and the reference potential line 15, and the capacitance of the variable capacitance unit 9. And the input impedance of the current-voltage converter 5b constitute a low-pass filter. As a result, the reference signal Vref can be given the same band limitation as the light amount signal Smon.

  The level hold unit 6b includes a peak detection unit 61b and a bottom detection unit 62b. The peak detector 61b detects the peak level of the reference signal Vref, and provides the peak level reference voltage Vpk indicating the peak level to the error amplifier 17a via the switch element 10c. The bottom detector 62b detects the bottom level of the reference signal Vref, and provides the bottom level reference voltage Vbt indicating the bottom level to the error amplifier 17b via the switch element 10d. Note that the peak level of the reference signal Vref corresponds to the low level of the transmission signals Sp and Sn (that is, the low level of the signal light P), and the bottom level of the reference signal Vref is the high level of the transmission signals Sp and Sn. (That is, the high level of the signal light P).

  The control unit 16 has both a function as a gain control unit and a function as a capacity control unit in the present embodiment. That is, the control unit 16 provides gain control signals ZTSEL1 and ZTSEL2 for changing the transimpedance of the common base type TIA circuit to the current-voltage conversion units 5a and 5b. Further, the control unit 16 provides capacity control signals BW_AD <b> 1 to BW_AD <b> 3 for changing the capacity of the variable capacity unit 9 to the variable capacity unit 9. The gain control signals ZTSEL1 and ZTSEL2 are set according to individual variations in the light emission characteristics of the laser diode 21. Further, the capacitance control signals BW_AD1 to BW_AD3 are set according to individual variations in the internal capacitance of the photodiode 22 and the capacitance of the noise removing capacitor.

  Moreover, the control part 16 controls the conduction | electrical_connection state of each switch element 10a-10f. The control unit 16 has an output terminal electrically connected to the control terminals of the switch elements 10a and 10c, and sends a switch control signal RST_P to the control terminals of the switch elements 10a and 10c. In addition, the control unit 16 has an output terminal electrically connected to the control terminals of the switch elements 10b and 10d, and sends a switch control signal RST_B to the control terminals of the switch elements 10b and 10d. The control unit 16 has an output end electrically connected to the control terminals of the switch elements 10e and 10f, and sends a switch control signal LDOFF to the control terminals of the switch elements 10e and 10f. In each of the switch elements 10a to 10f, when the input logic to the control terminal is 0, the input end and the output end are in a conductive state. When the input logic to the control terminal is 1, the input terminal and the output terminal are in a non-conductive state, and the output terminal is short-circuited with the reference potential line 15.

  FIG. 4 is a circuit diagram showing in detail an example of the configuration of the modulation current generator 3 of the present embodiment. Referring to FIG. 4, the modulation current generator 3 includes a pair of transistors 31a and 31b, a pair of resistance elements 32a and 32b, and a modulation current source 33.

  The base of the transistor 31a is electrically connected to an input terminal for inputting the transmission signal Sp (positive phase) from the outside of the modulation current generator 3. The base of the transistor 31b is electrically connected to an input terminal for inputting the transmission signal Sn (reverse phase) from the outside of the modulation current generator 3. The emitters of both the transistors 31 a and 31 b are electrically connected to the reference potential line 15 via the modulation current source 33. The collector of the transistor 31a is electrically connected to the power supply potential line 14 via the resistance element 32a, and is also electrically connected to the cathode of the laser diode 21 (see FIG. 1). The collector of the transistor 31b is electrically connected to the power supply potential line 14 via the resistance element 32b, and is also electrically connected to the anode of the laser diode 21 (see FIG. 1). The modulation current source 33 has an input terminal for inputting the difference signal Sd_bt from the error amplifier 17b (see FIG. 1), and adjusts the amount of current so that the difference signal Sd_bt becomes small.

When the transmission signals Sp (positive phase) and Sn (reverse phase) that are differential signals are input to the modulation current generator 3 illustrated in FIG. 4, the transmission signal Sp is input to the base of the transistor 31a and the transmission signal Sn is input to the base of the transistor 31b. When the transmission signal Sp is input to the base of the transistor 31a, the modulation current Imod ₂ flows through the laser diode 21. The modulation current Imod ₂ has a current value defined by the modulation current source 33 and flows through the transistor 31a. Further, by transmitting signal Sn is input to the base of the transistor 31b, it flows modulation current Imod ₁ reverse phase to the modulation current Imod _2. The modulation current Imod ₁ is regulated by the modulation current source 33 to have the same amount of current as the modulation current Imod ₂ and flows through the transistor 31b.

  FIG. 5 is a circuit diagram showing in detail an example of the configuration of the current-voltage converter 5a of the present embodiment. Referring to FIG. 5, the current-voltage conversion unit 5 a includes transistors 51 and 52, constant current sources 53 and 54, and a variable resistance unit 57.

The transistor 51, the constant current source 53, and the variable resistor 57 constitute a common base type TIA circuit. In other words, the collector of the transistor 51 is electrically connected to the power supply potential line 14 via the variable resistance portion 57. The emitter of the transistor 51 is electrically connected to the cathode of the photodiode 22 and is also electrically connected to the reference potential line 15 via the constant current source 53. A predetermined common voltage V _CB is applied to the base of the transistor 51. With this configuration, a potential proportional to the amount of photocurrent Imon flowing through the photodiode 22 appears at the collector of the transistor 51. This proportionality coefficient (conversion gain) is determined by the resistance value of the variable resistance portion 57.

  The variable resistance unit 57 includes n columns (n ≧ 2, n = 3 in the present embodiment) of resistance elements 55a to 55c and (n−1) transistors 56b and 56c connected in parallel to each other. Transistors 56b and 56c are connected in series to resistance elements 55b and 55c, respectively. The transistors 56b and 56c are used as switching elements, and for example, MOS-FETs are preferably used. Specifically, one end of the resistance elements 55b and 55c is electrically connected to the power supply potential line 14, and the other end is electrically connected to one terminal of the sources and drains of the transistors 56b and 56c. The other terminal of the sources and drains of the transistors 56 b and 56 c is electrically connected to the collector of the transistor 51. The gates of the transistors 56b and 56c are electrically connected to the control unit 16, and receive gain control signals ZTSEL1 and ZTSEL2 from the control unit 16. The resistance value as the variable resistance unit 57 is changed by variously changing the conduction / non-conduction state between the source and drain of the transistors 56b and 56c in accordance with the gain control signals ZTSEL1 and ZTSEL2.

  The transistor 52 and the constant current source 54 constitute an emitter follower circuit. That is, since the above-described common base type TIA circuit has a large output impedance, the emitter follower circuit reduces the output impedance of the current-voltage converter 5a. Specifically, the base of the transistor 52 is electrically connected to the collector of the transistor 51, the collector of the transistor 52 is electrically connected to the power supply potential line 14, and the emitter of the transistor 52 is a constant current source 54. Is electrically connected to the reference potential line 15. Then, the potential at the emitter of the transistor 52 is output as the light amount signal Smon.

  As shown in FIG. 5, the optical module 2 of the present embodiment may further include a capacitor 23 for removing crosstalk noise connected in parallel to the photodiode 22.

  FIG. 6 is a block diagram illustrating an example of the configuration of the reference current generation unit 8 of the present embodiment. Referring to FIG. 6, the reference current generation unit 8 includes a low level reference voltage generation unit 81, a bias component generation unit 82, a high level reference voltage generation unit 83, a switch element 84, and a modulation component generation unit 85.

  The low level reference voltage generation unit 81 is a circuit for generating a reference voltage Vrefl indicating the low level magnitude of the reference current Iref. The magnitude of this reference voltage Vrefl is set to an appropriate value in advance. The bias component generation unit 82 converts the reference voltage Vrefl from the low level reference voltage generation unit 81 into a current, and outputs it as a current Irefl indicating the low level (bias component) of the reference current Iref.

  The high level reference voltage generation unit 83 is a circuit for generating a reference voltage Vrefh indicating the magnitude of the high level (the peak value of the modulation component) of the reference current Iref. The magnitude of the reference voltage Vrefh is set to an appropriate value in advance as in the case of the reference voltage Vrefl. The modulation component generation unit 85 converts the reference voltage Vrefh from the high level reference voltage generation unit 83 into a current, and modulates the current with the transmission signals Sp and Sn. The modulation component generator 85 outputs the current Irefh thus generated as a modulation component of the reference current Iref. The current Irefh and the current Irefl are overlapped with each other and provided as the reference current Iref from the reference current generator 8 to the current-voltage converter 5b (see FIG. 1).

  The high level reference voltage generator 83 is connected to the input terminal of the switch element 84, and the modulation component generator 85 is connected to the output terminal of the switch element 84. A control terminal of the switch element 84 is connected to the control unit 16 and receives a switch control signal RST_B from the control unit 16. In the switch element 84, when the input logic to the control terminal is 0, the input end and the output end are in a conductive state. When the input logic to the control terminal is 1, the input terminal and the output terminal are in a non-conductive state, and the output terminal is short-circuited with the reference potential line 15. With this configuration, when the logic of the switch control signal RST_B is 1, the modulation component (current Irefh) of the reference current Iref is not generated, and only the bias component (current Irefl) is output from the reference current generator 8 as the reference current Iref. The Further, when the logic of the switch control signal RST_B is 0, a modulation component (current Irefh) of the reference current Iref is generated, and is output from the reference current generation unit 8 as the reference current Iref together with the bias component (current Irefl).

  FIG. 7 is a circuit diagram showing in detail an example of the configuration of the variable capacitance section 9 of the present embodiment. FIG. 7 also shows in detail an example of the configuration of the modulation component generation unit 85 and an example of the configuration of the current-voltage conversion unit 5b. Among these, since the configuration of the current-voltage conversion unit 5b is the same as the configuration of the current-voltage conversion unit 5a described above, the description thereof is omitted.

  Referring to FIG. 7, the variable capacitance unit 9 is connected to capacitors (capacitance elements) 91 a to 91 c and capacitors 91 a to 91 c in m columns (m ≧ 2, in this embodiment, m = 3) connected in parallel to each other. N transistors 92a to 92c connected in series. The transistors 92a to 92c are used as switching elements, and for example, MOS-FETs are preferably used. Specifically, one ends of the capacitors 91a to 91c are electrically connected to the signal line 18 that connects the bias component generation unit 82, the modulation component generation unit 85, and the current-voltage conversion unit 5b. The other ends of the capacitors 91a to 91c are electrically connected to one terminal of the sources and drains of the transistors 92a to 92c, respectively. The other terminal of the sources and drains of the transistors 92 a to 92 c is electrically connected to a constant potential line such as the reference potential line 15. The gates of the transistors 92a to 92c are connected to the control unit 16 and receive capacitance control signals BW_AD1 to BW_AD3 from the control unit 16, respectively. The capacitance value as the variable capacitance unit 9 is changed by variously changing the conduction / non-conduction state between the source and drain of the transistors 92a to 92c according to the capacitance control signals BW_AD1 to BW_AD3.

For example, as shown in FIG. 7, the modulation component generation unit 85 includes a transistor 86, a pair of transistors 87 a and 87 b, and a constant current source 88. The transmission signal Sp is input to the base of the transistor 87a, and the transmission signal Sn is input to the base of the transistor 87b. The emitters of both the transistors 87 a and 87 b are electrically connected to the reference potential line 15 through a constant current source 88. The collector of the transistor 87a is electrically cascode-connected to the emitter of the transistor 86. The collector of the transistor 87b is electrically connected to the current / voltage converter 5b via the signal line 18. A bias component generation unit 82 is also connected to the signal line 18. The constant current source 88 has an input terminal for inputting the reference voltage Vrefh from the high level reference voltage generation unit 83, and defines the amount of current according to the reference voltage Vrefh. The collector of the transistor 86 is electrically connected to the power supply potential line 14, and a predetermined common voltage V _CB is applied to the base of the transistor 86. The transistor 86 is provided (as a pair) corresponding to the transistor 51 of the current-voltage converter 5b.

  The operation of the light emitting element driving circuit 1a having the above configuration will be described with reference to FIGS. 8A to 8E are respectively the reference voltage Vrefl (FIG. 8A) in the low level reference voltage generation unit 81 (see FIG. 6) and the high level reference voltage generation unit 83 (see FIG. 6). Time of the reference voltage Vrefh (FIG. 8B), the reference signal Vref (FIG. 8C), the peak level reference voltage Vpk (FIG. 8D), and the bottom level reference voltage Vbt (FIG. 8E) It is a graph which shows an example of a waveform. Each of FIGS. 9A to 9C includes a switch control signal LDOFF output from the control unit 16 (FIG. 9A), a switch control signal RST_B (FIG. 9B), and a switch control signal. It is a graph which shows an example of the time waveform of RST_P (FIG.9 (c)). Each of FIGS. 10A to 10D includes a light amount signal Smon (FIG. 10A), a peak level signal Spk (FIG. 10B), a bottom level signal Sbt (FIG. 10C), 11 is a graph showing an example of a time waveform of the drive current Id (FIG. 10D).

First, the reference voltage Vrefl from the low level reference voltage generator 81 rises at time _{t 1} (FIG. 8 (a)). At this time, the switch control signal RST_B from the control unit 16 is logic 1 (FIG. 9B), and the switch element 84 (FIG. 6) is not connected, so only the bias component (Irefl) is used as the reference current Iref. Is output from the reference current generator 8. Therefore, only the bias component Vrefb is output from the reference signal generator 13 as the reference signal Vref as shown in FIG. In the present embodiment, since the reference current Iref and the reference signal Vref are reversed from each other in the current-voltage converter 5b to form a complementary time waveform, the actual reference signal Vref is biased from the power supply voltage Vcc (Vrefb). The voltage value is reduced only by this.

  At this stage, since the modulation component is not included in the reference signal Vref, the peak level reference voltage Vpk and the bottom level reference voltage Vbt converge to the same level as the bias component Vrefb of the reference signal Vref (FIG. 8D). ), FIG. 8 (e)).

Subsequently, the switch control signal LDOFF and RST_P from the control unit 16 at time _{t 2} is switched to a logic 0 (FIG. 9 (a), the FIG. 9 (b)). That is, the switch elements 10a, 10c, 10e, and 10f (see FIG. 1) are connected. As a result, the peak level reference voltage Vpk is input to the error amplifier 17a, and the difference signal Sd_pk from the error amplifier 17a is input to the bias current source 4, thereby forming a closed loop circuit relating to the bias current Ibias. Then, the supply of the bias current Ibias is started by the bias current source 4 so that the difference signal Sd_pk becomes small (that is, the magnitude of the peak level signal Spk approaches the peak level reference voltage Vpk). Since the current value is adjusted, the current value of the bias current Ibias approaches the magnitude (Irefl) of the reference current Iref (FIGS. 10B and 10D).

Subsequently, at time _{t 3} after a predetermined time has elapsed from time _{t 2, the} switch control signal RST_B from the control unit 16 is switched to a logic 0 (FIG. 9 (c)). That is, the switch elements 10b and 10d (FIG. 1) and 84 (FIG. 6) are connected.

  At this time, the reference current generator 8 outputs not only the bias component (Irefl) but also the modulation component (Irefh) as the reference current Iref. Therefore, the reference signal generator 13 outputs the modulation component Vrefm as the reference signal Vref in addition to the bias component Vrefb as shown in FIG. 8C. That is, the bottom level reference voltage Vbt indicating the bottom level of the reference signal Vref is the sum of the peak value of the modulation component Vrefm and the bias component Vrefb as shown in FIG.

  Further, the bottom level reference voltage Vbt is input to the error amplifier 17b, and the difference signal Sd_bt from the error amplifier 17b is input to the modulation current generation unit 3, so that a closed loop circuit relating to the modulation current Imod is formed. Then, the modulation current generator 3 starts supplying the modulation current Imod so that the difference signal Sd_bt becomes small (that is, the magnitude of the bottom level signal Sbt approaches the bottom level reference voltage Vbt). Current value is adjusted. As a result, the magnitude of the drive current Id obtained by combining the bias current Ibias and the modulation current Imod is controlled according to the magnitude of the reference current Iref (FIG. 10D).

  The effect which the light emitting element drive circuit 1a of this embodiment has is demonstrated. FIGS. 11A and 11B are diagrams for explaining the effect of the light emitting element driving circuit 1a. FIG. 11A is a circuit diagram showing the current-voltage converter 5a of the light emitting element driving circuit 1a. FIG. 11B is a circuit diagram showing an example of a configuration of a current-voltage conversion circuit having a negative feedback type TIA circuit as a comparative example. Note that the capacitance component Cin shown in the circuit diagrams of FIGS. 11A and 11B includes the capacitance of the noise removing capacitor 23 (see FIG. 5) provided in parallel with the photodiode 22 and the parasitic characteristics of the photodiode 22. A combined capacity with an internal capacity such as a capacity is shown.

  Here, the current-voltage conversion circuit 7 having the negative feedback type TIA circuit shown in FIG. 11B will be described. The current-voltage conversion circuit 7 includes transistors 71 and 72, resistance elements 73a to 73c, and a variable resistance unit 74. The collector of the transistor 71 is connected to the power supply potential line 14 via the resistance element 73 a, and the emitter is connected to the reference potential line 15. The base of the transistor 71 is connected to the cathode of the photodiode 22 and is connected to the emitter of the transistor 72 via the variable resistance unit 74. The collector of the transistor 72 is connected to the power supply potential line 14 via the resistance element 73b, the base is connected to the collector of the transistor 71, and the emitter is connected to the reference potential line 15 via the resistance element 73c. ing.

  In the current-voltage conversion circuit 7, when the photocurrent Imon flowing through the photodiode 22 passes through the variable resistor 74, a base current proportional to the current amount of the photocurrent Imon flows out from the base of the transistor 71, thereby Shows a potential Vb having an opposite phase to the base current. A potential having a phase opposite to that of the potential Vb appears at the collector of the transistor 71, a potential having the same phase as the potential Vb appears on the collector side of the transistor 72, and a potential opposite to the potential Vb appears on the emitter side. The emitter potential having a phase opposite to that of the potential Vb of the transistor 72 is negatively fed back via the variable resistor 74, thereby operating as a negative feedback transimpedance amplifier. The collector potential of the transistor 72 is taken out as a light amount signal Smon. The proportionality coefficient (conversion gain) of the current-voltage conversion circuit 7 is determined by the resistance value of the variable resistance unit 74.

The current-voltage conversion unit 5a of this embodiment shown in FIG. 11A is a common base type as described above, and the input impedance Zin and the transimpedance ZT are expressed by the following equations (1) and (2), respectively. expressed. Further, the cutoff frequency fc by the input impedance Zin and the capacitance component Cin is expressed by the following mathematical formula (3). In Equations (1) to (3), g _m is the mutual conductance of the transistor 51, and RTIA is the resistance value of the variable resistance unit 57.

As shown in the above equation (1), in the current-voltage conversion unit 5a of the present embodiment, the input impedance Zin is dependent only on the transconductance g _m of the transistor 51, determined according to the emitter current flowing through the transistor 51 . Accordingly, the cutoff frequency fc, will depend only on the capacitive component Cin and transconductance g _m As shown in the above equation (3). Further, as shown in the above equation (2), the transimpedance ZT of the current-voltage converter 5a is determined only by the resistance value RTIA of the variable resistor 57. Therefore, even if the resistance value RTIA of the variable resistor 57 is changed in order to change the transimpedance ZT, the cutoff frequency fc hardly changes. Conversely, even if the capacitance component Cin is changed to change the cutoff frequency fc, the transimpedance ZT hardly changes.

In contrast, in the current-voltage conversion circuit 7 having the negative feedback type TIA circuit shown in FIG. 11B, the input impedance Zin and the transimpedance ZT are expressed by the following equations (4) and (5), respectively. The Further, the cutoff frequency fc by the input impedance Zin and the capacitance component Cin is expressed by the following mathematical formula (6). In Equations (4) to (6), Ao is an open loop gain, and RTIA is a resistance value of the variable resistance unit 74.

  As shown in the above equation (4), in the current-voltage conversion circuit 7 having the negative feedback type TIA circuit, the input impedance Zin depends on the open loop gain Ao. Further, as shown in the above equation (5), the transimpedance ZT of the current-voltage conversion circuit 7 depends on the resistance value RTIA and the open loop gain Ao of the variable resistance unit 74. Accordingly, the cutoff frequency fc depends on the capacitance component Cin and the transimpedance ZT as shown in the above equation (6). For this reason, if the resistance value RTIA of the variable resistance unit 74 is changed to change the transimpedance ZT, the cutoff frequency fc also changes.

  Originally, the common base type TIA circuit (FIG. 11A) easily picks up noise as compared with the negative feedback type TIA circuit (FIG. 11B), and thus is not suitable for the light emitting element driving circuit. However, as described above, the present inventors have made the transimpedance resistance element (variable resistance portion 57) and the capacitance component Cin (the internal capacitance of the photodiode 22 and the capacitance of the parallel capacitor 23) of the common base type TIA circuit. Since the low-pass filter is not configured and the transimpedance ZT is changed, the high-frequency cut-off frequency fc hardly changes. Therefore, the level detection accuracy in the level hold unit 6 does not depend on the transimpedance ZT, and the signal is accurately obtained. It was found that the light intensity and extinction ratio can be controlled.

  That is, according to the light emitting element driving circuit 1a and the optical transmitter according to the present embodiment, the current-voltage conversion unit 5a includes the common base type TIA circuit, and the transimpedance ZT (variable resistance unit 57) is variable. The conversion gain at the time of converting the photocurrent Imon from the photodiode 22 into the light amount signal Smon can be adjusted, and the influence on the frequency characteristics (cutoff frequency fc) due to the adjustment of the conversion gain can be reduced. This also facilitates band correction when adjusting the conversion gain in accordance with the characteristics of the individual laser diodes 21 and keeps the bandwidth constant.

  Further, as in the present embodiment, the light emitting element driving circuit 1a is provided with gain control signals ZTSEL1 and ZTSEL2 for changing the transimpedance ZT of the common base type TIA circuit to the current-voltage converters 5a and 5b. It is preferable to include a control unit (control unit 16). Thereby, the conversion gain of the current-voltage conversion part 5a can be easily changed according to the dispersion | variation in the characteristic of the laser diode 21. FIG.

  Further, as in the present embodiment, the reference signal generation unit 13 includes a reference current generation unit 8 and a current-voltage conversion unit 5b, and the current-voltage conversion unit 5b is a common base having a variable transimpedance like the current-voltage conversion unit 5a. By having the type TIA circuit, when the conversion characteristics such as the gain of the current-voltage converter 5a change due to a temperature change or the like, the conversion characteristics of the current-voltage converter 5b also change in the same way, so that the change in characteristics can be offset . Further, by having the variable capacitance unit 9 between the signal line 18 connecting the reference current generation unit 8 and the current-voltage conversion unit 5 b and the reference potential line 15, the frequency of the light amount signal Smon caused by the internal capacitance of the photodiode 22. Frequency characteristics similar to the characteristics can be given to the reference signal Vref, and variations in internal capacitance of the individual photodiodes 22 can be easily handled. Therefore, according to the light emitting element drive circuit 1a of the present embodiment, the amount of drive current Id can be controlled with higher accuracy.

  Further, as in the present embodiment, the light emitting element driving circuit 1a is provided with a capacitance control unit (control unit 16) for providing the variable capacitance unit 9 with capacitance control signals BW_AD1 to BW_AD3 for changing the capacitance of the variable capacitance unit 9. ). Thereby, the capacitance value of the variable capacitance section 9 can be easily changed according to the variation in the internal capacitance of the photodiode 22.

(Second Embodiment)
FIG. 12 is a block diagram showing a configuration of a light emitting element driving circuit 1b according to another preferred embodiment of the present invention. A light-emitting element driving circuit 1b shown in the figure is a circuit for driving the optical module 2, and constitutes the optical transmitter according to the present embodiment together with the optical module 2. In addition, since the structure of the optical module 2 of this embodiment is the same as that of the said 1st Embodiment, detailed description is abbreviate | omitted.

  The light emitting element driving circuit 1 b includes a modulation current generation unit 103 and a bias current generation unit 104. The modulation current generation unit 103 and the bias current generation unit 104 constitute a drive current generation unit 119 in the present embodiment. The modulation current generator 103 is electrically connected to the cathode of the laser diode 21 via the capacitor 102. The bias current generation unit 104 is electrically connected to the cathode of the laser diode 21 via the inductor 101. In the present embodiment, the anode of the laser diode 21 is electrically connected to the power supply potential line 14.

  The bias current generator 104 supplies a forward bias current Ibias that is a bias component of the drive current Id supplied to the laser diode 21. The bias current generation unit 104 has a control end 104a. As will be described later, the magnitude of the bias current Ibias is adjusted according to the difference signal Sd_ave provided to the control terminal 104a. The input terminal 103a of the modulation current generator 103 is electrically connected to the signal input terminal 115 of the light emitting element driving circuit 1b. The signal input terminal 115 is supplied with a transmission signal Sin from an external signal generator (not shown). The modulation current generator 103 receives the transmission signal Sin through the input terminal 103a, generates a modulation current Imod corresponding to the transmission signal Sin at the output terminal 103b, and superimposes it on the bias current Ibias. The modulation current Imod is a modulation component of the drive current Id supplied to the laser diode 21 and has a frequency equal to the data transmission speed of the transmission signal Sin. The modulation current generator 103 further has a control terminal 103c. As will be described later, the amplitude of the modulation current Imod is adjusted according to the difference signal Sd_amp provided to the control terminal 103c. The inductor 101 prevents the AC modulation current Imod from flowing into the bias current generation unit 104, and the capacitor 102 prevents the DC bias current Ibias from flowing into the modulation current generation unit 103.

  The cathode of the photodiode 22 is electrically connected to the power supply potential line 14 for applying a reverse bias voltage to the photodiode 22. The anode of the photodiode 22 is electrically connected to the input terminal 105 a of the current-voltage conversion unit 105. The current-voltage conversion unit 105 is a first current-voltage conversion unit in the present embodiment, and has a circuit configuration similar to that of the current-voltage conversion unit 5a of the first embodiment. That is, the current-voltage conversion unit 105 includes a common base type TIA circuit (see FIG. 5), and is configured to be able to change the transimpedance of the common base type TIA circuit. The current-voltage converter 105 receives the photocurrent Imon from the photodiode 22 and converts it into a light amount signal Smon that is a voltage signal. This light quantity signal Smon is generated at the output terminal 105 b of the current-voltage converter 105.

  Further, the current-voltage conversion unit 105 has a control end 105c. The control end 105c is electrically connected to the control unit 130. The transimpedance of the current-voltage conversion unit 105 is changed according to the gain control signal ZTSEL provided from the control unit 130 to the control terminal 105c.

  The output terminal 105 b is electrically connected to the input terminal 106 a of the average value detection unit 106. The average value detector 106 receives the light amount signal Smon from the current-voltage converter 105 and detects the average value of the light amount signal Smon. This average value is equivalent to the average value of the photocurrent Imon from the photodiode 22, and therefore equivalent to the average value of the signal light intensity of the laser diode 21. The average value detection unit 106 generates an output voltage corresponding to the detected average value at the output terminal 106b.

  The output terminal 106b is electrically connected to the non-inverting input terminal 108a of the error amplifier 108. Therefore, the output voltage from the average value detection unit 106 is provided to the non-inverting input terminal 108a. The inverting input terminal 108 b of the error amplifier 108 is electrically connected to the reference voltage generation unit 107. The reference voltage generation unit 107 generates a predetermined reference voltage and provides it to the inverting input terminal 108b. The error amplifier 108 generates a voltage signal (difference signal Sd_ave) corresponding to the difference between the output voltage from the average value detection unit 106 and the reference voltage from the reference voltage generation unit 107 at the output terminal 108c. The output terminal 108 c is connected to the control terminal 104 a of the bias current generator 104, and the difference signal Sd_ave is provided to the control terminal 104 a of the bias current generator 104.

  The bias current generation unit 104 adjusts the magnitude of the bias current Ibias according to the difference signal Sd_ave from the error amplifier 108. Specifically, the bias current generation unit 104 performs negative feedback control on the magnitude of the bias current Ibias so that the difference signal Sd_ave is stabilized. Thus, the light emitting element driving circuit 1b adjusts the bias current Ibias according to the average value of the signal light intensity of the laser diode 21, and stabilizes the average intensity of the signal light P.

  The output terminal 105 b of the current-voltage conversion unit 105 is further electrically connected to the amplitude detection unit 110 via the low-pass filter 109. The low-pass filter 109 filters the light amount signal Smon generated by the current-voltage conversion unit 105 and limits the frequency band of the light amount signal Smon. The low-pass filter 109 has a predetermined cutoff frequency, and mainly attenuates a component having a frequency higher than the cutoff frequency in the light amount signal Smon. In the present embodiment, the low pass filter 109 has a cutoff frequency that is lower than the data transmission rate of the transmission signal Sin.

  The light amount signal Smon filtered by the low pass filter 109 is supplied to the input terminal 110 a of the amplitude detector 110. The amplitude detector 110 detects the amplitude of the light quantity signal Smon. An output voltage corresponding to the detected amplitude is generated at the output terminal 110b of the amplitude detector 110. The output terminal 110b is electrically connected to the non-inverting input terminal 111a of the error amplifier 111. Therefore, the output voltage from the amplitude detector 110 is provided to the non-inverting input terminal 111a.

  The inverting input terminal 111 b of the error amplifier 111 is electrically connected to the output terminal 114 b of the amplitude detection unit 114. The input end 114 a of the amplitude detection unit 114 is electrically connected to the output end 112 b of the reference amplitude generation unit 112 via the low pass filter 113. The input terminal 112a of the reference amplitude generator 112 is electrically connected to the signal input terminal 115 of the light emitting element driving circuit 1b.

  The reference amplitude generator 112 receives the transmission signal Sin through the signal input terminal 115 and generates the reference signal Vref at the output terminal 112b. This reference signal Vref is a pulse signal corresponding to the transmission signal Sin. That is, the reference signal Vref has an amplitude corresponding to the amplitude of the transmission signal Sin. The reference signal Vref has a rising edge and a falling edge corresponding to the vibration of the transmission signal Sin. The reference signal Vref indicates the same data as the transmission signal Sin. For example, the reference amplitude generation unit 112 may generate the reference signal Vref by amplifying the transmission signal Sin.

  The low-pass filter 113 has substantially the same frequency characteristics and cut-off frequency as the low-pass filter 109. The low-pass filter 113 filters the reference signal Vref and limits the frequency band of the reference signal Vref. The low-pass filter 113 mainly attenuates a component having a frequency higher than the cutoff frequency in the reference signal Vref. The reference signal Vref filtered by the low pass filter 113 is provided to the input terminal 114a of the amplitude detector 114. The amplitude detector 114 detects the amplitude of the reference signal Vref. An output voltage corresponding to the detected amplitude is generated at the output end 114 b of the amplitude detector 114. As a result, the output voltage of the amplitude detector 114 is provided to the inverting input terminal 111 b of the error amplifier 111.

  The error amplifier 111 generates a voltage signal (difference signal Sd_amp) corresponding to the difference between the output voltages of the amplitude detectors 110 and 114 at the output terminal 111c. The output end 111 c is electrically connected to the control end 103 c of the modulation current generation unit 103, and the difference signal Sd_amp is provided to the control end 103 c of the modulation current generation unit 103.

  The modulation current generator 103 adjusts the amplitude of the modulation current Imod according to the difference signal Sd_amp. Specifically, the modulation current generator 103 performs negative feedback control on the amplitude of the modulation current Imod so that the difference signal Sd_amp is stabilized. As described above, the light emitting element driving circuit 1b adjusts the amplitude of the modulation current Imod according to the amplitude of the signal light intensity of the laser diode 21, and stabilizes the extinction ratio of the signal light P. Thereby, the unstable operation | movement of the laser diode 21 resulting from the individual dispersion | variation in the laser diode 21 and a temperature fluctuation can be prevented.

  According to the light emitting element driving circuit 1b and the optical transmitter according to the present embodiment, the current-voltage conversion unit 105 includes the common base type TIA circuit, and the transimpedance is variable, whereby the light emitting element driving of the first embodiment is performed. Similar to the circuit 1a, the conversion gain when the photocurrent Imon from the photodiode 22 is converted into the light amount signal Smon can be adjusted, and the influence on the frequency characteristics (cutoff frequency) due to the adjustment of the conversion gain can be reduced. This also facilitates band correction when adjusting the conversion gain in accordance with the characteristics of the individual laser diodes 21 and keeps the bandwidth constant.

  Further, in the light emitting element driving circuit 1 b according to the present embodiment, the low-pass filter 109 is installed between the current-voltage conversion unit 105 and the amplitude detection unit 110. Since the low-pass filter 109 limits the band of the output voltage signal (light quantity signal Smon) of the current-voltage converter 105, crosstalk noise in the light quantity signal Smon due to the drive current Id is reduced. Therefore, the amplitude detector 110 can accurately detect the amplitude of the light quantity signal Smon. As a result, the amplitude of the signal light intensity of the laser diode 21 can be accurately monitored, and the amplitude of the modulation current Imod can be adjusted according to the amplitude to appropriately stabilize the extinction ratio of the signal light P.

(First modification)
FIG. 13 is a block diagram showing a configuration of a light emitting element driving circuit according to a modification of the second embodiment. The light emitting element driving circuit 1c includes a filter control unit 116 in addition to the components of the light emitting element driving circuit 1b shown in FIG. Hereinafter, differences between the light emitting element driving circuit 1c of the present modification and the light emitting element driving circuit 1b of the second embodiment will be described.

  In this modification, the cut-off frequencies of the low-pass filters 109 and 113 are variable. The filter control unit 116 controls the cutoff frequency of the low-pass filters 109 and 113. The low-pass filters 109 and 113 have control ends 109c and 113c, respectively, and the control ends 109c and 113c are electrically connected to the output end 116b of the filter control unit 116. The filter control unit 116 sends control signals from the output end 116b to the low-pass filters 109 and 113, and sets their cutoff frequencies. Data transmission rate information Dtr of the transmission signal Sin is sent from an external circuit to the input terminal 116a of the filter control unit 116.

  The filter control unit 116 controls the cutoff frequencies of the low-pass filters 109 and 113 to be lower than the data transmission rate according to the data transmission rate information Dtr. In addition, the filter control unit 116 sets the cutoff frequencies of the low-pass filters 109 and 113 to the same value. As a result, even if the data transmission rate of the transmission signal Sin changes, the cutoff frequencies of the low-pass filters 109 and 113 do not exceed the data transmission rate. Therefore, crosstalk noise can be reduced regardless of the data transmission speed, and the extinction ratio of the signal light P can be satisfactorily stabilized. In addition, the cutoff frequency can be optimized according to the data transmission rate. As a result, the light emitting element driving circuit 1c can operate under a wide range of data transmission speeds.

(Second modification)
FIG. 14 is a block diagram showing a configuration of a light emitting element driving circuit according to another modification of the second embodiment. The light emitting element drive circuit 1d according to the present modification includes a reference current generation unit 121, a current-voltage conversion unit 122, and a variable capacitance unit 123 instead of the reference amplitude generation unit 112 in the light emission element drive circuit 1b. The reference current generator 121, the current-voltage converter 122, and the variable capacitor 123 constitute a reference signal generator in the present embodiment, and are electrically connected to each other.

  The reference current generation unit 121 has an input terminal 121a that is electrically connected to the signal input terminal 115 of the light emitting element driving circuit 1d. The reference current generator 121 receives the transmission signal Sin through the signal input terminal 115, and generates a reference current corresponding to the transmission signal Sin at the output terminal 121b. This reference current is a pulse signal corresponding to the transmission signal Sin. That is, the reference current has an amplitude corresponding to the amplitude of the transmission signal Sin. The reference current has a rising edge and a falling edge corresponding to the vibration of the transmission signal Sin. The reference current indicates the same data as the transmission signal Sin.

  The output terminal 121 b of the reference current generation unit 121 is electrically connected to the input terminal 122 a of the current / voltage conversion unit 122. The current-voltage conversion unit 122 is a second current-voltage conversion unit in the present embodiment, and has a circuit configuration similar to that of the current-voltage conversion unit 105. That is, the current-voltage converter 122 includes a common base type TIA circuit (see FIG. 5), and is configured to be able to change the transimpedance of the common base type TIA circuit. The current-voltage converter 122 receives the reference current generated by the reference current generator 121 and converts it into a reference signal Vref that is a voltage signal. The reference signal Vref is generated at the output terminal 122b of the current-voltage converter 122.

  The current-voltage conversion unit 122 has a control end 122c. The control end 122c is electrically connected to the output end 130a of the control unit 130. The transimpedance of the current-voltage conversion unit 122 is changed according to a gain control signal ZTSEL provided from the output terminal 130a of the control unit 130 to the control terminal 122c.

  An output end 122 b of the current-voltage conversion unit 122 is electrically connected to the low-pass filter 113. The low-pass filter 113 receives the reference signal Vref from the current-voltage converter 122 and filters it. Thereafter, the amplitude detector 114 detects the amplitude of the reference signal Vref and generates an output voltage corresponding to the amplitude. This output voltage is provided to the inverting input terminal 111 b of the error amplifier 111. The error amplifier 111 generates a difference signal Sd_amp corresponding to the difference between the output voltages of the amplitude detectors 110 and 114 at the output terminal 111c. The amplitude of the modulation current Imod in the modulation current generator 103 is negative feedback controlled in accordance with the difference signal Sd_amp from the error amplifier 111, thereby stabilizing the extinction ratio of the signal light P.

  A variable capacitor 123 is connected between the signal line connecting the reference current generator 121 and the current-voltage converter 122 and the reference potential line 15. That is, one end of the variable capacitance unit 123 is electrically connected to a signal line that connects the output end 121 b of the reference current generation unit 121 and the input end 122 a of the current-voltage conversion unit 122. The other end of the variable capacitor 123 is electrically connected to the reference potential line 15. The variable capacitor 123 is provided to correct the influence of the capacitance of the photodiode 22 on the operation of the current-voltage converter 105.

  The variable capacitance unit 123 is electrically connected to the output end 130b of the control unit 130 so that the capacitance value can be changed according to the capacitance control signal BW_AD output from the output end 130b of the control unit 130. It is configured. For example, the variable capacitor 123 may have the same configuration as the variable capacitor 9 of the first embodiment shown in FIG.

  The light emitting element driving circuit 1d further includes an average value detection unit 117. The average value detection unit 117 preferably has the same configuration as the average value detection unit 106. An input terminal 117 a of the average value detection unit 117 is electrically connected to the low-pass filter 113, and an output terminal 117 b is electrically connected to the inverting input terminal 108 b of the error amplifier 108. The error amplifier 108 generates a difference signal Sd_ave corresponding to the difference between the output voltages from the average value detection units 106 and 117 at the output terminal 108c. The bias current generation unit 104 performs negative feedback control on the magnitude of the bias current Ibias according to the difference signal Sd_ave. Thereby, the average light intensity of the signal light P is stabilized.

  According to the light emitting element driving circuit 1d of the present modification, in addition to the effects of the light emitting element driving circuit 1b of the second embodiment, the following effects can be further obtained. That is, like the light emitting element driving circuit 1 d of this modification, the reference signal generation unit includes the reference current generation unit 121 and the current-voltage conversion unit 122, and the current-voltage conversion unit 122 is similar to the current-voltage conversion unit 105. By having the impedance-variable common base type TIA circuit, when the conversion characteristic such as the gain of the current-voltage conversion unit 105 changes due to a temperature change or the like, the conversion characteristic of the current-voltage conversion unit 122 also changes similarly. Fluctuations can be offset. In addition, since the variable capacitance unit 123 is provided between the signal line connecting the reference current generation unit 121 and the current-voltage conversion unit 122 and the reference potential line 15, the frequency characteristic of the light amount signal Smon caused by the internal capacitance of the photodiode 22 is obtained. The same frequency characteristic as that of the reference signal Vref can be applied to the reference signal Vref, and variations in internal capacitance of the individual photodiodes 22 can be easily handled. Therefore, according to the light emitting element drive circuit 1d of the present modification, the amount of drive current Id can be controlled with higher accuracy.

  Further, as in the present modification, the light emitting element drive circuit 1 d includes a capacitance control unit (control unit 130) for providing the variable capacitance unit 123 with a capacitance control signal BW_AD for changing the capacitance of the variable capacitance unit 123. It is preferable to provide. Thereby, the capacitance value of the variable capacitance unit 123 can be easily changed according to the variation in the internal capacitance of the photodiode 22.

(Third Modification)
FIG. 15 is a block diagram showing a configuration of a light emitting element driving circuit according to another modification of the second embodiment. The light emitting element drive circuit 1e of the present modification has a structure in which the current-voltage converter and the low-pass filter in the second modification are replaced with a single circuit portion. That is, the light emitting element driving circuit 1e includes the current / voltage converter 105 and the low-pass filter 109 in FIG. 14 as the current-voltage converter 124 with a filter function, and the current-voltage converter 122 and the low-pass filter 113 as the current-voltage converter 125 with a filter function. Each has a structure replaced.

  These current-voltage converters 124 and 125 with a filter function have the same circuit structure, and therefore have substantially the same operating characteristics. Further, the current-voltage conversion units 124 and 125 with a filter function include a common base type TIA circuit, and are configured to be able to change the transimpedance of the common base type TIA circuit. The input terminal 124 a of the current-voltage converter 124 with a filter function is electrically connected to the anode of the photodiode 22, and the output terminal 124 b is an input terminal 106 a of the average value detection unit 106 and an input terminal of the amplitude detection unit 110. 110a is electrically connected. Further, the input terminal 125a of the current-voltage conversion unit with filter function 125 is electrically connected to the output terminal 121b of the reference current generation unit 121, and the output terminal 125b includes the input terminal 117a and the amplitude of the average value detection unit 117. The detector 114 is electrically connected to the input end 114a.

  Moreover, the current-voltage converters 124 and 125 with a filter function have control ends 124c and 125c, respectively. The control ends 124 c and 125 c are electrically connected to the output end 130 a of the control unit 130. The transimpedance of the current-voltage conversion units 124 and 125 with the filter function is changed according to the gain control signal ZTSEL provided from the output terminal 130a of the control unit 130 to the control terminals 124c and 125c.

  The operations of the current-voltage converters 124 and 125 with a filter function are the same as the operations of the current-voltage converter and the low-pass filter in the second modification. Therefore, the light emitting element drive circuit 1e of this modification has the same effect as the light emitting element drive circuit 1d of the second modification.

(Fourth modification)
FIG. 16 is a block diagram showing a configuration of a light emitting element driving circuit according to another modification of the second embodiment. The light emitting element driving circuit 1f of this modification is a differential AC coupled light emitting element driving circuit. That is, the light emitting element drive circuit 1f includes a differential modulation current generation unit 126 instead of the modulation current generation unit 103 (FIG. 12) of the second embodiment. The modulation current generator 126 has two output ends 126b and 126d. A modulation current Imod ₁ having the same phase as that of the transmission signal Sin is output from _{one of the} output terminals 126b and 126d, and a modulation current Imod ₂ having a phase opposite to that of the transmission signal Sin is output from the other.

In the present modification, the anode of the laser diode 21 is electrically connected to the power supply potential line 14 via the inductor 101a, and is electrically connected to the output end 126b of the modulation current generator 126 via the capacitor 102a. Connected to. The cathode of the laser diode 21 is electrically connected to the bias current generation unit 104 via the inductor 101b, and is also electrically connected to the output terminal 126c of the modulation current generation unit 126 via the capacitor 102b. The difference between the modulation currents Imod ₁ and Imod ₂ output from the two output terminals 126b and 126c is superimposed on the bias current Ibias and supplied to the laser diode 21 as the drive current Id. As a result, the laser diode 21 generates signal light P corresponding to the transmission signal Sin. The modulation current generator 126 further has a control terminal 126c. The amplitudes of the modulation currents Imod ₁ and Imod ₂ are adjusted according to the difference signal Sd_amp provided to the control terminal 126c.

  The light emitting element drive circuit 1f processes the photocurrent Imon from the photodiode 22 using the same circuit structure as the light emitting element drive circuit 1b of the second embodiment, and controls the light intensity of the signal light P. Therefore, the light emitting element driving circuit 1f has the same effect as the light emitting element driving circuit 1b. As described above, the present invention can also be applied to a differential light emitting element driving circuit, and in this case, the same effect as that of a non-differential light emitting element driving circuit can be obtained.

  The light-emitting element driving circuit and the optical transmitter according to the present invention are not limited to the above-described embodiments and modifications, and various modifications can be made. For example, although the light emitting element driving circuit of the differential AC coupling circuit is described in the embodiment of FIG. 1, the light emitting element driving circuit according to the present invention may be a single-phase AC coupling circuit or a DC coupling type. It is good. The embodiment of FIG. 16 is a modification of the embodiment of FIG. 12 to a differential AC coupling circuit, but the light emitting element driving circuit shown in the embodiments of FIGS. 13 to 15 is modified to a differential type. May be.

FIG. 1 is a block diagram showing a configuration of a first embodiment of a light emitting element driving circuit according to the present invention. FIG. 2A is a diagram illustrating an example of output characteristics of a laser diode. FIG. 2B is a graph showing an example of the time waveform of the signal light output from the laser diode. FIG. 2C is a graph showing an example of a time waveform of the drive current necessary for outputting the signal light shown in FIG. FIG. 3 is a graph showing an example of (a) a time waveform of a photocurrent from a photodiode and (b) an example of a time waveform of a light amount signal corresponding to the photocurrent in the first embodiment. FIG. 4 is a circuit diagram illustrating in detail an example of the configuration of the modulation current generation unit of the first embodiment. FIG. 5 is a circuit diagram showing in detail an example of the configuration of the current-voltage conversion unit of the first embodiment. FIG. 6 is a block diagram illustrating an example of the configuration of the reference current generation unit according to the first embodiment. FIG. 7 is a circuit diagram showing in detail an example of the configuration of the variable capacitance section of the first embodiment. FIG. 8 shows (a) a reference voltage Vrefl in the low level reference voltage generator, (b) a reference voltage Vrefh in the high level reference voltage generator, (c) a reference signal Vref, (d) a peak level reference voltage Vpk, and ( e) A graph showing an example of a time waveform of the bottom level reference voltage Vbt. FIG. 9 is a graph showing an example of time waveforms of (a) the switch control signal LDOFF output from the control unit, (b) the switch control signal RST_B, and (c) the switch control signal RST_P. FIG. 10 is a graph showing examples of time waveforms of (a) the light amount signal Smon, (b) the peak level signal Spk, (c) the bottom level signal Sbt, and (d) the drive current Id. FIG. 11A is a circuit diagram illustrating a current-voltage conversion unit of the light emitting element driving circuit according to the first embodiment. FIG. 11B is a circuit diagram showing an example of a configuration of a current-voltage conversion circuit having a negative feedback type TIA circuit as a comparative example. FIG. 12 is a block diagram showing a configuration of the second embodiment of the light-emitting element driving circuit according to the present invention. FIG. 13 is a block diagram showing a configuration of a light emitting element driving circuit according to the first modification. FIG. 14 is a block diagram showing a configuration of a light emitting element driving circuit according to a second modification. FIG. 15 is a block diagram showing a configuration of a light emitting element driving circuit according to a third modification. FIG. 16 is a block diagram showing a configuration of a light emitting element driving circuit according to a fourth modification.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1a-1f ... Light emitting element drive circuit, 2 ... Optical module, 3 ... Modulation current generation part, 4 ... Bias current source, 5a, 5b ... Current voltage conversion part, 6a, 6b ... Level hold part, 8 ... Reference current generation part , 9 ... Variable capacitance unit, 13 ... Reference signal generation unit, 16 ... Control unit, 19 ... Drive current generation unit, 21 ... Laser diode, 22 ... Photodiode, BW_AD1 to BW_AD3 ... Capacitance control signal, Ibias ... Bias current, Imod _1, Imod 2 _... modulation current, Imon ... photocurrent, P ... signal light, RST_B, RST_P ... switch control signal, Smon ... light quantity signal, Sp, Sn ... transmission signal, Vref ... reference signal, ZTSEL1, ZTSEL2 ... gain control signal .

Claims (5)

A light emitting element driving circuit for generating signal light corresponding to a transmission signal in a light emitting element and controlling the signal light based on a photocurrent from a light receiving element for detecting the signal light;
A first current-voltage converter that converts the photocurrent from the light receiving element into a light amount signal that is a voltage signal;
A reference signal generation unit that generates a reference signal that is a voltage signal corresponding to the transmission signal;
Drive that generates a drive current modulated according to the transmission signal, adjusts the amount of the drive current according to a difference between the light amount signal and the reference signal, and supplies the drive current to the light emitting element A current generator and
The light-emitting element driving circuit, wherein the first current-voltage conversion unit includes a common base type transimpedance amplifier circuit, and the transimpedance of the common base type transimpedance amplifier circuit is variable.   The light emission according to claim 1, further comprising: a gain control unit that provides a gain control signal for changing the transimpedance of the common base type transimpedance amplifier circuit to the first current-voltage conversion unit. Element drive circuit. The reference signal generator is
A reference current generator for generating a reference current corresponding to the transmission signal;
A second current-voltage converter that converts the reference current into the reference signal;
A variable capacitance section connected between a signal line connecting the reference current generation section and the second current-voltage conversion section and a constant potential line;
The second current-voltage conversion unit includes a common base type transimpedance amplifier circuit, and the transimpedance of the common base type transimpedance amplifier circuit is variable. Light emitting element driving circuit.   The light emitting element drive circuit according to claim 3, further comprising a capacitance control unit that provides a capacitance control signal for changing a capacitance of the variable capacitance unit to the variable capacitance unit. An optical transmitter that outputs signal light according to a transmission signal,
A light emitting element for generating the signal light;
A light receiving element for detecting the signal light from the light emitting element;
A light emitting element driving circuit for supplying a driving current modulated in accordance with the transmission signal to the light emitting element,
The light emitting element driving circuit comprises:
A current-voltage converter that converts the photocurrent from the light receiving element into a light amount signal that is a voltage signal;
A reference signal generation unit that generates a reference signal that is a voltage signal corresponding to the transmission signal;
A drive current generator that generates the drive current, adjusts a current amount of the drive current according to a difference between the light amount signal and the reference signal, and supplies the drive current to the light emitting element;
An optical transmitter, wherein the current-voltage conversion unit includes a common base type transimpedance amplifier circuit, and the transimpedance of the common base type transimpedance amplifier circuit is variable.

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