JP4677858B2 - Light emitting element driving circuit and optical transmitter - Google Patents

Description

  The present invention relates to a light emitting element driving circuit and an optical transmitter used for optical communication.

  A peak detection type light emitting element driving circuit for stabilizing the signal light intensity and the extinction ratio of the light emitting element is known. For example, a semiconductor laser driving circuit disclosed in Patent Document 1 detects the average level of the output current from a monitor photodiode in an AC coupling laser driving circuit, and detects the bias current supplied to the laser diode. The size is controlled according to the average level. This drive circuit controls the modulation degree of the bias current according to the average level and high level (peak value) of the output current from the monitor photodiode.

  Further, in the conventional light emitting element driving circuit for DC coupling, the low level (bottom value) of the output current from the light receiving element (monitoring photodiode) is detected, and the magnitude of the bias current is determined according to the low level. Control. Further, the high level (peak value) of the output current from the monitor photodiode is detected, and the magnitude of the modulation current is controlled according to the high level.

JP-A-4-13979

  The present inventors have studied a peak detection type light emitting element driving circuit capable of stabilizing the signal light intensity and the extinction ratio of the light emitting element in any of AC coupling and DC coupling. Here, a problem when the peak detection type light emitting element driving circuit for DC coupling is used for AC coupling will be described.

  FIG. 12 shows the bias current Ibias and the modulation current Imod, the signal light intensity PH on the high level, and the low level in (a) the light emitting element driving circuit for DC coupling and (b) the light emitting element driving circuit for AC coupling. It is a figure which shows the correlation with the signal light intensity PL of the side.

  In the DC-coupled laser drive circuit (FIG. 12A), when the magnitude of the bias current Ibias is changed, the signal light intensity PH on the high level side is changed along with the signal light intensity PL on the low level side. On the other hand, even if the magnitude of the modulation current Imod is changed, only the signal light intensity PH on the high level side changes, and the signal light intensity PL on the low level side does not change. Therefore, as described above, if the magnitude of the bias current Ibias is controlled according to the low-level signal light intensity PL and the magnitude of the modulation current Imod is controlled according to the high-level signal light intensity PH, the signal Since the light intensity PL and the bias current Ibias can be stabilized independently, the signal light intensity PH (modulation current Imod) can also be stabilized after the signal light intensity PL is stabilized.

  However, in the AC coupled laser driving circuit (FIG. 12B), when the magnitudes of the modulation currents Imod (+) and Imod (−) are changed, both the signal light intensities PL and PH change. End up. Therefore, the signal light intensity PL on the low level side cannot be stabilized independently based on the bias current Ibias, and the signal light intensity PH on the high level side cannot be stabilized independently based on the modulation current Imod. There is a possibility that the light intensity and the extinction ratio cannot be controlled stably.

  The present invention has been made in view of the above problems, and a peak detection type light-emitting element driving circuit capable of stably controlling the signal light intensity and the extinction ratio in any of AC coupling and DC coupling, and optical transmission. The purpose is to provide a vessel.

  In order to solve the above problems, the light emitting element driving circuit according to the present invention causes the light emitting element to generate signal light corresponding to the transmission signal, and controls the signal light based on the photocurrent from the light receiving element that detects the signal light. A light emitting element driving circuit, a bias unit for supplying a bias current to a light emitting element, a modulating unit for supplying a modulation current corresponding to a transmission signal to a light emitting element, and a current for converting a photocurrent from the light receiving element into a voltage signal A voltage conversion unit; a first level detection unit that detects a first level of the voltage signal corresponding to the low level of the signal light; and a second level that detects a second level of the voltage signal corresponding to the high level of the signal light. 2 level detection units, the bias unit adjusts the amount of bias current according to the difference between the first level and the first reference level, and the modulation unit includes the second level and the second level. Change depending on the difference from the reference level Adjusting the amount of current, the bias portion after a lapse of a predetermined time from the start of the supply of the bias current, the modulation unit is characterized by starting the supply of the modulation current.

  The bias unit adjusts the current amount of the bias current according to the first level corresponding to the low level of the signal light, and the modulation unit adjusts the current amount of the modulation current according to the second level corresponding to the high level of the signal light. As described in the section “Problems to be Solved by the Invention”, the signal light intensity and the extinction ratio can be stably controlled in the case of DC coupling. There is a possibility that it cannot be controlled stably. In contrast, in the above-described light emitting element driving circuit, the modulation unit starts supplying the modulation current after a predetermined time has elapsed since the bias unit started supplying the bias current. That is, if the modulation current is not supplied for a predetermined time after the bias unit starts supplying the bias current, a low level change due to the change in the modulation current does not occur. Therefore, since the low level can be controlled and stabilized independently during this period, the high level can also be stabilized by the modulation unit starting to supply the modulation current after the low level is stabilized (that is, after a predetermined time has elapsed). . Thus, according to the above-described light emitting element driving circuit, the signal light intensity and the extinction ratio can be stably controlled not only during DC coupling but also during AC coupling.

  In the light emitting element driving circuit, each of the first and second level detection units is connected in series with a capacitive element for holding the first and second levels of the voltage signal. The switching element may be switched to a non-conductive state after the modulation unit starts to supply the modulation current. Thereby, after the signal light intensity and the extinction ratio are stably controlled, the time constants in the first and second level detection units can be changed more greatly, and the signal light intensity and the extinction ratio can be further stabilized against noise and the like. .

  The light emitting element driving circuit detects a reference signal generation unit that generates a reference signal corresponding to the transmission signal, a first level of the reference signal corresponding to the low level of the signal light, and sets the first level to the first level. A first reference level generation unit that outputs the first reference level, and a second level of the reference signal corresponding to the high level of the signal light, and outputs the second level as the second reference level. And a second reference level generation unit. As a result, the first and second reference levels can be changed in accordance with changes in the drive current to the light emitting element, so that the signal light intensity and the extinction ratio can be controlled more accurately.

  An optical transmitter according to the present invention is an optical transmitter that outputs signal light according to a transmission signal, a light emitting element that generates signal light, a light receiving element that detects signal light from the light emitting element, and a transmission A light emitting element driving circuit that supplies a modulation current corresponding to a signal and a bias current to the light emitting element, the light emitting element driving circuit including a bias unit that generates a bias current, a modulation unit that generates a modulation current, and a light receiving element A current-voltage converter for converting the photocurrent from the light into a voltage signal, a first level detector for detecting a first level of the voltage signal corresponding to the low level of the signal light, and a high level of the signal light A second level detection unit that detects a second level of the voltage signal, and the bias unit adjusts a current amount of the bias current according to a difference between the first level and the first reference level, The modulation unit has a second level and a second level. Adjusting the current amount of the modulation current in accordance with the difference between the reference level, the bias portion after a lapse of a predetermined time from the start of the supply of the bias current, the modulation unit is characterized by starting the supply of the modulation current. In this optical transmitter, the modulation unit starts supplying modulation current after a predetermined time has elapsed since the bias unit started supplying bias current. Therefore, similarly to the above-described light emitting element driving circuit, the signal light intensity and the extinction ratio can be stably controlled not only during DC coupling but also during AC coupling.

  According to the light emitting element driving circuit or the optical transmitter of the present invention, the signal light intensity and the extinction ratio can be stably controlled regardless of whether the coupling is AC coupling or DC coupling.

  Hereinafter, preferred embodiments of a light emitting element driving circuit and an optical transmitter according to the present invention will be described in detail with reference to the drawings. In the description of the drawings, the same or corresponding parts are denoted by the same reference numerals, and redundant description is omitted.

  FIG. 1 is a block diagram showing a configuration of a light emitting element driving circuit 1 which is a preferred embodiment of the present invention. A light-emitting element driving circuit 1 shown in the figure is a circuit for driving the optical module 2 and constitutes the optical transmitter according to the present embodiment together with the optical module 2. The optical module 2 has a small package configuration such as TOSA (Transmitter Optical Sub-Assembly). The optical module 2 detects (monitors) a light emitting element such as a laser diode 21 that generates the signal light P and the signal light P from the laser diode 21 (more precisely, light from the light reflection end face of the laser diode 21). And a light receiving element such as a diode 22. The laser diode 21 receives the drive current Id including the modulation current Imod and the bias current Ibias according to the transmission signals Sp and Sn from the outside from the light emitting element drive circuit 1, and the signal light P that is the laser light according to the drive current Id. Is generated.

Here, FIG. 2A is a diagram illustrating an example of output characteristics of the laser diode 21. 2A, the horizontal axis indicates the forward current (LD drive current) flowing through the laser diode 21, and the vertical axis indicates the light output intensity of the laser diode 21. FIG. Further, the graph G1 in the figure shows the output characteristics when the temperature of the laser diode 21 is a certain value T ₁ [° C.], and the graph G2 shows the temperature of the laser diode 21 at T ₂ (> T ₁ ) [° C. ] Shows the output characteristics. Ith ₁ in the graph G1 and Ith ₂ in the graph G2 indicate the threshold current values of the laser diode 21 at the temperatures T ₁ and T ₂ , respectively.

FIG. 2B is a graph showing an example of a time waveform of the signal light P output from the laser diode 21. The signal light intensity PL at the low level and the signal light intensity PH at the high level are alternately displayed. The waveform that repeats is shown. FIG. 2C is a graph showing an example of a time waveform of the drive current Id necessary for outputting the signal light P shown in FIG. In FIG. 2C, a graph G3 is an example of a time waveform of the drive current Id required at the temperature T ₁ [° C.], and a graph G4 shows the drive current Id required at the temperature T ₂ [° C.]. It is an example of a time waveform. Note that Ibias ₁ in FIG. 2C indicates a bias current value when the laser driving circuit is AC-coupled to the laser diode. In addition, Ibias ₂ in FIG. 2C indicates a bias current value when the laser drive circuit is DC coupled to the laser diode. The drive current Id is formed by convolution of the modulation current Imod with the bias current Ibias ₁ (or Ibias ₂ ).

As shown in FIG. 2A, the output characteristics of the laser diode 21 vary greatly with changes in temperature. That is, as the element temperature of the laser diode 21 increases, the laser oscillation threshold current value increases, and a larger drive current is required to obtain a predetermined light output intensity. Therefore, in order to keep the signal light intensity PH and the extinction ratio (PH / PL) constant regardless of the element temperature of the laser diode 21, as shown in graphs G3 and G4 in FIG. 2C, the bias current Ibias ₁ (Ibias ₂₎ and it is necessary to control in accordance with the magnitude of the modulation current Imod in the light output intensity. Therefore, the light emitting element driving circuit 1 controls the signal light intensity PH and the extinction ratio (PH / PL) of the signal light P based on the photocurrent Imon from the photodiode 22.

  The light emitting element driving circuit 1 of the present embodiment is a so-called AC coupled laser driving circuit. The light emitting element driving circuit 1 includes a modulation current generator 3, a bias current source 4, current-voltage converters 5a and 5b, level hold units 6a and 6b, error amplifiers 7a and 7b, a reference current generator 8, a controller 9, and The switch elements 10a to 10f are included.

The modulation current generation unit 3 is a modulation unit in this embodiment, and is a circuit for supplying modulation currents Imod ₁ and Imod ₂ corresponding to the transmission signals Sn and Sp to the laser diode 21. Specifically, the modulation current generator 3 includes two input terminals for inputting transmission signals Sp (positive phase) and Sn (negative phase), which are differential signals including transmission data, from the outside, and the transmission signal Sn. and two output terminals for outputting to the laser diode 21 modulated current Imod ₂ in accordance with the modulation current Imod ₁ and the transmission signal Sp corresponding to. The output terminal for outputting the modulation current Imod ₁ is connected to the anode of the laser diode 21 via a capacitor (capacitance element) 11a for cutting low frequency components. The output terminal for outputting the modulation current Imod ₂ is connected to the cathode of the laser diode 21 via a capacitor (capacitance element) 11b for cutting low frequency components.

The modulation current generator 3 further has an input terminal for inputting a difference signal Sd_bt from an error amplifier 7b described later. The modulation current generator 3 adjusts the modulation currents Imod ₁ and Imod ₂ so that the magnitude of the difference signal Sd_bt becomes small (that is, the magnitude of the bottom level signal Sbt described later approaches the bottom level reference voltage Vbt). Adjust the size.

  The bias current source 4 is a bias unit in this embodiment, and is a circuit for supplying a forward bias current Ibias to the laser diode 21. Specifically, one end of the bias current source 4 is connected to the cathode of the laser diode 21 via an inductor 12b for cutting high frequency components. The other end of the bias current source 4 is connected to a reference potential line 15 such as a GND line. The anode of the laser diode 21 is connected to the power supply potential line 14 via an inductor 12a for cutting high frequency components. A power supply voltage Vcc is supplied to the power supply potential line 14. The bias current source 4 further has an input terminal for inputting a difference signal Sd_pk from an error amplifier 7a described later. The bias current source 4 adjusts the magnitude of the bias current Ibias so that the magnitude of the difference signal Sd_pk becomes small (that is, the magnitude of a peak level signal Spk described later approaches the peak level reference voltage Vpk). .

  In the present embodiment, the bias current source 4 is connected between the cathode of the laser diode 21 and the reference potential line 15, but the bias current source is connected between the anode of the laser diode and the power supply potential line 14. It may be connected. Further, when the light emitting element driving circuit 1 is DC coupled to the laser diode 21, for example, the anode of the laser diode 21 may be directly connected to the power supply potential line 14 without being connected to the modulation current generating unit 3.

  The current-voltage conversion unit 5a is a circuit for converting the photocurrent Imon from the photodiode 22 into a voltage signal Smon. Here, FIG. 3 is a graph showing an example of a time waveform of the photocurrent Imon from the photodiode 22 and an example of a time waveform of the voltage signal Smon corresponding to the photocurrent Imon in this embodiment. is there. As shown in FIGS. 3A and 3B, the current-voltage converter 5a of the present embodiment outputs a peak level (first level) as a voltage signal Smon corresponding to the low level of the photocurrent Imon, A bottom level (second level) is output as a voltage signal Smon corresponding to the high level of the photocurrent Imon. Therefore, the photocurrent Imon and the voltage signal Smon are reversed from each other and have complementary time waveforms.

  The level hold unit 6a includes a peak detection unit 61a and a bottom detection unit 62a. The peak detector 61a is a first level detector in the present embodiment, and is a circuit for detecting the peak level of the voltage signal Smon corresponding to the low level of the signal light P. The peak detector 61a generates a peak level signal Spk indicating the peak level of the voltage signal Smon, and provides it to the error amplifier 7a via the switch element 10a. The bottom detector 62a is a second level detector in the present embodiment, and is a circuit for detecting the bottom value (second level) of the voltage signal Smon corresponding to the high level of the signal light P. . The bottom detection unit 62a generates a bottom level signal Sbt indicating the bottom level of the voltage signal Smon, and provides the bottom level signal Sbt to the error amplifier 7b via the switch element 10b.

  The error amplifier 7a amplifies the difference between the peak level signal Spk from the peak detection unit 61a and a peak level reference voltage Vpk (first reference level) from the peak detection unit 61b described later, and provides it to the bias current source 4 It is a circuit for doing. Specifically, the error amplifier 7a includes an operational amplifier, the non-inverting input terminal of the error amplifier 7a is connected to the peak detection unit 61a, and the inverting input terminal of the error amplifier 7a is connected to the peak detection unit 61b. ing. The output terminal of the error amplifier 7a is connected to the bias current source 4 through the switch element 10e. The error amplifier 7 a generates a difference signal Sd_pk indicating a difference between the peak level signal Spk and the peak level reference voltage Vpk, and provides the difference signal Sd_pk to the bias current source 4.

  Further, the error amplifier 7b amplifies a difference between a bottom level signal Sbt from the bottom detection unit 62a and a bottom level reference voltage Vbt (second reference level) from a bottom detection unit 62b described later, and generates a modulation current generation unit. 3 is a circuit for providing the data to 3. Specifically, the error amplifier 7b includes an operational amplifier, the non-inverting input terminal of the error amplifier 7b is connected to the bottom detection unit 62a, and the inverting input terminal of the error amplifier 7b is connected to the bottom detection unit 62b. ing. The output terminal of the error amplifier 7b is connected to the modulation current generator 3 through the switch element 10f. The error amplifier 7 b generates a difference signal Sd_bt that indicates a difference between the bottom level signal Sbt and the bottom level reference voltage Vbt, and provides the difference signal Sd_bt to the modulation current generator 3.

  The reference current generator 8 constitutes the reference signal generator 13 in the present embodiment together with a current-voltage converter 5b described later. The reference current generator 8 is a circuit for generating a reference current Iref corresponding to the transmission signals Sp and Sn from the outside. This reference current Iref is a pulse current corresponding to the transmission signals Sp and Sn. That is, the reference current Iref includes the same data component as the transmission signals Sp and Sn, and has a rise and a fall corresponding to the vibration of the transmission signals Sp and Sn. For example, the reference current generator 8 generates a modulation component by performing voltage-current conversion on the transmission signals Sp and Sn, and superimposes the modulation component on a predetermined bias component, thereby adjusting the target value of the photocurrent Imon. A reference current Iref can be generated.

  The current-voltage converter 5b is a circuit for converting the reference current Iref from the reference current generator 8 into a reference signal Vref that is a voltage signal. The function and internal configuration of the current-voltage converter 5b are the same as those of the current-voltage converter 5a described above. That is, the current-voltage converter 5b of the present embodiment outputs a peak level (first reference level) as the reference signal Vref corresponding to the low level of the reference current Iref, and the reference signal corresponding to the high level of the reference current Iref. The bottom level (second reference level) is output as Vref. Therefore, the reference current Iref and the reference signal Vref are reversed from each other and have a complementary time waveform.

  The level hold unit 6b includes a peak detection unit 61b and a bottom detection unit 62b. The peak detector 61b is a first reference level generator in the present embodiment. The peak detector 61b detects the peak level of the reference signal Vref and generates a peak level reference voltage Vpk indicating the peak level. Note that the peak level of the reference signal Vref corresponds to the low level of the transmission signals Sp and Sn (that is, the low level of the signal light P). The peak detector 61b provides the peak level reference voltage Vpk to the error amplifier 7a via the switch element 10c.

  Moreover, the bottom detection part 62b is a 2nd reference level production | generation part in this embodiment. The bottom detector 62b detects the bottom level of the reference signal Vref and generates a bottom level reference voltage Vbt indicating the bottom level. Note that the bottom level of the reference signal Vref corresponds to the high level of the transmission signals Sp and Sn (that is, the high level of the signal light P). The bottom detector 62b provides the bottom level reference voltage Vbt to the error amplifier 7b via the switch element 10d.

  The control unit 9 is a circuit for controlling the conduction state of the switch elements 10a to 10f. The control unit 9 has an output terminal connected to the control terminals of the switch elements 10a and 10c, and sends a switch control signal RST_P to the control terminals of the switch elements 10a and 10c. The control unit 9 has an output terminal connected to the control terminals of the switch elements 10b and 10d, and sends a switch control signal RST_B to the control terminals of the switch elements 10b and 10d. The control unit 9 has an output terminal connected to the control terminals of the switch elements 10e and 10f, and sends a switch control signal LDOFF to the control terminals of the switch elements 10e and 10f. In each of the switch elements 10a to 10f, when the input logic to the control terminal is 0, the input end and the output end are in a conductive state. When the input logic to the control terminal is 1, the input terminal and the output terminal are in a non-conductive state, and the output terminal is short-circuited with the reference potential line 15.

  FIG. 4 is a circuit diagram showing in detail an example of the configuration of the modulation current generator 3 of the present embodiment. Referring to FIG. 4, the modulation current generator 3 includes a pair of transistors 31a and 31b, a pair of resistance elements 32a and 32b, and a modulation current source 33.

  The base of the transistor 31a is electrically connected to an input terminal for inputting the transmission signal Sp (positive phase) from the outside of the modulation current generator 3. The base of the transistor 31b is electrically connected to an input terminal for inputting the transmission signal Sn (reverse phase) from the outside of the modulation current generator 3. The emitters of both the transistors 31 a and 31 b are electrically connected to the reference potential line 15 via the modulation current source 33. The collector terminal of the transistor 31a is electrically connected to the power supply potential line 14 through the resistance element 32a and is also electrically connected to the cathode of the laser diode 21 (see FIG. 1). The collector terminal of the transistor 31b is electrically connected to the power supply potential line 14 via the resistance element 32b, and is also electrically connected to the anode of the laser diode 21 (see FIG. 1). The modulation current source 33 has an input terminal for inputting the difference signal Sd_bt from the error amplifier 7b (see FIG. 1), so that the difference signal Sd_bt becomes small (that is, the magnitude of the bottom level signal Sbt). The amount of current is adjusted so that the voltage approaches the bottom level reference voltage Vbt.

When the transmission signals Sp (positive phase) and Sn (reverse phase) that are differential signals are input to the modulation current generator 3 illustrated in FIG. 4, the transmission signal Sp is input to the base of the transistor 31a and the transmission signal Sn is input to the base of the transistor 31b. When the transmission signal Sp is input to the base of the transistor 31a, the modulation current Imod ₂ flows through the laser diode 21. The modulation current Imod ₂ has a current value defined by the modulation current source 33 and flows through the transistor 31a. Further, by transmitting signal Sn is input to the base of the transistor 31b, it flows modulation current Imod ₁ reverse phase to the modulation current Imod _2. The modulation current Imod ₁ is regulated by the modulation current source 33 to have the same amount of current as the modulation current Imod ₂ and flows through the transistor 31b.

  FIG. 5 is a circuit diagram showing an example of the configuration of the peak detectors 61a and 61b and the bottom detectors 62a and 62b of the present embodiment. Referring to FIG. 5, the peak detectors 61 a and 61 b include a capacitor (capacitance element) 63 for holding the peak level of the voltage signal Smon and the reference signal Vref. The capacitor 63 is connected between the signal line 66 and the reference potential line 15. Then, a current based on the voltage signal Smon (reference signal Vref) is supplied to the signal line 66 by a rectifying element such as a diode or a transistor, and electric charges are accumulated in the capacitor 63, whereby the potential of the signal line 66 becomes the voltage signal Smon ( The peak level of the reference signal Vref) is maintained. The peak detector 61a (61b) outputs the potential of the signal line 66 thus held to the error amplifier 7a as a peak level signal Spk (peak level reference voltage Vpk).

  The peak detectors 61a and 61b further include a resistance element 64 and a transistor 65. The transistor 65 is a switch element in the present embodiment, and for example, a MOS-FET can be used. The resistance element 64 and the transistor 65 are provided for adjusting the time constants of the peak detectors 61 a and 61 b and are connected in parallel to the capacitor 63. Specifically, the resistance element 64 and the transistor 65 are connected in series between the signal line 66 and the reference potential line 15. For example, one end of the resistance element 64 is connected to the signal line 66 and the other end is connected to one of the source and drain of the transistor 65. The other of the source and the drain of the transistor 65 is connected to the reference potential line 15. The gate of the transistor 65 is connected to the control unit 9 and receives a control signal TCNT from the control unit 9.

  When the source and drain of the transistor 65 are turned on by the control signal TCNT, the charge accumulated in the capacitor 63 flows to the reference potential line 15 through the resistance element 64, so that the capacitor 63 is discharged earlier. Conversely, when the source and drain of the transistor 65 are turned off, the time required for discharging the capacitor 63 becomes longer. Thus, by switching the connection state of the transistor 65, the discharge time of the capacitor 63 (that is, the time constant of the peak detection units 61a and 61b) can be changed.

  The bottom detection units 62a and 62b include a capacitor (capacitance element) 67 for holding the bottom level of the voltage signal Smon and the reference signal Vref. The capacitor 67 is connected between the power supply potential line 14 and the signal line 70. The signal line 70 is supplied with a current by the voltage signal Smon (reference signal Vref) by a rectifying element such as a diode or a transistor, and the electric charge is accumulated in the capacitor 67, whereby the potential of the signal line 70 is changed to the voltage signal Smon ( The reference signal Vref) is held at the bottom level. The bottom detector 62a (62b) outputs the potential of the signal line 70 thus held to the error amplifier 7b as a bottom level signal Sbt (bottom level reference voltage Vbt).

  The bottom detection units 62a and 62b further include a resistance element 68 and a transistor 69. The transistor 69 is a switch element in the present embodiment, and for example, a MOS-FET can be used. The resistance element 68 and the transistor 69 are provided to adjust the time constants of the bottom detection units 62 a and 62 b and are connected in parallel to the capacitor 67. Specifically, the resistance element 68 and the transistor 69 are connected in series between the power supply potential line 14 and the signal line 70. For example, one end of the resistance element 68 is connected to the signal line 70 and the other end is connected to one of the source and drain of the transistor 69. The other of the source and the drain of the transistor 69 is connected to the power supply potential line 14. The gate of the transistor 69 is connected to the control unit 9 and receives a control signal TCNT from the control unit 9.

  When the source and drain of the transistor 69 are turned on by the control signal TCNT, the charge accumulated in the capacitor 67 flows to the power supply potential line 14 through the resistance element 68, so that the capacitor 67 is discharged earlier. Conversely, when the source and drain of the transistor 69 are turned off, the time required for discharging the capacitor 67 becomes longer. Thus, by switching the connection state of the transistor 69, the discharge time of the capacitor 67 (that is, the time constants of the bottom detection units 62a and 62b) can be changed.

  FIG. 6 is a block diagram illustrating an example of the configuration of the reference current generation unit 8 of the present embodiment. Referring to FIG. 6, the reference current generation unit 8 includes a low level reference voltage generation unit 81, a bias component generation unit 82, a high level reference voltage generation unit 83, a switch element 84, and a modulation component generation unit 85.

  The low level reference voltage generation unit 81 is a circuit for generating a reference voltage Vrefl indicating a low level magnitude of the reference current Iref. The magnitude of this reference voltage Vrefl is set to an appropriate value in advance. The bias component generation unit 82 converts the reference voltage Vrefl from the low level reference voltage generation unit 81 into a current, and outputs the current as a current Irefl indicating the low level (bias component) of the reference current Iref.

  The high level reference voltage generation unit 83 is a circuit for generating a reference voltage Vrefh indicating the magnitude of the high level (the peak value of the modulation component) of the reference current Iref. The magnitude of the reference voltage Vrefh is set to an appropriate value in advance as in the case of the reference voltage Vrefl. The modulation component generation unit 85 converts the reference voltage Vrefh from the high level reference voltage generation unit 83 into a current, and modulates the current with the transmission signals Sp and Sn. The modulation component generator 85 outputs the current Irefh thus generated as a modulation component of the reference current Iref. The current Irefh and the current Irefl are overlapped with each other and provided as the reference current Iref from the reference current generator 8 to the current-voltage converter 5b (see FIG. 1).

  The high level reference voltage generator 83 is connected to the input terminal of the switch element 84, and the modulation component generator 85 is connected to the output terminal of the switch element 84. The control terminal of the switch element 84 is connected to the control unit 9 and receives a switch control signal RST_B from the control unit 9. In the switch element 84, when the input logic to the control terminal is 0, the input end and the output end are in a conductive state. When the input logic to the control terminal is 1, the input terminal and the output terminal are in a non-conductive state, and the output terminal is short-circuited with the reference potential line 15. With this configuration, when the logic of the switch control signal RST_B is 1, the modulation component (current Irefh) of the reference current Iref is not generated, and only the bias component (current Irefl) is output from the reference current generator 8 as the reference current Iref. The Further, when the logic of the switch control signal RST_B is 0, a modulation component (current Irefh) of the reference current Iref is generated, and is output from the reference current generation unit 8 as the reference current Iref together with the bias component (current Irefl).

  The operation at the start of transmission of the light emitting element driving circuit 1 having the above configuration will be described with reference to FIGS. 7A to 7E respectively show the reference voltage Vrefl (FIG. 7A) in the low level reference voltage generation unit 81 (see FIG. 6) and the high level reference voltage generation unit 83 (see FIG. 6). Time of the reference voltage Vrefh (FIG. 7B), the reference signal Vref (FIG. 7C), the peak level reference voltage Vpk (FIG. 7D), and the bottom level reference voltage Vbt (FIG. 7E) It is a graph which shows an example of a waveform. 8A to 8D are respectively a switch control signal LDOFF output from the control unit 9 (FIG. 8A), a switch control signal RST_B (FIG. 8B), and a switch control signal RST_P. It is a graph which shows an example of the time waveform of (FIG.8 (c)) and control signal TCNT (FIG.8 (d)). Each of FIGS. 9A to 9D includes a voltage signal Smon (FIG. 9A), a peak level signal Spk (FIG. 9B), a bottom level signal Sbt (FIG. 9C), 10 is a graph showing an example of a time waveform of the drive current Id (FIG. 9D).

First, rises the reference voltage Vrefl from the low level reference voltage generating unit 81 at time _{t 1} (FIG. 7 (a)). At this time, the switch control signal RST_B from the control unit 9 is logic 1 (FIG. 8 (b)), and the switch element 84 (FIG. 6) is not connected, so only the bias component (Irefl) is used as the reference current Iref. Is output from the reference current generator 8. Therefore, only the bias component Vrefb is output from the reference signal generator 13 as the reference signal Vref as shown in FIG. In the present embodiment, since the reference current Iref and the reference signal Vref are reversed to form a complementary time waveform in the current-voltage converter 5b, the actual reference signal Vref is biased (Vrefb) from the power supply voltage Vcc. The voltage value is reduced only by this.

  At this stage, since the modulation component is not included in the reference signal Vref, the peak level reference voltage Vpk and the bottom level reference voltage Vbt converge to the same level as the bias component Vrefb of the reference signal Vref (FIG. 7 (d). ), FIG. 7 (e)).

Subsequently, the switch control signal LDOFF and RST_P from the control unit 9 at time _{t 2} is switched to a logic 0 (FIG. 8 (a), the Figure 8 (b)). That is, the switch elements 10a, 10c, 10e, and 10f (see FIG. 1) are connected. As a result, the peak level reference voltage Vpk is input to the error amplifier 7a, and the difference signal Sd_pk from the error amplifier 7a is input to the bias current source 4, thereby forming a closed loop circuit relating to the bias current Ibias. Then, the supply of the bias current Ibias is started by the bias current source 4 so that the difference signal Sd_pk becomes small (that is, the magnitude of the peak level signal Spk approaches the peak level reference voltage Vpk). Since the current value is adjusted, the current value of the bias current Ibias approaches the magnitude (Irefl) of the reference current Iref (FIGS. 9B and 9D).

Subsequently, at time _{t 3} after a predetermined time has elapsed from time _{t 2, the} switch control signal RST_B from the control unit 9 is switched to a logic 0 (FIG. 8 (c)). That is, the switch elements 10b and 10d (FIG. 1) and 84 (FIG. 6) are connected.

  At this time, the reference current generator 8 outputs not only the bias component (Irefl) but also the modulation component (Irefh) as the reference current Iref. Therefore, the reference signal generator 13 outputs the modulation component Vrefm as the reference signal Vref in addition to the bias component Vrefb as shown in FIG. 7C. That is, the bottom level reference voltage Vbt indicating the bottom level of the reference signal Vref is the sum of the peak value of the modulation component Vrefm and the bias component Vrefb as shown in FIG.

  Further, the bottom level reference voltage Vbt is input to the error amplifier 7b, and the difference signal Sd_bt from the error amplifier 7b is input to the modulation current generation unit 3, so that a closed loop circuit relating to the modulation current Imod is formed. Then, the modulation current generator 3 starts supplying the modulation current Imod so that the difference signal Sd_bt becomes small (that is, the magnitude of the bottom level signal Sbt approaches the bottom level reference voltage Vbt). Current value is adjusted. As a result, the magnitude of the drive current Id obtained by combining the bias current Ibias and the modulation current Imod is controlled according to the magnitude of the reference current Iref (FIG. 9D).

Then, at time _{t 4,} the control signal TCNT from the control unit 9 is switched to a logic 0 (FIG. 8 (d)). That is, the drain-source state of the transistors 65 and 69 (see FIG. 5) of the peak detectors 61a and 61b and the bottom detectors 62a and 62b becomes non-conductive, and the peak detectors 61a and 61b and the bottom detectors 62a and 62b The time constant increases.

  The effects of the light emitting element driving circuit 1 according to the present embodiment described above will be described. Here, FIG. 10 is a control block diagram of the light emitting element driving circuit 1. FIG. 11 is a control block diagram when a circuit similar to the light emitting element driving circuit 1 is DC coupled to the laser diode 21.

10 and 11, each element represents the following numerical value.
Ith (T): threshold current value [A] of the laser diode 21 at the temperature T [° C.]
ηLD (T): slope efficiency [W / A] of the laser diode 21 at a temperature T [° C.]
ηPD: Photoelectric conversion efficiency of photodiode 22 [A / W]
RTIA: Gain [V / A] in the current-voltage converters 5a and 5b
FP (s): Conversion function [V / V] in the peak detectors 61a and 61b
FB (s): Conversion function [V / V] in the bottom detectors 62a and 62b
G _bias (s): gain [V / V] in the error amplifier 7a
G _mod (s): gain [V / V] in the error amplifier 7b
Gm: Conductance [A / V] in the bias current source 4 and the modulation current source 33
GAC: Current loss FAC (s) due to AC coupling (capacitors 11a and 11b): Filter effect (low frequency cut) due to AC coupling (capacitors 11a and 11b)
IrefL: Low level [A] of the reference current Iref
IrefH: High level of the reference current Iref [A]
Here, s in the above parentheses is jω, indicating that it is a function of frequency.

また、図１０に示す制御ブロック線図において、ハイレベル側の信号光強度ＰＨについての伝達関数は以下の式（２）で表される。

In the control block diagram shown in FIG. 10, the transfer function for the low-level signal light intensity PL is expressed by the following equation (1).

In the control block diagram shown in FIG. 10, the transfer function for the high-level signal light intensity PH is expressed by the following equation (2).

また、図１１に示す制御ブロック線図において、ハイレベル側の信号光強度ＰＨについての伝達関数は以下の式（４）で表される。

In the control block diagram shown in FIG. 11, the transfer function for the low-level signal light intensity PL is expressed by the following equation (3).

In the control block diagram shown in FIG. 11, the transfer function for the high-level signal light intensity PH is expressed by the following equation (4).

  The second term on the right side in Equation (4) indicates that the behavior of the signal light intensity PL on the low level side affects (interfers) the behavior of the signal light intensity PH on the high level side. That is, in the case of a DC-coupled drive circuit, as shown in part B of FIG. 11, for controlling the signal light intensity PH on the high level side from the closed loop for controlling the signal light intensity PL on the low level side. Interference with the closed loop occurs. However, as shown in Expression (3), the behavior of the signal light intensity PH on the high level side does not affect the behavior of the signal light intensity PL on the low level side. That is, as shown in FIG. 12A, in the DC-coupled laser driving circuit, the magnitude of the bias current Ibias is controlled in accordance with the low-level signal light intensity PL, and the high-level signal light is controlled. If the magnitude of the modulation current Imod is controlled according to the intensity PH, the signal light intensity PL and the bias current Ibias can be stabilized independently. Therefore, after the signal light intensity PL is stabilized, the signal light intensity PH (modulation current Imod) is stabilized. ) Can also be stabilized.

  However, the second term on the right-hand side in equations (1) and (2) represents the mutual closed-loop interference. That is, in the case of an AC-coupled drive circuit, as shown in part A of FIG. 10, a closed loop for controlling the signal light intensity PL on the low level side and a signal light intensity PH on the high level side are controlled. And the closed loop of each other interfere with each other. As shown in FIG. 12B, in the AC-coupled laser driving circuit, when the modulation currents Imod (+) and Imod (−) are changed, both the signal light intensities PL and PH are changed. Indicates that will change. Accordingly, it can be seen that the signal light intensity PH and PL cannot be stabilized independently as they are, and the signal light intensity and the extinction ratio (PH / PL) may not be stably controlled.

  In order to solve this problem, in the light emitting element driving circuit 1 of the present embodiment, the operation start timing of each closed loop is controlled by the control unit 9, and the bias current source 4 starts supplying the bias current Ibias. After a predetermined time has elapsed, the modulation current generator 3 starts supplying the modulation current Imod. Since the closed loop on the signal light intensity PH side does not function unless the modulation current Imod is supplied for a predetermined time after the bias current source 4 starts supplying the bias current Ibias, the closed loop on the signal light intensity PL side receives interference. Absent. Therefore, the signal light intensity PL can be independently controlled and stabilized during this period. Then, after the signal light intensity PL is stabilized (that is, after a predetermined time has elapsed), the modulation current generator 3 starts supplying the modulation current Imod. By doing so, it becomes a constant term, and the signal light intensity PH can be stabilized without depending on PL. As described above, according to the light emitting element driving circuit 1 of the present embodiment, the signal light intensity PH and the extinction ratio (PH / PL) can be stably controlled not only during DC coupling but also during AC coupling.

  Such an effect of the light emitting element driving circuit 1 is particularly remarkable in the following case. That is, the capacitor 63 of the peak detectors 61a and 61b shown in FIG. 5 performs a discharging operation when the operation starts, and the capacitor 67 of the bottom detectors 62a and 62b performs a charging operation when the operation starts. In such a case, since the discharge operation is slower than the charge operation, it takes time to stabilize the closed loop circuit for controlling the bias current Ibias including the peak detection units 61a and 61b. If the above-described interference occurs in the closed loop circuit, the time required for stabilization of the closed loop circuit is further increased. On the other hand, according to the light emitting element driving circuit 1 of the present embodiment, the closed loop circuit including the peak detectors 61a and 61b is stabilized first, so that mutual interference between the closed loops can be avoided and the stabilization can be performed quickly and reliably. Can be

  As shown in FIG. 5, the peak detector 61 a includes a capacitor 63 for maintaining the peak level of the voltage signal Smon, and a resistor element 64 that is connected in series with each other and connected in parallel to the capacitor 63. It is preferable that the transistor 65 is switched to a non-conductive state after the modulation current generator 3 starts to supply the modulation current Imod. Similarly, the bottom detection unit 62 a includes a capacitor 67 for maintaining the bottom level of the voltage signal Smon, and a resistance element 68 and a transistor 69 that are connected in series with each other and connected in parallel to the capacitor 67. Then, after the modulation current generator 3 starts supplying the modulation current Imod, the transistor 69 is preferably switched to a non-conductive state. Thus, after the signal light intensity and the extinction ratio are stably controlled, the time constants in the peak detection unit 61a and the bottom detection unit 62a can be changed more greatly, and the signal light intensity and the extinction ratio can be further stabilized against noise and the like. .

  Further, as in the present embodiment, the light emitting element driving circuit 1 includes a reference signal generation unit 13 that generates the reference signal Vref corresponding to the transmission signals Sp and Sn, a peak detection unit 61b that detects the level of the reference signal Vref, and It is preferable to include a bottom detection unit 62b. As a result, the peak level reference voltage Vpk and the bottom level reference voltage Vbt can be changed according to the change in the drive current Id to the laser diode 21, so that the signal light intensity and the extinction ratio can be controlled with higher accuracy.

  The light emitting element driving circuit and the optical transmitter according to the present invention are not limited to the above-described embodiments, and various modifications are possible. For example, in the above embodiment, an AC-coupled light-emitting element driving circuit is described, but the light-emitting element driving circuit according to the present invention may be a DC-coupled type.

FIG. 1 is a block diagram showing a configuration of a light emitting element driving circuit according to a preferred embodiment of the present invention. FIG. 2A is a diagram showing output characteristics of the laser diode. FIG. 2B is a graph showing an example of the time waveform of the signal light output from the laser diode. FIG. 2C is a graph showing an example of a time waveform of the drive current necessary for outputting the signal light shown in FIG. FIG. 3A is a graph showing an example of a time waveform of the photocurrent from the photodiode. FIG. 3B is a graph showing an example of a time waveform of the voltage signal. FIG. 4 is a circuit diagram showing in detail an example of the configuration of the modulation current generator. FIG. 5 is a circuit diagram illustrating an example of the configuration of the peak detection unit and the bottom detection unit. FIG. 6 is a block diagram illustrating an example of the configuration of the reference current generation unit 8 of the present embodiment. FIG. 7 shows an example of time waveforms of (a) reference voltage Vrefl, (b) reference voltage Vrefh, (c) reference signal Vref, (d) peak level reference voltage Vpk, and (e) bottom level reference voltage Vbt. It is a graph. FIG. 8 is a graph showing an example of time waveforms of (a) switch control signal LDOFF, (b) switch control signal RST_B, (c) switch control signal RST_P, and (d) control signal TCNT. FIG. 9 is a graph showing an example of time waveforms of (a) voltage signal Smon, (b) peak level signal Spk, (c) bottom level signal Sbt, and (d) drive current Id. FIG. 10 is a control block diagram of the light emitting element driving circuit. FIG. 11 is a control block diagram when a circuit similar to the light emitting element driving circuit is DC coupled to the laser diode. FIG. 12 shows the bias current and modulation current, the high-level signal light intensity, and the low-level signal in (a) the DC coupling light-emitting element driving circuit and (b) the AC coupling light-emitting element driving circuit. It is a figure which shows the correlation with light intensity.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Light emitting element drive circuit, 2 ... Optical module, 3 ... Modulation current generation part, 4 ... Bias current source, 5a, 5b ... Current-voltage conversion part, 6a, 6b ... Level hold part, 7a, 7b ... Error amplifier, 8 Reference current generator, 9 Control unit, 10a to 10f Switch element, 11a, 11b Capacitor, 12a, 12b Inductor, 13 Reference signal generator, 21 Laser diode, 22 Photodiode, 31a, 31b ... transistor, 32a, 32b ... resistive element, 33 ... modulation current source, 61a, 61b ... peak detector, 62a, 62b ... bottom detector.

Claims (4)

A light emitting element driving circuit for generating signal light corresponding to a transmission signal in a light emitting element and controlling the signal light based on a photocurrent from a light receiving element for detecting the signal light;
A bias unit for supplying a bias current to the light emitting element;
A modulator for supplying a modulation current corresponding to the transmission signal to the light emitting element;
A current-voltage converter that converts the photocurrent from the light receiving element into a voltage signal;
A first level detector that detects a first level of the voltage signal corresponding to a low level of the signal light;
A second level detection unit for detecting a second level of the voltage signal corresponding to the high level of the signal light,
The bias unit adjusts a current amount of the bias current according to a difference between the first level and a first reference level;
The modulation unit adjusts a current amount of the modulation current according to a difference between the second level and a second reference level;
The light emitting element driving circuit according to claim 1, wherein the modulation unit starts supplying the modulation current after a predetermined time has elapsed since the bias unit started supplying the bias current. Each of the first and second level detectors is
A capacitive element for holding each of the first and second levels of the voltage signal;
A resistor element and a switch element connected in series with each other and connected in parallel to the capacitor element;
2. The light emitting element drive circuit according to claim 1, wherein the switch element is switched to a non-conductive state after the modulation unit starts supplying the modulation current. 3. A reference signal generator for generating a reference signal corresponding to the transmission signal;
A first reference level generation unit that detects a first level of the reference signal corresponding to a low level of the signal light and outputs the first level as the first reference level;
A second reference level generation unit that detects a second level of the reference signal corresponding to a high level of the signal light and outputs the second level as the second reference level; The light emitting element drive circuit according to claim 1 or 2. An optical transmitter that outputs signal light according to a transmission signal,
A light emitting element for generating the signal light;
A light receiving element for detecting the signal light from the light emitting element;
A light emitting element driving circuit for supplying a modulation current corresponding to the transmission signal and a bias current to the light emitting element, and
The light emitting element driving circuit comprises:
A bias unit for generating the bias current;
A modulation unit for generating the modulation current;
A current-voltage converter that converts the photocurrent from the light receiving element into a voltage signal;
A first level detector that detects a first level of the voltage signal corresponding to a low level of the signal light;
A second level detection unit for detecting a second level of the voltage signal corresponding to the high level of the signal light,
The bias unit adjusts a current amount of the bias current according to a difference between the first level and a first reference level;
The modulation unit adjusts a current amount of the modulation current according to a difference between the second level and a second reference level;
The optical transmitter, wherein the modulation unit starts supplying the modulation current after a predetermined time has elapsed since the bias unit started supplying the bias current.

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