JP2006182254A - Electric power steering device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an electric power steering device capable of accurately detecting a motor electric current by a simple structure, and applying an inexpensive microcomputer. <P>SOLUTION: The electric power steering device comprises a non-connection brush-less motor 12 having a stator installed with mutually independent armature windings Lu-Lw of a plurality of N phases; a steering torque detecting means 3 detecting steering torque inputted to a steering system; inverter circuits 34u to 34w of a plurality of N pieces, to which both ends of the armature windings Lu-Lw of the non-connection brush-less motor 12 are connected, and individually supplying driving signals to the armature windings; current detecting means 17u to 17w disposed on either one of a grounding side or a power source side of each inverter circuit 34u to 34w; and a drive controlling portion 15 drive controlling the inverter circuits on the basis of a winding current detected by the current detecting means 17u to 17w and the steering torque detected by the detecting means 3. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to an electric power steering apparatus using a wireless brushless motor that is independent without connecting armature windings of a stator.

  In general, an electric power steering apparatus uses a permanent magnet as a motor for generating a steering assist force for a steering system, and an N-phase (N is an integer of 3 or more) armature winding is Y-connected or Δ A wired N-phase brushless motor is used, and this N-phase brushless motor is driven and controlled according to the steering torque detected by the steering torque detecting means by the motor drive circuit.

  In such a motor drive circuit, generally, an inverter circuit having a switching element twice the number of phases and driven by a pulse width modulation (PWM) signal is connected to the N-phase electronic winding of the N-phase brushless motor. The position of the rotor of the brushless motor is detected by the position sensor, the current flowing through each armature winding is detected by the current detection circuit, the rotor position detected by the position sensor, and each armature detected by the current detection circuit A control circuit for driving the inverter circuit based on the winding current and the current target value is provided.

  Here, as the current detection circuit for detecting the current flowing through each armature winding, for example, both ends of a current detection resistor inserted in at least N-1 connection lines between the inverter circuit and the N-phase brushless motor. A configuration is known in which a voltage is detected, the voltage between both ends is converted into a digital signal by an A / D converter, and input to a microcomputer that drives and controls the inverter circuit (see, for example, Patent Document 1).

  As another current detection method, as shown in FIG. 23, the electric power steering apparatus includes three switching elements Tr1 and Tr3 that supply current to the U-phase to W-phase armature windings of the three-phase brushless motor 100, and A three-phase inverter circuit 101 having an upper arm constituted by Tr5 and a lower arm constituted by three switching elements Tr2, Tr4 and Tr6, and each switching element Tr2 constituting the lower arm of the inverter circuit 101, Detected by current detection operational amplifiers OPu, OPv, and OPw to which voltages at both ends of current detection resistors Ru, Rv, and Rw inserted between Tr4 and Tr6 and the ground are input, a steering torque sensor 102, and a vehicle speed sensor 103, respectively. And the operational torque OPu, O are input to the A / D conversion input terminal. The microcomputer 104 to which the detection voltages of v and OPw are input, and the gate drive circuit 105 that drives the three-phase inverter circuit 100 by inputting the duty command value calculated by the pulse width modulation (PWM) function of the microcomputer 104 The motor current detected by the current detection operational amplifiers OPu to OPw by the A / D conversion function in the microcomputer 104 is converted into a digital signal by starting A / D conversion by a trigger signal from the pulse width modulation function. It is configured to read.

Furthermore, a configuration is also known in which a shunt resistor is inserted on the ground side of the three-phase inverter circuit, and a current flowing through the shunt resistor is detected by a current detection circuit and input to a microcomputer (for example, Patent Document 2). reference).
JP 2002-238293 A (page 3 to page 4, FIG. 1) JP-A-12-350490 (page 4, FIG. 1)

However, in the conventional example described in Patent Document 1, since current detection means is provided in at least two phases of the connection lines between the inverter circuit and the three-phase brushless motor, timing control for A / D conversion is performed. However, there is an unsolved problem that the potential variation of the shunt resistor for current detection is large and the configuration of the current detection circuit becomes complicated.
Further, in the conventional example shown in FIG. 23, current detection resistors Ru to Rw for individually detecting currents of respective phases are inserted in the inverter circuit 100, and voltages at both ends of these current detection resistors Ru to Rw are obtained. Although the current detection operational amplifiers OPu to OPw individually detect currents flowing in different directions through the current detection resistors Ru to Rw, the positive and negative motor currents are the maximum output dynamic range of the current detection operational amplifiers OPu to OPw. In order to secure values obtained by adding margins to the values ± Imax, it is necessary to set a wide range. As a result, the bit rate of the current detection value, that is, the motor current amount A / bit per bit of the A / D converter is increased, and the unresolved that the roughness of the ionization detection accuracy adversely affects the control of the electric power steering apparatus. There are challenges.

Further, in the conventional example described in Patent Document 2, the PWM signal is output in order from the phase with the higher duty value of the PWM signal of the upper arm, which is not described in the publication, and the PWM output of each phase. The current flowing through the shunt resistor is shifted regularly by shifting the PWM output timing by shifting the predetermined start time required for the A / D converter to sample and hold the A / D converter. By generating a timing at which the phase current value and the phase current value coincide with each other and performing A / D conversion at that timing, at least two phase currents can be detected with one shunt resistor, and the remaining one phase current is Iu + Iv + Iw = 0. The circuit configuration itself is the simplest, but the generation of the PWM signal is special.応可 not is known, there is an unsolved problem that it is impossible to apply the inexpensive microcomputer.
Accordingly, the present invention has been made paying attention to the unsolved problems of the above-described conventional example, and can detect a motor current with high accuracy with a simple configuration, and an inexpensive microcomputer can be applied. An object of the present invention is to provide a simple electric power steering device.

  In order to achieve the above object, an electric power steering apparatus according to a first aspect of the present invention is independent of a rotor provided with a permanent magnet and a plurality of N-phase armature windings facing each other without connecting them to each other. A non-connection type brushless motor having a stator disposed therein, steering torque detection means for detecting a steering torque input to the steering system, and individually connected to both ends of each armature winding of the non-connection type brushless motor An inverter circuit, a current detection means arranged on either the ground side or the power supply side of each inverter circuit, a winding current detected by the current detection means, and a steering torque detected by the steering torque detection means And a drive control section for controlling the drive of each inverter circuit based on the above.

  The electric power steering apparatus according to claim 2 of the present invention is the electric power steering apparatus according to claim 1, wherein the current detection means is for current detection inserted in either the ground side or the power supply side of each inverter circuit. The drive control unit includes A / D conversion means for sampling and A / D-converting the voltage between the terminals detected by the current detection means. The sampling timing of the D conversion means is determined based on the duty ratio of the pulse width modulation signal supplied to each armature winding.

  Furthermore, in the electric power steering apparatus according to claim 3 of the present application, in the invention according to claim 2, switching of the sampling timing of the A / D conversion means is performed with a predetermined point across the point where the duty ratio of the pulse width modulation signal is 50%. The width is set to have a hysteresis characteristic.

  The electric power steering apparatus according to claim 4 is the electric power steering apparatus according to claim 1, wherein the current detection means is a current detection resistor interposed between the ground side and the power supply side of each inverter circuit. The drive control unit has A / D conversion means for sampling and A / D converting the voltage between the terminals detected by the current detection means, and the A / D The sampling timing of the conversion means is determined for each armature winding based on the direction and magnitude of the drive current.

  Still further, according to a fifth aspect of the present invention, there is provided an electric power steering apparatus according to the fourth aspect of the invention, wherein the sampling timing of the A / D conversion means is changed to a predetermined width across a point where the drive current of the armature winding is zero. It is characterized by having a hysteresis characteristic.

  According to the present invention, each armature winding of a wireless brushless motor having a rotor in which a permanent magnet is disposed and a stator in which a plurality of N-phase armature windings are independently disposed without being connected to each other. Current detection of each armature winding is individually detected by current detection means provided on either the power supply side or the ground side in an inverter circuit individually connected to both ends of each armature winding. By adjusting the timing when this detection current is A / D converted, it is possible to detect a current value that is almost absolute without including current direction information, and to reduce the dynamic range to detect the current. An effect is obtained in that the current detection accuracy can be improved by reducing the bit rate of the value.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is an overall configuration diagram showing an embodiment in which the present invention is applied to an electric power steering apparatus. In the figure, reference numeral 1 denotes a steering wheel, which is applied to the steering wheel 1 from a driver. A steering force is transmitted to a steering shaft 2 having an input shaft 2a and an output shaft 2b. The steering shaft 2 has one end of the input shaft 2a connected to the steering wheel 1 and the other end connected to one end of the output shaft 2b via a steering torque sensor 3 as steering torque detecting means.

  The steering force transmitted to the output shaft 2 b is transmitted to the lower shaft 5 via the universal joint 4 and further transmitted to the pinion shaft 7 via the universal joint 6. The steering force transmitted to the pinion shaft 7 is transmitted to the tie rod 9 via the steering gear 8 and steers steered wheels (not shown). Here, the steering gear 8 is configured in a rack and pinion type having a pinion 8a connected to the pinion shaft 7 and a rack 8b meshing with the pinion 8a, and the rotational motion transmitted to the pinion 8a is linearly moved by the rack 8b. It has been converted to movement.

A steering assist mechanism 10 for transmitting a steering assist force to the output shaft 2b is connected to the output shaft 2b of the steering shaft 2. The steering assist mechanism 10 includes a reduction gear 11 connected to the output shaft 2b and a wireless motor 12 as an electric motor that generates a steering assist force connected to the reduction gear 10.
The steering torque sensor 3 detects the steering torque applied to the steering wheel 1 and transmitted to the input shaft 2a. For example, the steering torque sensor 3 is a torsion bar (not shown) in which the steering torque is interposed between the input shaft 2a and the output shaft 2b. It is configured to convert to a torsional angular displacement and detect the torsional angular displacement with a potentiometer. The steering torque sensor 3, as shown in FIG. 2, when the steering torque input is zero, a predetermined neutral voltage V ₀ becomes, when the right turn from this state, the neutral voltage V ₀ in accordance with the increase of the steering torque When the steering torque is turned to the left from a state where the steering torque is zero, a torque detection value T that is a voltage that decreases from the neutral voltage V _{0 as the} steering torque increases is output.

  Further, as shown in FIG. 3, the wireless motor 12 has a rotating shaft 24 rotatably supported on a housing 21 via a pair of bearings 22 and 23. A rotor core 27 formed by laminating a plurality of disk-shaped electromagnetic steel plates 25 and 26 is mounted around the rotating shaft 24 between the pair of bearings 22 and 23, and the outer peripheral surface of the rotor core 27 is The rotor magnet 28 is fixed. The rotor magnet 28 uses a segment magnet as a permanent magnet for generating a field. Further, as shown in FIG. 4, on the outside of the rotor magnet 18, a cylindrical magnet that prevents the rotor magnet 28 from being scattered and displaced by bringing a flange portion 29 a formed at one end into contact with the end surface of the rotor magnet 28. A cover 29 is provided. The rotating shaft 24, the rotor core 27, the rotor magnet 28, and the magnet cover 29 constitute the rotor 20.

  A stator 31 is disposed in the housing 21 so as to face the rotor 20 in the radial direction, an annular stator core 32 fixed to the inner peripheral surface of the housing 21, and an electric machine wound around the stator core 32. And an exciting coil 33 as a child winding. As shown in FIG. 5, the exciting coil 33 is composed of, for example, three-phase exciting coils Lu, Lv and Lw, and these exciting coils Lu to Lw are wound independently without being connected to each other. It is a connection type (open type) brushless motor wiring, and inverter circuits 34u, 34v and 34w are connected between both ends of each excitation coil Lu, Lv and Lw, and drive currents Iu, Iv and Iw are individually supplied. .

  As shown in FIG. 5, the inverter circuit 34j (j = u, v, w) has four switching elements Trj1 to Trj4 configured by, for example, N-channel MOSFETs, and the switching elements Trj1 and Trj2 are connected in series. The series circuit in which the switching elements Trj3 and Trj4 are connected in series is connected in parallel to form an H-bridge circuit Hj. The connection point of the switching elements Trj1 and Trj3 of the H bridge circuit Hj is connected to the battery B via the relay RY, the connection point of the switching elements Trj2 and Trj4 is grounded via the shunt resistor Rj for current detection, and further switching is performed. A connection point between the elements Trj1 and Trj2 is connected to one terminal tja of the excitation coil Lj in the wireless brushless motor 12, and a connection point between the switching elements Trj3 and Trj4 is connected to the other terminal tjb of the excitation coil Lj. Note that a flywheel diode D is connected in the forward direction between the source and drain of each of the switching elements Trj1 to Trj4.

  The PWM (pulse width modulation) signal Pj1 output from the drive control circuit 15 is supplied to the switching elements Trj1 and Trj4 of each inverter circuit 34j, and the PWM (pulse width modulation) is supplied from the drive control circuit 15 to the switching elements Trj2 and Trj3. A PWM (Pulse Width Modulation) signal Pj2 having an opposite phase to that of the signal Pj1, that is, an on / off inversion is supplied.

Here, as shown in FIG. 6, the equivalent circuit of each exciting coil Lu, Lv, and Lw has a resistance R ₀ ′, an inductance L ₀ ′, a counter electromotive voltage eu () between the terminals tua and tub. = Ω × Kt ′ × sin (ωt)) are arranged in series, the terminal voltage Vua of the terminal tua is Vua = V ₀ × sin (ωt + α), and the terminal voltage Vub of the terminal tub is Vub = V ₀ × sin (Ωt−π + α), the inter-terminal voltage Vuab is Vuab = 2 × V ₀ × sin (ωt + α), and the phase current Iu is Iu = I ₀ ′ × sin (ωt).

Similarly, for the exciting coil Lv, a resistor R ₀ ′, an inductance L ₀ ′, and a counter electromotive voltage eu (= ω × Kt ′ × sin (ωt−2π / 3)) are arranged in series between the terminals tva and tvb. The terminal voltage Vva of the terminal tva is Vva = V ₀ × sin (ωt−2π / 3 + α), the terminal voltage Vvb of the terminal tu is Vvb = V ₀ × sin (ωt−2π / 3−π + α), The voltage Vvab between terminals is Vvab = 2 × V ₀ × sin (ωt−2π / 3 + α), and the phase current Iv is Iv = I ₀ ′ × sin (ωt−2π / 3).

Similarly, for the exciting coil Lw, a resistor R ₀ ′, an inductance L ₀ ′, and a counter electromotive voltage eu (= ω × Kt ′ × sin (ωt−4π / 3)) are arranged in series between the terminals twa and twb. The terminal voltage Vwa of the terminal twa is Vwa = V ₀ × sin (ωt−4π / 3 + α), the terminal voltage Vwb of the terminal tu is Vwb = V ₀ × sin (ωt−4π / 3−π + α), The voltage Vwab between terminals is Vwab = 2 × V ₀ × sin (ωt−4π / 3 + α), and the phase current Iw is Iw = I ₀ ′ × sin (ωt−4π / 3).

In addition, the motor constant is designed to be either a motor constant of a conventional Y-connection motor, a motor constant of a conventional Δ-connection motor, or a unique motor constant that satisfies the required performance, and is a three-phase connectionless system. A brushless motor is configured.
Here, the magnetized magnet of the rotor 20 is magnetized so that the induced voltage waveform of the wireless motor 12 becomes a pseudo-rectangular wave in which the third harmonic and the fifth harmonic are superimposed on a sine wave, as will be described later. A winding method of the stator 31 is set.

  Further, a phase detector 35 of the rotor 20 is disposed in the vicinity of the one bearing 22. The phase detection unit 35 includes an annular phase detection permanent magnet 36 attached to the rotary shaft 24 and a phase detection element 37 that faces the permanent magnet 36 and is fixed to the housing 21 side. Since the motor 12 is a brushless motor that does not include a mechanical commutator (brush and commutator), the phase detection unit 35 detects the phase of the rotor 20 and controls the excitation coil according to the phase under the control of the drive circuit 15. This is for energizing 33. A resolver, an encoder, or the like can also be used as the phase detection unit.

Then, the steering torque detection value T output from the steering torque sensor 3 is input to the drive control circuit 15 to which power is supplied from the battery B via the ignition switch IG as shown in FIG.
The drive control circuit 15 includes a vehicle speed detection value V detected by the vehicle speed sensor 16 in addition to the torque detection value T, and excitation coils Lu to Lw of the wireless brushless motor 12 detected by the motor current detection units 17u to 17w. The motor currents Iau to Iaw flowing through the motor and the phase detection signal of the rotor 20 detected by the phase detection unit 34 are input.

  Here, each of the motor current detectors 17u, 17v, and 17w is connected between the connection points of the switching elements Tru2 and Tru4 of the inverter circuits 34u, 34v, and 34w, the connection point of the Trv2 and Trv4, and the connection point of the Trw2 and Trw4 and the ground. And shunt resistors Ru, Rv, and Rw as current detection resistors, and operational amplifiers OPu, OPv, and OPw that detect voltages between the terminals. In the operational amplifiers OPu to OPw, the voltages across the shunt resistors Ru to Rw are output as motor currents Iau to Iaw having a value that becomes Vref when the amplitude based on the reference voltage Vref, that is, the motor current is “0”.

  As shown in FIGS. 5 and 7, the drive control circuit 15 includes a microcomputer 18 having an A / D conversion input terminal that performs A / D conversion on an input signal, and a PWM duty output from the microcomputer 18. The command values Du, Dv, and Dw are inputted, and PWM signals Pu1, Pv1, Pw1 having duty ratios corresponding to the PWM duty command values Du, Dv, and Dw for the switching elements Tru1 to Trw4 of the inverter circuits 34u, 34v, and 34w and their The FET gate drive circuit 19 outputs PWM signals Pu2, Pv2, and Pw2 in which on / off is inverted. Here, the FET gate drive circuit 19 has a PWM pulse generation up / down counter constituted by a software counter provided therein, and a triangular wave formed by the count value of this counter and duty command values Du to Dw. Based on this, PWM signals Pu1 to Pw1 and Pu2 to Pw2 are formed.

  The microcomputer 18 receives the motor currents Iau to Iaw detected by the motor current detectors 17u to 17w and the steering torque detection value T output from the steering torque sensor 3 to the A / D conversion input terminal. ing. Further, the vehicle speed detection value V detected by the vehicle speed sensor 16 and the phase detection signal detected by the phase detection unit 34 are converted into the electrical angle θ by the electrical angle conversion unit 50 and input to the other input terminals of the microcomputer 18. At the same time, the motor angular velocity ω calculated by differentiating the electrical angle θ by the motor angular velocity converter 51 as a rotation speed detector is input.

  In the microcomputer 18, a central processing unit (CPU) 18 a that executes arithmetic processing, a ROM 18 b that stores a processing program for arithmetic processing executed by the central processing unit 18 a, and values required in the arithmetic process of the central processing unit 18 a and At least a RAM 18c for storing calculation results is provided, and the steering control process shown in FIG. 7 is executed by the central processing unit 18a, and the current detection process shown in FIG. 10 is executed.

As shown in FIG. 10, in the steering control process, first, in step S1, the detected torque value T detected by the steering torque sensor 3 is read, and then the process proceeds to step S2, where the neutral voltage V ₀ is obtained from the detected torque value T. The steering torque Ts (= T−V ₀ ) is calculated by subtraction. Next, the process proceeds to step S3, where the vehicle speed detection value V detected by the vehicle speed sensor 16 is read, and then the process proceeds to step S4, where the steering assist command value calculation shown in FIG. 11 is performed based on the steering torque Ts and the vehicle speed detection value V. the map is referenced to calculate the steering assist command value I _T to be the motor current command value.

Here, the steering assist command value calculation map, as shown in FIG. 8, the horizontal axis represents the steering torque detection value T, the vertical axis represents the steering assist command value I _T, and the vehicle speed detecting value V as a parameter characteristic A straight line portion L1 that is configured by a line diagram and extends at a relatively gentle gradient regardless of the vehicle speed detection value V until the steering torque Ts increases in the positive direction from “0” and reaches the first set value Ts1. When the steering torque Ta increases from the first set value Ts1, the straight portions L2 and L3 extending with a relatively gentle gradient and the steering torque detection value Ts are the first in the state where the vehicle speed detection value V is relatively fast. Straight line portions L4 and L5 that are parallel to the horizontal axis in the vicinity of the second set value Ts2 that is larger than the set value Ts1, and in the state where the vehicle speed detection value V is slow, the straight line portions L6 and L7 having a relatively large gradient, Greater slope than straight lines L6 and L7 Four characteristic lines composed of line parts L8 and L9, a straight line part L10 having a larger gradient than the straight line part L8, and straight line parts L11 and L12 extending in parallel with the horizontal axis from the end of the straight line parts L9 and L10 Similarly, when the steering torque Ts increases in the negative direction, four characteristic lines to be pointed with the above and the origin are formed.

Next, the process proceeds to step S5, where the motor acceleration ω calculated by the motor angular velocity conversion unit 51 is read, and then the process proceeds to step S6, where the motor angular velocity ω is multiplied by the inertia gain K _i to accelerate / decelerate the motor inertia. The inertia compensation value I _i (= K _i · ω) for inertia compensation control for obtaining the steering feeling without inertia is calculated by removing the torque from the steering torque Ts, and the absolute value of the steering assist command value I _T is calculated. Is multiplied by the friction coefficient gain K _f to eliminate the influence of the friction of the power transmission unit and the electric motor on the steering force, and the friction compensation value I _f (= K _f · | I _T |) for friction compensation control Is calculated. Here, the sign of the friction compensation value _If is determined on the basis of the sign of the steering torque Ts and the steering direction signal for determining whether the steering is increased / returned based on the steering torque Ts.

Next, the process proceeds to step S7, where the steering torque Ts is differentiated to calculate a center response improvement command value Ir for ensuring stability in the assist characteristic dead zone and compensating for static friction, and then the process proceeds to step S8. The calculated inertia compensation value I _i , friction compensation value _If and center response improvement command value Ir are added to the steering assist command value I _T to obtain the steering assist compensation value I _T ′ (= I _T + I _i + I _f + Ir). After calculating, the process proceeds to step S9.

In this step S9, the motor electrical angle θ converted by the electrical angle conversion unit 50 is read, and then the process proceeds to step S10, where U to W shown in FIGS. 9A to 9C based on the motor electrical angle θ. Referring to the phase current calculation map, U to W phase current command values Iu to Iw are calculated.
Here, as shown in FIGS. 9A to 9C, the phase current calculation map is formed as a trapezoidal quasi-rectangular wave in which the third and fifth harmonics are superimposed on a sine wave and the corners are rounded. The relationship between the phase current command values Iu to Iw having the same waveform as the induced voltage waveform of the armature windings Lu to Lw of the wireless brushless motor 12 and the electrical angle θ is expressed, and each phase current command value Iu to Iw is expressed. Are 120 ° out of phase with each other.

Next, the process proceeds to step S11, the phase assist target value I _T ′ is multiplied by the phase current command values Iu to Iw to calculate the phase current target values I _TU ^{* to} I _TW ^* , and then the process proceeds to step S12. Then, after reading the digital motor currents Idu to Idw obtained by A / D converting the motor currents Iau to Iaw read from the motor current detection units 17u to 17w stored in the RAM by the current detection process described later, the process proceeds to step S13. To do.

In step S13, motor currents Idu to Idw are subtracted from phase current target values I _TU ^{* to} I _TW ^* to calculate current deviations ΔIu to ΔIw, and then the process proceeds to step S14, and the following (1) to (3 ) To calculate voltage command values Vv to Vw.
Vu = Kp × ΔIu + Ki∫ΔIudt (1)
Vv = Kp × ΔIv + Ki∫ΔIvdt (2)
Vw = Kp × ΔIw + Ki∫ΔIwdt (3)
Here, Kp is a proportional gain, and Ki is an integral gain.

Next, the process proceeds to step S15, and after performing a voltage limiting process for limiting each of the voltage command values Vu to Vw calculated in step S14 with the positive and negative battery voltages ± Vb, the process proceeds to step S16.
In step S16, U to W-phase duty command values Du to Dw are calculated by performing calculations of the following equations (4) to (6) based on the voltage command values Vu to Vw whose voltages are limited.
Du = 50 + (Vu / 2Vb) × 100 (4)
Dv = 50 + (Vv / 2Vb) × 100 (5)
Dw = 50 + (Vw / 2Vb) × 100 (6)

Next, the process proceeds to step S17, where the duty command values Du to Dw calculated in step S16 are output to the gate drive circuit 19, and then the process returns to step S1.
The process of FIG. 7 corresponds to the drive control unit.
Further, the microcomputer 18 executes a current detection process shown in FIG. 10 in which motor currents Iau to Iaw having voltage values detected by the motor current detection units 17u to 17w are calculated as digital values.

As shown in FIG. 10, this current detection process is executed as a timer interruption process every predetermined time, for example, 250 usec. First, in step S31, the duty command value Dj (j = u, v, w) is read and it is determined whether or not this duty command value Dj is less than the hysteresis lower limit threshold D _{S1 in the} vicinity of 50%, and when Dj <D _S1 . the process proceeds to step S32, below the location flag FD hysteresis lower threshold D _S1 duty ratio indicating whether the duty ratio is present above the or the hysteresis upper threshold D _S2 is present below the hysteresis limit threshold D _S1 The process proceeds to step S33 after resetting it to “0” indicating that it exists on the side.

In step S33, the count value N of the PWM pulse generation up / down counter that forms a triangular wave for performing pulse width modulation (PWM) constituted by a software counter provided in the gate drive circuit 19 is read, and then in step S34. It is determined whether or not the count value N has reached the maximum value N _MAX indicating the upper vertex of the triangular wave. If N <N _MAX , the process returns to step S33, and if N = N _MAX , step S35. Migrate to

  In this step S35, the motor current Iaj input from the motor current detection unit 17j is read, and then the process proceeds to step S36 to execute an A / D conversion process for converting the motor current Iaj into a digital value, and then to step S37. Then, the net digital motor current is calculated by subtracting the reference voltage Vref of the operational amplifier OPj of the motor current detector 17j from the digital motor current calculated by the A / D conversion process, and then the process proceeds to step S38. The calculated net digital motor current is negatively signed when the presence position flag FD = "0" and positively signed when the presence position flag FD = "1", and the digital motor current Idj is obtained. Then, the process proceeds to step S39, the calculated digital motor current Idj is stored in the RAM, and then the timer interrupts Exit the sense to return to the predetermined main program.

When the determination result in step S31 is Dj ≧ D _S1 , the process proceeds to step S40, and the hysteresis upper limit threshold D _S2 in which the duty command value Dj is set to a value larger than this when the duty ratio is near 50%. If Dj> D _S2 , the process proceeds to step S42. If Dj ≦ D _S2 , the process proceeds to step S41, and the presence position flag FD is reset to “0”. If it is set to “1”, the process proceeds to step S33. If the presence position flag FD is reset to “0”, the process proceeds to step S43.

In step S42, the presence position flag FD is set to “1” indicating that the duty ratio is 100% from the hysteresis upper limit threshold D _S2 , and then the process proceeds to step S43.
In this step S43, the count value N of the PWM pulse generating up / down counter described above is read, and then the process proceeds to step S44 to check whether the count value N has reached the minimum value 0 indicating the lower vertex of the triangular wave. If N> 0, the process returns to step S43. If N = 0, the process proceeds to step S35.
The processing of FIG. 10 and the motor current detection units 17u to 17w correspond to current detection means.

Next, the operation of the first embodiment will be described.
Now, the vehicle is stopped with the ignition switch IG turned off, the power is not supplied to the drive control circuit 15, and the unconnected motor 12 is also supplied with current to the armature windings Lu to Lw. It is assumed that it has stopped.
In this state, when the ignition switch IG is turned on, power is supplied from the battery B to the drive control circuit 15 and the microcomputer 18 of the drive control circuit 15 is activated to turn on the relay RY. As a result, power from the battery B is supplied to the inverter circuits 34u to 34w, and the steering torque sensor 3, the vehicle speed sensor 16, and the position detection unit 35 are activated.

For this reason, the central processing unit 18a of the microcomputer 18 starts the processing shown in FIGS. 7 and 10 after performing a predetermined initialization process, and the FET gate driving circuit 19 uses a PWM pulse constituted by a software counter. The generation counter is activated.
At this time, in the current detection process of FIG. 10, “0” is stored as an initial value of the digital motor currents Idu to Idw in the RAM 18 c in the initialization process, and the presence position flag FD is reset to “0”.

In this state, the steering wheel 1 is not steered, the steering torque detection value T detected by the steering torque sensor 3 is the voltage V ₀ , and the vehicle speed V detected by the vehicle speed sensor 16 when the vehicle is stopped is also It is assumed that it is “0”.
When the steering control process shown in FIG. 10 is executed by the central processing unit 18a of the microcomputer 18 in this state, the steering torque detection value T is the voltage V ₀ , so the steering torque Ts calculated in step S2 is “0”. next, since the vehicle is also "0" vehicle speed detection value V have stopped, steering assist command value I _T is also "0", which is calculated with reference to the control map of FIG. 11, each compensation value I _i, Since _If and Ir are also “0”, the steering assist compensation value I _T ′ is also “0”.

  At this time, if the phase of the rotor 20 detected by the phase detection unit 35 of the wireless motor 12 is supplied to the electrical angle conversion unit 50 and the electrical angle θ at this time is, for example, 0 °, FIG. The phase current command value Iu for the U phase calculated with reference to the phase current command value calculation maps shown in (a) to (c) is “0”, and the phase current command value Iv for the V phase is the phase current command value Iu. On the other hand, since the phase is delayed by 120 °, −Imax is obtained, and the phase current command value Iw of the W phase is + Imax because the phase is advanced by 120 ° with respect to the phase current command value Iu.

The phase current target values I _TU ^* , I _TV ^*, and I _TW ^* are calculated by multiplying the phase current command values Iu, Iv, and Iw and the steering assist command value I _T. (Step S11).
Further, since the digital motor currents Idu, Idv, and Idw stored in the RAM 18c are also “0” as the initial values, the current deviations ΔIu, ΔIv, and ΔIw are also “0”, and the voltage calculated based on them. The command values Vu, Vv, and Vw are also “0”, the duty command values Du, Dv, and Dw are all 50%, and 50% duty command values Du, Dv, and Dw are output to the FET gate drive circuit 19. .

  For this reason, the on / off ratios of the PWM signals Pu1, Pv1, Pw1 and Pu2, Pv2, Pw2 output from the FET gate drive circuit 19 are substantially equal. For example, when viewed in the inverter circuit 34u, the switching elements Tru1 and Tru4 are turned on. Since the time when the switching element Tru2 and the switching element Tru3 are turned on is equal and these are alternately performed, no average current flows through the electronic winding Lu, and the other inverter circuits 34v and 34w similarly. Since the average current does not flow through the armature windings Lv and Lw, the non-connection type brushless motor 12 maintains the stopped state.

When the wireless brushless motor 12 is stopped, the count value N of the PWM pulse generating up / down counter in the FET gate drive circuit 19 is changed from “0” to N at a constant PWM cycle as shown in FIG. The up-count up to _MAX and the down-count from N _MAX to “0” are repeated, and the PWM signals Paj and Pbj output from the gate drive circuit 19 based on the 50% duty command value Dj are shown in FIG. As shown in (c) and (c), the pulses are in the opposite phase with the same on / off ratio in synchronization with the PWM period of the PWM pulse generating up / down counter.

For this reason, the actual motor current Imj flowing in the armature winding Lj of the wireless brushless motor 12 increases slightly from negative to positive at “0” as shown in FIG. The state of slightly decreasing from positive to negative is repeated.
At this time, the motor current Iaj output from the operational amplifier OPj of the motor current detector 17 repeats a state of increasing from negative to positive in the vicinity of the reference voltage Vref as shown in FIG.

For this reason, when the current detection process of FIG. 10 is executed, the duty command value Dj is 50%, which is larger than the hysteresis lower limit threshold D _{S1 and} smaller than the hysteresis upper limit threshold D _S2. Then, the process proceeds to step S41, and the existence position flag FD is reset to “0”. Therefore, the process proceeds to step S43, the count value N of the PWM pulse generation counter is read, and the count value N becomes “0”. In step S35, the motor current Iaj output from the operational amplifier OPj is read (step S35), A / D conversion processing is performed on the read motor current Iaj to calculate a digital motor current (step S36). The net motor current is calculated by subtracting the reference voltage Vref from (step S37), Based on the lag FD value calculated digital motor current Idj performing a negative sign with (step S38), and stores it in the RAM (step S39). For this reason, the calculated digital motor current Idj also maintains substantially “0”.

When the driver performs a stationary operation to steer the steering wheel 1 to the right, for example, from the stop state of the wireless brushless motor 12 when the vehicle is stopped, the steering torque sensor 3 responds to the steering torque of the driver accordingly. The detected steering torque value T becomes higher than the voltage V ₀ , and the steering torque Ts becomes a large positive value.
Therefore, the steering assist command value I _T calculated with reference to the steering assist command value calculation map of FIG. 8 becomes a relatively large positive value, and the steering assist is obtained by adding the compensation values I _i , I _f , Ir to this. The compensation value I _T ′ is calculated (step S8), and this is multiplied by the positive phase current command values Iu, Iv and Iw calculated with reference to the phase current calculation maps shown in FIGS. Therefore, the phase current target values I _TU ^* , I _TV ^*, and I _TW ^* with the amplitude as the steering assist command value I _T are calculated (step S11).

At this time, since the A / D converted digital motor currents Idu, Idv and Idw maintain “0”, the current deviations ΔIu, ΔIv and ΔIw are the phase current target values I _TU ^* , I _TV ^* and I _TW. ^* Is calculated as it is, and based on this, relatively large voltage command values Vu, Vv, and Vw are calculated. When the voltage command values Vu, Vv, and Vw at this time exceed the battery voltage + Vb, the battery voltage It is limited to + Vb (step S15).

  Since the duty command values Du, Dv, and Dw are calculated based on the limited voltage command values Vu, Vv, and Vw, the duty command values Du, Dv, and Dw are calculated. Since this is output to the gate drive circuit 19, the PWM signal Paj output from the gate drive circuit 19 is longer in the ON period than in the OFF period, for example, as shown in FIG. As shown in FIG. 12C, the off section becomes longer than the on section. For this reason, in the inverter circuit 34j, as shown in FIG. 13, a motor current flows from the switching element Trj1 to the ground through the terminal tja, the armature winding Lj, the terminal tjb, and the switching element Trj4, and the wireless brushless motor 12 is rotated clockwise, for example. Is driven to rotate. Therefore, an auxiliary steering force corresponding to the target auxiliary steering torque Tt based on the steering torque T is generated by the wireless motor 12, and this auxiliary steering force can be transmitted to the steering shaft 2 via the reduction gear 11. The driver can perform light steering.

Then, as shown in FIG. 12, when the duty command value Dj exceeds the hysteresis upper limit threshold D _S2 and is close to 100%, the motor current flowing in the armature winding Lj is Imj as shown in FIG. Assuming that the time when flowing in the direction from the terminal tja to the terminal tjb is positive, the PWM signal Paj repeatedly increases from a positive predetermined value in the ON section with a gentle slope and gradually decreases in the OFF section. As shown in FIG. 12E, the motor current Iaj output from the operational amplifier OPj of the current detection unit 17j increases from a high value corresponding to the motor current Imj with respect to the reference voltage Vref in the ON period of the PWM signal Paj. When the PWM signal Paj is inverted from the on state to the off state, the increase starts from a symmetrical voltage across the reference voltage Vref with respect to the voltage at that time. When PWM signal Paj is inverted from the OFF state to the ON state, repeating to start increasing from symmetrical voltage across the reference voltage Vref relative to the voltage at that time.

  In this state, in the current detection process of FIG. 10, when the count value N of the PWM pulse generation counter becomes “0”, the motor current Iaj is read from the motor current detection unit 17j, and the read motor current Iaj Since the digital motor current Idj is calculated by subtracting and signing the reference voltage Vref after the A / D conversion processing, the digital motor current Idj is the actual armature winding Lj shown in FIG. This is a positive value equal to the motor current Imj flowing through and is stored in the RAM 18c.

The duty command values Du, Dv, and Dw calculated in step S16 in the steering control process of FIG. 7 change due to the rotation of the motor.
For example, the PWM signal Paj output from the FET gate drive circuit 19 has an off period longer than the on period as shown in FIG. 14 (b), and conversely, the PWM signal Pbj has an on period as shown in FIG. 14 (c). It becomes longer than the off section. For this reason, in the inverter circuit 34j, as shown in FIG. 15, the motor current flows from the switching element Trj3 to the ground through the terminal tjb, the armature winding Lj, the terminal tja, and the switching element Trj2, and the unconnected brushless motor 12 is driven to rotate. The Therefore, an auxiliary steering force corresponding to the target auxiliary steering torque Tt based on the steering torque T is generated by the wireless motor 12, and this auxiliary steering force can be transmitted to the steering shaft 2 via the reduction gear 11. The driver can perform light steering.

As shown in FIG. 14, when the duty command value Dj is below the hysteresis upper limit threshold D _S1 and is close to 0%, the motor current Imj flowing through the armature winding Lj is as shown in FIG. Since the current Imj flows from the terminal tjb to the terminal tja, the PWM signal Pbj repeatedly decreases from a negative predetermined value in the on period with a gentle gradient and gradually increases in the off period, so that the motor current detector 17j As shown in FIG. 14E, the motor current Iaj output from the operational amplifier OPj increases from a high value corresponding to the absolute value of the motor current Imj with respect to the reference voltage Vref in the ON period of the PWM signal Pbj. When the PWM signal Pbj is inverted from the on state to the off state, the increase starts from a symmetrical voltage across the reference voltage Vref with respect to the voltage at that time. When PWM signal Pbj is inverted from the OFF state to the ON state, repeating to start increasing from symmetrical voltage across the reference voltage Vref relative to the voltage at that time.

In this state, since Dj <D _S1 in the current detection process of FIG. 10, the process proceeds from step S31 to step S32, the existence position flag FD is reset to “0”, and the count value of the PWM pulse generation counter When N reaches the maximum value N _MAX , the motor current Iaj is read from the motor current detector 17j, the A / D conversion process is performed on the read motor current Iaj, and the net motor current is calculated. Since the digital motor current Idj is calculated by performing negative signing based on the presence position flag FD value, the digital motor current Idj is the motor current Imj flowing in the actual armature winding Lj shown in FIG. This is a negative value equal to and stored in the RAM 18c.

Furthermore, the duty command value Dj may be larger than the hysteresis lower limit threshold Ds1 and smaller than the hysteresis upper limit threshold Ds2 due to the rotation of the motor.
As described above, when the duty command value Dj changes from the state in which the duty command value Dj is higher than the hysteresis upper limit threshold value Ds2 to the state in which the duty command value Dj is lower than the hysteresis upper limit threshold value Ds2, in the current detection process shown in FIG. When the hysteresis upper limit threshold value D _S2 is exceeded, the existence position flag FD is set to “1”. Therefore, when the duty command value Dj becomes equal to or less than the hysteresis upper limit threshold value D _S2 , steps S31 to S40 are performed. Then, the process proceeds to step S41, and the presence position flag FD is set to “1”. Therefore, the process proceeds to step S33, which triggers an A / D conversion process in case the motor current Imj is in the negative direction. The count value N of the PWM pulse generation counter is set to the maximum value N _MAX . change.

Since the trigger timing is changed before the duty command value Dj is reduced to 50%, a value smaller than the reference voltage Vref is temporarily A / D converted to calculate the digital motor current Idj.
However, since the digital motor current Idj at this time has a duty ratio in the vicinity of 50%, the timing for reading the motor current Idj can be sufficiently secured, and since the encoding is performed based on the presence position flag FD value, it is again performed. A digital motor current Idj equal to the motor current Imj can be calculated.

Similarly, when the duty ratio Dj is smaller than the hysteresis lower limit threshold D _S1 , the existence position flag FD is reset to “0”. From this state, when the duty command value Dj becomes equal to or higher than the hysteresis lower limit threshold D _S1 , FIG. In the process, the process proceeds from step S31 to step S40 to step S41, and the existence position flag FD is reset to “0”. Therefore, the process proceeds to step S43, and the trigger timing for the A / D conversion process generates the PWM pulse. When the count value N of the counter becomes “0”, the digital motor current Idj is calculated by temporarily A / D converting a value smaller than the reference voltage Vref. Since the encoding is performed based on the digital motor current Idj equal to the actual motor current Rukoto can.

  As described above, in the current detection process of FIG. 10, since the hysteresis characteristic is provided for the duty command value Dj, the vehicle is traveling straight, and the steering force transmitted from the driver to the steering wheel 1 is small. When the command value Dj changes slightly in the vicinity of 50%, it is possible to reliably prevent hunting from occurring at the trigger timing for the A / D conversion process, and to ensure a stable steering state. .

  Moreover, the operational amplifier OPj of the motor current detector 17j amplifies the voltage across the terminals of the shunt resistor Rj inserted between the connection point of the switching elements Trj2 and Trj4 and the ground, and detects it as a change with respect to the reference voltage Vref. Since the motor current Iaj output from the operational amplifier OPj does not include information indicating the direction of the current flowing through the armature winding Lj of the wireless brushless motor 12, the output dynamic range of the operational amplifier OPj is A value obtained by adding upper and lower margins to a voltage range from a voltage slightly lower than the reference voltage Vref to a maximum voltage corresponding to the motor maximum current when the trigger timing of the A / D conversion processing of the reference voltage Vref is changed. Expressed in motor current amount A / bit per bit when A / D conversion processing is possible. It is possible to improve nearly double the conventional current detection accuracy by reducing the appropriate bit rate, it is possible to apply inexpensive microcomputer.

For this reason, good steering assist control can be performed without affecting the steering assist control of the electric power steering apparatus, and the drive control circuit can be configured at low cost.
Further, in this embodiment, the motor is not one in which one end or both ends of the excitation coil are connected to each other like the conventional Y-connection type motor or Δ-connection type motor, but each excitation coil Lu to Lw forming a three-phase brushless motor. Is a non-wired brushless motor 12 wound independently without being connected to each other, and therefore it is possible to individually control energization with each of the exciting coils Lu to Lw. A pseudo rectangular wave current including harmonics can be energized without any limitation. Therefore, the motor current waveform is a quasi-rectangular wave that is wide and rounded with respect to a sine wave similar to the counter electromotive voltage waveform.

For this reason, since the output of the wireless motor 12 is output = current × voltage = torque × rotational speed, the effective value is remarkably improved as compared with the case of using a sine wave back electromotive force and drive current. Therefore, a large output can be obtained and a constant output without torque ripple can be obtained.
On the other hand, in the conventional connection type brushless motor, the counter electromotive voltage waveform can be a pseudo rectangular wave substantially the same as that of the present embodiment shown in FIG. Since the third-order harmonic component cannot flow, the current waveform becomes a pseudo rectangular wave having a narrower width than the pseudo rectangular wave of the present embodiment shown in FIG. 17B, as shown in FIG. Although the area becomes smaller and the effective value is better than that of the sine wave, it will be lower than that of this embodiment, and the output will be reduced accordingly.

Further, by using the wireless brushless motor 12, the inverter circuits 34 u, 34 v and 34 w are respectively connected to both ends of each exciting coil, and both ends of these exciting coils Lu, Lv and Lw are driven in opposite phases, thereby As described above, the inter-terminal voltages Vuab, Vvab, and Vwab of each exciting coil are expressed by the following equations (7), (8), and (9).
Vun = 2 × V ₀ × sin (ωt + α) (7)
Vvn = 2 × V ₀ × sin (ωt−2π / 3 + α) (8)
Vwn = 2 × V ₀ × sin (ωt−2π / 3 + α) (9)

On the other hand, in the case of the Y-connection motor having the same configuration, as shown in FIG. 18, an equivalent circuit is such that the voltage Vn at the neutral point where one end of each of the excitation coils Lu, Lv and Lw is connected to each other is Therefore, the inter-terminal voltages Vun, Vvn, and Vwn of the respective excitation coils Lu, Lv, and Lw are expressed by the following formulas (10), (11), and (12).
Vun = V ₀ × sin (ωt + α) (10)
Vvn = V ₀ × sin (ωt−2π / 3 + α) (11)
Vwn = V ₀ × sin (ωt−2π / 3 + α) (12)

  For this reason, taking the exciting coil Lu as an example, the terminal voltages Vua and Vub and the inter-terminal voltage Vuab of the unconnected motor 12 according to the present invention are as shown in FIG. The terminal voltage Vu, terminal voltage Vv, inter-terminal voltage Vuv, and neutral point voltage Vn are as shown in FIG. On the other hand, the inter-terminal voltages Vuab, Vvab, Vwab of the wireless motor 12 according to the present invention are as shown in FIG. 20A, and the coil end voltages Vun, Vvn, Vwn in the case of the conventional Y-wired motor are shown in FIG. 20 (b).

  As can be seen from FIGS. 19 and 20, when comparing the voltage amplitudes that can be applied to both ends of the exciting coil, the unconnected motor 12 has the same effect as when the Y-connected motor is driven with twice the power supply voltage. Is obtained. Accordingly, when the battery voltage Vb is the same, the drive voltage of the exciting coils Lu to Lw can be improved in the wireless motor. Therefore, when the steering wheel 1 is steered rapidly, there is no shortage of voltage and optimal steering assistance is achieved. Force can be generated and smooth steering can be performed.

  In the above embodiment, as shown in FIG. 16, the case where the hysteresis characteristic for changing the trigger timing of the A / D conversion process is set based on the duty command values Du to Dw has been described. As shown in FIG. 21, the hysteresis lower limit threshold value set with “0” sandwiched between the digital currents Idu to Idw calculated in the previous process stored in the RAM 18c, as shown in FIG. Hysteresis characteristics may be given based on Ih and the hysteresis upper threshold + Ih.

  In this case, the current detection process executed by the central processing unit 18a of the microcomputer 18 may be changed as shown in FIG. That is, in the current detection process of FIG. 22, the processes of steps S31 and S40 are omitted in the process of FIG. 10 described above, and instead, the digital motor current Idj calculated during the previous current detection process stored in the RAM 18c. Is smaller than the hysteresis lower limit threshold value −Ih. When Idj <−Ih, the process proceeds to step S32, and when Idj ≧ −Ih, the process proceeds to step S51 and stored in the RAM 18c. It is determined whether or not the digital motor current Idj exceeds the hysteresis upper limit threshold value + Ih. When Idj ≦ + Ih, the process proceeds to step S39, and when Idj> + Ih, the process proceeds to step S52. Except that, the same processing as in FIG. 0 and the corresponding process given the same step numbers in the detailed description thereof will be omitted.

In the current detection process of FIG. 22, when the motor current Idj stored in the RAM 18c at the time of the previous process is smaller than the hysteresis lower limit threshold −Ih, the existence position flag FD is reset to “0” and then the PWM pulse generation counter counts. When the value N reaches the maximum value N _MAX , the motor current Iaj is read from the motor current detector 17j and A / D conversion processing is performed. Then, the net motor current is calculated, and a positive or negative sign is added to the digital value. The motor current Idj is stored in the RAM 18c. Conversely, when the motor current Idj stored in the RAM 18c exceeds the hysteresis upper limit threshold value + Ih, the presence position flag FD is set to "1" and then the PWM pulse generation counter is counted. When the value N becomes the minimum value 0, the motor current Iaj is output from the motor current detector 17j. , The A / D conversion process is performed, the net motor current is calculated, the sign of this is negated, and the digital motor current Idj is stored in the RAM 18c, and the motor current Idj stored in the RAM 18c is -Ih ≦ When Idj ≦ + Ih, the process proceeds to step S41. When the presence position flag FD is reset to “0”, the process proceeds to step S43. When the existence position flag FD is set to “1”, the process proceeds to step S33. Migrate to

For this reason, the same operational effects as those of the above-described embodiment can be obtained only by changing the determination criterion of the trigger timing of the A / D conversion process to the digital motor current Idj.
In the above embodiment, the motor current detectors 17u to 17w have been described between the connection points of the switching elements Trj2 and Trj4 in the inverter circuits 34u to 34w and the ground, but the present invention is not limited to this. A shunt resistor Ru to Rw is inserted between the connection point of the switching elements Trj1 and Trj3 in each of the inverter circuits 34u to 34w and the positive side of the battery B, and the voltage across the shunt resistor Ru to Rw is an operational amplifier. Even if it detects by OPu-OPw, the effect similar to the said embodiment can be acquired.

  Furthermore, in the above-described embodiment, the case where the FET gate drive circuit 19 has the PWM pulse generation up / down counter configured as a software counter has been described. However, the present invention is not limited to this, and the PWM configured by hardware is used. A pulse generation up / down counter may be used, and further, a triangular wave generator having another configuration capable of notifying the central processing unit 18a of the microcomputer 18 of the upper and lower vertices of the triangular wave may be applied.

Furthermore, in the above-described embodiment, the case where the microcomputer 18 and the FET gate drive circuit 19 are separate has been described. However, the present invention is not limited to this, and the FET gate is connected to the central processing unit 18a of the microcomputer 18. The function of the drive circuit 19 may be provided.
Furthermore, in the above-described embodiment, the case where the induced voltage waveform and the drive current waveform of the wireless motor are the same pseudo-rectangular wave shape has been described. However, the present invention is not limited to this. Even if only the amplitude is changed without changing the phase and shape of the current waveform, the same effect as the above embodiment can be obtained.

In the above embodiment, the case where the pseudo-rectangular wave is formed by superimposing the third and fifth harmonics on the sine wave has been described. However, the present invention is not limited to this. Waves may be superimposed in an arbitrary combination, or a current waveform of only a sine wave without harmonics superimposed may be used. In this case, the phase current calculation map may be a three-phase sine wave.
Furthermore, in the above embodiment, the case where the drive control circuit 15 has a simple configuration has been described. However, the present invention is not limited to this, and the vector control d and q component components can be obtained using the excellent characteristics of vector control. After determining the current command value, the current command value is converted into each phase current command value corresponding to each exciting coil Lu to Lw, thereby calculating the phase current target values I _TU ^* , I _TV ^* and I _TW ^* . Alternatively, all may be performed by vector control.

Furthermore, in the above-described embodiment, the case where the armature windings Lu to Lw of the wireless brushless motor 12 are arranged to correspond to the Y-connected motor has been described. However, the present invention is not limited to this. A winding arrangement corresponding to the Δ-connection type motor can also be adopted.
Furthermore, in the above-described embodiment, the case where the present invention is applied to a non-wired three-phase brushless motor has been described. However, the present invention is not limited to this. It can also be applied to brushless motors or other motors.

1 is a system configuration diagram illustrating a first embodiment when the present invention is applied to an electric power steering apparatus. It is a characteristic view which shows the output characteristic of the steering torque detection value output from a steering torque sensor. It is sectional drawing which shows a non-connection type motor. It is a perspective view which shows the rotor of FIG. It is a block diagram which shows the drive circuit of a non-connection type motor. It is a circuit diagram which shows the equivalent circuit of a non-connection type motor. It is a flowchart which shows an example of the steering control processing procedure performed with the central processing unit of a microcomputer. It is a characteristic diagram which shows a steering assistance command value calculation map. It is a characteristic diagram which shows a phase current command value calculation map. It is a flowchart which shows an example of the electric current detection process procedure performed with the central processing unit of a microcomputer. It is a time chart supplied to description of an electric current detection process when a duty command value is 50%. It is a time chart with which it uses for description of the electric current detection process when a duty command value exceeds 50%. It is explanatory drawing which shows the electric current direction in an inverter circuit when a duty command value exceeds 50%. It is a time chart with which it uses for description of an electric current detection process when a duty command value is less than 50%. It is explanatory drawing which shows the electric current direction in an inverter circuit when a duty command value is less than 50%. It is explanatory drawing which shows the hysteresis characteristic of the A / D conversion process trigger timing of the motor current based on a duty command value. It is a time chart which shows a motor current waveform and a counter electromotive voltage waveform. It is a circuit diagram which shows the equivalent circuit of the conventional Y connection type motor. It is a characteristic diagram which shows the terminal voltage of a non-connection type motor, and the terminal voltage of a Y-connection type motor. It is a characteristic diagram which shows the coil both-ends voltage of a non-connection motor, and the coil both-ends voltage of a Y connection motor. It is explanatory drawing which shows the hysteresis characteristic of the A / D conversion process trigger timing of the motor current based on a digital motor current. It is a flowchart which shows the other example of the electric current detection processing procedure performed with the central processing unit of a microcomputer. It is a block diagram which shows a prior art example.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Steering wheel, 2 ... Steering shaft, 3 ... Steering torque sensor, 8 ... Steering gear, 10 ... Steering assist mechanism, 12 ... Wireless motor, 15 ... Drive control circuit, B ... Battery, IG ... Ignition key, 16 ... Vehicle speed sensor, 17u to 17w ... Motor current detector, Ru to Rw ... Shunt resistor, OPu to OPw ... Operational amplifier, 18 ... Microcomputer, 19 ... FET gate drive circuit, 20 ... Rotor, 21 ... Housing, 27 ... Rotor core , 28 ... rotor magnet, 31 ... stator, 33 ... exciting coil, Lu, Lv, Lw ... three-phase exciting coil, 34u-34w ... inverter circuit, 35 ... phase detector

Claims (5)

  A wireless brushless motor having a rotor provided with a permanent magnet and a stator arranged independently of each other without connecting a plurality of N-phase armature windings facing the rotor, and input to the steering system And a plurality of N inverter circuits for connecting the both ends of each armature winding of the wireless brushless motor and individually supplying a drive signal to each armature winding. And current detection means arranged on either the ground side or the power supply side of each inverter circuit, the winding current detected by the current detection means, and the steering torque detected by the steering torque detection means An electric power steering apparatus comprising: a drive control unit that drives and controls each inverter circuit.   The current detection means is configured to detect a voltage between terminals of a current detection resistor inserted on either the ground side or the power supply side of each inverter circuit, and the drive control unit is configured to detect the current detection A / D conversion means for sampling and A / D converting the voltage between the terminals detected by the means, and the sampling timing of the A / D conversion means is the duty of the pulse width modulation signal supplied to each armature winding The electric power steering apparatus according to claim 1, wherein the electric power steering apparatus is determined based on a ratio.   3. The sampling timing of the A / D conversion means is set so as to have a hysteresis characteristic of a predetermined width across a point where the duty ratio of the pulse width modulation signal is 50%. Electric power steering device.   The current detection means is configured to detect a voltage between terminals of a current detection resistor inserted in either one of a ground side and a power supply side of each inverter circuit, and the drive control unit is configured to detect the current detection means. A / D conversion means for sampling the voltage between the terminals detected in step A / D conversion, and the sampling timing of the A / D conversion means is the direction and magnitude of the drive current for each armature winding. The electric power steering apparatus according to claim 1, wherein the electric power steering apparatus is determined based on   5. The sampling timing of the A / D conversion means is set so as to have a hysteresis characteristic of a predetermined width across a point where the drive current of the armature winding is zero. Electric power steering device.

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