JP2006149146A - Drive controller of connectionless motor and electric power steering system employing it - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To generate a predetermined torque by sustaining the driving of a brushless motor even upon occurrence of abnormal turn-on of a switching element, line-to-line fault or line-to-ground fault of a motor harness, or the like, at the drive control section of an inverter, or the like. <P>SOLUTION: The drive controller of a connectionless motor comprises a connectionless brushless motor having a rotor arranged with a permanent magnet, and a stator arranged with armature windings of a plurality of N phases oppositely to the rotor and independently from each other without connecting, an inverter circuit connected individually across each armature winding and supplying a drive signal thereto, a section performing drive control of the inverter circuit, a section for detecting current abnormality of each armature winding, and an abnormal time control section for driving the connectionless brushless motor while suppressing brake force being generated therein when current abnorality of one armature winding is detected at the current abnormality detecting section. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a drive control device for a wireless motor that is independent without connecting armature windings of a stator, and an electric power steering device using the drive control device.

In general, an electric power steering apparatus uses a brushed DC motor or a brushless DC motor that uses a permanent magnet for a motor as a motor that generates a steering assist force for a steering system. The drive is controlled.
Here, in the case of a brushed DC motor, when one of the switching elements constituting the motor drive circuit continues to be on abnormally, the switching element in which both ends of the brushed DC motor BM are abnormally on and the normal switching element By forming a closed loop circuit connected via flywheel diodes inserted in parallel, the DC motor is rotated by an external force, so that a current due to the induced electromotive force flows to the closed loop circuit, and this current is DC Since this acts as an electromagnetic brake for braking the motor, in order to prevent this, a switch means is interposed between the motor driving means and the motor, and the switch means is detected when an on abnormality of the switching element is detected. Is known to return to the manual steering state by turning it off (for example, patent documents) Reference 1).

In addition, by detecting an abnormality in which one of the excitation coils of the brushless motor is in a non-conductive state, the drive current flowing through the brushless motor is reduced compared to the normal time without disconnecting the brushless motor from the steering system. It is also known to continue applying steering assist torque (see, for example, Patent Document 2).
Japanese Examined Patent Publication No. 7-96387 (first page, FIG. 1) JP-A-10-181617 (first page, FIG. 2)

  However, in the conventional example described in Patent Document 1, a switch such as a relay circuit is provided between the motor driving means and the motor in order to cut off the closed loop circuit formed when the switching element of the motor driving circuit is abnormally on. When the switch means is opened, the drive control of the DC motor is stopped when the switch means is opened, so that the steering assist force that has been generated by the DC motor is not applied. In particular, in a large vehicle, there is an unsolved problem that a large steering force is required to steer the steering wheel, which places a heavy burden on the driver.

  Further, in the conventional example described in Patent Document 2, the motor drive can be continued for the switching element OFF abnormality of the motor drive circuit. However, in the case of the switching element ON abnormality or the motor harness, There is an unresolved problem that when a fault abnormality occurs, a closed loop circuit is formed and an electromagnetic brake is generated, so that the generation of braking torque cannot be avoided.

  Therefore, the present invention has been made paying attention to the unsolved problems of the above-mentioned conventional example, and when a switching control element ON abnormality, a motor harness power supply fault, a ground fault, etc. occur in a drive control unit such as an inverter. However, it is an object of the present invention to provide a control device for a wireless motor capable of generating a predetermined torque by continuously driving a brushless motor and an electric power steering device using the same.

  To achieve the above object, a drive control device for a wireless motor according to claim 1 connects a rotor provided with a permanent magnet and a plurality of N-phase armature windings facing each other to the rotor. A wireless brushless motor having a stator arranged independently, an inverter circuit individually connected to both ends of each armature winding and supplying a drive signal to each armature winding, and the inverter circuit A drive control unit that controls the drive, an abnormality detection unit that individually detects current / voltage abnormality of each armature winding, and a current / voltage abnormality of one armature winding detected by the abnormality detection unit And an abnormal time control unit that drives the wireless brushless motor while suppressing the braking force generated by the wireless brushless motor.

  According to a second aspect of the present invention, there is provided a drive control device for a wireless motor, wherein a rotor having a permanent magnet and a plurality of N-phase armature windings are arranged independently of each other without being connected to each other. A wireless brushless motor having a stator provided thereon, an inverter circuit individually connected to both ends of each armature winding and supplying a drive signal to each armature winding, and a drive for driving and controlling the inverter circuit A control unit, an abnormality detection unit for individually detecting current / voltage abnormality of each armature winding, and when the abnormality detection unit detects current / voltage abnormality of one armature winding, Abnormality control unit that drives a brushless motor while suppressing braking force generated by the wireless brushless motor, a rotation speed detection unit that detects the rotation speed of the wireless brushless motor, and the abnormality detection circuit At A motor speed suppression unit that suppresses the rotational speed of the wireless brushless motor when a motor rotational speed detected by the rotational speed detection unit is equal to or higher than a set speed when a current / voltage abnormality of the slave winding is detected; It is characterized by having.

Further, in the invention according to claim 1 or 2, the drive control device for a wireless motor according to claim 3 is characterized in that the abnormality control unit is configured to detect current / voltage abnormality of one armature winding at the abnormality detection unit. When this is detected, only the drive control of the drive element of the inverter circuit corresponding to the armature winding is stopped.
Furthermore, the drive control apparatus for a wired motor according to a fourth aspect is the invention according to any one of the first to third aspects, wherein the abnormality detection unit includes an abnormality of a drive element constituting the inverter circuit and the inverter. An abnormality of the motor harness between the circuit and the armature winding of the wireless brushless motor is detected.

Still further, the drive control device for a wireless motor according to a fifth aspect is the invention according to any one of the first to fourth aspects, wherein the drive control unit includes a phase for each armature winding of the wireless motor. A control signal for the inverter circuit is formed based on a current command value corrected by superimposing a harmonic component on the current command value.
According to a sixth aspect of the present invention, there is provided an electric power steering apparatus using the drive control device for a wireless motor according to any one of the first to fifth aspects.

  Furthermore, an electric power steering apparatus according to a seventh aspect connects a steering torque detector for detecting a steering torque, a rotor provided with a permanent magnet, and a plurality of N-phase armature windings facing each other. A wireless brushless motor having a stator arranged independently and generating a steering assist force for the steering system, and each armature winding individually connected to both ends of each armature winding An inverter circuit that supplies a drive signal to the motor, a drive control unit that drives and controls the inverter circuit based on the steering torque detected by the steering torque detection unit, and current and voltage abnormalities of the armature windings are individually detected. And a braking force generated by the non-connection type brushless motor when the abnormality detection unit detects a current / voltage abnormality of one armature winding. It is characterized in that it comprises a abnormality control unit for driving while suppressing.

  Furthermore, an electric power steering apparatus according to an eighth aspect of the present invention includes a steering torque detector that detects a steering torque, a rotor provided with a permanent magnet, and a plurality of N-phase armature windings facing each other. A wireless brushless motor having a stator arranged independently without being connected, an inverter circuit that is individually connected to both ends of each armature winding and supplies a drive signal to each armature winding; A drive control unit that drives and controls the inverter circuit based on the steering torque detected by the steering torque detection unit, an abnormality detection unit that individually detects current / voltage abnormality of each armature winding, and the abnormality detection unit When an abnormal current / voltage abnormality is detected in one armature winding, the wireless brushless motor is driven while suppressing the braking force generated by the wireless brushless motor. A control unit, a rotation speed detection unit for detecting the rotation speed of the wireless brushless motor, and a motor detected by the rotation speed detection unit when the abnormality detection circuit detects a current / voltage abnormality of the armature winding. A motor speed suppression unit that suppresses the rotation speed of the wireless brushless motor when the rotation speed is equal to or higher than a set speed is provided.

  Still further, the drive control device for a wireless motor according to claim 9 is the invention according to claim 7 or 8, wherein the drive control unit is configured to output a phase current command value for each armature winding based on the steering torque. A phase current command value calculation unit for calculating the phase current, a motor current detection unit for detecting a phase current of each armature winding, and a drive current for each armature winding based on the phase current command value and the phase current. And a current control unit for controlling.

  An electric power steering apparatus according to a tenth aspect of the present invention includes, in the invention according to the ninth aspect, an electric angle detection circuit that detects an electric angle of the wireless motor, and the phase current command value calculation unit includes A phase current command value calculation unit for calculating a phase current command value corresponding to a counter electromotive voltage including harmonics corresponding to each armature winding of the wireless motor based on an electrical angle; and And a phase current target value calculation unit for calculating a phase current target value for each armature winding based on the steering torque detection value.

  According to the present invention, each electronic winding of a wireless brushless motor having a rotor having a permanent magnet and a stator in which a plurality of N-phase armature windings are independently arranged without being connected to each other. When an abnormality detection circuit detects individual current abnormalities and detects current and voltage abnormalities such as a power fault and ground fault in one armature winding, the controller at the time of abnormality detects a wireless brushless motor. Since driving is performed while suppressing the braking force due to the current that flows due to the induced electromotive force generated in the armature winding that has caused each current / voltage abnormality, it is possible to output drive torque even in the state where current / voltage abnormality occurs The effect is obtained.

Also, in addition to the control unit at the time of abnormality, the generation of braking force due to the induced electromotive force is suppressed by suppressing the rotation speed of the wireless brushless motor when the rotation speed of the wireless brushless motor is equal to or higher than the set speed. Thus, it is possible to secure the driving torque.
Furthermore, when a current abnormality occurs in one armature winding, the remaining normal armature windings are superimposed with secondary, third harmonic, etc. harmonic components, and the normal armature windings are pseudo-rectangular. By flowing the wave current, it is possible to reduce the pulsation of the driving torque.

  Furthermore, when an electric power steering device is configured using the drive control device for a wireless motor having the above-described effects, a current / voltage abnormality occurs in one electronic winding of the wireless brushless motor. However, since a steering assist torque can be generated with a wireless brushless motor and transmitted to the steering system, a large steering assist force does not fluctuate when a current / voltage abnormality occurs, without causing a sense of incongruity. The effect that the steering assist control can be continued is obtained.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is an overall configuration diagram showing an embodiment in which the present invention is applied to an electric power steering apparatus. In the figure, reference numeral 1 denotes a steering wheel, which is applied to the steering wheel 1 from a driver. A steering force is transmitted to a steering shaft 2 having an input shaft 2a and an output shaft 2b. The steering shaft 2 has one end of the input shaft 2a connected to the steering wheel 1 and the other end connected to one end of the output shaft 2b via a steering torque sensor 3 as steering torque detecting means.

  The steering force transmitted to the output shaft 2 b is transmitted to the lower shaft 5 via the universal joint 4 and further transmitted to the pinion shaft 7 via the universal joint 6. The steering force transmitted to the pinion shaft 7 is transmitted to the tie rod 9 via the steering gear 8 and steers steered wheels (not shown). Here, the steering gear 8 is configured in a rack and pinion type having a pinion 8a connected to the pinion shaft 7 and a rack 8b meshing with the pinion 8a, and the rotational motion transmitted to the pinion 8a is linearly moved by the rack 8b. It has been converted to movement.

A steering assist mechanism 10 for transmitting a steering assist force to the output shaft 2b is connected to the output shaft 2b of the steering shaft 2. The steering assist mechanism 10 includes a reduction gear 11 connected to the output shaft 2b and a wireless motor 12 as an electric motor that generates a steering assist force connected to the reduction gear 10.
The steering torque sensor 3 detects the steering torque applied to the steering wheel 1 and transmitted to the input shaft 2a. For example, the steering torque sensor 3 is a torsion bar (not shown) in which the steering torque is interposed between the input shaft 2a and the output shaft 2b. It is configured to convert to a torsional angular displacement and detect the torsional angular displacement with a potentiometer. The steering torque sensor 3, as shown in FIG. 2, when the steering torque input is zero, a predetermined neutral voltage V ₀ becomes, when the right turn from this state, the neutral voltage V ₀ in accordance with the increase of the steering torque When the steering torque is turned to the left from a state where the steering torque is zero, a torque detection value T that is a voltage that decreases from the neutral voltage V _{0 as the} steering torque increases is output.

  Further, as shown in FIG. 3, the wireless motor 12 has a rotating shaft 24 rotatably supported on a housing 21 via a pair of bearings 22 and 23. A rotor core 27 formed by laminating a plurality of disk-shaped electromagnetic steel plates 25 and 26 is mounted around the rotating shaft 24 between the pair of bearings 22 and 23, and the outer peripheral surface of the rotor core 27 is The rotor magnet 28 is fixed. The rotor magnet 28 uses a segment magnet as a permanent magnet for generating a field. Further, as shown in FIG. 4, on the outside of the rotor magnet 18, a cylindrical magnet that prevents the rotor magnet 28 from being scattered and displaced by bringing a flange portion 29 a formed at one end into contact with the end surface of the rotor magnet 28. A cover 29 is provided. The rotating shaft 24, the rotor core 27, the rotor magnet 28, and the magnet cover 29 constitute the rotor 20.

  A stator 31 is disposed in the housing 21 so as to face the rotor 20 in the radial direction, an annular stator core 32 fixed to the inner peripheral surface of the housing 21, and an electric machine wound around the stator core 32. And an exciting coil 33 as a child winding. As shown in FIG. 5, the exciting coil 33 is composed of, for example, three-phase exciting coils Lu, Lv and Lw, and these exciting coils Lu to Lw are wound independently without being connected to each other. It is a connection type (open type) brushless motor wiring, and inverter circuits 34u, 34v and 34w are connected between both ends of each excitation coil Lu, Lv and Lw, and drive currents Iu, Iv and Iw are individually supplied. .

  As shown in FIG. 5, the inverter circuit 34j (j = u, v, w) has four switching elements Trj1 to Trj4 configured by, for example, N-channel MOSFETs, and the switching elements Trj1 and Trj2 are connected in series. The series circuit in which the switching elements Trj3 and Trj4 are connected in series is connected in parallel to form an H-bridge circuit Hj. The connection point of the switching elements Trj1 and Trj3 of the H-bridge circuit Hj is connected to the battery B via the relay circuit RY, the connection point of the switching elements Trj2 and Trj4 is grounded via the shunt resistor Rj for current detection, The connection point of the switching elements Trj1 and Trj2 is connected to one terminal tja of the excitation coil Lj in the wireless brushless motor 12, and the connection point of the switching elements Trj3 and Trj4 is connected to the other terminal tjb of the excitation coil Lj. . Note that a flywheel diode D is connected in the forward direction between the source and drain of each of the switching elements Trj1 to Trj4.

The PWM (pulse width modulation) signal Pj1 output from the drive control circuit 15 is supplied to the switching elements Trj1 and Trj4 of each inverter circuit 34j, and the PWM (pulse width modulation) is supplied from the drive control circuit 15 to the switching elements Trj2 and Trj3. A PWM (Pulse Width Modulation) signal Pj2 having an opposite phase to that of the signal Pj1, that is, an on / off inversion is supplied.
Here, as shown in FIG. 6, the equivalent circuit of each exciting coil Lu, Lv, and Lw has a resistance R ₀ ′, an inductance L ₀ ′, a counter electromotive voltage eu () between the terminals tua and tub. = Ω × Kt ′ × sin (ωt)) are arranged in series, the terminal voltage Vua of the terminal tua is Vua = V ₀ × sin (ωt + α), and the terminal voltage Vub of the terminal tub is Vub = V ₀ × sin (Ωt−π + α), the inter-terminal voltage Vuab is Vuab = 2 × V ₀ × sin (ωt + α), and the phase current Iu is Iu = I ₀ ′ × sin (ωt).

Similarly, for the exciting coil Lv, a resistor R ₀ ′, an inductance L ₀ ′, and a counter electromotive voltage eu (= ω × Kt ′ × sin (ωt−2π / 3)) are arranged in series between the terminals tva and tvb. The terminal voltage Vva of the terminal tva is Vva = V ₀ × sin (ωt−2π / 3 + α), the terminal voltage Vvb of the terminal tu is Vvb = V ₀ × sin (ωt−2π / 3−π + α), The voltage Vvab between terminals is Vvab = 2 × V ₀ × sin (ωt−2π / 3 + α), and the phase current Iv is Iv = I ₀ ′ × sin (ωt−2π / 3).

Similarly, for the exciting coil Lw, a resistor R ₀ ′, an inductance L ₀ ′, and a counter electromotive voltage eu (= ω × Kt ′ × sin (ωt−4π / 3)) are arranged in series between the terminals twa and twb. The terminal voltage Vwa of the terminal twa is Vwa = V ₀ × sin (ωt−4π / 3 + α), the terminal voltage Vwb of the terminal tu is Vwb = V ₀ × sin (ωt−4π / 3−π + α), The voltage Vwab between terminals is Vwab = 2 × V ₀ × sin (ωt−4π / 3 + α), and the phase current Iw is Iw = I ₀ ′ × sin (ωt−4π / 3).

In addition, the motor constant is designed to be either a motor constant of a conventional Y-connection motor, a motor constant of a conventional Δ-connection motor, or a unique motor constant that satisfies the required performance, and is a three-phase connectionless system. A brushless motor is configured.
Here, the magnetized magnet of the rotor 20 is magnetized so that the induced voltage waveform of the wireless motor 12 becomes a pseudo-rectangular wave in which the third harmonic and the fifth harmonic are superimposed on a sine wave, as will be described later. A winding method of the stator 31 is set.

  Further, a phase detector 35 of the rotor 20 is disposed in the vicinity of the one bearing 22. The phase detection unit 35 includes an annular phase detection permanent magnet 36 attached to the rotary shaft 24 and a phase detection element 37 that faces the permanent magnet 36 and is fixed to the housing 21 side. Since the motor 12 is a brushless motor that does not include a mechanical commutator (brush and commutator), the phase detection unit 35 detects the phase of the rotor 20 and controls the excitation coil according to the phase under the control of the drive circuit 15. This is for energizing 33. A resolver, an encoder, or the like can also be used as the phase detection unit.

  Then, the steering torque detection value T output from the steering torque sensor 3 is input to the drive control circuit 15 to which power is supplied from the battery B via the ignition switch IG as shown in FIG. The drive control circuit 15 includes a vehicle speed detection value V detected by the vehicle speed sensor 16 in addition to the torque detection value T, and excitation coils Lu to Lw of the wireless brushless motor 12 detected by the motor current detection units 17u to 17w. The motor currents Idu to Idw flowing through the motor and the phase detection signal of the rotor 20 detected by the phase detection unit 34 are input. Here, each of the motor current detectors 17u to 17w is configured to detect the voltage between the terminals of the shunt resistors Ru to Rw by the operational amplifier OP.

  Further, the voltages of the motor harnesses MH1 to MH6 connecting between the inverter circuits 34u to 34w and the terminals tua to twb of the armature windings Lu to Lw of the unconnected brushless motor 12, that is, both ends of the unconnected brushless motor 12 are connected. An abnormality detection circuit 41 that individually detects voltages is provided, and an abnormality detection signal AS output from the abnormality detection circuit 41 is input to the drive control circuit 15.

  As shown in FIG. 7, the abnormality detection circuit 41 includes an adder circuit 42u configured by connecting the other ends of resistors R1 and R2 having one ends connected to the motor harnesses MH1 and MH2, and motor harnesses MH3 and MH4. An adder circuit 42v configured by connecting the other ends of the resistors R3 and R4 having one end connected to each other and a resistor circuit R5 and the other end of the resistors R5 and R6 having one ends connected to the motor harnesses MH5 and MH6 are connected to each other. Switching circuit 42w, and between the resistors R1 to R6 of each of the adding circuits 42u to 42w and the motor harnesses MH1 to MH6, the battery voltage Vb is detected as the on-resistance of the switching element and the resistance of the power supply / ground fault Compared with the motor excitation coil resistance value, the voltage is divided by high impedance resistors RH1 and RH2 and a voltage of Vb / 2 is applied as a bias voltage. And a voltage dividing circuit constituted by a parallel circuit of a resistor Rd and a capacitor Cf connected between the adding output to be filtered and the ground. And the filter circuit 44.

  The reason why the abnormality detection circuit 41 can detect the ON abnormality of the switching elements Tru1 to Trw4 and the abnormality of the motor harnesses MH1 to MH6 of the inverter circuits 34u to 34w is described below. As shown in FIGS. 8B and 8C, the PWM signal Pw1 supplied to the switching elements Trw1 and Trw4 and the PWM signal Pw2 supplied to the switching elements Trw2 and Trw3 are sections in which the PWM signal Pw1 is on. The PWM signal Pw2 is in the off state, and a dead time Td is provided between the two signals to avoid the on state at the same time. At this time, when the duty ratios of the PWM signals Pw1 and Pwu2 are Pw1> Pw2, as shown in FIGS. 8B and 8C, the battery B is connected to the inverter circuit 34w as shown in FIG. Current flows from the positive terminal to the ground through the switching element Trw1, the terminal twa, the exciting coil Lw, the terminal twb, and the switching element Trw4.

  For this reason, for example, the terminal voltage Vwa at the terminal twa of the exciting coil Lw of the wireless brushless motor 12 detected by the motor harness MH5 is approximately about the interval in which the PWM signal Pw1 is on as shown in FIG. The battery voltage Vb [V] is precisely the voltage Vb−Ron × Im obtained by subtracting the voltage Ron × Im obtained by multiplying the on-resistance Ron of the switching element Tru1 by the motor current Im from the battery voltage Vb, and the PWM signal Pw1 is turned off from the on state. It becomes −Vf which becomes a negative value slightly in the dead time Td section switched to the state, and is about 0 [V] in the section in which the subsequent PWM signal Pw2 maintains the on state, precisely −Ron × Im, and the next dead When the PWM signal Pw1 returns to the on state after becoming -Vf which becomes a negative value again in the time Td interval, the battery is again discharged. The voltage Vb exactly repeated that the Vb-Ron × Im.

  On the other hand, the terminal voltage Vwb at the terminal twb of the exciting coil Lw of the unconnected brushless motor 12 detected by the motor harness MH6 is a section in which the PWM signal Pw2 is in an ON state as shown in FIG. The battery voltage is exactly the voltage obtained by adding the battery voltage Vb to the voltage Ron × Im obtained by multiplying the on-resistance Ron by the motor current Im, and in the dead time Td section in which the PWM signal Pw2 is switched from the on state to the off state, It becomes a value obtained by adding a small voltage Vf to the battery voltage Vb, and after that, it is about 0 [V] accurately Ron × Im in a section where the PWM signal Pw1 is kept on.

  For this reason, when the switching elements Trw1 to Trw4 are normal and the motor harnesses MH5 and MH6 are in a normal state where no power supply fault and no ground fault have occurred, the addition circuit 42w adds the terminal voltages Vwa and Vwb to the abnormality detection circuit 41. As shown in FIG. 8F, the output addition voltage Vws matches the battery voltage Vb regardless of the duty ratio of the PWM signals Pw1 and Pw2.

However, as shown in FIG. 9A, when an ON abnormality occurs in which the switching element Trw4 of the inverter circuit 34w continues to be on, the switching element Trw1 is normal with respect to the terminal voltage Vwa. As shown in FIG. 8D, the battery voltage Vb [V] and 0 [V] are alternately changed as in FIG. 8D, but the terminal voltage Vwb is shown in FIG. As shown, it is fixed at approximately 0 [V].
For this reason, as shown in FIG. 9F, the terminal voltage Vwa appears as it is in the divided voltage Vws output from the adding circuit 42w and divided.
Accordingly, it is possible to accurately detect the terminal voltage abnormality by calculating the average value of the added voltage or measuring the level change of the waveform.

  As shown in FIGS. 5 and 7, the drive control circuit 15 includes a microcomputer 18 having an A / D conversion input terminal that performs A / D conversion on an input signal, and a PWM duty output from the microcomputer 18. The command values Du, Dv, and Dw are inputted, and PWM signals Pu1, Pv1, Pw1 having duty ratios corresponding to the PWM duty command values Du, Dv, and Dw for the switching elements Tru1 to Trw4 of the inverter circuits 34u, 34v, and 34w and their The FET gate drive circuit 19 outputs PWM signals Pu2, Pv2, and Pw2 in which on / off is inverted.

  The microcomputer 18 receives the motor currents Idu to Idw detected by the motor current detectors 17 u to 17 w at its A / D conversion input terminals, and detects the steering torque detection value T and abnormality detected from the steering torque sensor 3. Addition voltages Vus to Vws output from the circuit 41 are input. Further, the vehicle speed detection value V detected by the vehicle speed sensor 16 and the phase detection signal detected by the phase detection unit 34 are converted into the electrical angle θ by the electrical angle conversion unit 50 and input to the other input terminals of the microcomputer 18. At the same time, the motor angular velocity ω calculated by differentiating the electrical angle θ by the motor angular velocity converter 51 as a rotation speed detector is input.

The microcomputer 18 executes the steering control process shown in FIG. 10 and the abnormality detection process shown in FIG.
As shown in FIG. 10, in the steering control process, first, in step S1, the detected torque value T detected by the steering torque sensor 3 is read, and then the process proceeds to step S2, where the neutral voltage V ₀ is obtained from the detected torque value T. The steering torque Ts (= T−V ₀ ) is calculated by subtraction. Next, the process proceeds to step S3, where the vehicle speed detection value V detected by the vehicle speed sensor 16 is read, and then the process proceeds to step S4, where the steering assist command value calculation shown in FIG. 11 is performed based on the steering torque Ts and the vehicle speed detection value V. the map is referenced to calculate the steering assist command value I _T to be the motor current command value.

Here, the steering assist command value calculation map, as shown in FIG. 11, the horizontal axis represents the steering torque detection value T, the vertical axis represents the steering assist command value I _T, and the vehicle speed detecting value V as a parameter characteristic A straight line portion L1 that is configured by a line diagram and extends at a relatively gentle gradient regardless of the vehicle speed detection value V until the steering torque Ts increases in the positive direction from “0” and reaches the first set value Ts1. When the steering torque Ta increases from the first set value Ts1, the straight portions L2 and L3 extending with a relatively gentle gradient and the steering torque detection value Ts are the first in the state where the vehicle speed detection value V is relatively fast. Straight line portions L4 and L5 that are parallel to the horizontal axis in the vicinity of the second set value Ts2 that is larger than the set value Ts1, and in the state where the vehicle speed detection value V is slow, the straight line portions L6 and L7 having a relatively large gradient, Gradient magnitude from straight line parts L6 and L7 Four characteristic lines composed of straight portions L8 and L9, a straight portion L10 having a larger gradient than the straight portion L8, and straight portions L11 and L12 extending in parallel with the horizontal axis from the ends of the straight portions L9 and L10 Similarly, when the steering torque Ts increases in the negative direction, four characteristic lines to be pointed with the above and the origin are formed.

Next, the process proceeds to step S5, where the motor acceleration ω calculated by the motor angular velocity conversion unit 51 is read, and then the process proceeds to step S6, where the motor angular velocity ω is multiplied by the inertia gain K _i to accelerate / decelerate the motor inertia. The inertia compensation value I _i (= K _i · ω) for inertia compensation control for obtaining the steering feeling without inertia is calculated by removing the torque from the steering torque Ts, and the absolute value of the steering assist command value I _T is calculated. Is multiplied by the friction coefficient gain K _f to eliminate the influence of the friction of the power transmission unit and the electric motor on the steering force, and the friction compensation value I _f (= K _f · | I _T |) for friction compensation control Is calculated. Here, the sign of the friction compensation value _If is determined on the basis of the sign of the steering torque Ts and the steering direction signal for determining whether the steering is increased / returned based on the steering torque Ts.

Next, the process proceeds to step S7, where the steering torque Ts is differentiated to calculate a center response improvement command value Ir for ensuring stability in the assist characteristic dead zone and compensating for static friction, and then the process proceeds to step S8. The calculated inertia compensation value I _i , friction compensation value _If and center response improvement command value Ir are added to the steering assist command value I _T to obtain the steering assist compensation value I _T ′ (= I _T + I _i + I _f + Ir). After calculating, the process proceeds to step S9.

In step S9, the motor electrical angle θ converted by the electrical angle conversion unit 50 is read, and then the process proceeds to step S10, where U to W shown in FIGS. 12A to 12C based on the motor electrical angle θ. Referring to the phase current calculation map, U to W phase current command values Iu to Iw are calculated.
Here, as shown in FIGS. 12A to 12C, the phase current calculation map is formed as a trapezoidal quasi-rectangular wave in which the third and fifth harmonics are superimposed on a sine wave and the corners are rounded. The relationship between the phase current command values Iu to Iw having the same waveform as the induced voltage waveform of the armature windings Lu to Lw of the wireless brushless motor 12 and the electrical angle θ is expressed, and each phase current command value Iu to Iw is expressed. Are 120 ° out of phase with each other.

Next, the process proceeds to step S11, the phase assist target value I _T ′ is multiplied by the phase current command values Iu to Iw to calculate the phase current target values I _TU ^{* to} I _TW ^* , and then the process proceeds to step S12. The motor currents Idu to Idw are read from the motor current detectors 17u to 17w, and then the process proceeds to step S13 where the motor currents Idu to Idw are subtracted from the phase current target values I _TU ^{* to} I _TW ^* to obtain a current deviation ΔIu. After calculating ~ ΔIw, the process proceeds to step S14.

In step S14, voltage command values Vv to Vw are calculated by performing PI calculations of the following equations (1) to (3).
Vu = Kp × ΔIu + Ki∫ΔIudt (1)
Vv = Kp × ΔIv + Ki∫ΔIvdt (2)
Vw = Kp × ΔIw + Ki∫ΔIwdt (3)
Here, Kp is a proportional gain, and Ki is an integral gain.

Next, the process proceeds to step S15, and after performing a voltage limiting process for limiting each of the voltage command values Vu to Vw calculated in step S14 with the positive and negative battery voltages ± Vb, the process proceeds to step S16.
In step S16, U to W-phase duty command values Du to Dw are calculated by performing calculations of the following equations (4) to (6) based on the voltage command values Vu to Vw whose voltages are limited.
Du = 50 + (Vu / 2Vb) × 100 (4)
Dv = 50 + (Vv / 2Vb) × 100 (5)
Dw = 50 + (Vw / 2Vb) × 100 (6)

  Next, the process proceeds to step S17, where it is determined whether or not an abnormality flag AF set in an abnormality detection process to be described later is set to a value other than “0”, and the abnormality flag AF is set to “0”. If there is no abnormality, the motor drive system of the inverter circuits 34u to 34w, the motor harnesses MH1 to MH6, and the armature windings Lu to Lw is determined to be normal, and the process proceeds to step S18, and the duty command value calculated in step S16 After Du to Dw are output to the gate drive circuit 19, the process returns to step S1.

  On the other hand, when the determination result in step S17 is “1” to “3” other than “0”, the inverter circuits 34u to 34w, the motor harnesses MH1 to MH6, and the motors of the armature windings Lu to Lw. It is determined that an abnormality has occurred in the drive system, and the process proceeds to step S19. When the abnormality flag AF is “1”, it is determined that the U-phase drive system is abnormal, and the U-phase PWM signals Pu1 and Pu2 are detected. After outputting a PWM signal output stop command for stopping the output to the gate drive circuit 19, the process proceeds to step S20.

  In this step S20, it is determined whether or not the absolute value | ω | of the motor angular speed ω exceeds a preset value ωs that determines that the preset electric angle section in which the drive torque cannot be generated is generated. When ω ≦ ωs, it is determined that the low-speed rotation region is capable of generating drive torque in all electrical angle sections, and the process proceeds to step S21, and is set in step S16 for normal two phases. The duty command value is output to the gate drive circuit 19 and then the process returns to step S1. When | ω |> ωs, it is determined that the vehicle is in the high-speed steering region, and the process proceeds to step S22. The phase duty command value is limited to a symmetrical duty command value range DL to DH across 50% corresponding to the vicinity of the maximum speed in the low-speed steering area, and the limited duty command value is gated. And then returns the output to the dynamic circuit 19 in the step S1.

In the processing of FIG. 10, the processing of step S1 to step S18 corresponds to the drive control unit, the processing of steps S19 to S21 corresponds to the abnormality control unit, and the processing of steps S20 and S22 corresponds to the motor speed suppression unit. is doing.
Further, the microcomputer 18 executes the abnormality detection process shown in FIG. 13 for detecting an abnormality due to the inverter circuits 34u to 34w, the switching harness abnormality, the motor harnesses MH1 to MH6 and the armature windings Lu to Lw and the ground fault. .

As shown in FIG. 13, this abnormality detection process is executed as a timer interruption process at a predetermined time, for example, every 10 msec. First, in step S31, the current added voltage Vus (n) ˜ Vws (n) is read, and then the process proceeds to step S32, where the moving average calculation of the following formulas (7) to (9) is performed based on the read addition voltages Vus (n) to Vws (n). The values Vusm (n) to Vwsm (n) are calculated.
Vusm (n) = (1-a) Vusm (n-1) + a · Vus (n) (7)
Vvsm (n) = (1-a) Vvsm (n-1) + a.Vvs (n) (8)
Vwsm (n) = (1-a) Vwsm (n-1) + a.Vws (n) (9)
Here, n = 1, 2, 3..., Initial value Vusm (0) = Vvsm (0) = Vwsm (0) = Vb, a is a data weighting factor, and 0 <a <1, Vusm (n -1) is the previous moving average value.

  Next, the process proceeds to step S33, where it is determined whether or not the absolute value of the value (Vusm (n) −Vb) obtained by subtracting the battery voltage Vb from the moving average value Vusm (n) exceeds a preset threshold value Vms. , | Vusm (n) −Vb |> Vms, the ON abnormality in which the switching elements Tru1 to Tru4 of the inverter circuit 34u are fixed to the ON state in the U phase, the motor harnesses MH1 and MH2, and the armature winding Lu It is determined that the U-phase drive system abnormality has occurred, and the process proceeds to step S34, the abnormality flag AF is set to “1” indicating that the U-phase abnormality has occurred, and the timer interrupt process is terminated. Then, the program returns to the predetermined main program.

  If the determination result in step S33 is | Vusm (n) −Vb | ≦ Vms, it is determined that the U-phase drive system is normal, the process proceeds to step S35, and the battery is calculated from the moving average value Vvsm (n). It is determined whether or not the absolute value of the value obtained by subtracting the voltage Vb (Vvsm (n) −Vb) exceeds a preset threshold value Vms, and when | Vvsm (n) −Vb |> Vms, Step is determined as an abnormality in the V-phase drive system in which the switching element Trv1 to Trv4 of the inverter circuit 34v is fixed to the on state, the motor harnesses MH3 and MH4 and the armature winding Lv have a power fault and a ground fault. The process proceeds to S36, where the abnormality flag AF is set to "2" indicating that the V phase is abnormal, and then the timer interruption process is terminated and the process returns to the predetermined main program.

  Further, when the determination result in step S37 is | Vvsm (n) −Vb | ≦ Vms, it is determined that the V-phase drive system is normal, and the process proceeds to step S37, where the battery is calculated from the moving average value Vwsm (n). It is determined whether or not the absolute value of the value obtained by subtracting the voltage Vb (Vwsm (n) −Vb) exceeds a preset threshold value Vms, and when | Vwsm (n) −Vb |> Vms, Step is determined as an abnormality of the W-phase drive system in which the switching element Trw1 to Trw4 of the inverter circuit 34v is fixed to the ON state, the motor harnesses MH5 and MH6 and the armature winding Lw have a power supply fault and a ground fault. The process proceeds to S38, the abnormality flag AF is set to “3” indicating that the W phase is abnormal, the timer interruption process is terminated, and the process returns to the predetermined main program.

Furthermore, when the determination result in step S37 is | Vwsm (n) −Vb | ≦ Vms, it is determined that all of the U-phase drive system to the W-phase drive system are normal, and the process proceeds to step S39. Then, after resetting the abnormality flag AF to “0”, the timer interrupt process is terminated and the process returns to the predetermined main program.
The processing of FIG. 13 and the abnormality detection circuit 41 correspond to the abnormality detection unit.

Next, the operation of the first embodiment will be described.
Now, the vehicle is stopped and the unconnected motor 12 is also stopped, the steering wheel 1 is not being steered, and the steering torque detection value T detected by the steering torque sensor 3 is the voltage V ₀ . Shall.
When the steering control process shown in FIG. 10 is executed by the microcomputer 18 in this state, since the steering torque detection value T is the voltage V ₀ , the steering torque Ts calculated in step S2 becomes “0”, and the vehicle stops. since then the vehicle speed detection value V even if it is "0", the steering assist command value I _T is also "0", which refer to the calculated control map of FIG. 11, each compensation value I _i, also I _f and Ir Since it is “0”, the steering assist compensation value I _T ′ is also “0”.

At this time, if the phase of the rotor 20 detected by the phase detection unit 35 of the wireless motor 12 is supplied to the electrical angle conversion unit 50 and the electrical angle θ at this time is, for example, 0 °, FIG. The phase current command value Iu for the U phase calculated with reference to the phase current command value calculation maps shown in (a) to (c) is “0”, and the phase current command value Iv for the V phase is the phase current command value Iu. On the other hand, since the phase is delayed by 120 °, −Imax is obtained, and the phase current command value Iw of the W phase is + Imax because the phase is advanced by 120 ° with respect to the phase current command value Iu.
The phase current target values I _TU ^* , I _TV ^*, and I _TW ^* are calculated by multiplying the phase current command values Iu, Iv, and Iw and the steering assist command value I _T. (Step S11).

  Further, since the motor currents Idu, Idv and Idw detected by the motor current detection units 17u, 17v and 17w are also “0”, the current deviations ΔIu, ΔIv and ΔIw are also “0”, and are calculated based on these. The voltage command values Vu, Vv and Vw to be executed are also “0”, the duty command values Du, Dv and Dw are all 50%, and the wireless brushless motor 12 and its drive system are normal. 50% duty command values Du, Dv and Dw are output to the gate drive circuit 19.

  Therefore, the on / off ratios of the PWM signals Pu1, Pv1, Pw1 and Pu2, Pv2, Pw2 output from the gate drive circuit 19 are substantially equal. For example, when viewed in the inverter circuit 34u, the switching elements Tru1 and Tru4 are in the on state. Is equal to the time when the switching elements Tru2 and Tru3 are turned on, and these are alternately performed. Therefore, no current flows through the electronic winding Lu, and the other inverter circuits 34v and 34w are similarly armatured. Since no current flows through the windings Lv and Lw, the non-connection type brushless motor 12 maintains the stopped state.

When the driver performs, for example, a so-called stationary operation in which the steering wheel 1 is steered to the right, for example, from the stop state of the wireless brushless motor 12 when the vehicle is stopped, the steering torque sensor 3 changes the steering torque of the driver accordingly. The corresponding steering torque detection value T becomes higher than the voltage V ₀ , and the steering torque Ts becomes a large positive value.
Therefore, the steering assist command value I _T calculated with reference to the steering assist command value calculation map of FIG. 11 becomes a relatively large positive value, and the steering assist is obtained by adding the compensation values I _i , I _f , Ir to this. A compensation value I _T ′ is calculated (step S8), and this is multiplied by positive phase current command values Iu, Iv and Iw calculated with reference to the phase current calculation maps shown in FIGS. Therefore, the phase current target values I _TU ^* , I _TV ^*, and I _TW ^* with the amplitude as the steering assist command value I _T are calculated (step S11).

At this time, since the motor currents Idu, Idv, and Idw maintain “0”, the phase deviation target values I _TU ^* , I _TV ^*, and I _TW ^* are calculated as they are for the current deviations ΔIu, ΔIv, and ΔIw. The positive and relatively large voltage command values Vu, Vv and Vw are calculated based on this, and if the voltage command values Vu, Vv and Vw at this time exceed the battery voltage + Vb, they are limited to the battery voltage + Vb. (Step S15).

  Since the duty command values Du, Dv, and Dw are calculated based on the limited voltage command values Vu, Vv, and Vw, the duty command values Du, Dv, and Dw are larger than 50%, and this is a gate drive. Since it is output to the circuit 19, as shown in FIG. 8 (a), the inverter circuit 34u to 34w causes the current to phase 120 ° from the switching element Trj1 to the ground through the terminal tja, the armature winding Lj, the terminal tjb, and the switching element Trj4. Is different from the induced voltage waveform of the wireless brushless motor 12, and a pseudo-rectangular wave current in a trapezoidal wave state with rounded corners in which the third and fifth harmonics are superimposed on a sine wave flows and is not connected The brushless motor 12 is driven to rotate clockwise, for example. Therefore, an auxiliary steering force corresponding to the target auxiliary steering torque Tt based on the steering torque T is generated by the wireless motor 12, and this auxiliary steering force can be transmitted to the steering shaft 2 via the reduction gear 11. The driver can perform light steering.

Similarly, if the driver has left steering state outright laid the steering wheel 1, since the steering assist command value I _T is a negative value, the duty command value calculated in step S16 in the steering control process in FIG. 10 Du, Dv, and Dw are smaller than 50% and close to 0%, and the current flowing through the exciting coils Lu to Lw flows in the opposite direction to that described above, so that the wireless brushless motor 12 is driven in reverse rotation, for example, counterclockwise. Is done.

  On the other hand, in the abnormality detection process shown in FIG. 13, the average values Vusm (n) to Vwsm (n) are read by reading the addition voltages Vus to Vws output from the abnormality detection circuit 41 at predetermined time intervals and performing a moving average process on these. ) And the inverter circuits 34u to 34w, the motor harnesses MH1 to MH6, and the exciting coils Lu to Lw are normal, as described above, the terminal voltages Vja and Vjb of the inverter circuit 34j are the duty command values. ON / OFF is alternately repeated at a duty ratio according to Du to Dw, and the added value of both terminal voltages Vja and Vjb depends on the duty ratio of PWM signals Pj1 and Pj2, as shown in FIG. 8 (f). It almost coincides with the battery voltage Vb.

  Therefore, the absolute value | Vjsm (n) −Vb | obtained by subtracting the battery voltage Vb from the moving average value Vjsm (n) is substantially “0”, which is smaller than the threshold value Vms. In step S33, the process proceeds to step S39 through steps S35 and S37, and the abnormality flag AF is reset to "0" indicating normal.

However, as described above, when the steering wheel 1 is steered to the right in the stationary state, the switching element Trw4 of the inverter circuit 34w becomes an on abnormality that continues to be on even if the PWM signal Pw2 is off. The terminal voltage Vwb always continues at substantially the ground potential, that is, the potential obtained by multiplying the on-resistance R ₀ of the switching element Trw 4 by the motor current Im, so that the addition voltage output from the addition circuit 42 w of the abnormality detection circuit 41 As shown in FIG. 9F, Vws repeats substantially the battery voltage Vb and the substantially zero ground voltage, and the moving average value Vwsm (n) of the added voltage Vws becomes significantly smaller than the battery voltage Vb. Therefore, in step S33 in FIG. 13, | Vusm (n) −Vb |> Vms is established, the process proceeds to step S34, and the abnormality flag AF is set to “1”.

Therefore, the process proceeds from step S17 to step S19 in the steering control process of FIG. 10, and the PWM signal output stop for stopping the output of the PWM signals Pw1 and Pw2 for driving the U-phase inverter circuit 34w to the gate drive circuit 19 is stopped. Outputs a command. For this reason, the drive of the inverter circuit 34w is stopped and the energization control with respect to the exciting coil Lw is stopped.
However, for the remaining two normal V-phase and W-phase inverter circuits 34u and 34v, the duty command values Du and Dv calculated in step S16 are set in the state where the wireless brushless motor 12 is in the low-speed rotation region. The output of the PWM signals Pu1, Pu2 and Pv1, Pv2 based on them is continued.

  At this time, the U-phase current, the V-phase current and the W-phase current energized in the exciting coils Lu to Lw are those in which the U-phase current, the V-phase current and the W-phase current are sinusoidal for the sake of simplicity. Then, as shown in FIG. 14A, for the W-phase current in which an abnormality has occurred, the flywheel diode D-terminal twa in parallel with the switching element Trw2, the armature winding Lw, the terminal twb, and the switching element Trw4- Since a closed loop of the flywheel diode D is formed, a current due to an induced electromotive force flows along with the rotation of the rotor 20, and a motor braking torque in a direction to brake the motor is generated by this current.

  However, when the wireless brushless motor 12 is rotationally driven in the low-speed steering region, the motor braking torque generated in the abnormal W phase is small as shown by the curve La in FIG. The motor drive torque generated in the normal U phase and V phase pulsates as shown by the curve Ln in FIG. 14B, but there is no electrical angle section in which the drive torque cannot be generated, and the steering wheel 1 vibrates. However, the steering assist force can be sufficiently exerted.

  However, in the state where the wireless brushless motor 12 is driven in the high-speed turning region where the steering wheel 1 is steered suddenly, when the switching element Trw4 in the W-phase inverter 34w becomes abnormal as in the above case, The W-phase current based on the induced electromotive force generated by the closed loop formed in the inverter 34w has a negative amplitude as shown in FIG. 15 (a). As a result, the W-phase current is increased as shown in FIG. 15 (b). As a result, the motor braking torque generated by the motor is increased, thereby canceling out the motor driving torque generated by the normal U phase and V phase, and there is an electrical angle section in which the driving torque cannot be exhibited. This will cause the driver to feel uncomfortable.

  For this reason, in this embodiment, when the motor angular velocity ω of the wireless brushless motor 12 is detected and the absolute value of the motor angular velocity ω exceeds the set value ωs corresponding to the motor angular velocity in the vicinity of the upper limit of the low-speed steering region. In addition, in the process of FIG. 10, the process proceeds from step S20 to step S22, and the normal U-phase and V-phase duty command values Du and Dv at that time are changed to the duty command that becomes the motor angular velocity near the upper limit of the low-speed turning region. By limiting the value to a range of symmetrical duty command values DL to DH across 50%, the wireless brushless motor 12 is prevented from rotating in the high-speed turning area and is driven in the low-speed turning area. To continue the steering assist force generation state.

  Further, when the grounding of the motor harness MH6 or the excitation coil Lw can be detected in the same manner as described above, when the switching element Tj1 on the battery power supply side is abnormally turned on, the terminal voltage Vja is substantially equal to the battery voltage Vb. Therefore, since the terminal voltage Vjb repeats substantially the ground voltage and the battery voltage Vb, the added voltage repeats the battery Vb and a state twice that of the battery Vb, and the moving average value Vjsm (n) Becomes higher than the battery voltage Vb, and | Vjsm (n) −Vb |> Vms can be determined as an abnormal state. Furthermore, when an abnormality occurs in another U phase or V phase, it can be detected in the same manner as described above.

Further, as in the above-described embodiment, by performing drive control of the wireless brushless motor 12 without using vector control, the arithmetic processing can be simplified and a simple drive control unit can be formed.
Further, in this embodiment, the motor is not one in which one end or both ends of the excitation coil are connected to each other like the conventional Y-connection type motor or Δ-connection type motor, but each excitation coil Lu to Lw forming a three-phase brushless motor. Is a non-wired brushless motor 12 wound independently without being connected to each other, and therefore it is possible to individually control energization with each of the exciting coils Lu to Lw. A pseudo rectangular wave current including harmonics can be energized without any limitation. Therefore, the motor current waveform is a quasi-rectangular wave that is wide and rounded with respect to a sine wave similar to the counter electromotive voltage waveform.

  For this reason, since the output of the wireless motor 12 is output = current × voltage = torque × rotational speed, the effective value is remarkably improved as compared with the case of using a sine wave back electromotive force and drive current. It is possible to obtain a large output, a constant output with no torque ripple, and the motor drive torque when an abnormality occurs in the drive system of the inverter circuit, the motor harness, and the armature winding. The pulsation can be reduced compared to the case where the motor current waveform is a sine wave.

  On the other hand, in the conventional connection-type brushless motor, the counter electromotive voltage waveform can be a pseudo rectangular wave substantially the same as in the present embodiment, but a third harmonic component is passed through the armature winding of the motor. Therefore, the current waveform becomes a narrow pseudo-rectangular wave as shown in FIG. 16A, and the area becomes smaller and the effective value is better than the sine wave, but it is lower than that of this embodiment. Therefore, the output also decreases accordingly.

The reason why the third harmonic component cannot flow through the armature winding in the conventional connection type brushless motor is as follows.
That is, in a three-phase motor in which the excitation coil is Y-connected or Δ-connected, it is necessary to apply a voltage to each terminal of each excitation coil in order to pass a current to each phase. Each terminal voltage for flowing a primary component (sine wave) current is as follows.

Iu = I ₀ sin θ Vu = V ₀ sin θ
Iv = I ₀ sin (θ-2π / 3) Vv = V ₀ sin (θ-2π / 3)
Iw = I ₀ sin (θ + 2π / 3) Vw = V ₀ sin (θ + 2π / 3)
And each terminal voltage for applying the electric current which superimposed the 3rd harmonic is as follows.
Vu = V ₀ sin θ + V ₁ sin 3θ
Vv = V ₀ sin (θ-2π / 3) + V ₁ sin {3 (θ-2π / 3)}
= V ₀ sin (θ-2π / 3) + V ₁ sin (3θ-2π)
= V ₀ sin (θ-2π / 3) + V ₁ sin 3θ
Vw = V ₀ sin (θ + 2π / 3) + V ₁ sin {3 (θ + 2π / 3)}
= V ₀ sin (θ + 2π / 3) + V ₁ sin (3θ + 2π)
= V ₀ sin (θ + 2π / 3) + V ₁ sin 3θ

In this way, since the same voltage of V ₁ sin 3θ is applied to each terminal, the third harmonic current cannot flow in each of the U, V, and W phases. Similarly, in a five-phase motor, a fifth harmonic current cannot flow in each phase.
In order to calculate the phase current command value Iu~Iw obtained by superposing the third order harmonic and the fifth harmonic sine wave, a steering assist command value I _T input, the electrical angle of the non-connection type motor 12 Based on these, each phase current target value Iu to Iw is calculated so as to satisfy an output equation (output = torque × rotational speed = current × voltage) by internal calculation.

By the way, in order to maximize the performance of the wireless motor, it is necessary to superimpose the third harmonic on the sine wave. However, in the conventional vector control, the current of the 3n-order component can be flowed by the three-phase motor. As is impossible, the current command value of the 3n-phase component cannot be calculated by the vector control having the conventional 2-phase / 3-phase conversion.
Therefore, since the output of the motor is torque × rotational speed = current × voltage, the pseudo rectangular wave in which the third harmonic and the fifth harmonic are superimposed on the sine wave of the back electromotive force waveform and the current waveform is as follows. expressed.

Eu = E1 * sin (θ) + E3 * sin (3 * θ) + E5 * sin (5 * θ)
Ev = E1 * sin (θ-2 / 3 * PI) + E3 * sin (3 * (θ-2 / 3 * PI)) + E5 * sin (5 * (θ-2 / 3 * PI))
Ew = E1 * sin (θ + 2/3 * PI) + E3 * sin (3 * (θ + 2/3 * PI)) + E5 * sin (5 * (θ + 2/3 * PI))
Iu ＝ I1 * sin (θ) + I3 * sin (3 * θ) + I5 * sin (5 * θ)
Iv = I1 * sin (θ-2 / 3 * PI) + I3 * sin (3 * (θ-2 / 3 * PI)) + I5 * sin (5 * (θ-2 / 3 * PI))
Iw = I1 * sin (θ + 2/3 * PI) + I3 * sin (3 * (θ + 2/3 * PI)) + I5 * sin (5 * (θ + 2/3 * PI))
On the other hand, the motor output is

となる。出力が一定値であればトルクリップルがない。つまり、cos(6θ)を含む項が“０”になればトルクリップルのない一定出力が得られる。

It becomes. If the output is constant, there is no torque ripple. That is, when the term including cos (6θ) becomes “0”, a constant output without torque ripple can be obtained.

Since the counter electromotive voltage waveform is determined at the time of designing the motor, the first, third, and fifth order components E1, E3, and E5 of the counter electromotive voltage are known.
Therefore, if the 1st, 3rd and 5th order components I1, I3 and I5 of the current are determined so that the term including cos (6θ) is “0”, a phase current command value without torque ripple can be calculated.
The amplitude of the 1,3,5-order component I1, I3, I5 of the phase current command value is determined by the steering assist command value I _T, the ratio I1: I3: I5 is the back electromotive force 1,3,5-order component E1 , E3, E5 and cos (6θ) can be obtained in advance from the condition for setting the term including “0”.

As a specific example, FIG. 16B is an example in which a current waveform without torque ripple is obtained in advance with respect to the counter electromotive voltage waveform including the third and fifth harmonics shown in FIG. It is the same waveform as the characteristic line memorize | stored in the phase current command value calculation map shown to 12 (a)-(c).
More specifically, the phase current target value that satisfies the condition for setting the term including cos (6θ) to “0”, E1 · E3 + E3 · I3 + E5 · I1 = 0, is not uniquely determined.
Therefore, the current waveform and the counter electromotive voltage waveform are constrained under the condition that they have the same shape.
That is, I1 = aE1, I3 = aE3, I5 = aE5,
aE1 ・ E5 + aE3 ・ E3 + aE5 ・ E1 = 0
E3 ² = -2E1 ・ E5

If the motor counter electromotive voltage waveform is designed so as to satisfy the above relational expression and the same current waveform is output, a phase current target value without torque ripple can be obtained.
Since the back electromotive voltages E1 to E5 are proportional to the motor rotational speed ω, the equation of the motor output can be expressed as follows.
T = K1 / I1 + K3 / I3 + K5 / I5
Here, K1 to K5 are constants obtained in advance by the above-described procedure.

  As described above, in the first embodiment, the counter electromotive voltage waveform and the drive current waveform of the exciting coils Lu to Lw of the wireless motor 20 can both be a pseudo rectangular wave including the third harmonic, and the effective value can be It is possible to improve and obtain a large output. That is, since the third harmonic has a coefficient next to the first order component when the pseudo-rectangular wave is expanded in the Fourier series, the efficiency for raising the effective value by superimposing the third harmonic on the sine wave Will be the best, and a large output can be obtained.

Further, by using the wireless brushless motor 12, the inverter circuits 34 u, 34 v and 34 w are respectively connected to both ends of each exciting coil, and both ends of these exciting coils Lu, Lv and Lw are driven in opposite phases, thereby As described above, the inter-terminal voltages Vuab, Vvab, and Vwab of each exciting coil are expressed by the following equations (10), (11), and (12).
Vun = 2 × V ₀ × sin (ωt + α) (10)
Vvn = 2 × V ₀ × sin (ωt−2π / 3 + α) (11)
Vwn = 2 × V ₀ × sin (ωt−2π / 3 + α) (12)

On the other hand, in the case of a Y-connection motor having the same configuration, as shown in FIG. 17, an equivalent circuit has a neutral point voltage Vn at which one ends of the excitation coils Lu, Lv, and Lw are connected to each other, and Vn = 0 (V). Therefore, the inter-terminal voltages Vun, Vvn, and Vwn of the excitation coils Lu, Lv, and Lw are expressed by the following equations (13), (14), and (15).
Vun = V ₀ × sin (ωt + α) (13)
Vvn = V ₀ × sin (ωt−2π / 3 + α) (14)
Vwn = V ₀ × sin (ωt−2π / 3 + α) (15)

  Therefore, taking the exciting coil Lu as an example, the terminal voltages Vua and Vub and the inter-terminal voltage Vuab of the unconnected motor 12 according to the present invention are as shown in FIG. The terminal voltage Vu, terminal voltage Vv, inter-terminal voltage Vuv, and neutral point voltage Vn are as shown in FIG. On the other hand, the inter-terminal voltages Vuab, Vvab, Vwab of the unconnected motor 12 according to the present invention are as shown in FIG. 19A, and the coil end voltages Vun, Vvn, Vwn in the case of the conventional Y-connected motor are shown in FIG. As shown in 19 (b).

  As is apparent from FIGS. 18 and 19, when comparing the voltage amplitudes that can be applied to both ends of the exciting coil, the wireless motor 12 has the same effect as when the Y-wired motor is driven with twice the power supply voltage. Is obtained. Accordingly, when the battery voltage Vb is the same, the drive voltage of the exciting coils Lu to Lw can be improved in the wireless motor. Therefore, when the steering wheel 1 is steered rapidly, there is no shortage of voltage and optimal steering assistance is achieved. Force can be generated and smooth steering can be performed.

Similarly, in the case of a conventional Δ-connection type motor, an equivalent circuit is as shown in FIG. 20, and the inter-terminal voltages Vuv, Vvw, and Vwu are √3 times that of the Y-connection type motor, but the inter-terminal current is 1 / √3, and the coil current of the exciting coils Lu to Lw of the wireless motor 12 according to the present invention can effectively use the specified current as shown in FIG. Iuv, Ivw, Iwu and phase currents Iu, Iv, Iw are as shown in FIG. 21 (b), and each phase current Iu, Iv, Iw becomes 1 / √3 of the specified current. An effect equivalent to that when a Δ connection motor is energized with a motor current of √3 times can be obtained. As a result, since the wireless motor can improve the coil current of the exciting coil, the torque can be increased.
Therefore, when the Y-connection type motor and the Δ-connection type motor are equivalently converted, they can be expressed by the relational expressions shown in Table 1 below.

  Using this relational expression, the motor output and current characteristics when the equivalently exchanged Y-connection motor and Δ-connection motor are non-connection motors are the motor constants of the Y-connection motor. As shown in FIG. 22, the motor output characteristic when the motor is changed to the maximum rotation speed of the non-connection motor is controlled by the maximum current as shown by the solid line as compared to the rotation speed of the Y-connection motor shown by the broken line. As the torque decreases from the torque, the amount of increase in the rotational speed increases, and the rotational speed can be improved.

In addition, the motor output characteristics when the motor constant is changed to a non-wired motor with the motor constant of the Δ-connected motor, as shown in FIG. 23, the torque characteristics of the Δ-connected motor shown by the broken line are as follows. As the rotational speed decreases from the maximum rotational speed, the increase in torque increases, and the torque can be improved accordingly.
Furthermore, as shown in FIG. 24, the motor output characteristics when the motor is changed to a non-connection motor with an intermediate motor constant between the Y-connection motor and the Δ-connection motor, as shown in FIG. 24, In the case of a wireless motor, both rotational speed and torque can be improved as shown by the solid line.

For this reason, when the non-wired motor 12 is applied to the electric power steering apparatus, a desired motor output characteristic can be obtained by setting the motor constant according to the required characteristic.
In the above embodiment, the case where the energization control of the excitation coils Lu to Lw of the wireless brushless motor 12 is performed using the inverter circuits 34 u to 34 w has been described. However, the present invention is not limited to this. As shown in FIG. 5, even if the inverter circuits 34a and 34b are individually connected to both ends of the exciting coils Lu to Lw, the same effect as that of the above embodiment can be obtained.

  In the above embodiment, the case where the addition circuits 42u to 42w, the bias circuit 43, and the voltage divider / filter circuit 44 are provided as the abnormality detection circuit 41 has been described. However, the voltage divider / filter circuit 44 is omitted, Instead of this, the value of the threshold value Vms of the abnormality detection process of FIG. 13 may be changed. Further, when the abnormality detection at the time of starting is not performed, the bias circuit 43 may be omitted.

Furthermore, in the above-described embodiment, the case where the moving average value of the addition voltage Vjs is calculated in the abnormality detection process of FIG. 13 is described, but the present invention is not limited to this, and a predetermined number of addition voltages are averaged. A simple averaging process may be performed.
Furthermore, in the above embodiment, in the abnormality detection process, the amount of change for each timer interrupt cycle (sampling cycle) of the added voltage Vjs is calculated without calculating the average value, and this amount of change is greater than or equal to a predetermined value. Sometimes, it may be determined that there is an abnormality.

Furthermore, in the above-described embodiment, the case where the induced voltage waveform and the drive current waveform of the wireless motor are the same pseudo-rectangular wave shape has been described. However, the present invention is not limited to this. Even if only the amplitude is changed without changing the phase and shape of the current waveform, the same effect as that of the first embodiment can be obtained.
In the above embodiment, the case where the pseudo-rectangular wave is formed by superimposing the third and fifth harmonics on the sine wave has been described. However, the present invention is not limited to this. Waves may be superimposed in an arbitrary combination, or a current waveform of only a sine wave without harmonics superimposed may be used. In this case, the phase current calculation map may be a three-phase sine wave.

Furthermore, although the case where the drive control unit has a simple configuration has been described in the above embodiment, the present invention is not limited to this, and the current of the vector control d and q components using the excellent characteristics of vector control is not limited thereto. After the command value is determined, this current command value is converted into each phase current command value corresponding to each exciting coil Lu to Lw, thereby calculating the phase current target values I _TU ^* , I _TV ^* and I _TW ^* . Alternatively, all may be performed by vector control.

Furthermore, in each of the above embodiments, the case where the present invention is applied to a non-wired three-phase brushless motor has been described. However, the present invention is not limited to this, and a plurality of N (N is an integer of 3 or more) phases. It can also be applied to other brushless motors or other motors.
In each of the above embodiments, the case where the present invention is applied to the electric power steering apparatus has been described. However, the present invention is not limited to this, and the present invention is applied to any apparatus having another drive motor. Can do.

1 is a system configuration diagram illustrating a first embodiment when the present invention is applied to an electric power steering apparatus. It is a characteristic view which shows the output characteristic of the steering torque detection value output from a steering torque sensor. It is sectional drawing which shows a non-connection type motor. It is a perspective view which shows the rotor of FIG. It is a block diagram which shows the drive circuit of a non-connection type motor. It is a circuit diagram which shows the equivalent circuit of a non-connection type motor. It is a block diagram which shows the abnormality detection circuit of a non-connection type motor. It is a time chart with which it uses for description of operation | movement of the abnormality detection circuit when an inverter circuit is normal. It is a time chart with which it uses for description of operation | movement of the abnormality detection circuit at the time of ON abnormality of an inverter circuit. It is a flowchart which shows an example of the steering control processing procedure performed with a microcomputer. It is a characteristic diagram which shows a steering assistance command value calculation map. It is a characteristic diagram which shows a phase current command value calculation map. It is a flowchart which shows an example of the abnormality detection process procedure performed with a microcomputer. It is a time chart which shows the phase current and torque in a low speed steering area | region. It is a time chart which shows the phase current and torque in a high-speed steering area | region. It is a time chart which shows a motor current waveform and a counter electromotive voltage waveform. It is a circuit diagram which shows the equivalent circuit of the conventional Y connection type motor. It is a characteristic diagram which shows the terminal voltage of a non-connection type motor, and the terminal voltage of a Y-connection type motor. It is a characteristic diagram which shows the coil both-ends voltage of a non-connection motor, and the coil both-ends voltage of a Y connection motor. It is a circuit diagram which shows the equivalent circuit of the conventional (DELTA) connection type motor. It is a characteristic diagram which shows the coil current of a non-connection type motor, and the phase current and coil current of (DELTA) connection type motor. It is a characteristic diagram which shows a motor characteristic at the time of setting the motor constant of a non-connection type motor to the motor constant of a Y connection type motor. It is a characteristic diagram which shows a motor characteristic at the time of setting the motor constant of a non-connection type motor to the motor constant of (DELTA) connection type motor. It is a characteristic diagram which shows a motor characteristic at the time of setting the motor constant of a non-connection type motor to the middle of the motor constant of a Y connection type motor and a (DELTA) connection type motor. It is a circuit diagram which shows the other example of an inverter circuit.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Steering wheel, 2 ... Steering shaft, 3 ... Steering torque sensor, 8 ... Steering gear, 10 ... Steering assist mechanism, 12 ... Wireless motor, 15 ... Drive control circuit, B ... Battery, IG ... Ignition key, 16 DESCRIPTION OF SYMBOLS ... Vehicle speed sensor, 17u-17w ... Motor current detection part, 18 ... Microcomputer, 19 ... Gate drive circuit, 20 ... Rotor, 21 ... Housing, 27 ... Rotor core, 28 ... Rotor magnet, 31 ... Stator, 33 ... Excitation coil, Lu, Lv, Lw ... three-phase excitation coils, 34u to 34w, 34a, 34b ... inverter circuit, 35 ... phase detector, 41 ... abnormality detection circuit, 42u-42w ... addition circuit, 43 ... bias circuit, 44 ... partial pressure Also filter circuit, 50 ... electric angle converter, 51 ... motor angular velocity converter

Claims (10)

  Wireless brushless motor having a rotor having a permanent magnet and a stator in which a plurality of N-phase armature windings are arranged independently of each other without being connected to each other, and each armature winding An inverter circuit that is individually connected to both ends of the wire and supplies a drive signal to each armature winding, a drive control unit that drives and controls the inverter circuit, and current and voltage abnormality of each armature winding And detecting the current / voltage abnormality of one armature winding by the abnormality detection unit, the non-connection type brushless motor is subjected to a braking force generated by the non-connection type brushless motor. A drive control device for a wireless motor, comprising: an abnormality control unit that drives while suppressing.   Wireless brushless motor having a rotor having a permanent magnet and a stator in which a plurality of N-phase armature windings are arranged independently of each other without being connected to each other, and each armature winding An inverter circuit that is individually connected to both ends of the wire and supplies a drive signal to each armature winding, a drive control unit that drives and controls the inverter circuit, and current and voltage abnormality of each armature winding And detecting the current / voltage abnormality of one armature winding by the abnormality detection unit, the non-connection type brushless motor is subjected to a braking force generated by the non-connection type brushless motor. When the abnormality control unit that drives while suppressing, the rotation speed detection unit that detects the rotation speed of the wireless brushless motor, and the abnormality detection circuit detects the current / voltage abnormality of the armature winding, the rotation speed A drive control device for a wireless motor, comprising: a motor speed suppressing portion that suppresses the rotational speed of the wireless brushless motor when the motor rotational speed detected at the exit is equal to or higher than a set speed. .   When the abnormality detection unit detects a current / voltage abnormality of one armature winding, the abnormality control unit stops only drive control of the drive element of the inverter circuit corresponding to the armature winding. The drive control device for a wireless motor according to claim 1, wherein the drive control device is configured as described above.   The abnormality detection unit is configured to detect an abnormality of a drive element constituting the inverter circuit and an abnormality of a motor harness between the inverter circuit and an armature winding of the wireless brushless motor. The drive control apparatus for a non-wired motor according to any one of claims 1 to 3.   The drive control unit is configured to form a control signal for the inverter circuit based on a current command value corrected by superimposing a harmonic component on a phase current command value for each armature winding of the wireless motor. The drive control device for a wireless motor according to any one of claims 1 to 4, wherein the drive control device is a wired motor.   An electric power steering device using the drive control device for a wireless motor according to any one of claims 1 to 5.   A steering torque detector for detecting steering torque, a rotor having a permanent magnet disposed therein, and a stator that is disposed independently of each other without connecting a plurality of N-phase armature windings facing the rotor; A wireless brushless motor that generates a steering assist force for a steering system, an inverter circuit that is individually connected to both ends of each armature winding and supplies a drive signal to each armature winding, and the steering torque A drive control unit that drives and controls the inverter circuit based on the steering torque detected by the detection unit, an abnormality detection unit that individually detects current / voltage abnormality of each armature winding, and one abnormality detection unit An abnormality control unit that drives the wireless brushless motor while suppressing the braking force generated by the wireless brushless motor when an armature winding current / voltage abnormality is detected; Electric power steering apparatus according to claim.   A steering torque detector for detecting a steering torque; a rotor having a permanent magnet; and a stator that is arranged independently of the rotor so as to face the rotor without connecting a plurality of N-phase armature windings to each other. Based on a connectionless brushless motor, an inverter circuit that is individually connected to both ends of each armature winding and supplies a drive signal to each armature winding, and the steering torque detected by the steering torque detector A drive control unit that drives and controls the inverter circuit, an abnormality detection unit that individually detects current / voltage abnormality of each armature winding, and an abnormality detection unit that detects current / voltage abnormality of one armature winding And detecting the rotation speed of the wireless brushless motor and an abnormal time control unit that drives the wireless brushless motor while suppressing the braking force generated by the wireless brushless motor. When a current / voltage abnormality of the armature winding is detected by the rotation speed detection unit and the abnormality detection circuit, when the motor rotation speed detected by the rotation speed detection unit is equal to or higher than a set speed, the wireless brushless motor An electric power steering apparatus comprising: a motor speed suppressing unit that suppresses the rotation speed.   The drive control unit includes a phase current command value calculation unit that calculates a phase current command value for each armature winding based on the steering torque, and a motor current detection unit that detects a phase current of each armature winding. And an electric current steering unit that controls a driving current for each armature winding based on the phase current command value and the phase current. . An electrical angle detection circuit that detects an electrical angle of the wireless motor, and the phase current command value calculation unit is configured to generate a harmonic corresponding to each armature winding of the wireless motor based on the electrical angle. A phase current command value calculation unit for calculating a phase current command value corresponding to a counter electromotive voltage including the phase current command value and a phase current target value for each armature winding based on the phase current command value and the steering torque detection value The electric power steering apparatus according to claim 9, further comprising: a phase current target value calculation unit that performs the operation.

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